WO1995016303A1

WO1995016303A1 - Improved quadrature modulator

Info

Publication number: WO1995016303A1
Application number: PCT/US1994/008040
Authority: WO
Inventors: Paul Anthony Denny
Original assignee: National Semiconductor Corporation
Priority date: 1993-12-06
Filing date: 1994-07-20
Publication date: 1995-06-15

Abstract

In a quadrature modulation system which includes an adder for adding the phase shifted signals and an amplifier, a low pass or band pass filter is interposed between the adder and amplifier to eliminate the unwanted sideband associated with third order distortion in the amplifier. This unwanted sideband is located at a frequency equal to the carrier frequency less three times the data transmit frequency. Thus, it is proportionally very close to the carrier frequency and cannot be easily filtered by conventional techniques.

Description

IMPROVED QUADRATURE MODULATOR

FIELD OF THE INVENTION

This invention relates to an improved quadrature modulator for use in single sideband wireless communications systems.

BACKGROUND OF THE INVENTION

It is well known in wireless communication to mix (multiply) a data signal with a carrier signal. The data signal could be a voice signal. These two signals are illustrated in the frequency domain in Fig. 1A, where f_c represents the frequency of the carrier signal and f_s represents the frequency of the data signal. (It will be understood that the data signal is normally a band of frequencies, but for simplicity it is represented here by a single frequency f_s.) Mixing the two signals results in the spectrum illustrated in Fig. IB, which shows the carrier frequency f_c, an upper sideband at a frequency f_h = f_c + f_s, and a lower sideband at a frequency f_j_ = f - f . This flows from the following mathematical equation:

sin (c-±-t) x sin (ω₂t) = sin (ω_λ - ω₂)t + sin (ω± + ω₂)t (1)

where ω_x and ω₂ represent two angular frequencies and t represents time. Since the same information is contained in the upper and lower sidebands, one of the sidebands is often filtered out, leaving a single sideband transmission. One way to achieve this filtering is through a quadrature modulation system, which has as its basis the following mathematical equation: [ sin α^t) x s in ( ω₂t ) ] + [ cos ( ω₁t ) x ( cos ( ω₂t) ]

= cos ( ω_λ - ω₂ ) t ( 2 )

Since the cosine represents the sine shifted by 90 degrees, this means in effect that if a duplicate signal, phase shifted by 90 degrees, is mixed with a duplicate carrier, also phase shifted by 90 degrees, and then added to the original modulated signal, the upper sideband will be eliminated. The quantity on the righthand side of the equation cos (ω - _ω2)t, is the same as sin (ω_λ - ω₂)t in equation, (1) phase-shifted by 90 degrees. The transformations required by equation (2) are accomplished in. an arrangement illustrated in Fig. 2A, which is typically less expensive than attempting to filter one of the sidebands with a conventional L-C filter. In Fig. 2A, SI represents the data signal which is mixed with a carrier CI of frequency f_c in a mixer 20. CI is shifted by 90 degrees in a phase shifter 21 to form a signal designated as CQ, and SI is phase-shifted 90 degrees in a phase shifter 22 to form a signal designated as SQ. The signals SQ and CQ are mixed in a mixer 23. The output of mixers 20 and 23 are summed in an adder 24 and amplified in an amplifier 25. The output of amplifier 25 is connected to a transmit antenna 26. The junction between adder 24 and amplifier 25 is designated as node A, and the junction between amplifier 25 and antenna 26 is designated as node B.

Since the respective outputs of mixers 20 and 23 are typically currents, adder 23 may be simply a wire junction where the current signals are summed in accordance with Kirchoff's law.

The circuit shown in Fig. 2A cancels the upper sideband. The lower sideband may be canceled by using the following equation: [ sin (ω_αt) x cos (ω₂t) ] + [ cos (ω₂t) x sin (ω₂t) ]

= sin (ω_α + ω₂) t (3 )

The corresponding circuit is illustrated in Fig. 2B. In reality, the signals I, Q and f_c are not perfect sinusoidal waveforms. Amplifiers, for example, may be nonlinear and may be driven at levels where their outputs are clipped. In many cases, these signals may closely approximate a square wave. This leads to the creation of odd harmonics, with the sideband being on opposite sides of successive harmonics. For the circuit of Fig. 2A this is illustrated in Fig. 3, where the sideband of the fundamental frequency f_c is at a frequency f_c - f_s, the sideband for the third order harmonic is at a frequency 3f_c + f_s, the sideband for the fifth order harmonic is at a frequency 5f_c - f_s, and so forth.

Amplifier 25 normally creates what is referred to as third order distortion. It is known that this creates an unwanted sideband at a frequency f_u which is separated by 3f_B from the carrier frequency, f_c and is on the opposite side of the carrier frequency from the uncanceled sideband.

This can be derived briefly as follows. Third order distortion refers to the cubing of a signal. If signals having frequencies ω-^ and ω are cubed (i.e., (cos ω± + cos ω₂t)³) , one of the terms that emerges is cos ( 2ω - ω₂)t. If U_j represents the lower sideband f₁ = f_c - f and ω₂ represent the sideband near the third harmonic 3f_c + f_s, then the following results:

cos (2ω_x - ω₂)t = cos [2 (f_c - f_s) - (3f_c + f_s]t

This can be simplified to:

cos (2f_c - 2f_s - 3f_c - f_s)t = cos (-f_c - 3f_s)t The signal represented by cos (-f_c - 3f_s)t is in reality the same as the signal cos (f_c + 3f_s)t phase-shifted by 180 degrees.

Thus, cubing the sidebands near first and third harmonics, respectively, yields an unwanted sideband f_u = f_c + 3f_s which is illustrated in Fig. IC. Since f_c is typically on the order of 2000 MHz, and f_s is typically on the order of 1 MHz, it is practically impossible to filter out the unwanted sideband. While the unwanted sideband may be eliminated by making sure that all of the amplifiers and other components are operating within their linear ranges, this reduces the power efficiency of the system and limits the possibility of obtaining noise suppression from the carrier frequency ports of the mixer and output stage.

SUMMARY OF THE INVENTION

In accordance with this invention, the unwanted sideband which is located at a separation equal to 3f_s from the carrier frequency in a quadrature modulation system is eliminated by including a filter between the adder and the amplifier. The filter may be either a low pass or a band pass filter. The parameters of the filter are set, in accordance with known techniques, to eliminate the sidebands associated with the third and higher order harmonics.

BRIEF DESCRIPTION OF THE DRAWINGS

Figs. 1A and IB illustrate the results of mixing a data signal with a carrier signal. Fig. IC illustrates the position of the unwanted sideband which occurs from third order distortion in the amplifier.

Figs. 2A and 2B illustrate conventional quadrature modulation systems.

Fig. 3 illustrates the sidebands of the fundamental and third and fifth order harmonics.

SUBSTITUTE SHEET {RULE 26) Fig. 4 illustrates a quadrature modulation system in accordance with this invention.

Fig. 5 illustrates the frequency response of the filter in the arrangement of Fig. 4. Fig. 6 illustrates a circuit diagram for the mixers and adder shown in Fig. 4.

Figs. 7A-7D illustrate waveforms useful in explaining the operation of the circuitry shown in Fig. 6.

Figs. 8A and 8B illustrate circuit diagrams of a low pass filter.

Fig. 8C illustrates a circuit diagram of a band pass filter.

DESCRIPTION OF THE INVENTION

Fig. 4 illustrates an embodiment in accordance with the invention. The quadrature modulation system is similar to that illustrated in Fig. 2. A filter 40, however, has been included at node A between adder 24 and amplifier 25. Filter 40 may be a low pass filter having a frequency response illustrated by curve 50 in Fig. 5 or a band pass filter having a frequency response illustrated by curve 51 in Fig. 5. As is evident from Fig. 5, filter 40 should block all frequencies above a frequency that is less than or equal to the frequency of the sideband associated with the third order harmonic (3f_c) . Connecting filter 40 between adder 24 and amplifier 25 allows filter 40 to be a relatively inexpensive filter as compared with the sharp, expensive filter that would be required to filter out the unwanted sideband at f_u = f_c + 3f_s with a filter connected at the output of amplifier 25. Fig. 6 illustrates a circuit diagram of mixers 20 and 23, which are of a design known to those skilled in the art. Using mixer 20 for illustration (mixer 23 is identical) , a current source 200 is connected to a pair of current paths which extend through a resistor 201A and a transistor 202A, and through a resistor 201B and a

SUBSTITUTE SHEET {RULE 26) transistor 202B, respectively. The base of transistor 202A is connected to the data signal SI; the base of transistor 202B is grounded.

The collectors of transistors 202A and 202B are connected to two additional pairs of current paths. Thus, the collector of transistor 202A is connected to current paths containing transistors 203A and 203B, and the collector of transistor 202B is connected to current paths containing transistors 204A and 204B. The bases of transistors 203A and 204B are connected to the carrier signal CI; the bases of transistors 203B and 204A are grounded.

The operation of mixer 20 is illustrated in Figs. 7A- 7D. The output of current source 200 is represented by I, which is split into currents I_a and I_b through transistors 202A and 202B, respectively. By Kirchoff's law, 1 = 1 + I_b. Fig. 7B shows an illustrative signal SI that is applied to the base of transistor 202A, and Fig. 7A shows the resulting currents I_a and I_b. For example, as SI increases, transistor 202A conducts more current (I_a) and conversely transistor 202B conducts less current (I_b) .

Transistors 203A, 203B, 204A and 204B operate essentially as switching transistors, since there are no resistors comparable to resistors 201A and 201B connected in series with them. When carrier signal CI is high

(above ground) transistor 203A conducts practically all of current I_a through a resistor 206, and transistor 204B conducts practically all of current I_b through a resistor 205. 205 and 206 are load resistors that convert the output current to an output voltage. 201A and 201B are "degeneration" resistors that cause transistors 202A and 202B to switch linearly with input voltage SI. Conversely, when carrier signal CI is low (below ground) transistor 203B conducts practically all of current I_a through resistor 205 and transistor 204A conducts practically all of current I_b through resistor 206. This is illustrated in Figs. 7C and 7D, which show that as signal CI passes across the zero voltage level, the current directed through resistor 206 (denoted by the solid line) and resistor 205 (denoted by the dashed line) 5 shifts abruptly between I_a and I_b (Fig. 7A) . The current signals shown in Fig. 7D are reflected as voltages across resistors 205 and 206.

Mixer 23 is similar to mixer 20 except that the inputs are the quadrature signals SQ and CQ. Mixer 23

10 contains transistors 209A, 209B, 210A and 210B which correspond to transistors 203A, 203B, 204A and 204B, respectively, in mixer 20. Adder 24 (Fig. 4) includes lines 207 and 208. Line 207 ties together the collectors of transistors 203A, 204A, 209B and 210B, and line 208

15 ties together the collectors of transistors 203B, 204B, 209A and 210A. Thus, mixers 20 and 23 have differential outputs (i.e., the currents through nodes 211 and 212 for mixer 20, and the currents through nodes 213 and 214 for mixer 23) , and these currents are summed on lines 207 and

20 208. The voltages at nodes a and b represent the summed differential outputs of mixers 20 and 23.

Filter 30 may be either a low pass filter or a band pass filter. Figs. 8A and 8B illustrate circuit diagrams for a low pass filter showing the connections to nodes a,

25 b and c in Fig. 6. Fig. 8C illustrates a circuit diagram for a band pass filter. The filters shown in Figs. 8A-8C are simple, straightforward circuits that will be familiar to persons skilled in the art. Numerous alternative forms of low pass and band pass filters may be used for filter

30 40, provided that they filter out substantially all frequencies above 3f_c.

While specific circuits and embodiments have been described, it will be understood that the scope of this invention is to be limited only by the following claims.

Claims

CLAIMSI claim:

1. A quadrature modulation system comprising: a first mixer for mixing a first data signal and a first carrier signal, said first carrier signal having a frequency f_c; a second mixer for mixing a second data signal and a second carrier signal, said second carrier signal being related to said first carrier signal by a phase difference of 90 degrees; an adder having inputs connected to the outputs of said first and second mixers; an amplifier connected to the output of said adder; and a filter interposed between said adder and said amplifier, said filter adapted to block signals having frequencies equal to or greater than approximately 3f_c.

2. The quadrature modulation system of Claim 1 wherein said filter is a low pass filter.

3. The quadrature modulation system of Claim 1 wherein said filter is a band pass filter.

4. The quadrature modulation system of Claim 1 wherein said adder comprises a line for summing currents generated by said first and second mixers.

5. The quadrature modulation system of Claim 1 wherein each of said mixers has a differential output.

6. A method of filtering out a sideband in a quadrature modulation system, said method comprising the steps of:

(a) mixing a carrier signal CI and a data signal SI;

(b) mixing a quadrature carrier signal CQ and a quadrature data signal SQ;

(c) adding the signal produced by step (a) to the signal produced by step (b) ;

(d) filtering the signal produced by step (c) to remove frequencies greater than 3f_c, wherein f_c is the frequency of carrier signals CI and SI.

7. The method of Claim 6 comprising the further step of amplifying the signal produced by step (d) .