WO1992008297A1 - An apparatus and method for varying a signal in a transmitter of a transceiver - Google Patents

An apparatus and method for varying a signal in a transmitter of a transceiver Download PDF

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Publication number
WO1992008297A1
WO1992008297A1 PCT/US1991/007012 US9107012W WO9208297A1 WO 1992008297 A1 WO1992008297 A1 WO 1992008297A1 US 9107012 W US9107012 W US 9107012W WO 9208297 A1 WO9208297 A1 WO 9208297A1
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WO
WIPO (PCT)
Prior art keywords
signal
varying
transfer function
transceiver
transmitter
Prior art date
Application number
PCT/US1991/007012
Other languages
French (fr)
Inventor
Stephen V. Cahill
Original Assignee
Motorola, Inc.
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Motorola, Inc. filed Critical Motorola, Inc.
Publication of WO1992008297A1 publication Critical patent/WO1992008297A1/en
Priority to GB9213179A priority Critical patent/GB2254973B/en

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/005—Control of transmission; Equalising
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03C—MODULATION
    • H03C3/00—Angle modulation
    • H03C3/38—Angle modulation by converting amplitude modulation to angle modulation
    • H03C3/40—Angle modulation by converting amplitude modulation to angle modulation using two signal paths the outputs of which have a predetermined phase difference and at least one output being amplitude-modulated
    • H03C3/406—Angle modulation by converting amplitude modulation to angle modulation using two signal paths the outputs of which have a predetermined phase difference and at least one output being amplitude-modulated using a feedback loop containing mixers or demodulators
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/32—Modifications of amplifiers to reduce non-linear distortion
    • H03F1/3241—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits
    • H03F1/3247—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits using feedback acting on predistortion circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
    • H04L27/20—Modulator circuits; Transmitter circuits
    • H04L27/2032—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner
    • H04L27/2053—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases
    • H04L27/206—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers
    • H04L27/2067—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers with more than two phase states
    • H04L27/2071—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers with more than two phase states in which the data are represented by the carrier phase, e.g. systems with differential coding

Definitions

  • the present invention relates generally to radio systems having a transmitter and a receiver and, more particularly, to an apparatus and method for linearization and gain control of the transmitter's power amplifier in a TDM system by selectively sampling the amplifier's output signal with the receiver, processing the signal to determine a correction value, applying the correction value to the amplifier's input signal or gain stage to vary the output signal.
  • a radio communications system is comprised, at a minimum, of a transmitter and a receiver.
  • the transmitter and the receiver are interconnected by a radio-frequency channel to permit transmission of an information signal therebetween.
  • a transceiver will generally include both a receiver and a transmitter.
  • the transmitter portion of the transceiver will generally include a radio-frequency (RF) power amplifier for increasing the power of the transmitted signal.
  • RF power amplifiers generally have nonlinear transfer function characteristics relating their input and output signals over a portion of their output power operating range. This nonlinearity appears as an input-level-dependent gain over a portion of the operating range of input level.
  • an RF channel is shared (time-division- multiplexed) among several subscribers attempting to access the radio system in certain ones of various time-division- multiplexed time slots. This permits transmission of more than one signal at the same frequency, using the sequential time-sharing of a single channel by several radio telephones.
  • the time slots are arranged into periodically repeating frames, thus, a radio communication of interest may be periodically discontinuous wherein unrelated signals are interleaved with signals transmitted in other time slots.
  • USDC is called ⁇ /4 - shift quadrature differential phase-shift keying (QDPSK).
  • QDPSK ⁇ /4 - shift quadrature differential phase-shift keying
  • a ⁇ /4 - shift QDPSK modulation scheme speech signals are encoded into a serial data stream.
  • the serial data stream is demultiplexed into two secondary data streams and processed to generate discrete samples in time of the in-phase (I) and quadrature (Q) signal components of a QDPSK constellation.
  • the discrete signal sample is used in conventional digital signal processing (DSP), operations.
  • Linear modulation schemes, such as ⁇ /4 - shift QDPSK generally have narrow bandwidths and non-constant signal envelopes. The narrow bandwidth optimizes the efficiency of the radio frequency spectrum.
  • nonlinear RF power amplifiers introduce distortion components which tend to spread the spectrum thus eliminating any spectrum frequency advantage.
  • Low pass filtering of the transmit signal to achieve the narrow bandwidth causes the signal envelope to vary thus eliminating the full use of the amplifier's linear region.
  • the nonlinear operating region may also be used. If the nonlinear region is utilized, the amplifier could operate at a higher output level and then be limited by operating supply voltage, bias current or heat dissipation. A daunting challenge then is to provide a transmitter RF power amplifier which is both linear and power efficient.
  • USDC modulation also results in variation of the transmitted signal envelope, which results in erroneous measurement of average signal power by the peak-detecting diode detector.
  • One previous approach taken to correct the problems associated with non-linear transmit power amplifiers uses a pre-defined complex-valued correction look-up table. In-phase (I) and quadrature (Q) components of the modulation signal are used as pointers into the look-up table to determine a gain correction pair. The gain correction pair is applied to the I and Q component signal values prior to amplification to pre- distort the transmit signal. Predistorting the transmit signal cancels the distortion added by the RF power amplifier caused by nonlinearity. The result is a transmitted output signal nearly linear with the input signal over the amplifier's non linear region.
  • the correction factor is applied to the input signal resulting in an effective linear output signal from the power amplifier.
  • Problems associated with this approach include a delay caused by filtering the input signal before it is coupled to the transmitter amplifier which creates instability in the loop, and the need to maintain controlled phase shift through the transmitter amplifier in spite of variations in load impedance, drive level and power supply voltage.
  • the transceiver includes an antenna, from which the transmitter of the transceiver transmits a first signal from the antenna and in which a second signal is induced.
  • a selector selects between the first signal and the second signal and when the first signal is selected a selected signal is produced.
  • the selected signal is processed to produce a processed signal which is used to produce the varied first signal.
  • FIG. 1 is a block diagram of a transceiver including a transmitter, a receiver and a portion of a signal processor which may employ the present invention.
  • FIG. 2 is a block diagram of a portion of a receiver demodulator showing switched alternate receiving paths.
  • FIG. 3 is a block diagram of a signal processor which may employ the present invention.
  • FIG. 4 is a graph showing three function curves relating an input signal to an output signal of a power amplifier.
  • the preferred embodiment of the present invention samples the output of the transmitter's power amplifier with the receiver during a transmit time slot in a TDM signal.
  • the demodulated power amplifier output signal is compared with the input to the modulator to create the look-up table correction values.
  • the correction values are applied to each input signal level of the modulator.
  • This table is updated as the transmitter operates to correct for changes in the amplifier's transfer function characteristics.
  • the path from the transmitter's power amplifier through the receiver and back to the power amplifier determines a feedback path for linearizing the power amplifier thus permitting efficient operation.
  • FIG. 1 shows a block diagram of a transceiver including a transmitter, a receiver and a signal processor portion.
  • a typical received signal a received time slot in a TDM signal, is coupled through an antenna 101 into a receiver bandpass filter 103.
  • the filtered response 104 is quadrature demodulated in the receiver quadrature demodulator 105.
  • the demodulated signal is composed of In-phase (I) and Quadrature-phase (Q) components.
  • the I and Q quadrature components are coupled to a digital signal processor 107 through a sampling Analog-to-Digital (A/D) converter 109.
  • A/D Analog-to-Digital
  • the transmitter quadrature modulator 111 combines the I"IN and Q"IN signal components into a transmitter excitation signal 112.
  • the excitation signal 112 is amplified with power amplifier 115 and further coupled to the antenna 101 through a transmitter bandpass filter 117.
  • the receiver bandpass filter 103 and the transmitter bandpass filter 117 each pass a different frequency range to isolate the receiver and transmitter portions of the transmitter.
  • a coupler 119 couples a portion of the output signal of the power amplifier 115 to the receiver quadrature demodulator 105 through the attenuator 120.
  • the purpose of the attenuator 120 is to reduce the signal level out of the coupler to levels within the dynamic range of the receiver in a controlled manner.
  • a receiver control signal 121 from the digital signal processor 107 configures the receiver quadrature demodulator 105 to receive a signal either from the antenna 101 or the output of the transmit amplifier 115. Similar receiver circuitry in the receiver quadrature demodulator 105 is used to demodulate signals from both sources.
  • FIG. 2 shows a portion of the receiver quadrature demodulator which selects between two candidate signals 104 and 106 depending on the state of the receiver control signal 121.
  • the receiver control signal 121 activates switch 405 to select whether the received signal is coupled from the antenna 101 or from the transmitter amplifier 115.
  • switch contact 410 When the switch contact 410 is coupled to terminal 406, mixer 407 and local oscillator 409 convert the transmit signal 106 from power amplifier 115 to the normal IF frequency processed by mixers 401 and 403.
  • mixer 417 and local oscillator 415 convert a received carrier signal 104 from the antenna 101 to the normal IF frequency processed by mixers 401 and 403.
  • Separate local oscillators 409 and 415 are used because of different transmit and receive frequencies.
  • the state of the receiver control signal 121 is determined using timing information recovered from the normal receive signal.
  • the receiver control signal 121 determines which signal will be received based upon the operating position of the transmit and receive time slots in a TDM signal for the transceiver.
  • the receive control signal 121 instructs the receiver quadrature demodulator 105 to receive a carrier signal 104 from the antenna 101.
  • the receive control signal 121 instructs the receiver quadrature demodulator 105 to receive the transmit signal from the power amplifier 115.
  • the receiver quadrature demodulator 105 receives the power amplifier's output signal 106 from the coupler 119 coupled through the attenuator 120.
  • the receiver quadrature demodulator 105 demodulates the output signal into its quadrature components which are further coupled to the digital signal processor 107 through an A/D converter 109 resulting in component signals I ou t and Q ou t.
  • Component signals, I ou t and Q 0 ut.. are processed in the DSP 107 to adjust the input signal of the power amplifier 115.
  • the adjusted input signals, I'IN and Q ⁇ N are coupled from the DSP 107 to the D/A converter 113 as I"IN and Q"IN. and further to the transmitter quadrature modulator 111.
  • the transmitter quadrature modulator 111 modulates the quadrature signal components, I"IN and Q"IN > into a transmitter excitation signal which is further amplified by the power amplifier 115.
  • the input signal level to the power amplifier 115 is adjusted with a correction value which results in an output signal that is linear with the original input signal before adjustment.
  • the output of the power amplifier 115 is coupled through a transmitter bandpass filter 117 into the antenna 101.
  • the gain control signal, H determined by the DSP 107, is coupled through the D/A converter 123 to the power amplifier 115.
  • the gain nonlinearity of power amplifier 115 is compensated for by adjusting the input signal level presented to the power amplifier 115 by modulator 111 based on what the energy of the input signal is, to compensate for the gain nonlinearity of the power amplifier.
  • the nonlinearity of the power amplifier may be described as a variation of gain with the input signal level.
  • the correction value is determined by sampling the output signal of power amplifier 115.
  • the input and sampled signals in DSP 107 update the correction value used for each input signal level.
  • Factors that cause operation in the amplifiers nonlinear region include large-signal gain variations, saturation and cutoff effects, temperature, input signal level variations, and the disturbance of internal bias points by reflected energy at the transmit amplifier output due to the voltage standing wave ratio (VSWR) of the antenna. It is also desirable to chose to operate the amplifier in the non-linear region because the amplifier is more efficient in the non-linear region.
  • VSWR voltage standing wave ratio
  • an efficient power amplifier 115 may be applied to both fixed and mobile transceivers using a power amplifier. Specific advantages in a mobile or portable transceiver include: longer talk time, smaller battery size, cooler operation and increased reliability.
  • FIG. 3 there is shown a block diagram of a signal processor which may employ the present invention.
  • the following text describes the general process of how the amplifier's output signal, IoUT and Qou ⁇ » is processed in the DSP 107 to determine an adjusted input signal, I'IN and Q'IN-
  • the receiver selector 215 determines the state of the receiver control signal 121.
  • the receiver selector 215 makes its decision based on the time state of the TDMA received channel.
  • the data source 207 provides the Ii n and Qi n quadrature component signals for transmitting.
  • the TDM time clock 206 synchronizes the data source's transmitting activities and the selection of modes for the dual-mode receiver quadrature demodulator 105 of FIG. 1.
  • the quadrature data is coupled to the linearizer 211 as I ⁇ n and Q ⁇ n through a low pass filter 213.
  • the linearizer 211 has four outputs: Pi n and K(Pi n ) coupled to the inverse transfer function determiner 205 and I'i n and Q'i n coupled to the D/A Converter 113.
  • the linearizer 211 determines Pi n from input signals, IlN and QIN, looks up a K(PIN), a correction value, in the look ⁇ up table and multiplies _i n and Qj n each by the K(PIN) which results in a gain-adjusted transmitter excitation input signal, I'in and Q'in, coupled to modulator 111.
  • the adjusted input signal, I'i n and Q' ⁇ n provides an amplified signal at the output of the transmit amplifier 115 which is linear with the input signal Ii n and Q ⁇ n .
  • the lookup table consists of correction factors which apply to each input signal level, to result in a net linear gain relationship between input signal level and output signal level. This table is updated during each transmit time slot of the power amplifier 115 as the transceiver operates, so that changes in amplifier transfer function characteristics are corrected for soon after they occur.
  • the look-up table has as many entries as is necessary to correct for variations in power amplifier gain accurately. In the preferred embodiment, a table of 100 entries corrects for amplifier gain variations over a 50dB range of signal power, in 0.5dB steps. Since there is a single entry in the table for each input signal level, the look-up table is substantially smaller than the complex-valued correction look-up table cited previously.
  • the look-up table is periodically updated by the inverse transfer function determiner 205 to reflect possible changes in the power amplifier's distortion characteristics. For instance, the linearity of the amplifier is dependent on the load impedance presented to it by the antenna of the transceiver. This load impedance, in turn, is dependent on the proximity of metal objects to the antenna, thus as the transceiver moves it is desirable to update the look-up table.
  • the inverse transfer function dete ⁇ ninator 205 uses a coarse gain factor, H, the recovered transmit signals I ou t and Q 0 ut, Pin and K(Pin), the correction value for PIN signals from the linearizer in order to determine the inverse transfer function's optimum gain correction value, K'(Pi n ). This, in turn, is used to correct the current K(Pjn) value.
  • the look-up table is periodically updated with new correction factor entries as the nonlinearity of the gain of the power amplifier changes.
  • the receiver portion of the dual- mode receiver demodulates an attenuated sample of the power amplifier output signal.
  • Receiver quadrature, output signals, lout and Qout. are used to measure the transmitter gain for given gain-corrected transmit amplifier input signal level of (K(P in )2 ⁇ p in ) from
  • the transmit amplifier gain, G', for gain-corrected transmit amplifier input level, (K(Pi n ) 2 )(Pi n ) can be derived from:
  • G'(P ⁇ n ) is not a constant except for truly linear amplifiers.
  • An ideal linear gain, G is desired.
  • G is a known desired net gain
  • G'(Pi n ) is a measured value
  • the table of correction factors is applied to _i n and Qi n to generate corrected I'in and Q'i n values for the gain error (and hence distortion) occurring at that particular Pi n level.
  • the table is also iteratively updated to reflect changes in distortion, according to the following update equation:
  • K'(Pi n ) K(P in ) - (alpha)K'(PiN)
  • K"(Pi n ) is the updated gain adjustment value for a particular observed Pjure.
  • the constant alpha is a small error-correction factor.
  • the overall transmitter gain may be adjusted via a separate control signal applied to an adjustable gain stage prior to the transmitter amplifier.
  • the proper correction factor becomes:
  • H is the gain of the adjustable gain stage at the adjustable gain level for which K(Pj n ) is calculated; thus as errors in measured G'(Pi n ) are corrected on a per-sample basis by K(Pi n ), dependent on Pin, an overall course gain correction is applied via the constant H, which reduces the requirements on the modulation D/A converters for handling a wide range of output levels.
  • the look-up table may be substituted with a power series calculation.
  • the output signal from the inverse transfer function determinator, K"(Pin) is a correction to the coefficients of a gain adjustment power series equation.
  • a ⁇ -_ coefficients, k > 1 are the distortion terms of the transmit amplifier transfer function expressed as a power series. Knowing measured values of the vector X and corresponding values of & we can solve n equations in n unknowns to obtain the values a n to the desired order of non-linearity n. An inverse transfer function can then be derived by solving the system of equations:
  • the coefficients of the inverse transfer function can be corrected as the transmit amplifier's transfer function changes with operating conditions by calculating new coefficients periodically, and using a portion of the new coefficients to steer the current coefficient estimates toward their optimum values.
  • the value P ⁇ n is calculated for each new set of Ij n , Qi n values generated at the output of the modulation low-pass filters. This Pi n value is then applied to the estimated inverse transfer function to produce a value P'i n , which is the desired input power level required to produce the desired output power level when processed through the non ⁇ linear transmitter amplifier.
  • a correction factor is calculated:
  • FIG. 3 is a graph showing three curves relating the input energy level to the output energy level of the power amplifier.
  • the input energy level of the power amplifier is denoted by EI on the abscissa.
  • the output energy level of the power amplifier is denoted by EOUT on the ordinate.
  • the three curves on the graph represent a transfer function 305, an inverse transfer function 307, and an ideal linear transfer function 309 of the power amplifier 115.
  • the transfer function of the power ampUfier 115 will follow the transfer function curve 305.
  • the slope of the curve is linear throughout most of its operating region.
  • the graph shows nonlinearity beginning at a coordinate point 304. Above this transition point the power amplifier 115 no longer has a linear transfer function characteristic.
  • the slope of the transfer function curve 305 decreases whereby an incremental change in the level of the input signal, does not produce a corresponding incremental change in the level of the output signal.
  • the preferred embodiment of the present invention describes a process to determine the transfer function curve 305 and the inverse transfer function curve 307 for the power amplifier 115.
  • an inverse transfer function representing the difference between curves 307 and 309 can be determined from the transfer function 305.
  • the amplifier 115 operates within non-linear region to improve efficiency.
  • the effect of the inverse transfer function determinator 205 is to determine the coordinate point 311 on transfer function curve 305 and determine a second coordinate point 313 on the inverse transfer function curve 307. Each point is an equal distance from a third coordinate point 315 on the linear function 309. Other sets of equal distance points, not labeled, are also shown to indicate the relationship between the three curves.
  • An adjustment signal representing a measure of the ratio of point 313 on the image curve 307 to point 315 on the linear curve is multiplied by the input signals _i n and Qi n of the transmitter quadrature modulator 111 in order to adjust the transmit amplifier input signal resulting in a linear output signal at the coordinate point 315 on the linear transfer function curve 309.
  • the tr ⁇ msfer function and its inverse are determined to effectively linearize the transfer function of the power amplifier 115 over its nonlinear operating range.
  • an effective linear transfer function is determined, based on a measured signal collected at the output of the transmitter amplifier by a dual-mode receiver.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Physics & Mathematics (AREA)
  • Nonlinear Science (AREA)
  • Power Engineering (AREA)
  • Transmitters (AREA)
  • Amplifiers (AREA)

Abstract

An apparatus and method for linearizing the gain of a nonlinear power amplifier (115) in a TDM digital transceiver. The power amplifier's transfer function describes a relationship between its input signal and output signal. The output signal is selectively demodulated, substantially with existing receiver elements, during the transmit time slot of a TDM time frame. An inverse transfer function is determined from the nominal gain of the power amplifier (115) and the transfer function. A correction value, corresponding to the transfer function and its inverse, is determined in a vector table. The correction value is multiplied with the input signal resulting in an adjusted input signal. The amplified adjusted input signal produces a varied output signal that is linear relative to the input signal.

Description

AN APPARATUS AND METHOD FOR VARYING A SIGNAL IN A TRANSMITTER OF A TRANSCEIVER
Field of the Invention
The present invention relates generally to radio systems having a transmitter and a receiver and, more particularly, to an apparatus and method for linearization and gain control of the transmitter's power amplifier in a TDM system by selectively sampling the amplifier's output signal with the receiver, processing the signal to determine a correction value, applying the correction value to the amplifier's input signal or gain stage to vary the output signal.
Background of the Invention
A radio communications system is comprised, at a minimum, of a transmitter and a receiver. The transmitter and the receiver are interconnected by a radio-frequency channel to permit transmission of an information signal therebetween. A transceiver will generally include both a receiver and a transmitter. The transmitter portion of the transceiver will generally include a radio-frequency (RF) power amplifier for increasing the power of the transmitted signal. RF power amplifiers generally have nonlinear transfer function characteristics relating their input and output signals over a portion of their output power operating range. This nonlinearity appears as an input-level-dependent gain over a portion of the operating range of input level. Although the concept of RF power amplification in radio signal transmission is well understood, RF power amplification of Time Division Multiplex (TDM) signals presents new challenges to the land-mobile industry.
Increased usage of cellular communications systems has resulted, in many instances, in the full utilization of every available transmission channel of the frequency band allocated for cellular radiotelephone communications. In an alternative cellular system proposed to increase capacity in the United States, hereinafter called USDC for United States Digital Cellular, an RF channel is shared (time-division- multiplexed) among several subscribers attempting to access the radio system in certain ones of various time-division- multiplexed time slots. This permits transmission of more than one signal at the same frequency, using the sequential time-sharing of a single channel by several radio telephones. The time slots are arranged into periodically repeating frames, thus, a radio communication of interest may be periodically discontinuous wherein unrelated signals are interleaved with signals transmitted in other time slots. A particular linear modulation scheme chosen for
USDC is called π/4 - shift quadrature differential phase-shift keying (QDPSK). In a π/4 - shift QDPSK modulation scheme, speech signals are encoded into a serial data stream. The serial data stream is demultiplexed into two secondary data streams and processed to generate discrete samples in time of the in-phase (I) and quadrature (Q) signal components of a QDPSK constellation. The discrete signal sample is used in conventional digital signal processing (DSP), operations. Linear modulation schemes, such as π/4 - shift QDPSK, generally have narrow bandwidths and non-constant signal envelopes. The narrow bandwidth optimizes the efficiency of the radio frequency spectrum. Although linear modulation methods can achieve high spectrum efficiency, nonlinear RF power amplifiers introduce distortion components which tend to spread the spectrum thus eliminating any spectrum frequency advantage. Low pass filtering of the transmit signal to achieve the narrow bandwidth causes the signal envelope to vary thus eliminating the full use of the amplifier's linear region. To optimize the amplifier's efficiency the nonlinear operating region may also be used. If the nonlinear region is utilized, the amplifier could operate at a higher output level and then be limited by operating supply voltage, bias current or heat dissipation. A formidable challenge then is to provide a transmitter RF power amplifier which is both linear and power efficient.
Cellular systems commonly require the adjustment of output power to a number of discrete average values. It is desired to maintain the adjustment of average output power to a accurate value, in spite of the variation of amplifier gain with temperature, supply voltage and operating load. Traditionally this has been done using a temperature- compensated diode rectifier detecting output power, which produces a DC voltage proportional to the envelope peaks of the output power signal. The diode detector is not attractive for a USDC system since it produces an accurate measurement of power only over a relatively small range of signal power. The USDC system specifies a greater range of output power than can conventionally be processed by a diode detector. USDC modulation also results in variation of the transmitted signal envelope, which results in erroneous measurement of average signal power by the peak-detecting diode detector. One previous approach taken to correct the problems associated with non-linear transmit power amplifiers uses a pre-defined complex-valued correction look-up table. In-phase (I) and quadrature (Q) components of the modulation signal are used as pointers into the look-up table to determine a gain correction pair. The gain correction pair is applied to the I and Q component signal values prior to amplification to pre- distort the transmit signal. Predistorting the transmit signal cancels the distortion added by the RF power amplifier caused by nonlinearity. The result is a transmitted output signal nearly linear with the input signal over the amplifier's non linear region. This approach does not compensate for changes in the amplifier's nonlinear gain transfer function over time, due to temperature, supply voltage or operating load. In addition, for good linearization, the look-up table becomes unreasonably large. There are a large number of modulation I and Q signal pairs which have the same average power, and hence the same gain adjustment, so significant redundancy is present in the table. Another previous approach used to solve the problems associated with a non-linear power amplifier uses a cartesian coordinate negative feedback control system. This system uses sensitive continuous high bandwidth feedback loops to adjust transmit amplified modulation as it encounters the non- linearity of the power amplifier. The distorted output signal of the power amplifier is subtracted from the input signal to yield a distortion correction factor. The correction factor is applied to the input signal resulting in an effective linear output signal from the power amplifier. Problems associated with this approach include a delay caused by filtering the input signal before it is coupled to the transmitter amplifier which creates instability in the loop, and the need to maintain controlled phase shift through the transmitter amplifier in spite of variations in load impedance, drive level and power supply voltage.
Thus, it is desirable to accurately adjust the average output power over the range specified in the USDC system. It is also desirable to amplify radio frequency signals in a transmitter power amplifier which is both linear and power efficient but which does not require either a large memory of correction factors or delay-induced amplifier instability.
Summary of the Invention
An apparatus for varying a first signal generated in a transceiver is disclosed. The transceiver includes an antenna, from which the transmitter of the transceiver transmits a first signal from the antenna and in which a second signal is induced. A selector selects between the first signal and the second signal and when the first signal is selected a selected signal is produced. The selected signal is processed to produce a processed signal which is used to produce the varied first signal.
Brief Description of the Drawings
FIG. 1 is a block diagram of a transceiver including a transmitter, a receiver and a portion of a signal processor which may employ the present invention.
FIG. 2 is a block diagram of a portion of a receiver demodulator showing switched alternate receiving paths.
FIG. 3 is a block diagram of a signal processor which may employ the present invention.
FIG. 4 is a graph showing three function curves relating an input signal to an output signal of a power amplifier.
Detailed Description of a Preferred Embodiment
The preferred embodiment of the present invention samples the output of the transmitter's power amplifier with the receiver during a transmit time slot in a TDM signal. The demodulated power amplifier output signal is compared with the input to the modulator to create the look-up table correction values. The correction values are applied to each input signal level of the modulator. This table is updated as the transmitter operates to correct for changes in the amplifier's transfer function characteristics. The path from the transmitter's power amplifier through the receiver and back to the power amplifier determines a feedback path for linearizing the power amplifier thus permitting efficient operation.
FIG. 1 shows a block diagram of a transceiver including a transmitter, a receiver and a signal processor portion. A typical received signal, a received time slot in a TDM signal, is coupled through an antenna 101 into a receiver bandpass filter 103. The filtered response 104 is quadrature demodulated in the receiver quadrature demodulator 105. The demodulated signal is composed of In-phase (I) and Quadrature-phase (Q) components. The I and Q quadrature components are coupled to a digital signal processor 107 through a sampling Analog-to-Digital (A/D) converter 109. A typical transmitted signal, a transmit time slot in a TDM signal, originates in the digital signal processor 107 and is further coupled to a transmitter quadrature modulator 111 as I"IN and Q"IN signal components through a Digital-to- Analog (D/A) converter 113. The transmitter quadrature modulator 111 combines the I"IN and Q"IN signal components into a transmitter excitation signal 112. The excitation signal 112 is amplified with power amplifier 115 and further coupled to the antenna 101 through a transmitter bandpass filter 117. The receiver bandpass filter 103 and the transmitter bandpass filter 117 each pass a different frequency range to isolate the receiver and transmitter portions of the transmitter.
In the preferred embodiment of the present invention, a coupler 119 couples a portion of the output signal of the power amplifier 115 to the receiver quadrature demodulator 105 through the attenuator 120. The purpose of the attenuator 120 is to reduce the signal level out of the coupler to levels within the dynamic range of the receiver in a controlled manner. A receiver control signal 121 from the digital signal processor 107 configures the receiver quadrature demodulator 105 to receive a signal either from the antenna 101 or the output of the transmit amplifier 115. Similar receiver circuitry in the receiver quadrature demodulator 105 is used to demodulate signals from both sources.
FIG. 2 shows a portion of the receiver quadrature demodulator which selects between two candidate signals 104 and 106 depending on the state of the receiver control signal 121. The receiver control signal 121 activates switch 405 to select whether the received signal is coupled from the antenna 101 or from the transmitter amplifier 115. When the switch contact 410 is coupled to terminal 406, mixer 407 and local oscillator 409 convert the transmit signal 106 from power amplifier 115 to the normal IF frequency processed by mixers 401 and 403. Likewise, when switch contact 410 is coupled to terminal 408, mixer 417 and local oscillator 415 convert a received carrier signal 104 from the antenna 101 to the normal IF frequency processed by mixers 401 and 403. Separate local oscillators 409 and 415 are used because of different transmit and receive frequencies. Local oscillator 411 and 90° phase shifter 413 provide conventional receiver demodulating functions. The output of mixers 401 and 403, I and Q component signals, respectively, are coupled to A D converter 109. The advantage of using similar receiver circuitry allows additional hardware, otherwise necessary to receive the transmit amplifier output signal 106 to be eliminated. The state of the receiver control signal 121 is determined using timing information recovered from the normal receive signal. The receiver control signal 121 determines which signal will be received based upon the operating position of the transmit and receive time slots in a TDM signal for the transceiver. During the transceiver's receive time slot, the receive control signal 121 instructs the receiver quadrature demodulator 105 to receive a carrier signal 104 from the antenna 101. During the transceiver's transmit time slot, the receive control signal 121 instructs the receiver quadrature demodulator 105 to receive the transmit signal from the power amplifier 115.
During the transceiver's transmit time slot, the receiver quadrature demodulator 105 receives the power amplifier's output signal 106 from the coupler 119 coupled through the attenuator 120. The receiver quadrature demodulator 105 demodulates the output signal into its quadrature components which are further coupled to the digital signal processor 107 through an A/D converter 109 resulting in component signals Iout and Qout.
Component signals, Iout and Q0ut.. are processed in the DSP 107 to adjust the input signal of the power amplifier 115. The adjusted input signals, I'IN and Q ΓN, are coupled from the DSP 107 to the D/A converter 113 as I"IN and Q"IN. and further to the transmitter quadrature modulator 111. The transmitter quadrature modulator 111 modulates the quadrature signal components, I"IN and Q"IN> into a transmitter excitation signal which is further amplified by the power amplifier 115. The input signal level to the power amplifier 115 is adjusted with a correction value which results in an output signal that is linear with the original input signal before adjustment. The output of the power amplifier 115 is coupled through a transmitter bandpass filter 117 into the antenna 101. The gain control signal, H, determined by the DSP 107, is coupled through the D/A converter 123 to the power amplifier 115.
As described above, the gain nonlinearity of power amplifier 115 is compensated for by adjusting the input signal level presented to the power amplifier 115 by modulator 111 based on what the energy of the input signal is, to compensate for the gain nonlinearity of the power amplifier. The nonlinearity of the power amplifier may be described as a variation of gain with the input signal level. The correction value is determined by sampling the output signal of power amplifier 115. The input and sampled signals in DSP 107 update the correction value used for each input signal level.
Factors that cause operation in the amplifiers nonlinear region include large-signal gain variations, saturation and cutoff effects, temperature, input signal level variations, and the disturbance of internal bias points by reflected energy at the transmit amplifier output due to the voltage standing wave ratio (VSWR) of the antenna. It is also desirable to chose to operate the amplifier in the non-linear region because the amplifier is more efficient in the non-linear region.
Compensation for non-linearity permits a net increase in the efficiency of the power amplifier 115. Advantages of an efficient power amplifier 115 may be applied to both fixed and mobile transceivers using a power amplifier. Specific advantages in a mobile or portable transceiver include: longer talk time, smaller battery size, cooler operation and increased reliability.
Now referring to FIG. 3 there is shown a block diagram of a signal processor which may employ the present invention. The following text describes the general process of how the amplifier's output signal, IoUT and Qouτ» is processed in the DSP 107 to determine an adjusted input signal, I'IN and Q'IN-
The receiver selector 215 determines the state of the receiver control signal 121. The receiver selector 215 makes its decision based on the time state of the TDMA received channel.
The data source 207 provides the Iin and Qin quadrature component signals for transmitting. The TDM time clock 206 synchronizes the data source's transmitting activities and the selection of modes for the dual-mode receiver quadrature demodulator 105 of FIG. 1. The quadrature data is coupled to the linearizer 211 as I\n and Qιn through a low pass filter 213. The linearizer 211 has four outputs: Pin and K(Pin) coupled to the inverse transfer function determiner 205 and I'in and Q'in coupled to the D/A Converter 113.
The linearizer 211 determines Pin from input signals, IlN and QIN, looks up a K(PIN), a correction value, in the look¬ up table and multiplies _in and Qjn each by the K(PIN) which results in a gain-adjusted transmitter excitation input signal, I'in and Q'in, coupled to modulator 111. The adjusted input signal, I'in and Q'ιn, provides an amplified signal at the output of the transmit amplifier 115 which is linear with the input signal Iin and Qιn.
The lookup table consists of correction factors which apply to each input signal level, to result in a net linear gain relationship between input signal level and output signal level. This table is updated during each transmit time slot of the power amplifier 115 as the transceiver operates, so that changes in amplifier transfer function characteristics are corrected for soon after they occur.
For each discrete sample pair of Ijn and Qin, input signal power is calculated by the signal processor, after the low-pass filtering process is performed using, the following equation: Pin = din2 + Qin2)
PlN determines from the look-up table the level of Iin and Qιn should be adjusted to maintain a linear relationship between input and output of the power amplifier 115. The look-up table has as many entries as is necessary to correct for variations in power amplifier gain accurately. In the preferred embodiment, a table of 100 entries corrects for amplifier gain variations over a 50dB range of signal power, in 0.5dB steps. Since there is a single entry in the table for each input signal level, the look-up table is substantially smaller than the complex-valued correction look-up table cited previously.
The look-up table is periodically updated by the inverse transfer function determiner 205 to reflect possible changes in the power amplifier's distortion characteristics. For instance, the linearity of the amplifier is dependent on the load impedance presented to it by the antenna of the transceiver. This load impedance, in turn, is dependent on the proximity of metal objects to the antenna, thus as the transceiver moves it is desirable to update the look-up table.
In tiie preferred embodiment of the present invention, the inverse transfer function deteπninator 205 uses a coarse gain factor, H, the recovered transmit signals Iout and Q0ut, Pin and K(Pin), the correction value for PIN signals from the linearizer in order to determine the inverse transfer function's optimum gain correction value, K'(Pin). This, in turn, is used to correct the current K(Pjn) value.
The look-up table is periodically updated with new correction factor entries as the nonlinearity of the gain of the power amplifier changes. The receiver portion of the dual- mode receiver demodulates an attenuated sample of the power amplifier output signal. Receiver quadrature, output signals, lout and Qout.are used to measure the transmitter gain for given gain-corrected transmit amplifier input signal level of (K(Pin)2χpin) from
The output signal level: Pout = dout2 + Qout2). and
given a known attenuation D in the attenuator and coupler. The transmit amplifier gain, G', for gain-corrected transmit amplifier input level, (K(Pin)2)(Pin) can be derived from:
G'(Pin) = P0Ut/((D)(Pin)(K(Pin)2)
In general, G'(Pιn) is not a constant except for truly linear amplifiers. An ideal linear gain, G, is desired. Consider quantizing the expected range of Pin into a discrete set of values. Based on values of G'(Pin) obtained as above, for each of the observed values of Pin during the course of transmitter operation there exists a optimum correction factor K'(Pjn),
K'(Pin) = (G/G'(Pin))°-5
where G is a known desired net gain, and G'(Pin) is a measured value.
During normal operation, the table of correction factors is applied to _in and Qin to generate corrected I'in and Q'in values for the gain error (and hence distortion) occurring at that particular Pin level. The table is also iteratively updated to reflect changes in distortion, according to the following update equation:
K'(Pin) = K(Pin) - (alpha)K'(PiN)
Where K"(Pin) is the updated gain adjustment value for a particular observed Pj„. The constant alpha is a small error-correction factor. Thus each entry in the table is adjusted by a portion of the observed actual correction factor, when the input power Pin for that entry occurs in the modulation signal, to steer the entries in the correction table toward the actual correction factors, as the actual correction factors drift over time.
If it is found that K(Pin) differs greatly from 1, to the point that the adjusted I'i and Q'in values would fall outside the useful range of the modulation D/A converters, the overall transmitter gain may be adjusted via a separate control signal applied to an adjustable gain stage prior to the transmitter amplifier. In this case, the proper correction factor becomes:
(H)(K(Pin)) = (G/røPin))0-5
Where H is the gain of the adjustable gain stage at the adjustable gain level for which K(Pjn) is calculated; thus as errors in measured G'(Pin) are corrected on a per-sample basis by K(Pin), dependent on Pin, an overall course gain correction is applied via the constant H, which reduces the requirements on the modulation D/A converters for handling a wide range of output levels.
As an alternate embodiment of the present invention, the look-up table may be substituted with a power series calculation. For this alternate embodiment, the output signal from the inverse transfer function determinator, K"(Pin), is a correction to the coefficients of a gain adjustment power series equation.
It is possible to derive the gain correction values for modulation samples using an approximation to the transmit amplifier transfer function and a derived inverse transfer function. Instead of deriving a table-based set of correction factors, a pair of equation matrices are solved to return the coefficients of a correction power series, an inverse transfer function power series, which is applied to the gain of Iιn and Qin. to generate I'in and Q'in. For 2. and Xvectors of input and output signal energy, each entry in X being one of the set of values Pin corresponding to a value Pout measured at the output of the transmit amplifier, a corresponding entry in X, we have the system of equations:
aι(X) + a3(χ3) + a5(X5) + ... + an(χn) = £
Where the a\-_ coefficients, k > 1, are the distortion terms of the transmit amplifier transfer function expressed as a power series. Knowing measured values of the vector X and corresponding values of & we can solve n equations in n unknowns to obtain the values an to the desired order of non-linearity n. An inverse transfer function can then be derived by solving the system of equations:
bι(«2D) + b3(f(X))3 + b5(f-X))5 + ... + bniff∑W = (GXX)
where fQD is the power series producing X, above. This set of equations is solved for the coefficients b with a set of input values of X and measured resultants X, to produce a power series which, when applied to the input energy, produces a normalized output energy. To correct nominally for up to 5th order non-linearity, this requires solving two sets of equations, each 3 equations in 3 unknowns.
As with the table method, the coefficients of the inverse transfer function can be corrected as the transmit amplifier's transfer function changes with operating conditions by calculating new coefficients periodically, and using a portion of the new coefficients to steer the current coefficient estimates toward their optimum values. In the preferred embodiment, the value P{n is calculated for each new set of Ijn, Qin values generated at the output of the modulation low-pass filters. This Pin value is then applied to the estimated inverse transfer function to produce a value P'in, which is the desired input power level required to produce the desired output power level when processed through the non¬ linear transmitter amplifier. A correction factor is calculated:
K(Pin) = (Pin P'in)0-5
this then multiplies, in the same manner as the lookup-table entry in the preferred embodiment, Iin and Qin, in order to produce the desired P'jn power which will generate the desired linear-gained instantaneous output power, resulting in a net linear gain for the particular _in and Qin modulation signal pair.
FIG. 3 is a graph showing three curves relating the input energy level to the output energy level of the power amplifier. The input energy level of the power amplifier is denoted by EI on the abscissa. The output energy level of the power amplifier is denoted by EOUT on the ordinate.
The three curves on the graph represent a transfer function 305, an inverse transfer function 307, and an ideal linear transfer function 309 of the power amplifier 115. Generally, the transfer function of the power ampUfier 115 will follow the transfer function curve 305. The slope of the curve is linear throughout most of its operating region. The graph shows nonlinearity beginning at a coordinate point 304. Above this transition point the power amplifier 115 no longer has a linear transfer function characteristic. The slope of the transfer function curve 305 decreases whereby an incremental change in the level of the input signal, does not produce a corresponding incremental change in the level of the output signal. The preferred embodiment of the present invention describes a process to determine the transfer function curve 305 and the inverse transfer function curve 307 for the power amplifier 115. Using signal processing, an inverse transfer function representing the difference between curves 307 and 309 can be determined from the transfer function 305. By applying the inverse transfer function as a table of correction values to the transfer function curve 305, a related net linear transfer function curve 309 is determined. In the preferred embodiment, the amplifier 115 operates within non-linear region to improve efficiency. The effect of the inverse transfer function determinator 205 is to determine the coordinate point 311 on transfer function curve 305 and determine a second coordinate point 313 on the inverse transfer function curve 307. Each point is an equal distance from a third coordinate point 315 on the linear function 309. Other sets of equal distance points, not labeled, are also shown to indicate the relationship between the three curves.
An adjustment signal representing a measure of the ratio of point 313 on the image curve 307 to point 315 on the linear curve is multiplied by the input signals _in and Qin of the transmitter quadrature modulator 111 in order to adjust the transmit amplifier input signal resulting in a linear output signal at the coordinate point 315 on the linear transfer function curve 309. Thus, the trεmsfer function and its inverse are determined to effectively linearize the transfer function of the power amplifier 115 over its nonlinear operating range.
Thus, by adjusting the input signal to the power amplifier 115, an effective linear transfer function is determined, based on a measured signal collected at the output of the transmitter amplifier by a dual-mode receiver.

Claims

Claims
1. An apparatus for varying a first signal generated in a transceiver, the transceiver including an antenna from which the transmitter of the transceiver transmits the varied first signal and in which a second signal is induced, the apparatus comprising: means for selecting between the varied first signal and the second signal to produce a selected signal when the varied first signal is selected; means for processing said selected signal to produce a processed signal; and means for varying the first signal in response to said processed signal to produce the varied first signal.
2. An apparatus for varying the first signal in accordance with claim 1 wherein said means for selecting further comprises a receiver.
3. An apparatus for varying the first signal in accordance with claim 1 wherein said means for processing further comprises an inverse transfer function.
4. An apparatus for varying the first signal in accordance with claim 1 wherein said means for processing further comprises a correction value derived from a look-up table.
5. An apparatus for varying the first signal in accordance with claim 1 wherein said means for processing further comprises a correction value derived from a power series matrix.
6. An apparatus for varying the first signal in accordance with claim 1 wherein said means for varying further comprises adjusting the input signal.
7. An apparatus for varying the first signal in accordance with claim 1 wherein said means for varying further comprises adjusting the gain of the input signal.
8. A method for varying a first signal in a transceiver, the transceiver including an antenna from which the transmitter of the transceiver transmits the varied first signal and in which a second signal is induced, the method comprising the steps of: selecting between the varied first signal and the second signal to produce a selected signal when the first signal is selected; processing said selected signal to produce a processed signal: and varying the first signal, responsive to said processed step to produce the varied first signal.
9. An apparatus for varying an output signal of a transmitter in a transceiver, the transmitter of the transceiver including a power amplifier that amplifies the output signal for transmission from an antenna, the receiver of the transceiver receives a carrier signal induced in the antenna , the power amplifier having a transfer function describing a relationship between an input signal and the output signal, the apparatus comprising: means for selecting between the output signal and the carrier signal to produce a selected signal in the receiver when the output signal is selected; means for processing said selected signal and an inverse of the transfer function to produce a correction value; and means for adjusting the input signal in response to said correction value to vary the output signal.
10. An apparatus for linearizing an output signal of a transmitter in a transceiver operating in a TDM system, the transmitter of the transceiver including a power amplifier that amplifies the output signal for transmision antenna, the receiver of the transceiver receiving a carrier signal induced in the antenna, the power amplifier having a transfer function describing a relationship between an input signal and the output signal and a first operating point of the transfer function defined by a sample of the input signal and the output signal, the apparatus comprising: means for selecting between the output signal and the carrier signal to produce a selected signal in the receiver when the output signal is selected; means for determining a first operating point of the transfer fiinction associated with said selected signal; means for determining a second operating point of an inverse of the transfer function associated with said first operating point; means for determining a third operating point from said first and second operating points; means for determining a correction value responsive to the first, second and third operating points; means for adjusting the input signal responsive to said correction value to produce an adjusted input signal; and means for varying the output signal, responsive to said adjusted input signal to produce a linearized output signal.
PCT/US1991/007012 1990-10-24 1991-09-26 An apparatus and method for varying a signal in a transmitter of a transceiver WO1992008297A1 (en)

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GB2345599A (en) * 1998-12-23 2000-07-12 Nokia Mobile Phones Ltd A baseband predistorter for a transmitter with combined downconversion and demodulation in the feedback path
WO2001063791A3 (en) * 2000-02-23 2004-02-26 Scient Generics Ltd Transmitter and receiver circuit
WO2001063791A2 (en) * 2000-02-23 2001-08-30 Scientific Generics Limited Transmitter and receiver circuit
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GB2379109A (en) * 2001-08-21 2003-02-26 Ubinetics Ltd A predistorted mobile phone base station transmitter with reduced digital subsystem dynamic range requirements
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EP1499015A1 (en) * 2003-07-17 2005-01-19 Siemens Aktiengesellschaft Circuit and process for linearizing the characteristics of a GSM power amplifier
TWI425785B (en) * 2008-03-11 2014-02-01 Nec Corp Communication device, distortion compensation circuit, and distortion compensation method
WO2012040737A2 (en) 2010-09-24 2012-03-29 Intel Corporation Close-loop power amplifier pre-distortion correction
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CN108871381A (en) * 2017-04-27 2018-11-23 罗德施瓦兹两合股份有限公司 Signal calibration method, the purposes of method, the system and oscillograph of correction measuring signal
CN108871381B (en) * 2017-04-27 2022-03-11 罗德施瓦兹两合股份有限公司 Signal correction method, application of method, system for correcting measurement signal and oscilloscope

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GB2254973A (en) 1992-10-21
JPH05503408A (en) 1993-06-03
FR2669165A1 (en) 1992-05-15
GB2254973B (en) 1995-02-22
FR2669165B1 (en) 1995-07-21
CA2069476C (en) 1996-12-17
CA2069476A1 (en) 1992-04-25
GB9213179D0 (en) 1992-08-12

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