The present invention relates to analog-to-digital converters (ADCs) using oscillators.

Some types of analog-to-digital converters convert analog signals to digital signals using a voltage controlled oscillator. In such converters, generally an analog input voltage signal is fed to the VCO, and the digital output value is determined based on the output of the VCO, which may for example involve a counting of pulses output by the VCO.

The resolution of such analog-to-digital converters is limited by non-linearities of the VCO. To improve the resolution, analog feedback or digital calibration are sometimes used. Analog feedback uses a large silicon area and increases the power consumption. The reduction of the non-linearity by digital calibration may be not sufficient for some applications.

In the following, some embodiments of the present invention will be described in detail. It is to be understood that the following description is given only for the purpose of illustration and is not to be taken in a limiting sense. The scope of the invention is not intended to be limited by the embodiments described hereinafter with reference to the accompanying drawings, but is intended to be limited only by the appended claims and equivalents thereof.

It is also to be understood that in the following description of embodiments any direct connection or coupling between functional blocks, devices, components, circuit elements or other physical or functional units shown in the drawings or described herein could also be implemented by an indirect connection or coupling, i.e. a connection or coupling comprising one or more intervening elements. Furthermore, it should be appreciated that functional blocks or units shown in the drawings may be implemented as separate circuits in some embodiments, but may also be fully or partially implemented in a common circuit in other embodiments. In other words, the description of various functional blocks is intended to give a clear understanding of various functions performed in a device or system shown and is not to be construed as indicating that these functional blocks have to be implemented as separate physical units. For example, one or more functional units may be implemented by programming a processor like a single digital signal processor accordingly.

It is further to be understood that any connection which is described as being wire-based in the following specification may also be implemented as a wireless connection and vice versa unless noted to the contrary.

It should be noted that the drawings are provided to give an illustration of some aspects of the embodiments of the present invention and therefore are to be regarded as schematic only. In particular, the elements shown in the drawings are not necessarily to scale with each other, and the placement of various elements in the drawings is chosen to provide a clear understanding of the respective embodiment and is not to be construed as necessarily being a representation of the actual relative location of the various components and implementations according to embodiments of the invention.

The features of the various embodiments described herein may be combined with each other unless specifically noted otherwise. On the other hand, describing an embodiment with a plurality of features is not to be construed as indicating that all those features are necessary for practicing the present invention, as other embodiments may comprise less features and/or alternative features.

In the following, various embodiments of analog-to-digital converters (ADCs) will be described. Analog-to-digital converters generally are devices which convert one or more analog input signals, for example voltage signals or current signals, to one or more digital output signals.

Some embodiments described in the following comprise controllable oscillators. Controllable oscillators generally are oscillators which output one or more oscillating signals, also referred to as oscillations, the one or more signals having a frequency which is dependent on a control signal supplied to the controllable oscillator. One common class of controllable oscillators are voltage controlled oscillators (VCOs). It should be noted that such a voltage controllable oscillator may e.g. be converted to a current controlled oscillator by adding a current-to-voltage converter.

Turning now to the Figures, in FIG. 1 an analog-to-digital converter according to an embodiment is shown.

The apparatus shown in FIG. 1 comprises a first analog input 10 for receiving a first analog input signal IN1 and a second analog input 13 to receive a second analog input signal IN2. Signals IN1, IN2 may for example be voltage signals or current signals. In an embodiment, IN1 and IN2 are two part signals of a differential signal. In some embodiments, IN2 is the negative of IN1.

First analog input 10 is coupled to a first signal path 11, first signal path 11 comprising a first controllable oscillator 12. Controllable oscillator 12 is configured to generate one or more output signals the frequency of which depends on first analog input signal IN1. First signal path 11 is further configured to generate a first digital output signal OUT1 based on the output signals of first controllable oscillator 12.

Second analog input 13 is coupled to a second signal path 14 comprising a second controllable oscillator 15. Second controllable oscillator 15 is configured to output one or more output signals the frequency of which depends on second analog input signal IN2. Second signal path 14 is further configured to generate a second digital output signal OUT2 based on the signal(s) output from second controllable oscillator 15.

The apparatus of FIG. 1 further comprises a combiner 16 which combines first digital output signal 1 and second digital output signal 2, for example by subtraction or addition, and generates a further digital output signal OUT which constitutes the output signal of the analog-to-digital converter.

It is to be noted that the description of the above embodiments with two signal paths and two analog input signals is not to be construed as limiting, and other embodiments may comprise additional analog signal inputs and/or additional signal paths.

In FIG. 2, an analog-to-digital converter according to a further embodiment of the invention is shown.

The apparatus shown in FIG. 2 comprises a first analog signal input 20 and a second analog signal input 23 to receive a positive part signal V_inp and a negative part signal V_inn, respectively, of a differential input voltage signal. Furthermore, the apparatus of FIG. 2 comprises a first calibration input 21 and a second calibration input 24. A switch 22 is provided to switch between first analog input 20 and first calibration input 21, and a switch 25 is provided to switch between second analog signal input 23 and second calibration input 24.

Switch 22 is further coupled to an input of a first voltage-controlled oscillator (VCO) 26. In the embodiment of FIG. 2, first voltage-controlled oscillator 26 is configured to output 64 output signals having different phases, but the same frequency, the frequency being determined by the signal supplied via first switch 22.

In the embodiment of FIG. 2, one of the signals output by first VCO 26 is fed to a first path comprising a first asynchronous counter 27 which counts the periods of the signal received from first VCO 26. For example, first asynchronous counter 27 may count rising edges or falling edges of the signal received from first VCO 26. In other words, asynchronous counter 27 counts a number of full periods of the output signal. In the embodiment of FIG. 2, asynchronous counter 27 is a 6-bit counter outputting a 6-bit digital signal. This 6-bit digital signal is fed to a first 6-fold sample and hold unit 28 of the first path (one sample and hold circuit for each bit) clocked by a clock signal clk. The output of the first 6-fold sample and hold unit 28 represents upper 6 bits, numbered 6 to 11 in the embodiment of FIG. 2, of a first digital output signal phasep.

All 64 output signals of first VCO 26 are fed to a second path comprising a first 32-fold sample and hold unit 29, wherein two output signals are fed to each sample and hold circuit of 32-fold sample and hold unit 29, which is also clocked by clock signal clk. First 32-fold sample and hold unit 29 outputs a 32-bit output signal to a first thermometer to binary encoder 210 of the second path, which converts the 32-bit signal to a 6-bit value representing lower 6 bits, i.e. bits numbers 0 to 5, of first digital output signal phasep.

In this way, first 32-fold sample and hold unit 29 and first thermometer to binary encoder 28 basically determine a value corresponding to a fractional portion of a period of the output signals of VCO 26.

First VCO 26, first asynchronous counter 27, first 6-fold sample and hold unit 28, first 32-fold sample and hold unit 29 and first thermometer to binary encoder 210 form a first signal path comprising the above-explained first and second paths. A second signal path operating in the same manner on a signal received via second switch 25 is formed by a second VCO 211, a second asynchronous counter 212, a second 6-fold sample and hold unit 213, a second 32-fold sample and hold unit 214 and a second thermometer to binary encoder 215, which generates a second digital output signal phasen. The operation of the second signal path corresponds to the operation of the first signal path described above and will therefore not be described again.

First digital output signal phasep is fed to a positive input of a subtractor 216, and second digital output signal phasen is fed to a negative input of subtractor 216, which subtractor 216 generates as an output a 12-bit signal the difference phasep-phasen. This difference signal is fed to a first order differentiator 217 which outputs a 12-bit output signal freq in FIG. 2.

Signal freq via a switch 218 is fed to a digital distortion correction 219 using a lookup-table which digital distortion correction 219 may for example correct for non-linearities of first VCO 26 and second VCO 211. For calibration, switch 218 may be switched to feed signal freq to a distortion estimation unit 220.

Digital distortion correction unit 219 then outputs a 12-bit output signal out which is a digital representation of the analog differential input signal V_inp, V_inn.

It should be noted that any numbers given in the embodiment of FIG. 2 are merely given us an illustration. FIG. 2 represents a 12-bit analog-to-digital converter, but other bit widths are equally possible by adjusting e.g. the number of output signals of first VCO 26 and second VCO 211 and the number of sample and hold circuits in units 28, 29, 213 and 214 accordingly.

In the following, examples for implementations of some elements of some embodiments will be described with reference to FIGS. 3 to 5. It should be noted that the elements of FIGS. 1 and 2 are not restricted to the ones described in the following, but the following additional description serves merely illustrative purposes.

In some embodiments, oscillators like first VCO 26, second VCO 211 oscillator 12 and/or oscillator 15 may be implemented as a ring oscillator using a chain of inverters. For example, as shown in FIG. 3 a ring of 16 differential inverters 31 with local interpolation may be used, each inverter outputting four of the output signals of first VCO 26 or second VCO 211. As a matter of course, for ADCs other than a 12-bit ADC, the number of inverters may vary, and in other embodiments instead of inverters with interpolation invertors without interpolation with a corresponding increased number of inverters or oscillators other than ring oscillators may be used.

In the embodiment of FIG. 3, inverter 31 outputs output signals number 1, 2, 33 and 34, signals number 33 and 34 being the inverse of signals number 1 and 2. Inverter 32 outputs signals number 3, 4, 35 and 36, signals number 35 and 36 being the inverse of signals number 3 and 4, up to inverter 33, which outputs output signals number 31, 32, 63 and 0, signals 63 and 0 being the inverse of signals 31 and 32, respectively. In FIG. 3, the signal V_in is the respective analog input signal, e.g. V_inp or V_inn in the embodiment of FIG. 2.

An example implementation of interpolating inverters 31, 32 and/or 33 is shown in FIG. 4.

In FIG. 4, an inverter is shown comprising PMOS transistors 41, 42 and NMOS transistors 43, 44. The voltage V_in is fed to the gates of PMOS transistors 41 and 42. An input signal, which corresponds to an output signal of a previous inverter in the ring shown in FIG. 3, is fed to the gates of NMOS transistors 43 and 44, in for example corresponding to the signal output from the output marked with a “+” in FIG. 3 and in corresponds to the signal from the output marked with a “−” in FIG. 3. An output signal out and its inverted version out can be tapped at nodes between PMOS transistor 41 and NMOS transistor 44 for the signal out and between PMOS transistor 42 and NMOS transistor 43 for the signal out.

Furthermore, using resistors 45, 46, 47 and 48, the input signals and the output signals are used to form interpolated signals outi, outi as shown in FIG. 4.

For illustration, assuming that the circuit of FIG. 4 represents inverter 32 of FIG. 3, the signal in would correspond to signal number 2, the signal in would correspond to signal number 34, the signal outi would correspond to signal number 3, the signal outi would correspond to signal 35, the signal out would correspond to signal number 4 and the signal out would correspond to signal number 36.

As mentioned, the arrangement of FIG. 4 is only one implementation of an inverter which is usable in a ring up implementation, and other implementations, in particular implementations without interpolation, are also possible as explained above.

Next, with respect to FIG. 5 an example implementation of a sample and hold circuit e.g. of 32-fold sample and hold unit 214 of FIG. 2 will be explained. The example implementation of a sample and hold circuit shown in FIG. 5 comprises PMOS transistors 51 to 54 and 510 to 513 and 516, NMOS transistors 55 to 58 and 514, 515, 517 to 519. 50 denotes a positive supply voltage, for example VDD, and 59 denotes a negative supply voltage, for example VSS or ground.

clk, as in FIG. 2, denotes a clock signal. The input signals to the sample and hold circuit of FIG. 5 are denoted D and D, D being the inverse of D. For example, 32-fold sample and hold unit 214 of FIG. 2 may comprise 32 circuits as shown in FIG. 5, a first one of these circuits receiving output signals number 1 and 33 as shown in FIG. 3, a second one receiving output signals 2 and 34 as shown in FIG. 3 as D and D, respectively, etc.

The corresponding output signals are labeled Q and Q in FIG. 5.

Again, it is to be noted that the implementation shown in FIG. 5 serves merely as an example, and other sample and hold circuits known in the art may be used as well.

In a regular mode of operation of the embodiment of FIG. 2, as already mentioned switches 22, 25 and 218 are in the position shown in FIG. 2, and the analog input signal V_inp, V_inn is converted to a digital output signal out. In a calibration mode, switch 22 couples input 21 with first oscillator 26, switch 25 couples input 24 with second oscillator 211, and switch 21 couples distortion estimated unit 220 with the output of differentiator 217. This calibration which may compensate some non-linearities will be explained in the following in some more detail using FIGS. 6 and 7 as examples.

In the calibration mode, via input 21 and 24 a “DC sweep” is performed, i.e. a series of predetermined DC voltages is input. The thus applied predetermined input values are converted to signal freq as explained above, and distortion estimation unit 220 compares the signal freq with a target value which target value corresponds to the predetermined analog input value at the corresponding time and, in case the value of freq deviates from the target value, stores a corresponding correction value in a lookup-table which later in the regular mode of operation is used by digital distortion correction unit 219 to correct the output signal. This is performed for a predetermined number of correction points.

For example, in FIG. 6 a digital output value depending on the input signal, i.e. the difference of the input signals applied to inputs 21 and 24, for a specific implementation of an embodiment are shown. A solid line shows the result without correction, circles show the calibration points and a dashed line then shows the result with correction. Likewise, FIG. 7 shows the non-linearity for the implementation in units of the least significant bit (LSB, i.e. if the non-linearity is e.g. 10 LSB, this means that a digital code output deviates from the “correct” result by 10 LSB) with and without such a correction. As can be seen, using the correction the linearity is increased. However, as also can be seen from FIG. 6, the correction may also be omitted in some embodiments if the linearity without the correction is sufficient for a given application.

It is to be noted that the above-described embodiments serve only as examples and are not to be construed as limiting, as a plurality of variations and modifications are possible. Some of these modifications already have been discussed above, others will be explained in more detail below.

While in the embodiment of FIG. 2 separate inputs are provided for regular mode and for calibration, switches 22, 25 may also be omitted and the same inputs may be used for regular conversion or for calibration.

While in the embodiment of FIG. 2, voltage inputs have been used, in other embodiments current inputs together with or without current-to-voltage converters may also be used. As already mentioned, the bit width shown in FIG. 2 serves only as an example, and other bit widths for the converter are equally possible.

The number of calibration shown in FIG. 6 also serves only as an example, and any desired number of calibration points to obtain a desired accuracy of calibration may be used.