US7676161B2 - Modulation E-field based control of a non-linear transmitter - Google Patents
Modulation E-field based control of a non-linear transmitter Download PDFInfo
- Publication number
- US7676161B2 US7676161B2 US11/008,515 US851504A US7676161B2 US 7676161 B2 US7676161 B2 US 7676161B2 US 851504 A US851504 A US 851504A US 7676161 B2 US7676161 B2 US 7676161B2
- Authority
- US
- United States
- Prior art keywords
- signal
- bit
- optical
- dither
- field
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Active, expires
Links
- 230000003287 optical effect Effects 0.000 claims abstract description 160
- 238000000034 method Methods 0.000 claims abstract description 105
- 230000002194 synthesizing effect Effects 0.000 claims abstract description 6
- 230000006870 function Effects 0.000 claims description 98
- 238000004891 communication Methods 0.000 claims description 14
- 238000012546 transfer Methods 0.000 claims description 14
- 230000010287 polarization Effects 0.000 claims description 13
- 238000013139 quantization Methods 0.000 claims description 9
- 239000000654 additive Substances 0.000 claims description 8
- 230000000996 additive effect Effects 0.000 claims description 8
- 230000001419 dependent effect Effects 0.000 claims description 8
- 230000000694 effects Effects 0.000 claims description 8
- 230000001629 suppression Effects 0.000 claims description 6
- 238000001228 spectrum Methods 0.000 claims description 4
- 238000001514 detection method Methods 0.000 claims description 3
- 238000001914 filtration Methods 0.000 claims 3
- 238000012358 sourcing Methods 0.000 claims 1
- 230000004044 response Effects 0.000 description 15
- 230000008878 coupling Effects 0.000 description 14
- 238000010168 coupling process Methods 0.000 description 14
- 238000005859 coupling reaction Methods 0.000 description 14
- 230000015572 biosynthetic process Effects 0.000 description 6
- 239000006185 dispersion Substances 0.000 description 6
- 238000003786 synthesis reaction Methods 0.000 description 6
- 230000008859 change Effects 0.000 description 5
- 238000005284 basis set Methods 0.000 description 4
- 230000006735 deficit Effects 0.000 description 4
- 238000010586 diagram Methods 0.000 description 4
- 230000009977 dual effect Effects 0.000 description 4
- 230000032683 aging Effects 0.000 description 3
- 238000013459 approach Methods 0.000 description 3
- 230000008901 benefit Effects 0.000 description 3
- 238000004364 calculation method Methods 0.000 description 3
- 238000009472 formulation Methods 0.000 description 3
- 238000003780 insertion Methods 0.000 description 3
- 230000037431 insertion Effects 0.000 description 3
- 238000007620 mathematical function Methods 0.000 description 3
- 239000000203 mixture Substances 0.000 description 3
- 238000012545 processing Methods 0.000 description 3
- 230000005540 biological transmission Effects 0.000 description 2
- 238000006243 chemical reaction Methods 0.000 description 2
- 230000003595 spectral effect Effects 0.000 description 2
- 238000002834 transmittance Methods 0.000 description 2
- YBJHBAHKTGYVGT-ZKWXMUAHSA-N (+)-Biotin Chemical compound N1C(=O)N[C@@H]2[C@H](CCCCC(=O)O)SC[C@@H]21 YBJHBAHKTGYVGT-ZKWXMUAHSA-N 0.000 description 1
- 108091006146 Channels Proteins 0.000 description 1
- 238000010420 art technique Methods 0.000 description 1
- 230000002238 attenuated effect Effects 0.000 description 1
- 230000008033 biological extinction Effects 0.000 description 1
- 239000003990 capacitor Substances 0.000 description 1
- 238000013329 compounding Methods 0.000 description 1
- 238000010276 construction Methods 0.000 description 1
- 238000007796 conventional method Methods 0.000 description 1
- 238000009826 distribution Methods 0.000 description 1
- 238000011156 evaluation Methods 0.000 description 1
- 230000001747 exhibiting effect Effects 0.000 description 1
- 238000004519 manufacturing process Methods 0.000 description 1
- 230000000737 periodic effect Effects 0.000 description 1
- 230000010363 phase shift Effects 0.000 description 1
- 238000011084 recovery Methods 0.000 description 1
- 238000010079 rubber tapping Methods 0.000 description 1
- 238000005070 sampling Methods 0.000 description 1
- 230000011664 signaling Effects 0.000 description 1
- 230000035882 stress Effects 0.000 description 1
- 238000012360 testing method Methods 0.000 description 1
- 230000009466 transformation Effects 0.000 description 1
- 230000001052 transient effect Effects 0.000 description 1
- 238000011144 upstream manufacturing Methods 0.000 description 1
- FEPMHVLSLDOMQC-UHFFFAOYSA-N virginiamycin-S1 Natural products CC1OC(=O)C(C=2C=CC=CC=2)NC(=O)C2CC(=O)CCN2C(=O)C(CC=2C=CC=CC=2)N(C)C(=O)C2CCCN2C(=O)C(CC)NC(=O)C1NC(=O)C1=NC=CC=C1O FEPMHVLSLDOMQC-UHFFFAOYSA-N 0.000 description 1
Images
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
- H04B10/50—Transmitters
- H04B10/501—Structural aspects
- H04B10/503—Laser transmitters
- H04B10/505—Laser transmitters using external modulation
- H04B10/5053—Laser transmitters using external modulation using a parallel, i.e. shunt, combination of modulators
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
- H04B10/50—Transmitters
- H04B10/501—Structural aspects
- H04B10/503—Laser transmitters
- H04B10/505—Laser transmitters using external modulation
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
- H04B10/50—Transmitters
- H04B10/501—Structural aspects
- H04B10/503—Laser transmitters
- H04B10/505—Laser transmitters using external modulation
- H04B10/5055—Laser transmitters using external modulation using a pre-coder
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
- H04B10/50—Transmitters
- H04B10/501—Structural aspects
- H04B10/503—Laser transmitters
- H04B10/505—Laser transmitters using external modulation
- H04B10/5057—Laser transmitters using external modulation using a feedback signal generated by analysing the optical output
- H04B10/50572—Laser transmitters using external modulation using a feedback signal generated by analysing the optical output to control the modulating signal amplitude including amplitude distortion
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
- H04B10/50—Transmitters
- H04B10/501—Structural aspects
- H04B10/503—Laser transmitters
- H04B10/505—Laser transmitters using external modulation
- H04B10/5057—Laser transmitters using external modulation using a feedback signal generated by analysing the optical output
- H04B10/50575—Laser transmitters using external modulation using a feedback signal generated by analysing the optical output to control the modulator DC bias
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
- H04B10/50—Transmitters
- H04B10/58—Compensation for non-linear transmitter output
Definitions
- the present invention relates to optical signal transmitters for optical communications systems, and in particular to a system and methods for controlling a non-linear optical transmitter.
- the first basis set is a Cartesian coordinate system centered on the E-field origin.
- the total E-field E(t) is decomposed along the orthogonal Real (Re) and Imaginary (Im), or, equivalently, In-phase (I) and Quadrature (Q), directions.
- the second basis set is a polar coordinate system, again sharing its origin with that of the E-field vector.
- the E-field is decomposed into vector length (S) and phase angle ( ⁇ ) relative to the Re-direction.
- S vector length
- ⁇ phase angle
- These two basis sets are related by a non-linear transformation, in a manner well known in the art.
- the time-sequence of loci of the end-point of the E-field vector may be referred to as a trajectory of the E-field.
- a popular technique utilizes a laser 2 coupled to an external Electrical-to-Optical (E/O) converter 4 , as shown in FIG. 1 a.
- the laser 2 generates a narrow-band continuous wave (CW) optical carrier signal 6 having a desired wavelength.
- the E/O converter 4 operates to modulate the amplitude and/or phase of the carrier signal 6 to generate the optical communications signal 8 based on one or more drive signals S(t) generated by a driver circuit 10 based on an input data signal x(t).
- the E/O converter 4 is provided by a well known Mach-Zehnder (MZ) interferometer.
- MZ Mach-Zehnder
- Other types of E/O converters may be used, depending on the desired type of modulation.
- an electro-absorptive E/O converter (EAM) or a variable optical attenuator (VOA) may be used for amplitude modulation, whereas phase shifters are well known for implementing phase modulation schemes.
- the driver circuit 10 generates the drive signals S(t), primarily by scaling the input data signal x(t) to satisfy the voltage and current requirements of the E/O converter 4 .
- the input data signal x(t) is encoded in accordance with a desired modulation scheme (e.g. for quadrature encoding), and the resulting encoded data signal scaled to satisfy the voltage and current requirements of the E/O converter 4 .
- the format of the drive signal S(t) output from the driver circuit 10 is principally governed by the desired modulation scheme, and will typically take the form of a baseband (i.e., binary, direct current) signal; a coded (e.g. quadrature encoded) signal; or a modulated electrical (e.g. radio frequency) carrier signal.
- each drive signal may also be generated in the form of a differential signal pair ⁇ S(t), which provides certain advantages known in the art.
- the drive signals S(t) are directly modeled on the input data signal x(t), and represent the data to be modulated onto the CW carrier 6 . This is true even where an encoding scheme, such quadrature encoding, is used.
- FIGS. 1 a and 1 b illustrate a conventional transmitter in which an input data signal x(t) is transmitted using the well known On-Off Keying (OOK) modulation scheme.
- OLK On-Off Keying
- the driver 10 scales the input data signal x(t) to produce a differential pair of bi-state (that is two-level) baseband drive signals ⁇ S(t).
- the baseband drive signal pair ⁇ S(t) is then used to drive excursions of the E/O converter's (sinusoidal) amplitude response between maximum and minimum transmittance, as may be seen in FIG. 1 b.
- This operation yields an amplitude-modulated optical communications signal 8 having an optical E-field E O (t) which exhibits excursions of amplitude between two states reflecting the binary values of each bit of the input data signal x(t), as shown seen in FIG. 1 c, in which the optical E-field E O (t) is represented in the complex Re/Im plane.
- amplitude modulation of the CW carrier 6 in the above manner results in excursions of the optical E-field E O (t) between loci clustered about two points on the real (Re) axis.
- all of the E-field loci will be located on the real (Re) axis.
- the optical E-field E O (t) will also exhibit excursions in the imaginary (Im) direction due to phase chirp resulting from coupling of phase and amplitude responses of all real electro-optical devices.
- FIG. 1 d illustrates an arrangement in which the input data signal x(t) is encoded by an encoder block 12 prior to scaling.
- Various encoding schemes are known in the art.
- U.S. Pat. No. 6,522,439 (Price et al), teaches an arrangement in which the input data signal x(t) is split into a pair of parallel In-phase (I) and Quadrature (Q) signal components (i.e. the sine and cosine of the data signal x(t)), which are then modulated onto an electrical (RF) carrier and scaled to yield a corresponding pair of drive signals S I (t) and S Q (t).
- I In-phase
- Q Quadrature
- the drive signal 10 (in any of baseband, coded, or modulated electrical carrier formats) is supplied to the signal distorter 14 which imposes a dispersive function F[ ] on the drive signal S(t).
- the resulting distorted drive signal F[S(t)] is then supplied to the E/O converter 4 to generate a predistorted optical communications signal for transmission through the link.
- the signal distorter 12 is provided by an analog filter circuit (not shown) having a group delay characteristic selected to counteract chromatic dispersion of the link. Multiple filter circuits may be cascaded to compensate some other distortions.
- Schemmann et al. teach that the (typically squaring response) of an optical receiver can also be precompensated at the transmitter, by means of a suitable filter circuit within the signal distorter 12 .
- a common characteristic among all of the above-noted predistortion techniques is that analog filter circuits are used to distort an otherwise conventionally generated (and thus conventionally formatted) analog drive signal S(t).
- the premise is that distorting the drive signal S(t) will suitably distort the optical signal 8 in such a way as to offset distortions due to impairments of the optical link. While this approach is satisfactory for compensating linear distortions (such as chromatic dispersion, and receiver squaring effects) it cannot compensate non-linear impairments such as SPM and four-wave mixing. Quite apart from the limitations inherent to analog filter circuits, all of which are well known, compensation of non-linear distortions is complicated by the fact that all real electro-optical devices (i.e.
- a signal processor 16 receives the input data signal x(t) as an input, and uses a compensation function C[ ] to compute successive multi-bit In-phase and Quadrature values (E I (n) and E Q (n), respectively) representing successive loci of the end-point of a desired or target optical E-field vector.
- a linearizer 18 then uses the multi-bit (E I (n), E Q (n)) loci to synthesize a pair of multi-bit digital drive signals V R (n) and V L (n).
- the digital drive signals V x (n), in which x is an index identifying the involved branch of the signal path, are then converted into analog (RF) signals by respective high speed multi-bit Digital-to-Analog Converters (DACs) 20 , which are then amplified (and possibly band-pass filtered to remove out-of-band noise) to generate the drive signals S x (t) supplied to a complex E/O converter 22 .
- the digital drive signals V x (n) are computed such that the drive signals S x (t) supplied to the complex E/O converter 22 will yield an optical E-field E O (t) at the complex E/O converter output 24 that is a high-fidelity reproduction of the target E-field computed by the signal processor 16 .
- the signal processor 16 is capable of implementing any desired mathematical function, which means that the compensation function C[ ] can be selected to compensate any desired signal impairments, including, but not limited to, dispersion, Self-Phase Modulation (SPM), Cross-Phase Modulation (XPM), four-wave mixing and polarization dependent effects (PDEs) such as polarization dependent loss.
- the compensation function C[ ] can be dynamically adjusted for changes in the optical properties of the link, and component drift due to aging.
- the inherent flexibility of the mathematical function implemented by the signal processor 16 also implies that the signal processor 16 can be placed into a “test” mode, and used to generate (E I (n), E Q (n)) loci of a desired optical E-field vector independently (or even in the absence) of an input data signal x(t).
- the linearizer 18 can also implement any desired mathematical function, and thus can perform signal format conversion (i.e. from Cartesian to polar coordinates); compensate for non-linearities in the signal path between the linearizer 18 and the output 24 of the complex E/O converter 22 ; and perform various scaling and clipping operations to limit dynamic range requirements of electrical components downstream of the linearizer 18 (principally the DACs 20 ).
- each analog drive signal S x (t) is governed by that of the DACs 20 .
- each DAC 20 has a resolution of M-bits, where M is an integer, which yields excursions of each analog drive signal S x (t) between 2 M discrete levels.
- M is greater than 4.
- the complex E/O converter 22 will normally be provided as either a conventional dual branch MZ interferometer, or as nested MZ interferometers, as illustrated in FIG. 3 .
- the complex E/O converter 22 comprises a dual branch “main” MZ interferometer 26 , each branch of which comprises a respective branch MZ interferometer 28 R and 28 L.
- Each branch MZ interferometer 28 x (again, where x identifies the respective branch) is preferably driven by a respective differential signal pair ⁇ S x (t).
- Multi-bit digital generation of the drive signals S x (t) in this manner enables the optical transmitter to synthesize any desired E-field waveform at the output 24 of the complex E/O converter 22 .
- the linearizer 18 synthesizes the digital drive signals V x (n) based on a model of the target optical E-field (as opposed to the data signal being transmitted), it is possible to derive a mathematical representation of the entire data path between the signal processor 16 and the E/O converter output 24 , which enables phase and amplitude of the output E-field E O (t) to be independently controlled, even with significant coupling of phase and amplitude responses of the complex E/O converter 22 . This is an operational feature which is simply not possible in prior art transmitters.
- a known method for dynamically controlling an E/O converter 4 of the type described above with reference to FIGS. 1 and 2 is to implement one or more control loops using a dither signal inserted into the drive signal S (t).
- dither signals take the form of a low frequency sinusoidal analog signal that is added to the drive signals S(t), and detected in the optical signal at some point downstream of the E/O converter 4 .
- Differences (typically of amplitude) between the added and detected dither signals provide a direct indication of gain, from which other performance characteristics may be inferred.
- the frequency of the dither signal is typically selected to be low enough to avoid interference with data traversing the signal path, but high enough to avoid being attenuated by low-frequency cutoff.
- an object of the present invention is to provide methods and apparatus for controlling a transmitter capable of synthesizing an arbitrary optical E-field waveform remains highly desirable.
- an aspect of the present invention provides a method of synthesizing an optical signal.
- a multi-bit digital representation of a desired optical E-field is generated.
- the multi-bit digital representation has a resolution of N1-bits, where N1 is an integer greater than 2.
- At least two analog drive signals are synthesized based on the multi-bit digital representation. Each analog drive signal exhibits excursions between 2 M discrete states (i.e. has a resolution of M-bits), where M is an integer greater than 2.
- An electrical-to-optical (E/O) converter is driven using the analog drive signals to generate an output optical E-field at an output of the E/O converter.
- An error is detected between the output optical E-field and the desired complex E-field waveform, and at least one parameter adjusted so as to minimize the detected error.
- a further aspect of the present invention provides a method of controlling a non-linear analog optical transmitter comprising an electrical to optical (E/O) converter driven by a signal path having a multi-bit digital stage and an analog stage having at least one branch for generating a respective analog drive signal.
- E/O electrical to optical
- a selected component of a signal traversing the signal path is dithered using at least one respective digital dither signal.
- At least one artefact of the at least one dither signal is detected in an optical signal at an output of the E/O converter.
- At least one parameter of the transmitter is then adjusted based on the detection result.
- a still further aspect of the present invention provides a method of generating a dithered M-bit digital communications signal.
- an N 1 -bit signal is generated as a predetermined function of an N-bit communications signal and a dither signal, wherein a magnitude of a dither component of the N 1 -bit signal corresponding to the dither signal is less than a predetermined scaling factor, and wherein at least one of N and N 1 is greater than M.
- the N 1 -bit signal is divided by the predetermined scaling factor to generate a scaled signal, and the scaled signal quantized.
- An advantage of the present invention is that accurate synthesis of a desired E-field at the E/O converter output can be obtained independently of device calibration.
- FIGS. 1 a - 1 e schematically illustrate principal components and operation of an optical transmitter known in the prior art
- FIG. 2 schematically illustrates principal components of an optical transmitter implementing electrical precompensation of optical distortions known in the prior art
- FIGS. 3 a - 3 c schematically illustrates principal components of and operation of an optical transmitter implementing digital synthesis of an arbitrary complex optical E-field waveform, known from Applicant's co-pending U.S. patent applications Ser. No. 10/262,944, filed Oct. 3, 2002; Ser. No. 100/307,466 filed Dec. 2, 2002; and Ser. No. 10/405,236 filed Apr. 3, 2003, and International Patent Application No. PCT/CA03/01044 filed Jul. 11, 2003;
- FIG. 4 schematically illustrates principal components and operation of a control system in accordance with an embodiment of the present invention, for controlling the optical transmitter of FIGS. 3 a - 3 c;
- FIG. 5 illustrates a representative detector block usable in the control system of FIG. 4 ;
- FIGS. 6 a and 6 b are block diagrams schematically illustrating respective embodiments of a linearizer of the optical transmitter of FIG. 4 ;
- FIG. 7 is a block diagram schematically illustrating an embodiment of clipping and quantization function implemented within the optical transmitter of FIG. 4 ;
- FIG. 8 is a block diagram illustrating a portion of the E/O converter of FIG. 4 in greater detail.
- the present invention provides methods and apparatus for controlling a non-linear optical transmitter to accurately generate a desired optical E-field at the E/O converter output. Embodiments of the invention are described below, by way of example only, with reference to FIGS. 4-8 .
- FIG. 4 is a block diagram schematically illustrating a control system in accordance with the present invention, for controlling a flexible non-linear optical transmitter of the type described above with reference to FIG. 3 .
- the transmitter comprises a high speed “signal path” between the signal processor 16 and the output 24 of the complex E/O converter 22 .
- This signal path comprises a high speed, multi-bit digital stage 30 cascaded with an analog radio-frequency (RF) stage 32 , which, in turn, drives the complex E/O converter 22 .
- RF radio-frequency
- the multi-bit digital stage 30 includes the signal processor 16 , linearizer 18 and multi-bit DACs 20 x, where x is an index identifying a respective branch of the signal path.
- the analog stage 32 can usefully be considered as being divided into parallel branches 34 , each of which comprises a (fixed gain) low-noise amplifier 36 and a variable gain amplifier (VGA) 38 to scale the DAC output and thereby generate a respective RF drive signal S x (t) (which will preferably be a differential drive signal pair ⁇ S x (t)).
- each branch 34 may also include a band-pass filter 40 to attenuate out-of-band noise, and a DC-blocking capacitor 42 in order to prevent DC current drain from the DAC 20 .
- Each RF drive signal S x (t) is supplied to a respective branch MZ interferometer 28 x of the complex E/O converter 22 , so as to generate a corresponding branch optical signal E x (t).
- the branch optical signals E x (t) are combined at the E/O converter output 24 to produce the output optical signal E O (t) 8 .
- the high speed digital stage 30 of the signal path is preferably driven at a sample rate of at least double the expected bit rate of the input data signal x(t), in order to satisfy Nyquist's criteria for the input data signal x(t). This is primarily an operational consideration, which ensures that the output optical signal 8 will contain sufficient information to enable recovery of the input data signal x(t) at a receiver-end of the link. In principal, any sample rate may be used, although higher sample rates will be preferred.
- Each of the DACs 20 x is designed to provide digital-to-analog conversion at a resolution of M-bits, where (M) is greater than 2. The actual resolution chosen for the DACs is a balance between precision and cost.
- the complex optical E/O converter 22 is provided by a dual branch “main” Mach-Zehnder (MZ) interferometer 26 , having a respective “branch” MZ interferometer 28 x within each branch.
- MZ interferometers 28 x is independently driven by a respective one of the branch drive signals S x (t).
- the complex E/O converter 22 also includes a respective direct-current (DC) input port for each of the main and branch MZ interferometers 26 , 28 x, each of which supplies a bias signal to its respective MZ interferometer.
- DC direct-current
- FIG. 4 is suitable for implementing accurate synthesis of a desired output optical signal having a single polarization direction.
- this architecture can readily be extended to provide accurate optical synthesis in two orthogonal polarization directions.
- the entire signal path can be duplicated and run in parallel, with each signal path controlling a respective polarization direction.
- a preferred option would be to utilize a single high speed digital stage 30 to compute digital drive signals V x (n) for both polarization directions.
- This enables the signal processor 16 and linearizer 18 to also control the polarization state of the output optical signal E O (t), which facilitates compensation of, for example, polarization mode dispersion and polarization dependent loss.
- a control system in accordance with an embodiment of the present invention comprises a controller unit 44 , a feedback path 46 which samples the optical signal E O (t) at the E/O converter output 24 , and a feed-forward path 48 which samples the input data signal x(t).
- the controller unit 44 which may be provided as any suitable combination of hardware and software, implements a set of parallel control loops for controlling a variety of parameters of the signal path, such as: the target optical E-field (via the compensation function C[ ]); the digital drive signals (via the linearizer transfer function T[ ]), RF stage path gain (via the VGAs), and E/O converter bias.
- Each control loop involves injecting one or more dither signals into the signal path; detecting artefacts of these dither signals within the output optical signal E O (t); using the detected artefacts to compute one or more cost-function values that are indicative of an error between the target E-field and the actual E-field of the output optical signal; and, based on the results of the cost-function calculations, adjusting one or more parameters of the signal path so as to optimize transmitter performance, and thereby minimize the error.
- the feedback path 4 b comprises an optical coupler 50 , such as a conventional 20 dB coupler for sampling the output optical signal, and a detector block 52 for detecting predetermined artefacts within the sampled optical signal.
- the detector block 52 includes a P-Intrinsic-N (PIN) diode 54 which emits a current I PIN that is proportional to a power level of the sampled optical signal.
- PIN P-Intrinsic-N
- the PIN diode output is sampled by an Analog-to-Digital A/D converter 56 , and the sample values supplied to a set of normalized correlators 58 , each of which is controlled, in a manner known in the art, to detect signal components of a respective predetermined frequency.
- the output of each normalized correlator is proportional to the power level of the detected signal components, and is supplied to the controller unit 44 .
- low frequency components of the input data signal x(t) transiting the signal path can generate noise in a control loop.
- the feed-forward path 48 implements a decimation function 60 which approximates a time-integral of the input data signal x(t) over a predetermined period. This time integral provides information regarding the low-frequency content of the data signal x(t), and thus the noise in each control loop due to high speed data traversing the data path. This function enables the controller to estimate such low frequency components, and adjust the cost function computations accordingly.
- a “dither signal” can be any signal having a known frequency that is inserted into the signal path, and yields detectable artefacts in the output optical signal E O (t). These artefacts may take the form of optical power modulation at a frequency corresponding to one or more harmonics of a single dither, a beat of two or more dithers, and/or functions of these, as will be described in greater detail below.
- the dither signal can be composed of any of: a pure tone (i.e. a sinusoidal signal); a modulated tone; or a digital signal, which may be periodic (e.g. a clock signal), pseudo-random, or may contain data (e.g. control channel signalling), and can be inserted at any suitable point into the signal path.
- the dither signal frequency will be selected to avoid overlap with other dither signals (and/or their harmonics), and to avoid interference with input data x(t) traversing the signal path. Dither frequencies of 1 MHz and below are preferred. Representative dithers that are contemplated in the present invention include:
- Dithers a) and b) above can be implemented by suitable control of the signal processor 16 , and in particular by suitable selection of the parameters of the compensation function C[ ].
- Dithers c) and d) above can be implemented in the linearizer 18 .
- Additive and/or multiplicative variations of the digital drive signals V x (n) can be implemented by suitable selection of parameters of the linearizer transfer function T[ ]. It is worth noting that this operation can be implemented to independently insert dithers into the Re and Im components of the output optical signal E O (t). Since the dithers are orthogonal, the resulting artefacts in the output optical signal E O (t) will also be orthogonal, and thus will not interfere. Swapping between different transfer functions can be accomplished in various ways. For example, in embodiments in which the linearizer 18 is implemented as a Random-Access Memory Look-Up Table (RAM-LUT) 62 , as shown in FIGS.
- RAM-LUT Random-Access Memory Look-Up Table
- the dither signal may then be injected into the data path by selectively enabling one of the RAM-LUTs, as shown in FIG. 6 a, or alternatively by selecting the output of the one of the RAM-LUTs for further processing, as shown in FIG. 6 b.
- Dithers e) and f) are similar to conventional dithers, in that they are additive signals which vary the amplitude of the drive signals and/or bias signals in a conventional manner.
- the dither signals share the following characteristics. Firstly, the frequency of each dither is selected so that its harmonics will not interfere with any other dither (or its harmonics), or with data. Secondly, in order to avoid corrupting the output optical E-field, the magnitude of the effect of each dither is preferably selected to be less than a least significant bit (LSB) of the DACs 20 , at least during run-time.
- LSB least significant bit
- the insertion of such “sub-LSB” dithers is implemented by computing at least the digital drive signal values V R (n) and V L (n), including dithers, at a resolution of N bits, which is greater than the M-bit resolution of the DAC 20 .
- the N-bit drive signal values V R (n) and V L (n) are then scaled, by dividing the computed value by a predetermined scaling factor, and then the scaling result quantized and clipped to an M-bit value that is supplied to the DAC input.
- the scaling, quantization, and clipping operations may be implemented in various ways.
- these operations can be implemented as part of the linearizer transfer function T[ ], in which case these steps are purely logical, and may or may not be executed as discrete processing steps, as may be seen in FIGS. 6 a and 6 b.
- these operations can be implemented within the parallel data bus between the (N-bit) linearizer output, and the M-bit DAC input. An example of this latter approach is illustrated in FIG. 7 .
- Single-sided clipping and quantization can then be performed (in parallel) by truncating bits lying above the MSB and below the LSB of the DAC 20 , respectively.
- the drive signal values V x (n) can by computed by the linearizer 18 with a fixed offset equivalent to 1 ⁇ 2 of the LSB of the DAC 20 .
- This offset in combination with the thresholding function of the DAC 20 , converts the above quantization function (literally, truncation of bits lying below the LSB of the DAC 20 ) into a rounding operation.
- FIG. 7 presents a simplified view for illustration purposes only.
- a more practical system will accommodate both positive and negative drive signal values V x (n), and double-sided clipping (i.e. clipping of both positive and negative excursions exceeding respective maximum and minimum) limits will be performed. It is considered that implementation of such a system, based on the teaching provided herein, will be well within the purview of those skilled in the art.
- the controller unit 44 computes a cost function value which is indicative of an error between the actual output optical signal E-field E O (t) and the desired or target E-field.
- a low cost PIN detector 54 of the feed-back path 46 only detects optical power
- such a low cost PIN detector 54 will also tend to detect only the low-frequency portion of the optical signal E-field E O (t). Consequently, a direct comparison between actual and target E-fields is not possible, with a low cost, low speed PIN detector 54 of the type contemplated by the present invention.
- the overall system response is dependent on many variables, such a direct comparison does not always provide useful information as to which parameter should be changed to remove any detected error.
- each cost function defines an n-dimensional “control surface” which relates a set of one or more detectable artefacts of the output optical E-field E O (t) to a parameter of the signal path.
- the cost function value is controlled by adjusting the involved signal parameter, and is indirectly indicative of a respective feature of the output optical E-field E O (t).
- desired features of the output optical E-field E O (t) can be obtained by progressively adjusting the parameter (e.g. in a step-wise manner) to drive the cost function to a predetermined value which corresponds with a desired feature of the output optical signal.
- Representative optical signal features which can be controlled in this manner include, but are not necessarily limited to: spectral features such as carrier suppression; polarization state; and balance between the Re and Im components of the E-field E O (t).
- Representative path parameters contemplated in the present invention include: the target optical E-field (via the compensation function C[ ]); the digital drive signals V x (n) (via the linearizer transfer function T[ ]), RF stage path gain (via the VGAs 38 x ), and bias points of the main and branch MZ interferometers of the complex E/O converter 22 .
- cost functions are selected to be dependent upon a single parameter, so as to constrain degrees of freedom.
- cost functions are preferably defined in such a way as to be largely independent of device calibration. This may be accomplished through the formulation of the cost function itself and/or selection of a desired or target cost function value.
- the cost function can be formulated such that the desired optical E-field feature corresponds with a known value of the cost function, independently of device calibration.
- the cost function can be formulated such that the desired optical E-field feature corresponds with a local maximum, minimum or zero of the cost function. The location of these points on the cost function control surface will normally be independent of device calibration, even though the actual value (in the case of maxima and minima) will not be.
- device calibration variables are known, or can be calculated, and used in computing the cost function. Computation of device calibration variables relies on the fact that insertion of the dither into the multi-bit digital stage 30 of the signal path enables high precision control over the dither signal. This precision enables a correspondingly precise calculation of at least some device calibration variables, such as, for example, the coupling efficiency of each branch MZ interferometer 28 x (as will be described in greater detail below). Knowledge of device calibration variables in this manner(whether known in advance or calculated as described above) enables a path parameter to be controlled to yield a desired feature of the output optical signal which does not correspond with any of a maximum, minimum or zero of the cost function control surface.
- the purpose of the bias control loops is to drive each of the nested MZ interferometers 26 , 28 x to respective optimum bias points, so as to achieve desired optical waveform characteristics. While various techniques are known for controlling the bias point of an individual MZ interferometer driven by a bi-state drive signal, it can be shown that these techniques will not work satisfactorily in the case of nested main and branch MZ interferometers of the type illustrated in FIG. 4 . Nor will the conventional techniques work with an E/O converter (of any type) driven by high resolution drive signals S x (t) of the type contemplated in the present invention.
- the dither dx is characterized as a sinusoidal signal for convenience only. Any dither signal waveform may be used.
- the dithers can be either additive or multiplicative, as desired, and may be inserted either by the linearizer, or by varying the gain of the respective RF signal path (i.e. via the VGAs).
- the frequency ⁇ dx of each dither dx is selected to avoid overlap between respective harmonics of the two dithers.
- Quadrature control involves controlling the bias point of the branch MZ interferometer 26 by adjusting the bias voltage V dcQ .
- Each dither dx produces artefacts in the output optical signal, which can be detected as amplitude modulation of the optical power
- 2 at a frequency ⁇ beat
- the amplitude H beat of the beat component can be detected using a normalized correlator 58 in the detector block 52 of the feed-back path 46 .
- the bias voltages V dcR and V dcL are held constant during adjustment of bias voltage V dcQ .
- a target phase angle of ⁇ x of each branch MZ interferometer 28 x is known, or can be computed in advance, and it is desired to control the actual phase angles to the applicable target value.
- the branch MZ interferometer 26 is assumed to be maintained in quadrature, using the method described above.
- the amplitude of each of these harmonics can be detected using conventional normalized correlators 58 in the detector block 52 of the feed-back path 46 .
- each branch MZ interferometer 28 x can be computed as either
- the bias angle of the involved branch MZ interferometer 28 x can be determined by
- the phase angle of ⁇ x of each branch MZ interferometer 28 x is not known, or can not be computed in advance. For example, it may be desired to drive the complex E/O converter 22 to generate a carrier suppressed output optical signal E O (t). In this situation, it is desired to maximize the degree of carrier suppression, but the precise phase angles ⁇ x required to achieve this result may not be known.
- the main MZ interferometer 26 is maintained in quadrature, using, for example, the method described above.
- the first harmonics H R 1 and H L 1 provide a useful cost function for controlling the bias voltages V dcR and V dcL . It is a simple matter to implement a stepping function that incrementally adjusts the bias voltages V dcR and V dcL to drive the respective first harmonics H R 1 and H L 1 to a minimum value, and thereby achieve maximum carrier suppression.
- An alternative bias control loop can be implemented, utilizing the fact that, within the complex E/O converter 22 , the respective branch signals E R (t) and E L (t) are combined using a signal combiner 68 , as shown in FIG. 8 .
- the signal combiner 64 also has a (normally unused) second output, which carries the vector difference of the two branch signals.
- This second output can be tapped and supplied to the controller unit 44 via a respective second feedback path (not shown). It will be noted that tapping this second output increases device costs due to the added waveguide, photodetector and detector block, but does not reduce the optical energy in the “main” signal transmitted through the link.
- 2 E R (t) 2 ⁇ E L (t) 2 +2E R (t)E L (t)sin( ⁇ q).
- 2 E R (t) 2 ⁇ E L (t) 2 +2E R (t)E L (t)sin( ⁇ q).
- the bias voltage V DCQ can be driven to eliminate the mixing term 2E R (t)E L (t)sin( ⁇ q) from both
- the purpose of the System Balance control loop is to ensure that the branches of the optical transmitter (i.e. between the linearizer 18 and the output 24 of the complex E/O converter 22 ) are balanced. In practice, this means that equal values of the digital drive signals V R (n) and V L (n) produce equal optical power levels of the two branch optical signals E R (t) and E L (t).
- the coupling efficiencies p tx of the branch MZ interferometers 28 x can by computed using the methods described above. Since these computations are based on ratios of harmonic amplitudes, they are insensitive to path gain and E/O converter bias angle. Accordingly, it is possible to remove the effect of coupling efficiency p tx differences, and thereby isolate differences in the respective path gains G px .
- G px is the gain of the entire RF branch between the linearizer 18 and the complex E/O converter 22 .
- the purpose of the Common Gain control loop is to optimize the overall system gain of the optical transmitter (i.e. between the signal processor 16 and the output 24 of the complex E/O converter 22 ). In practice, this means that a change in the phase and/or amplitude of the target E-field produces an equal change in the output optical E-field E O (t).
- any residual error between the target and output optical E-field will be due the overall system gain of the optical transmitter between the signal processor 16 and the output 24 of the complex E/O converter 22 .
- a multiplicative low frequency binary (square wave) dither d(m, ⁇ ) is applied to the amplitude A of the target E-field E T (t).
- E T (t) ⁇ (1+d)Ae j ⁇ c t is applied to the amplitude A of the target optical E-field E T (t).
- the dither magnitude (m) may, for example, be 0.01.
- the dither d(m, ⁇ ) produces artefacts in the output optical signal E O (t), which can be detected as amplitude modulation of the optical power
- the peak-to-peak amplitude A d of the dither component can be detected using a normalized correlator 58 in the detector block 52 of the feed-back path 46 .
- the high speed data signal x(t) traversing the signal path appears a noise in each of the above-noted control loops.
- the spectral content of the data signal x(t) lying at the extreme low-end of the frequency spectrum is of particular concern.
- the feed-forward path 48 comprises a decimation function 60 , which counts the number of binary 1's in the high speed data signal x(t) during a predetermined time interval.
- This time interval may conveniently be selected to correspond with the period of a dither signal, but can be any desired number of bits. Since the high speed data signal x(t) is a serial binary signal, which steps between known voltage states, it will be appreciated that the integral over a selected time interval (i.e. number of bits) is directly proportional to the number of binary 1's and 0's received during that interval. Since the number of bits received during the interval is known, then this value can be determined by merely counting the number of binary 1's received during the interval in question.
- the decimation function repeatedly counts the number of binary 1's received within successive time intervals. Each count value is stored in memory, which thereby produces a time sequence of values representing a filtered version of the input data signal x(t). This time sequence of values is then analyzed by the controller unit to extract low-frequency component information, for example by computing a Fast Fourier Transform (FFT). This information provides an estimate the contribution that the data signal x(t) makes to the harmonic and beat amplitudes detected by the normalized correlators 58 of the feedback path 46 . This estimated contribution can then be subtracted from the detected harmonic and beat amplitudes, prior to computation of the respective cost function values used to drive each control loop.
- FFT Fast Fourier Transform
- decimation function approximates a low-pass filter for extracting low-frequency components of the input data stream x(t).
- this filter function could be omitted entirely, and the input data stream x(t) evaluated directly to extract low frequency components.
- the necessary evaluation i.e. FFT computation
- each dither signal has a respective frequency, which enables the detector to isolate corresponding signal components in the output optical signal E O .
- successful operation of the detector requires that the dither frequencies are selected to avoid interference between the dither signals and their respective harmonics. In general, this can be accomplished by selecting the dither frequencies such that no frequency corresponds to an harmonic of any other dither signal. In the case of symmetrical square-wave dither signals, only the odd harmonics need to be taken into account.
Abstract
Description
-
- the drive signals Sx(t)must be supplied to respective branches of the E/
O converter 22 with substantially zero phase and amplitude error; - generation of the drive signals must take into account the known response of the E/
O converter 22, as well as “component drift” due to changes in temperature, and aging; and - the E/O converter must be driven to an optimal bias point, which, for the complex E/
O converter 22 ofFIG. 3 , requires optimal bias settings of both branch MZ interferometers 28 x, and themain interferometer 26.
- the drive signals Sx(t)must be supplied to respective branches of the E/
-
- (a) An E-field vector inserted at a selected phase offset (e.g. ±45°) to the target E-field or at a selected frequency offset from the target E-field (e.g. as a narrow side-band);
- (b) A variation of the amplitude and/or phase of the target E-field, or, similarly, of the Real (Re) and imaginary (Im) components of the target E-field;
- (c) An additive or multiplicative variation of one or both of the digital drive signals;
- (d) Swapping between two or more different linearizer transfer functions T[ ];
- (e) a sinusoidal or digital variation of the RF path gain, via the VGAs; and
- (f) sinusoidal or digital variation of the E/O converter bias.
where Hxn is the detected amplitude of the nth harmonic. With the coupling efficiency computed as above, the bias angle of the involved branch MZ interferometer 28 x can be determined by
It should be noted that this computation of the bias angle θx is entirely independent of device calibration, and therefore provides a useful cost function for controlling the bias voltages VdcR and VdcL. It is a simple matter to implement a stepping function that incrementally adjusts each one of the bias voltages VdcR and VdcL to drive the respective bias angles of the branch MZ interferometers 28 x to the desired phase angle (i.e. θx=π/2)
Unknown Target Phase Angle
is directly proportional to (ptxSdx), where Sdx is the peak amplitude of the dither dx=Adx Cos(ωdxt) received by the respective branch MZ interferometer 28 x. That is, ptxSdx=ptxGpxAdx, where Gpx is the path gain between the point of insertion of the dither signal dx and the complex E/
which enables direct calculation of the system imbalance
from the detected first and second
to an optimum value (in this case
and thereby achieve system balance. It can be shown that the relative gain Grel can be used to balance the branches 28 of the E/O converter 4 in this manner, independently of common gain setting, bias error, and even extinction ratio.
Common Gain Control Loop
is directly proportional on the overall system gain. It can also be shown that, as the difference between the target and output optical E-fields goes to zero, the value of this ratio approaches a unique value (i.e.
independently of device calibration. Accordingly, it provides a useful cost function for controlling the common gain Gcom component of the VGA gains GR and GL. It is a simple matter to implement a stepping function that incrementally adjusts the common gain Gcom to drive
to the desired target value (i.e.
Feed-Forward Path
Claims (80)
Priority Applications (3)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US11/008,515 US7676161B2 (en) | 2004-12-10 | 2004-12-10 | Modulation E-field based control of a non-linear transmitter |
US11/067,011 US7787778B2 (en) | 2004-12-10 | 2005-02-28 | Control system for a polar optical transmitter |
US12/830,663 US8059970B2 (en) | 2004-12-10 | 2010-07-06 | Control system for a polar optical transmitter |
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US11/008,515 US7676161B2 (en) | 2004-12-10 | 2004-12-10 | Modulation E-field based control of a non-linear transmitter |
Related Child Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US11/067,011 Continuation-In-Part US7787778B2 (en) | 2004-12-10 | 2005-02-28 | Control system for a polar optical transmitter |
Publications (2)
Publication Number | Publication Date |
---|---|
US20060127102A1 US20060127102A1 (en) | 2006-06-15 |
US7676161B2 true US7676161B2 (en) | 2010-03-09 |
Family
ID=36584026
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US11/008,515 Active 2028-03-19 US7676161B2 (en) | 2004-12-10 | 2004-12-10 | Modulation E-field based control of a non-linear transmitter |
Country Status (1)
Country | Link |
---|---|
US (1) | US7676161B2 (en) |
Cited By (13)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US20090297153A1 (en) * | 2005-06-13 | 2009-12-03 | Broadband Royalty Corporation | Four quadrant linearizer |
US20100061738A1 (en) * | 2006-12-23 | 2010-03-11 | Telefonaktiebolaget Lm Ericsson (Publ) | Signal Processor for Compensating for Optical Fiber Chromatic Dispersion |
US20100189443A1 (en) * | 2009-01-23 | 2010-07-29 | Nortel Networks Limited | High speed signal generator |
US20100202785A1 (en) * | 2007-09-18 | 2010-08-12 | National Institute Of Information And Communications Technology | Quadrature amplitude modulation signal generating device |
US20110222850A1 (en) * | 2010-03-11 | 2011-09-15 | Nortel Networks Limited | Self test of a dual polarization transmitter |
US8103168B1 (en) * | 2006-11-09 | 2012-01-24 | Lockheed Martin Corporation | RF discrete time optical frequency translator |
US20130034188A1 (en) * | 2006-04-04 | 2013-02-07 | Apple Inc. | Signal Transmitter Linearization |
US20140029938A1 (en) * | 2012-07-26 | 2014-01-30 | Fujitsu Limited | Optical transmission system and method for monitoring polarization dependent characteristics of optical transmission line |
US20140147117A1 (en) * | 2012-11-28 | 2014-05-29 | Hitachi, Ltd. | Optical multilevel signal pre-equalization circuit, optical multilevel signal pre-equalization transmitter, and polarization-multiplexed pre-equalization transmitter |
US10014947B2 (en) | 2016-02-18 | 2018-07-03 | Ciena Corporation | Mitigation of electrical-to-optical conversion impairments induced at transmitter |
US10341022B2 (en) * | 2016-12-28 | 2019-07-02 | Zte Corporation | Optical pulse amplitude modulation transmission using digital pre-compensation |
US20200059301A1 (en) * | 2018-08-20 | 2020-02-20 | Photonic Technologies (Shanghai) Co., Ltd. | Pulse generation module, and optical communication transmitter system and non-linear equalizing method thereof |
US20230046863A1 (en) * | 2018-04-20 | 2023-02-16 | Neophotonics Corporation | Method and apparatus for bias control with a large dynamic range for mach-zehnder modulators |
Families Citing this family (40)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US7787778B2 (en) * | 2004-12-10 | 2010-08-31 | Ciena Corporation | Control system for a polar optical transmitter |
US7676161B2 (en) * | 2004-12-10 | 2010-03-09 | Nortel Networks Limited | Modulation E-field based control of a non-linear transmitter |
JP4422661B2 (en) * | 2005-08-31 | 2010-02-24 | 富士通株式会社 | Driving voltage setting method for differential quadrature phase shift modulator |
US20070116476A1 (en) * | 2005-11-18 | 2007-05-24 | Futurewei Technologies, Inc. | Method and apparatus for generating optical duo binary signals with frequency chirp |
US8160453B1 (en) | 2006-03-30 | 2012-04-17 | Rockstar Bidco, LP | Protection switching with transmitter compensation function |
JP5405716B2 (en) * | 2006-09-29 | 2014-02-05 | 富士通株式会社 | Optical transmitter |
JP5211528B2 (en) * | 2007-03-29 | 2013-06-12 | 富士通株式会社 | Optical modulation device and optical modulation system switching method |
US8131148B2 (en) * | 2008-09-16 | 2012-03-06 | Ciena Corporation | Optical transmitter error reduction using receiver feedback |
JP5476697B2 (en) | 2008-09-26 | 2014-04-23 | 富士通株式会社 | Optical signal transmitter |
JP2011022479A (en) * | 2009-07-17 | 2011-02-03 | Mitsubishi Electric Corp | Multi-value optical transmitter |
JP5195677B2 (en) | 2009-07-28 | 2013-05-08 | 富士通株式会社 | Optical signal transmitting apparatus and polarization multiplexed optical signal control method |
JP5793854B2 (en) * | 2010-11-24 | 2015-10-14 | 富士通株式会社 | COMMUNICATION SYSTEM, MEASUREMENT DEVICE, MEASUREMENT METHOD, AND CONTROL METHOD |
EP2709293A4 (en) * | 2011-05-13 | 2015-04-22 | Nec Corp | Synchronous signal transmission system, synchronous drive system for optical modulator, synchronous signal transmission method, and non-temporary computer-readable medium storing program therefor |
JP5868271B2 (en) * | 2012-06-26 | 2016-02-24 | 三菱電機株式会社 | Optical transmission device and optical transmission method |
WO2014181869A1 (en) * | 2013-05-09 | 2014-11-13 | 日本電信電話株式会社 | Optical modulator driver circuit and optical transmitter |
JP6229795B2 (en) * | 2013-07-31 | 2017-11-15 | 日本電気株式会社 | Signal generation apparatus and signal generation method |
US9281898B2 (en) * | 2014-02-19 | 2016-03-08 | Futurewei Technologies, Inc. | Mach-Zehnder modulator bias control for arbitrary waveform generation |
JP6354553B2 (en) * | 2014-12-02 | 2018-07-11 | 住友電気工業株式会社 | Bias control circuit and optical transmitter including the same |
JP6620409B2 (en) * | 2015-03-11 | 2019-12-18 | 富士通株式会社 | Optical transmitter, optical transmission system, and optical communication control method |
US10148363B2 (en) | 2015-12-08 | 2018-12-04 | Zte Corporation | Iterative nonlinear compensation |
US10148465B2 (en) * | 2015-12-08 | 2018-12-04 | Zte Corporation | Training assisted joint equalization |
JP6589659B2 (en) * | 2016-01-21 | 2019-10-16 | 富士通株式会社 | Transmission apparatus and transmission method |
US10042190B2 (en) * | 2016-06-10 | 2018-08-07 | Futurewei Technologies, Inc. | Second order detection of two orthogonal dithers for I/Q modulator bias control |
JP2018019150A (en) * | 2016-07-26 | 2018-02-01 | 富士通株式会社 | Optical transmitter, photoreceiver, optical communication system, optical transmission control method and optical transmission reception control method |
CN108667520B (en) * | 2017-03-31 | 2020-12-29 | 富士通株式会社 | Bias control device and method for optical transmitter modulator and optical transmitter |
JP7024234B2 (en) * | 2017-07-18 | 2022-02-24 | 富士通株式会社 | Optical transmitter and control method of optical transmitter |
GB201821175D0 (en) * | 2018-12-24 | 2019-02-06 | Leonardo Mw Ltd | An electro-optical modular |
US10892847B2 (en) | 2019-04-18 | 2021-01-12 | Microsoft Technology Licensing, Llc | Blind detection model optimization |
US10998982B2 (en) | 2019-04-18 | 2021-05-04 | Microsoft Technology Licensing, Llc | Transmitter for throughput increases for optical communications |
US10951342B2 (en) | 2019-04-18 | 2021-03-16 | Microsoft Technology Licensing, Llc | Throughput increases for optical communications |
US10897315B2 (en) | 2019-04-18 | 2021-01-19 | Microsoft Technology Licensing, Llc | Power-based decoding of data received over an optical communication path |
US10938485B2 (en) | 2019-04-18 | 2021-03-02 | Microsoft Technology Licensing, Llc | Error control coding with dynamic ranges |
US10911152B2 (en) | 2019-04-18 | 2021-02-02 | Microsoft Technology Licensing, Llc | Power-based decoding of data received over an optical communication path |
US10862591B1 (en) | 2019-04-18 | 2020-12-08 | Microsoft Technology Licensing, Llc | Unequal decision regions for throughput increases for optical communications |
US10742325B1 (en) * | 2019-04-18 | 2020-08-11 | Microsoft Technology Licensing, Llc | Power-based encoding of data to be transmitted over an optical communication path |
US10873393B2 (en) | 2019-04-18 | 2020-12-22 | Microsoft Technology Licensing, Llc | Receiver training for throughput increases in optical communications |
US11018776B2 (en) | 2019-04-18 | 2021-05-25 | Microsoft Technology Licensing, Llc | Power-based decoding of data received over an optical communication path |
US10911155B2 (en) | 2019-04-18 | 2021-02-02 | Microsoft Technology Licensing, Llc | System for throughput increases for optical communications |
US20200278589A1 (en) * | 2020-05-18 | 2020-09-03 | Intel Corporation | Optical architecture with hybrid on-silicon iii-v modulator |
CN114448500A (en) * | 2020-11-03 | 2022-05-06 | 富士通株式会社 | Phase frequency response measuring method and device |
Citations (56)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5148503A (en) | 1991-05-29 | 1992-09-15 | Crystal Technology, Inc | Apparatus and method for linearized cascade coupled integrated optical modulator |
EP0524758A2 (en) | 1991-07-23 | 1993-01-27 | AT&T Corp. | Distortion compensation for analog optical systems |
US5301058A (en) | 1990-12-31 | 1994-04-05 | Gte Laboratories Incorporated | Single sideband optical modulator for lightwave systems |
US5311346A (en) | 1992-06-17 | 1994-05-10 | At&T Bell Laboratories | Fiber-optic transmission polarization-dependent distortion compensation |
US5349312A (en) | 1993-05-28 | 1994-09-20 | Raytheon Company | Voltage variable attenuator |
US5408498A (en) | 1991-07-03 | 1995-04-18 | Sharp Kabushiki Kaisha | Serial-signal transmission apparatus |
US5416626A (en) | 1993-02-02 | 1995-05-16 | Northern Telecom Limited | Optically amplified transmission systems |
US5446574A (en) | 1993-03-26 | 1995-08-29 | Telefonaktiebolaget Lm Ericsson | System and method for dispersion compensation in fibre optic high speed systems |
US5513029A (en) | 1994-06-16 | 1996-04-30 | Northern Telecom Limited | Method and apparatus for monitoring performance of optical transmission systems |
US5579328A (en) | 1995-08-10 | 1996-11-26 | Northern Telecom Limited | Digital control of laser diode power levels |
US5761225A (en) | 1996-05-23 | 1998-06-02 | Litton Systems, Inc. | Optical fiber amplifier eled light source with a relative intensity noise reduction system |
US5892858A (en) | 1997-03-27 | 1999-04-06 | Northern Telecom Limited | Duobinary coding and modulation technique for optical communication systems |
US5949560A (en) | 1997-02-05 | 1999-09-07 | Northern Telecom Limited | Optical transmission system |
US5999258A (en) | 1997-06-26 | 1999-12-07 | Nortel Networks Corporation | Optical interference measurement method and system |
EP0971493A2 (en) | 1998-07-08 | 2000-01-12 | Fujitsu Limited | Chromatic dispersion and nonlinearity compensation in optical fibre communication |
US6067180A (en) | 1997-06-09 | 2000-05-23 | Nortel Networks Corporation | Equalization, pulse shaping and regeneration of optical signals |
US6115162A (en) | 1995-01-10 | 2000-09-05 | Harris Corporation | Double side band, carrier suppressed modulated coherent fiber optic link |
US6124960A (en) | 1997-09-08 | 2000-09-26 | Northern Telecom Limited | Transmission system with cross-phase modulation compensation |
US6128111A (en) | 1996-12-19 | 2000-10-03 | Nortel Networks Corporation | Monitoring of nonlinear effects |
WO2001003339A1 (en) | 1999-06-30 | 2001-01-11 | Koninklijke Philips Electronics N.V. | Pre-shaping laser modulation signals to increase modulation index |
US6205262B1 (en) | 1997-05-01 | 2001-03-20 | Alliance Fiber Optics Products, Inc. | Optical recirculation depolarizer and method of depolarizing light |
US6262834B1 (en) | 2000-02-23 | 2001-07-17 | The United States Of America As Represented By The Secretary Of The Navy | Wideband single sideband modulation of optical carriers |
US6278539B1 (en) * | 1998-11-25 | 2001-08-21 | Fujitsu Limited | Optical modulation apparatus and method of controlling |
US20010028760A1 (en) | 2000-03-03 | 2001-10-11 | Yaffe Henry H. | Methods and apparatus for compensating chromatic and polarization mode dispersion |
US6304369B1 (en) | 1999-07-29 | 2001-10-16 | Harmonic, Inc. | Method and apparatus for eliminating noise in analog fiber links |
WO2001091342A2 (en) | 2000-05-24 | 2001-11-29 | Purdue Research Foundation | Method and system for polarization control and polarization mode dispersion compensation for wideband optical signals |
US20020018268A1 (en) | 1998-11-04 | 2002-02-14 | Corvis Corporation Attn: Intellectual Property Department | Optical distortion compensation apparatuses, methods, and systems |
US20020024694A1 (en) | 2000-05-12 | 2002-02-28 | Newell Laurence J. | Control channel for an optical communications system utilizing frequency division multiplexing |
US6388786B1 (en) * | 1997-08-15 | 2002-05-14 | Nec Corporation | Method for generating duobinary signal and optical transmitter using the same method |
WO2002043340A2 (en) | 2000-11-22 | 2002-05-30 | Broadcom Corporation | Method and apparatus to identify and characterize nonlinearities in optical communications channels |
EP1223694A2 (en) | 2001-01-10 | 2002-07-17 | Fujitsu Limited | Dispersion compensating method, dispersion compensating apparatus and optical transmission system |
US20020106148A1 (en) | 1999-05-24 | 2002-08-08 | Philips Electronics North America Corporation | Optical communication with pre-compensation for odd order distortion in modulation and transmission |
US6441932B1 (en) | 1999-02-09 | 2002-08-27 | The United States Of America As Represented By The Secretary Of The Air Force | Intensity noise suppression using differential delay cancellation in external modulation links |
EP1237307A2 (en) | 2001-03-02 | 2002-09-04 | Fujitsu Limited | Receiving apparatus and method for detecting, measuring and compensating waveform degradation of received signal |
US6473214B1 (en) * | 1999-04-01 | 2002-10-29 | Nortel Networks Limited | Methods of and apparatus for optical signal transmission |
US6473013B1 (en) | 2001-06-20 | 2002-10-29 | Scott R. Velazquez | Parallel processing analog and digital converter |
US20020167693A1 (en) * | 2000-12-21 | 2002-11-14 | Quellan, Inc. | Increasing data throughput in optical fiber transmission systems |
US20030011847A1 (en) | 2001-06-07 | 2003-01-16 | Fa Dai | Method and apparatus for adaptive distortion compensation in optical fiber communication networks |
US6522438B1 (en) * | 1999-10-04 | 2003-02-18 | Lucent Technologies Inc. | High-speed optical duobinary modulation scheme |
US6559994B1 (en) | 1999-08-18 | 2003-05-06 | New Elite Technologies, Inc. | Optical fiber transmitter for long distance subcarrier multiplexed lightwave systems |
US6580532B1 (en) | 1999-01-28 | 2003-06-17 | California Institute Of Technology | Opto-electronic techniques for reducing phase noise in a carrier signal by carrier supression |
US6592274B2 (en) * | 2001-01-29 | 2003-07-15 | Stratalight Communications, Inc. | Transmission and reception of duobinary multilevel pulse-amplitude-modulated optical signals using finite-state machine-based encoder |
US6623188B1 (en) * | 2002-02-08 | 2003-09-23 | Optiuh Corporation | Dispersion tolerant optical data transmitter |
US20030198478A1 (en) * | 2002-04-23 | 2003-10-23 | Quellan, Inc. | Method and system for generating and decoding a bandwidth efficient multi-level signal |
US20040081470A1 (en) * | 2000-12-21 | 2004-04-29 | Robert Griffin | Optical communications |
US6970655B2 (en) * | 2001-03-02 | 2005-11-29 | Nec Corporation | Method and circuit for generating single-sideband optical signal |
US20060127102A1 (en) * | 2004-12-10 | 2006-06-15 | Nortel Networks Limited | Modulation E-field based control of a non-linear transmitter |
US7068948B2 (en) * | 2001-06-13 | 2006-06-27 | Gazillion Bits, Inc. | Generation of optical signals with return-to-zero format |
US7075695B2 (en) * | 2004-03-01 | 2006-07-11 | Lucent Technologies Inc. | Method and apparatus for controlling a bias voltage of a Mach-Zehnder modulator |
US7155134B2 (en) * | 2002-03-22 | 2006-12-26 | Agere Systems Inc. | Pulse amplitude modulated transmission scheme for optical channels with soft decision decoding |
US7190904B2 (en) * | 2001-09-26 | 2007-03-13 | International Business Machines Corporation | Wavelength modulation for optical based switching and routing |
US20070092263A1 (en) * | 1999-10-20 | 2007-04-26 | Agazzi Oscar E | Method, apparatus and system for high-speed transmission on fiber optic channel |
US7236707B2 (en) * | 2002-10-21 | 2007-06-26 | Main Street Ventures Llc | All-optical compression systems |
US7317877B2 (en) * | 2002-04-16 | 2008-01-08 | Broadwing Corporation | Optical communications systems, devices, and methods |
US20080025731A1 (en) * | 2002-12-03 | 2008-01-31 | Daniel Mahgerefteh | Versatile compact transmitter for generation of advanced modulation formats |
US7330666B1 (en) * | 2003-01-31 | 2008-02-12 | Ciena Corporation | Method and apparatus for controlling modulator phase alignment in a transmitter of an optical communications system |
-
2004
- 2004-12-10 US US11/008,515 patent/US7676161B2/en active Active
Patent Citations (57)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5301058A (en) | 1990-12-31 | 1994-04-05 | Gte Laboratories Incorporated | Single sideband optical modulator for lightwave systems |
US5148503A (en) | 1991-05-29 | 1992-09-15 | Crystal Technology, Inc | Apparatus and method for linearized cascade coupled integrated optical modulator |
US5408498A (en) | 1991-07-03 | 1995-04-18 | Sharp Kabushiki Kaisha | Serial-signal transmission apparatus |
EP0524758A2 (en) | 1991-07-23 | 1993-01-27 | AT&T Corp. | Distortion compensation for analog optical systems |
US5311346A (en) | 1992-06-17 | 1994-05-10 | At&T Bell Laboratories | Fiber-optic transmission polarization-dependent distortion compensation |
US5416626A (en) | 1993-02-02 | 1995-05-16 | Northern Telecom Limited | Optically amplified transmission systems |
US5446574A (en) | 1993-03-26 | 1995-08-29 | Telefonaktiebolaget Lm Ericsson | System and method for dispersion compensation in fibre optic high speed systems |
US5349312A (en) | 1993-05-28 | 1994-09-20 | Raytheon Company | Voltage variable attenuator |
US5513029A (en) | 1994-06-16 | 1996-04-30 | Northern Telecom Limited | Method and apparatus for monitoring performance of optical transmission systems |
US6115162A (en) | 1995-01-10 | 2000-09-05 | Harris Corporation | Double side band, carrier suppressed modulated coherent fiber optic link |
US5579328A (en) | 1995-08-10 | 1996-11-26 | Northern Telecom Limited | Digital control of laser diode power levels |
US5761225A (en) | 1996-05-23 | 1998-06-02 | Litton Systems, Inc. | Optical fiber amplifier eled light source with a relative intensity noise reduction system |
US6128111A (en) | 1996-12-19 | 2000-10-03 | Nortel Networks Corporation | Monitoring of nonlinear effects |
US5949560A (en) | 1997-02-05 | 1999-09-07 | Northern Telecom Limited | Optical transmission system |
US5892858A (en) | 1997-03-27 | 1999-04-06 | Northern Telecom Limited | Duobinary coding and modulation technique for optical communication systems |
US6205262B1 (en) | 1997-05-01 | 2001-03-20 | Alliance Fiber Optics Products, Inc. | Optical recirculation depolarizer and method of depolarizing light |
US6067180A (en) | 1997-06-09 | 2000-05-23 | Nortel Networks Corporation | Equalization, pulse shaping and regeneration of optical signals |
US5999258A (en) | 1997-06-26 | 1999-12-07 | Nortel Networks Corporation | Optical interference measurement method and system |
US6388786B1 (en) * | 1997-08-15 | 2002-05-14 | Nec Corporation | Method for generating duobinary signal and optical transmitter using the same method |
US6124960A (en) | 1997-09-08 | 2000-09-26 | Northern Telecom Limited | Transmission system with cross-phase modulation compensation |
EP0971493A2 (en) | 1998-07-08 | 2000-01-12 | Fujitsu Limited | Chromatic dispersion and nonlinearity compensation in optical fibre communication |
US20020018268A1 (en) | 1998-11-04 | 2002-02-14 | Corvis Corporation Attn: Intellectual Property Department | Optical distortion compensation apparatuses, methods, and systems |
US6362913B2 (en) * | 1998-11-25 | 2002-03-26 | Fujitsu Limited | Optical modulation apparatus and method of controlling optical modulator |
US6278539B1 (en) * | 1998-11-25 | 2001-08-21 | Fujitsu Limited | Optical modulation apparatus and method of controlling |
US6580532B1 (en) | 1999-01-28 | 2003-06-17 | California Institute Of Technology | Opto-electronic techniques for reducing phase noise in a carrier signal by carrier supression |
US6441932B1 (en) | 1999-02-09 | 2002-08-27 | The United States Of America As Represented By The Secretary Of The Air Force | Intensity noise suppression using differential delay cancellation in external modulation links |
US6473214B1 (en) * | 1999-04-01 | 2002-10-29 | Nortel Networks Limited | Methods of and apparatus for optical signal transmission |
US20020106148A1 (en) | 1999-05-24 | 2002-08-08 | Philips Electronics North America Corporation | Optical communication with pre-compensation for odd order distortion in modulation and transmission |
WO2001003339A1 (en) | 1999-06-30 | 2001-01-11 | Koninklijke Philips Electronics N.V. | Pre-shaping laser modulation signals to increase modulation index |
US6304369B1 (en) | 1999-07-29 | 2001-10-16 | Harmonic, Inc. | Method and apparatus for eliminating noise in analog fiber links |
US6559994B1 (en) | 1999-08-18 | 2003-05-06 | New Elite Technologies, Inc. | Optical fiber transmitter for long distance subcarrier multiplexed lightwave systems |
US6522438B1 (en) * | 1999-10-04 | 2003-02-18 | Lucent Technologies Inc. | High-speed optical duobinary modulation scheme |
US20070092263A1 (en) * | 1999-10-20 | 2007-04-26 | Agazzi Oscar E | Method, apparatus and system for high-speed transmission on fiber optic channel |
US6262834B1 (en) | 2000-02-23 | 2001-07-17 | The United States Of America As Represented By The Secretary Of The Navy | Wideband single sideband modulation of optical carriers |
US20010028760A1 (en) | 2000-03-03 | 2001-10-11 | Yaffe Henry H. | Methods and apparatus for compensating chromatic and polarization mode dispersion |
US20020024694A1 (en) | 2000-05-12 | 2002-02-28 | Newell Laurence J. | Control channel for an optical communications system utilizing frequency division multiplexing |
WO2001091342A2 (en) | 2000-05-24 | 2001-11-29 | Purdue Research Foundation | Method and system for polarization control and polarization mode dispersion compensation for wideband optical signals |
WO2002043340A2 (en) | 2000-11-22 | 2002-05-30 | Broadcom Corporation | Method and apparatus to identify and characterize nonlinearities in optical communications channels |
US20040081470A1 (en) * | 2000-12-21 | 2004-04-29 | Robert Griffin | Optical communications |
US20020167693A1 (en) * | 2000-12-21 | 2002-11-14 | Quellan, Inc. | Increasing data throughput in optical fiber transmission systems |
EP1223694A2 (en) | 2001-01-10 | 2002-07-17 | Fujitsu Limited | Dispersion compensating method, dispersion compensating apparatus and optical transmission system |
US6592274B2 (en) * | 2001-01-29 | 2003-07-15 | Stratalight Communications, Inc. | Transmission and reception of duobinary multilevel pulse-amplitude-modulated optical signals using finite-state machine-based encoder |
EP1237307A2 (en) | 2001-03-02 | 2002-09-04 | Fujitsu Limited | Receiving apparatus and method for detecting, measuring and compensating waveform degradation of received signal |
US6970655B2 (en) * | 2001-03-02 | 2005-11-29 | Nec Corporation | Method and circuit for generating single-sideband optical signal |
US20030011847A1 (en) | 2001-06-07 | 2003-01-16 | Fa Dai | Method and apparatus for adaptive distortion compensation in optical fiber communication networks |
US7068948B2 (en) * | 2001-06-13 | 2006-06-27 | Gazillion Bits, Inc. | Generation of optical signals with return-to-zero format |
US6473013B1 (en) | 2001-06-20 | 2002-10-29 | Scott R. Velazquez | Parallel processing analog and digital converter |
US7190904B2 (en) * | 2001-09-26 | 2007-03-13 | International Business Machines Corporation | Wavelength modulation for optical based switching and routing |
US6623188B1 (en) * | 2002-02-08 | 2003-09-23 | Optiuh Corporation | Dispersion tolerant optical data transmitter |
US7155134B2 (en) * | 2002-03-22 | 2006-12-26 | Agere Systems Inc. | Pulse amplitude modulated transmission scheme for optical channels with soft decision decoding |
US7317877B2 (en) * | 2002-04-16 | 2008-01-08 | Broadwing Corporation | Optical communications systems, devices, and methods |
US20030198478A1 (en) * | 2002-04-23 | 2003-10-23 | Quellan, Inc. | Method and system for generating and decoding a bandwidth efficient multi-level signal |
US7236707B2 (en) * | 2002-10-21 | 2007-06-26 | Main Street Ventures Llc | All-optical compression systems |
US20080025731A1 (en) * | 2002-12-03 | 2008-01-31 | Daniel Mahgerefteh | Versatile compact transmitter for generation of advanced modulation formats |
US7330666B1 (en) * | 2003-01-31 | 2008-02-12 | Ciena Corporation | Method and apparatus for controlling modulator phase alignment in a transmitter of an optical communications system |
US7075695B2 (en) * | 2004-03-01 | 2006-07-11 | Lucent Technologies Inc. | Method and apparatus for controlling a bias voltage of a Mach-Zehnder modulator |
US20060127102A1 (en) * | 2004-12-10 | 2006-06-15 | Nortel Networks Limited | Modulation E-field based control of a non-linear transmitter |
Non-Patent Citations (28)
Title |
---|
A. Mecozzi et al. "Cancellation of timing and Amplitude Jitter in Symmetric Links Using Highly Dispersed Pulses", IEEE Photonics Technology Letters, vol. 13, No. 5, May 2001. |
Adaptive Electronic Linearization of Fiber Optic Links, OFC 2003, vol. 2, pp. 477-480, Mar. 2003 Sadhwani et al. |
Automated Measurement of Polarization Mode Dispersion Using Jones Matrix Eigenanalysis, IEE PhotonicsTechnology Letters, vol. 4, No. 9, pp. 1066-1069, Sep. 1992, Heffner. |
Design of Broad-Band PMD Compensation Filters, IEEE Photonics Technology Letters, vol. 14, No. 8, Aug. 2002, A. Eyal et al. |
Dispersion Compensation by Active Predistorted Signal Synthesis, Journal of Lightwave Technology, vol. LT-3, No. 4, Aug. 1985, Thomas L. Koch and Rod C. Alferness. |
Dispersion Compensation with an SBS-Suppressed Fiber Phase Conjugator Using Synchronized Phase Modulation, OFC 2003, vol. 2, pp. 716-717, M. Tani. |
Electrical Signal Processing Techniques in Long-Haul Fiber-Optic Systems, 1990 IEEE-Transactions on Communications, vol. 38, No. 9, Jack H. Winters, et al. |
Feldhaus G., "Volterra Equalizer for Electrical Compensation of Dispersion and Fiber Nonlinearities", Journal of Optical Communicatinos, Fachverlag Schiele & Schon, Berlin, De, vol. 23, No. 3, Jun. 2002, pp. 82-84, XP001130377, ISSN: 0173-4911. |
H. Gysel et al. "Electrical Predistortion to Compensate for Combined Effect of Laser Chirp and Fibre Dispersion", Electronics Letters IEE Stevenage vol. 27, No. 5, Feb 1991. |
Henning Bulow, et al., "Dispersion Mitigation Using a Fiber-Bragg-Grating Sideband Filter and a Tunable Electronic Equalizer", Optical Society of America, 2000. |
High-Dynamic-Range Laser Amplitude and Phase Noise Measurement Techniques, IEEE Journal on Selected Topics in Quantum Electronics, vol. 7, No. 4, Jul./Aug. 2001, Ryan P. Sc. |
Hoon Kim, et al., "10 Gbit/s 177 km transmission over conventional singlemode fibre using a vestigial side-band modulation format" Electronics Letters, vol. 37, No. 25 Dec. 6, 2001 pp. 1533-1534. |
Kamoto, T. et al "An 8-bit 2-ns Monolithic DAC", IEEE Journal of Solid-State Circuits, Feb. 1988, vol. 23, No. 1. |
Lucas Illing, et al., "Shaping Current Waveforms for Direct Modulation of Semiconductor Lasers", Institute for Nonlinear Science, U.C. San Diego, 2003. |
M. Sieben, et al., "10Gbit/s optical single sideband system" Electronics Letters, vol. 33, No. 11, May 22, 1997, pp. 971-973. |
Measurement of High-Order Polarization Mode Dispersion, IEEE Photonics Technology Letters, vol. 12, No. 7, Jul. 2000, Yi Li et al. |
Mitigation of Dispersion-Induced Effects Using SOA in Analog Optical Transmission, IEEE Photonics Technology Letters, vol. 14, No. 8, Aug. 2002, Duk-Ho Jeon et al. |
P.M. Watts, et al., "Demonstration of Electrical Dispersion Compensation of Single Sideband Optical Transmission", London Communications Symposium 2003, University College Lon. |
P.S. Andre, et al., "Extraction of DFB Laser Rate Equation Parameters for Optical Simulation Pusposes", Conftele 1999 ISBN 972-98115-0-4. |
Performance of Smart Lightwave Receivers With Linear Equalization, Journal of Lightwave Technology, vol. 10, No. 8, Aug. 1992, John C. Cartledge, et al. |
Polarization Modulated Direct Detection Optical Transmission Systesm, Journal of Lightwave Technology, vol. 10, No. 12, Dec. 1992. |
Predistortion of Electroabsorption Modulators for Analog CATV Systems at 1.55 .m, Journal of Lightwave Technology, vol. 15, No. 9, Sep. 1997, Gordon C. Wilson et al. |
Predistortion of Electroabsorption Modulators for Analog CATV Systems at 1.55 •m, Journal of Lightwave Technology, vol. 15, No. 9, Sep. 1997, Gordon C. Wilson et al. |
Predistortion Techniques for Linearization of External Modulators, 1999 IEEE-Gordon Wilson, Lucent Technologies, NJ 07733, U.S.A. |
Ram Sadhwani, Adaptive CMOS Predistortion Linearizer for Fiber-Optic Links, Journal of Lightwave Technology, vol. 21, No. 12, Dec. 2003. |
Representation of Second-Order Polarisation Mode Dispersion, Electronics Letters, vol. 35, No. 19, Sep. 16, 1999, A. Eyal et al. |
Soliton Transmission Using Periodic Dispersion Compensation, Journal of Lightwave Technology, vol. 15, No. 10, Oct. 1997, Nicholas J. Smith et al. |
Theoretical Basis of Polarization Mode Dispersion Equalization up to the Second Order, Journal of Lightwave Technology, vol. 18, No. 4, Apr. 2000, Teruhiko Kudou et al. |
Cited By (26)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US8064777B2 (en) * | 2005-06-13 | 2011-11-22 | Arris Group, Inc. | Four quadrant linearizer |
US20090297153A1 (en) * | 2005-06-13 | 2009-12-03 | Broadband Royalty Corporation | Four quadrant linearizer |
US8737938B2 (en) * | 2006-04-04 | 2014-05-27 | Apple Inc. | Signal Transmitter linearization |
US20130034188A1 (en) * | 2006-04-04 | 2013-02-07 | Apple Inc. | Signal Transmitter Linearization |
US8103168B1 (en) * | 2006-11-09 | 2012-01-24 | Lockheed Martin Corporation | RF discrete time optical frequency translator |
US20100061738A1 (en) * | 2006-12-23 | 2010-03-11 | Telefonaktiebolaget Lm Ericsson (Publ) | Signal Processor for Compensating for Optical Fiber Chromatic Dispersion |
US8676060B2 (en) * | 2007-09-18 | 2014-03-18 | National Institute Of Information And Communications Technology | Quadrature amplitude modulation signal generating device |
US20100202785A1 (en) * | 2007-09-18 | 2010-08-12 | National Institute Of Information And Communications Technology | Quadrature amplitude modulation signal generating device |
US9130678B2 (en) | 2009-01-23 | 2015-09-08 | Ciena Corporation | High speed signal generator |
US9654220B2 (en) | 2009-01-23 | 2017-05-16 | Ciena Corporation | High speed signal generator |
US20100189443A1 (en) * | 2009-01-23 | 2010-07-29 | Nortel Networks Limited | High speed signal generator |
US8693876B2 (en) | 2009-01-23 | 2014-04-08 | Ciena Corporation | High speed signal generator |
US10148359B2 (en) | 2009-01-23 | 2018-12-04 | Ciena Corporation | High speed signal generator |
US8249467B2 (en) | 2010-03-11 | 2012-08-21 | Ciena Corporation | Self test of a dual polarization transmitter |
US20110222850A1 (en) * | 2010-03-11 | 2011-09-15 | Nortel Networks Limited | Self test of a dual polarization transmitter |
US20140029938A1 (en) * | 2012-07-26 | 2014-01-30 | Fujitsu Limited | Optical transmission system and method for monitoring polarization dependent characteristics of optical transmission line |
US9031403B2 (en) * | 2012-07-26 | 2015-05-12 | Fujitsu Limited | Optical transmission system and method for monitoring polarization dependent characteristics of optical transmission line |
US20140147117A1 (en) * | 2012-11-28 | 2014-05-29 | Hitachi, Ltd. | Optical multilevel signal pre-equalization circuit, optical multilevel signal pre-equalization transmitter, and polarization-multiplexed pre-equalization transmitter |
US9178617B2 (en) * | 2012-11-28 | 2015-11-03 | Hitachi, Ltd. | Optical multilevel signal pre-equalization circuit, optical multilevel signal pre-equalization transmitter, and polarization-multiplexed pre-equalization transmitter |
US10014947B2 (en) | 2016-02-18 | 2018-07-03 | Ciena Corporation | Mitigation of electrical-to-optical conversion impairments induced at transmitter |
US10608746B2 (en) | 2016-02-18 | 2020-03-31 | Ciena Corporation | Mitigation of electrical-to-optical conversion impairments induced at transmitter |
US10341022B2 (en) * | 2016-12-28 | 2019-07-02 | Zte Corporation | Optical pulse amplitude modulation transmission using digital pre-compensation |
US20230046863A1 (en) * | 2018-04-20 | 2023-02-16 | Neophotonics Corporation | Method and apparatus for bias control with a large dynamic range for mach-zehnder modulators |
US11870407B2 (en) * | 2018-04-20 | 2024-01-09 | Neophotonics Corporation | Method and apparatus for bias control with a large dynamic range for Mach-Zehnder modulators |
US20200059301A1 (en) * | 2018-08-20 | 2020-02-20 | Photonic Technologies (Shanghai) Co., Ltd. | Pulse generation module, and optical communication transmitter system and non-linear equalizing method thereof |
US10749606B2 (en) * | 2018-08-20 | 2020-08-18 | Photonic Technologies (Shanghai) Co., Ltd. | Pulse generation module, and optical communication transmitter system and non-linear equalizing method thereof |
Also Published As
Publication number | Publication date |
---|---|
US20060127102A1 (en) | 2006-06-15 |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
US7676161B2 (en) | Modulation E-field based control of a non-linear transmitter | |
US7787778B2 (en) | Control system for a polar optical transmitter | |
US10270535B1 (en) | Linearized optical digital-to-analog modulator | |
EP2381592B1 (en) | Optical transmitter | |
US7023601B2 (en) | Optical E-field modulation using a Mach-Zehnder interferometer | |
US6097525A (en) | Method for generating duobinary signal and optical transmitter using the same method | |
US9485032B2 (en) | Optical multilevel transmitter and optical transponder | |
GB2417333A (en) | Automatic bias control for an optical modulator | |
US7903981B2 (en) | Software-based electro-optic modulator bias control systems and methods | |
US11025339B2 (en) | Method for compensating channel distortions by pre-distortion of Mach-Zehnder modulators, based on symmetric imbalance | |
Rörich et al. | Optimal modulation index of the Mach-Zehnder modulator in a coherent optical OFDM system employing digital predistortion | |
US7680420B1 (en) | Optical E-field modulation using a directly driven laser | |
US20210096440A1 (en) | Optical modulator | |
CN114301521A (en) | Nonlinear predistortion method for microwave photon signal generation link | |
US20120194287A1 (en) | Radio frequency drive level control system and method for an electro-optic phase modulator | |
CN116210174A (en) | Self-calibration apparatus and method for in-phase and quadrature time skew and conjugation in a coherent transmitter |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
AS | Assignment |
Owner name: NORTEL NETWORKS LIMITED,CANADA Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:ROBERTS, KIM;HARLEY, JAMES;REEL/FRAME:017554/0274 Effective date: 20041202 Owner name: NORTEL NETWORK LIMITED, CANADA Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:ROBERTS, KIM;HARLEY, JAMES;REEL/FRAME:017554/0274 Effective date: 20041202 |
|
FEPP | Fee payment procedure |
Free format text: PAYOR NUMBER ASSIGNED (ORIGINAL EVENT CODE: ASPN); ENTITY STATUS OF PATENT OWNER: LARGE ENTITY |
|
STCF | Information on status: patent grant |
Free format text: PATENTED CASE |
|
AS | Assignment |
Owner name: CIENA LUXEMBOURG S.A.R.L.,LUXEMBOURG Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:NORTEL NETWORKS LIMITED;REEL/FRAME:024213/0653 Effective date: 20100319 Owner name: CIENA LUXEMBOURG S.A.R.L., LUXEMBOURG Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:NORTEL NETWORKS LIMITED;REEL/FRAME:024213/0653 Effective date: 20100319 |
|
AS | Assignment |
Owner name: CIENA CORPORATION,MARYLAND Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:CIENA LUXEMBOURG S.A.R.L.;REEL/FRAME:024252/0060 Effective date: 20100319 Owner name: CIENA CORPORATION, MARYLAND Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:CIENA LUXEMBOURG S.A.R.L.;REEL/FRAME:024252/0060 Effective date: 20100319 |
|
FPAY | Fee payment |
Year of fee payment: 4 |
|
AS | Assignment |
Owner name: DEUTSCHE BANK AG NEW YORK BRANCH, NEW YORK Free format text: SECURITY INTEREST;ASSIGNOR:CIENA CORPORATION;REEL/FRAME:033329/0417 Effective date: 20140715 |
|
AS | Assignment |
Owner name: BANK OF AMERICA, N.A., AS ADMINISTRATIVE AGENT, NO Free format text: PATENT SECURITY AGREEMENT;ASSIGNOR:CIENA CORPORATION;REEL/FRAME:033347/0260 Effective date: 20140715 |
|
FEPP | Fee payment procedure |
Free format text: PAYOR NUMBER ASSIGNED (ORIGINAL EVENT CODE: ASPN); ENTITY STATUS OF PATENT OWNER: LARGE ENTITY Free format text: PAYER NUMBER DE-ASSIGNED (ORIGINAL EVENT CODE: RMPN); ENTITY STATUS OF PATENT OWNER: LARGE ENTITY |
|
MAFP | Maintenance fee payment |
Free format text: PAYMENT OF MAINTENANCE FEE, 8TH YEAR, LARGE ENTITY (ORIGINAL EVENT CODE: M1552) Year of fee payment: 8 |
|
AS | Assignment |
Owner name: CIENA CORPORATION, MARYLAND Free format text: RELEASE BY SECURED PARTY;ASSIGNOR:DEUTSCHE BANK AG NEW YORK BRANCH;REEL/FRAME:050938/0389 Effective date: 20191028 |
|
AS | Assignment |
Owner name: BANK OF AMERICA, N.A., AS COLLATERAL AGENT, ILLINO Free format text: PATENT SECURITY AGREEMENT;ASSIGNOR:CIENA CORPORATION;REEL/FRAME:050969/0001 Effective date: 20191028 |
|
MAFP | Maintenance fee payment |
Free format text: PAYMENT OF MAINTENANCE FEE, 12TH YEAR, LARGE ENTITY (ORIGINAL EVENT CODE: M1553); ENTITY STATUS OF PATENT OWNER: LARGE ENTITY Year of fee payment: 12 |
|
AS | Assignment |
Owner name: CIENA CORPORATION, MARYLAND Free format text: RELEASE BY SECURED PARTY;ASSIGNOR:BANK OF AMERICA, N.A.;REEL/FRAME:065630/0232 Effective date: 20231024 |