US7031251B2 - Clipping distortion canceller for OFDM signals - Google Patents
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- US7031251B2 US7031251B2 US10/771,249 US77124904A US7031251B2 US 7031251 B2 US7031251 B2 US 7031251B2 US 77124904 A US77124904 A US 77124904A US 7031251 B2 US7031251 B2 US 7031251B2
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- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2614—Peak power aspects
- H04L27/2623—Reduction thereof by clipping
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- Orthogonal frequency division multiplexing is one of the technologies considered for 4G broadband wireless communications due to its robustness against multipath fading and relatively simple implementation compared to single carrier systems.
- OFDM transmitters utilize linear power amplifiers.
- One of the main drawbacks of using OFDM is the high cost of linear power amplifiers with high dynamic range.
- Such amplifiers are used because the OFDM signals have a high peak-to-average power ratio (PAPR), particularly since the OFDM signal will usually consists of a large number of subcarriers.
- PAPR peak-to-average power ratio
- the deliberate clipping of the OFDM signal before amplification is a simple and efficient way of controlling the PAPR.
- the clipping process is characterized by the clipping ratio (CR), defined as the ratio between the clipping threshold and the RMS level of the OFDM signal.
- CR clipping ratio
- Clipping is a non-linear process, which may lead to significant distortion and performance loss. In particular, clipping at the Nyquist sampling rate will cause all the clipping noise to fall in-band and suffers considerable peak re-growth after digital to analog (D/A) conversion.
- a method includes: (a) transforming a received orthogonal frequency division multiplexed (OFDM) signal from a transmission channel into the frequency domain, the OFDM signal having been subject to a clipping function prior to transmission in order to reduce the peak-to-average power ratio (PAPR); (b) recovering data symbols from the transformed OFDM signal, which include clipping noise; (c) estimating the clipping noise in the frequency domain based on the data symbols; and (d) subtracting the estimated clipping noise from the transformed OFDM signal.
- OFDM orthogonal frequency division multiplexed
- an apparatus includes: a receiver operable to receive an orthogonal frequency division multiplexed (OFDM) signal from a transmission channel, the OFDM signal having been subject to a clipping function prior to transmission in order to reduce the peak-to-average power ratio (PAPR); a frequency transform unit operable to transform the OFDM signal to the frequency domain; a decoding unit operable to recover data symbols from the frequency domain OFDM signal, which include clipping noise; a noise estimator operable to estimate the clipping noise in the frequency domain based on the data symbols; and a difference circuit operable to subtract the estimated clipping noise from the transformed OFDM signal.
- OFDM orthogonal frequency division multiplexed
- an apparatus includes a processor operating under the control of one or more software programs that cause the processor to carry out actions, including: (a) transforming a received orthogonal frequency division multiplexed (OFDM) signal from a transmission channel into the frequency domain, the OFDM signal having been subject to a clipping function prior to transmission in order to reduce the peak-to-average power ratio (PAPR); (b) recovering data symbols from the transformed OFDM signal, which include clipping noise; (c) estimating the clipping noise in the frequency domain based on the data symbols; and (d) subtracting the estimated clipping noise from the transformed OFDM signal.
- OFDM orthogonal frequency division multiplexed
- a storage medium contains one or more software programs that are operable to cause a processor executing the one or more software programs to carry out actions, including: (a) transforming a received orthogonal frequency division multiplexed (OFDM) signal from a transmission channel into the frequency domain, the OFDM signal having been subject to a clipping function prior to transmission in order to reduce the peak-to-average power ratio (PAPR); (b) recovering data symbols from the transformed OFDM signal, which include clipping noise; (c) estimating the clipping noise in the frequency domain based on the data symbols; and (d) subtracting the estimated clipping noise from the transformed OFDM signal.
- OFDM orthogonal frequency division multiplexed
- the methods and apparatus for controlling cache memories described thus far and/or described later in this document may be achieved utilizing suitable hardware, such as that shown in the drawings hereinbelow.
- suitable hardware such as that shown in the drawings hereinbelow.
- Such hardware may be implemented utilizing any of the known technologies, such as standard digital circuitry, analog circuitry, any of the known processors that are operable to execute software and/or firmware programs, one or more programmable digital devices or systems, such as programmable read only memories (PROMs), programmable array logic devices (PALs), any combination of the above, etc.
- PROMs programmable read only memories
- PALs programmable array logic devices
- the methods of the present invention may be embodied in a software program that may be stored on any of the known or hereinafter developed media.
- FIG. 1 is a block diagram illustrating a transmitter and a receiver incorporating a distortion cancellation feature in accordance with one or more aspects of the present invention
- FIGS. 2 a – 2 b are block diagrams illustrating a preferred approach to achieving a deliberately clipped OFDM signal, which may be employed in the transmitter and/or the receiver of FIG. 1 in accordance with one or more aspects of the present invention
- FIG. 3 is a graphical illustration of test results indicating the complementary cumulative density function (CCDF) versus the PAPR of a digitally clipped OFDM signal (with a clipping ratio of 1) in accordance with one or more aspects of the present invention
- FIG. 4 is a graphical illustration of test results indicating the packet error rate (PER) versus (E b /N 0 ) of the receiver of FIG. 1 over an AWGN channel including comparisons with the “signal reconstruction” approach;
- FIG. 5 is a graphical illustration of test results indicating the packet error rate (PER) versus (E b /N 0 ) of the receiver of FIG. 1 (employing deliberate clipping) over a Rayleigh fading channel including comparisons with the “signal reconstruction” approach;
- PER packet error rate
- E b /N 0 packet error rate
- FIG. 6 is a block diagram illustrating a preferred approach to achieving a repeatedly clipped OFDM signal, which may be employed in the transmitter and/or the receiver of FIG. 1 in accordance with one or more aspects of the present invention
- FIG. 7 is a graphical illustration of test results indicating the complementary cumulative density function (CCDF) versus the PAPR of a repeatedly clipped OFDM signal (with a clipping ratios of 1.5, 1.3, and 1.35) in accordance with one or more aspects of the present invention.
- CCDF complementary cumulative density function
- FIG. 8 is a graphical illustration of test results indicating the packet error rate (PER) versus (E b /N 0 ) of the receiver of FIG. 1 (employing repeated clipping) over a Rayleigh fading channel including comparisons with the “signal reconstruction” approach.
- PER packet error rate
- E b /N 0 packet error rate
- the present invention is directed to methods and apparatus for iteratively estimate and cancel the distortion caused by clipping noise at the receiver.
- experimentation and simulation has shown that the methods and apparatus of the present invention may be applied to clipped and filtered OFDM signals such that (for an IEEE 802.11a system) the PAPR can be reduced to as low as 4 dB while the system performance can be restored to within 1 dB of the non-clipped case with only moderate complexity increase and with substantially no bandwidth expansion.
- FIG. 1 is a block diagram of a system 100 including a transmitter 102 and a receiver 104 incorporating one or more distortion cancellation features in accordance with one or more aspects of the present invention.
- the system 100 is disclosed by way of block diagram to illustrate a logical partitioning of functional blocks, which may be considered as hardware elements, software routines, digital signal processing (DSP) routines, and/or straight method elements.
- DSP digital signal processing
- the functional partitioning is provided by way of example only, it being understood that many variations of partitioning are contemplated without departing from the spirit and scope of the present invention.
- the transmitter 102 preferably includes an encoder 106 , an interleaving/mapping element 108 , a clipping element 110 , an inverse frequency domain transform element 112 (such as an inverse fast Fourier transform, IFFT), and an output stage 114 (which may include among other things an antenna).
- an encoder 106 e.g., an encoder 106 , an interleaving/mapping element 108 , a clipping element 110 , an inverse frequency domain transform element 112 (such as an inverse fast Fourier transform, IFFT), and an output stage 114 (which may include among other things an antenna).
- IFFT inverse fast Fourier transform
- output stage 114 which may include among other things an antenna.
- the apparatus and/or process elements needed to achieve signal transmission over the air (or other transmission channel) preferably adhere to the requirements of the 802.11a standard (or protocol), e.g., with respect to operational frequency parameters, sub-carrier frequency parameters, power requirements, coding parameters, symbol definitions
- the encoder 106 is operable to improve system performance (e.g., lower the error rate) by adding information data (redundancy) to the input data bits. For example, if the input data bits into the encoder 106 are grouped segments of two bits long (e.g., 00, 01, 10, 11), the encoder may output encoded data bits (or coded data bits) of four bits in length (e.g., 1011, etc.). This has an advantageous effect at the receiver (which as will be discussed below includes a corresponding decoder 136 ). The decoder 136 takes the received data signals for the four bits (e.g., 1011), which will be influenced by noise, and make a decision as to what two information bits correspond to the received four bits (plus noise).
- the decoder 136 takes the received data signals for the four bits (e.g., 1011), which will be influenced by noise, and make a decision as to what two information bits correspond to the received four bits (plus noise).
- the decoder will output the proper two bits (e.g., 00). If the SNR is low, the decoder may output the wrong symbol in error. Any of the known (or hereinafter developed) encoders may be employed in connection with the present invention.
- the interleaving/mapping element 108 may be two separate functional elements, one for interleaving and one for mapping, although for simplicity they are illustrated in integral fashion.
- the interleaving function the coded data bits are interleaved (permutated) before being mapped to symbols.
- the major purpose for using interleaving is to combat the affects of multi-path fading channels.
- Any of the known (or hereinafter developed) interleaving techniques may be employed without departing from the spirit and scope of the present invention.
- an IEEE standard 802.11a block interleaver may be suitable for used in connection with implementing the present invention.
- the 802.11a interleaver is defined by a two-step permutation.
- the first permutation ensures that adjacent coded bits are mapped onto nonadjacent sub-carriers.
- the second permutation ensures that adjacent coded bits are mapped alternately onto less and more significant bits of the constellation and, thereby, long runs of low reliability (LSB) bits are avoided.
- LSB low reliability
- the interleaved data bits are then mapped into multi-level phase signals.
- the mapping may be in accordance with the M-QAM technique (quadrature amplitude modulation), such as 16-QAM, where bits 0000 are mapped to a complex signal point ⁇ 3 ⁇ 3i; bits 1001 are mapped to ⁇ 1+3i, etc.
- the mapping may be in accordance with the M-PSK technique (phase shift keying).
- M-QAM quadrature amplitude modulation
- M-PSK phase shift keying
- the clipping element 110 is operable to limit the PAPR of the OFDM signal before amplification and transmission.
- the clipping function is performed digitally.
- the time domain signal is preferably over-sampled by a factor greater than two and then the amplitude of the time domain signal samples are limited by a threshold A (i.e., they are clipped). Further details regarding the clipping element 110 will be discussed below.
- each modulated signal is assigned to a sub-carrier via the IFFT element 112 and transmitted over the transmission channel (such as air).
- a low-pass equivalent of an OFDM signal can be represented by the following equation:
- N is the number of sub-carriers
- f 0 is the sub-carrier spacing
- T is the symbol duration
- C k is the complex modulated symbol.
- the modulated symbols are obtained by mapping an encoded bit stream.
- the PAPR of the transmitted OFDM signal may be defined by the following equation:
- PAPR max 0 ⁇ t ⁇ T ⁇ ⁇ s ⁇ ( t ) ⁇ 2 P av , ( 2 )
- P av the average power of the transmitted symbol and the maximum is sought over the symbol duration. Note that the PAPR of equation (2) is defined for the average power P av measured after clipping and filtering.
- FIGS. 2 a – 2 b illustrate a deliberate clipping approach as illustrated in FIGS. 2 a – 2 b.
- FIG. 2 a illustrates a soft envelope limiter for an OFDM signal
- FIG. 2 b illustrates further details of the band pass filter (BPF) of FIG. 2 a .
- BPF band pass filter
- FIGS. 2 a – 2 b will not be described in detail herein as it is understood that one skilled in the art would know how to implement the clipping element illustrated.
- Further details concerning the clipping technique illustrated in FIGS. 2 a – 2 b may be found in H. Ochiai and H. Imai, “Performance Analysis of Deliberately Clipped OFDM Signals,” IEEE Transactions on Communications, Vol. 50, pp. 89–101 (January 2002), the entire disclosure of which is hereby incorporated by reference.
- the time domain signal may be over-sampled by a factor greater than two.
- IDFT inverse discrete Fourier transform
- IFFT inverse fast Fourier transform
- the clipping circuit 118 operates to limit the amplitude of the time domain signal samples via a threshold A.
- ⁇ overscore (s) ⁇ n be a clipped time sample with the phase left unchanged.
- ⁇ s _ n ⁇ ⁇ ⁇ s n ⁇ if ⁇ s n ⁇ ⁇ A A if ⁇ s n ⁇ > A . ( 4 )
- the FFT circuit 120 and the out of band removal circuit 122 operate to remove the out-of-band components resulting from clipping.
- DFT discrete Fourier transform
- C fast Fourier transform
- the resulting sequence is transmitted over the antenna 114 .
- the clipping process is repeated at the receiver 104 using the detected symbols. Thereafter, the frequency domain clipping noise is estimated and canceled.
- the receiver 104 operates in an iterative fashion to achieve this result.
- the receiver 104 preferably includes an antenna 130 , an FFT circuit 132 , a de-mapping and de-interleaving circuit 134 , a decoder 136 , an interleaving and mapping circuit 138 , a clipping element 140 , an attenuation circuit 142 , an adder 144 , an H circuit 154 , and another adder 156 .
- the FFT circuit 132 operates to convert the signals received from the antenna 130 into the frequency domain (for example using the discrete Fourier transform).
- the adder 156 operates to subtract an estimate of the clipping noise from the output of the FFT circuit 132 (which will be discussed later herein).
- the de-mapping and de-interleaving circuit 134 performs two basic functions. First, the received signal over each sub-carrier is demodulated (de-mapped) into signals of several bits (i.e., the reverse of the modulation/mapping process performed in the transmitter 102 ). This sequence of demodulated signals is then de-interleaved (i.e., the reverse of the interleaving process performed in the transmitter 102 ). This results in the original order prior to the interleaving process carried out in the transmitter 102 .
- the decoder circuit 136 is complementary to the encoder 106 of the transmitter 102 , which both adhere to the IEEE 802.11a standard.
- the decoder 136 takes the received data signals for the four bits (e.g., 1011), which will be influenced by noise, and make a decision as to what two information bits correspond to the received four bits (plus noise). If the signal to noise ratio (SNR) is high enough, the decoder will output the proper two bits (e.g., 00). If the SNR is low, the decoder may output the wrong symbol in error.
- SNR signal to noise ratio
- the clipping element 140 is substantially similar to the clipping element 110 of the transmitter 102 .
- the clipping element 140 preferably includes an IFFT circuit 146 , a clipping circuit 148 , an FFT circuit 150 and an out of band removal circuit 152 , which operate as discussed above with respect to the transmitter 102 .
- Simulations have been conducted using the methods and apparatus described above with respect to FIG. 1 . These simulations have been conducted for clipped and filtered OFDM signals over both AWGN and fading channels.
- the simulation model was designed to match IEEE Std. 802.11a.
- Decoding was carried out using a soft Viterbi algorithm. The system performance is measured based on the packet error rate (PER), where each packet consists of 16 OFDM symbols.
- the E b /N 0 is measured after signal clipping and filtering.
- CCDF complementary cumulative density function
- the simulated packet error rate (PER) performance of the receiver 104 over the AWGN channel is illustrated.
- the performance is compared with that of a receiver without clipping noise cancellation and to a receiver with signal reconstruction.
- the performance of a system without clipping is also provided.
- the performance gain over the non-cancellation case increases as SNR increases. This is because at high SNR the AWGN noise becomes relatively small and the clipping noise begins to dominate.
- the performance of the signal reconstruction approach is worse by about 1.5 dB as compared with the methods and apparatus of the present invention.
- FIG. 5 illustrates the simulated PER performance of the receiver 104 over a Rayleigh fading channel with an exponentially decaying power delay profile, with normalized delay spread equal to 2.
- the simulation results show that the clipping noise cancellation approach of the present invention can significantly restore the performance. Further, more than about two iterations yields diminishing benefit. The reason appears to be that there exist some OFDM symbols that are too badly damaged by clipping for the iterative process to converge. This performance gap may be further narrowed by combining the approach discussed above with bit or symbol interleaving methods. Since it is known at the transmitter 102 how the OFDM symbol is affected by clipping, the badly damaged symbols (according to some criterion) can be interleaved and re-clipped, which may result in less clipping noise.
- the clipping elements 110 and 140 may be implemented using the repeated clipping approach.
- the details of this clipping technique will not be discussed in great detail herein as they are well known to those skilled in the art. Further details on the repeated clipping technique may be found, however, in J. Armstrong, “Peak-to-Average Power Reduction for OFDM by Repeated Clipping and Frequency Domain Filtering,” Electronics Letters, Vol. 38, pp. 246–247 (February 2002), the entire disclosure of which is hereby incorporated by reference.
- the receiver 104 employing the repeated clipping technique has a similar structure as in FIG. 1 , except that clipping and filtering are repeated to match the transmitter 102 .
- the signals can be clipped to the same PAPR with less distortion. Without clipping noise cancellation, even with three or four times clipping and filtering the PAPR of the OFDM signal can be reduced only moderately (to 7 dB).
- the PAPR of the 64-subcarrier OFDM signal can be reduced to 4 dB and the clipping noise cancellation approach of the present invention restores the system performance to within 1 dB of the non-clipping case.
- FIG. 7 shows the PAPR distribution (CCDF) of the 64-subcarrier OFDM signals with this set of clipping ratios. It is noted that the PAPR is reduced to 4 dB, an 8 dB reduction compared to the non-clipped case.
- the clipping ratios used in the simulation has been chosen empirically.
- FIG. 8 shows the PER performance of the system with repeated clipping.
- various aspects of the present invention permit iterative distortion cancellation for clipped and filtered OFDM signals.
- the performance of a clipped and filtered OFDM system can be significantly improved with only moderate complexity increase at the receiver 104 .
- the PAPR of the transmitted signal can be significantly reduced with acceptable performance loss.
- the receiver 104 is particularly suitable for IEEE 802.11a wireless LAN systems since it allows signals to be significantly clipped with only slight performance degradation.
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Abstract
Description
where Pav is the average power of the transmitted symbol and the maximum is sought over the symbol duration. Note that the PAPR of equation (2) is defined for the average power Pav measured after clipping and filtering.
s n @s(nΔt)n=0, . . . , JN−1. (3)
{overscore (s)} n =αs n +d n n=0, . . . , JN−1, (5)
where the attenuation factor α is a function of the clipping ratio γ, defined as γ=A/√{square root over (Pin)}, with Pin the average signal power before clipping:
{overscore (C)} k =αC k +D k k=0, . . . , JN−1, (7)
where {Ck}k=0 JN−1 and {Dk}k=0 JN−1 are respectively, the DFT of {sn}n=0 JN−1and {dn}n=0 JN−1in equation (5). In particular, {Dk}k=0 JN−1 is the sequence representing the clipping noise in the frequency domain. The out of band removal circuit operates to remove the out-of-band components by processing only the in-band-components {{overscore (C)}k}k=0 N−1 through the IFFT circuit 112 (which may be an N-point IDFT).
Y k =H k(αC k +D k)+Z k k=0, . . . , N−1, (8)
where Hk is the complex channel gain of the k-th sub-carrier assumed to be perfectly known and Zk is AWGN.
G k =αĈ k +{circumflex over (D)} k k=0, . . . , N−1. (9)
{circumflex over (D)} k =G k −αĈ k k=0, . . . , N−1. (10)
Ŷ k =Y k −H k {circumflex over (D)} k k=0, . . . , N−1=αH k C k +H k(D k −{circumflex over (D)} k)+Z k, (11)
where (Dk−{circumflex over (D)}k) is the residual clipping noise and H is the transfer function of
Ŷ k (R) =Y k +H k ΔC k k=0, . . . , N−1. (12)
Ŷ k (R) =Y k −H k {circumflex over (D)} k+(1−α)H k Ĉ k =Ŷ k+(1−α)H k Ĉ k k=0, . . . , N−1. (13)
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WO2004073182A2 (en) | 2004-08-26 |
US7983144B2 (en) | 2011-07-19 |
US20040165524A1 (en) | 2004-08-26 |
WO2004073182A3 (en) | 2005-04-07 |
US20060250936A1 (en) | 2006-11-09 |
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