US6424940B1 - Method and system for determining gain scaling compensation for quantization - Google Patents

Method and system for determining gain scaling compensation for quantization Download PDF

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US6424940B1
US6424940B1 US09/512,386 US51238600A US6424940B1 US 6424940 B1 US6424940 B1 US 6424940B1 US 51238600 A US51238600 A US 51238600A US 6424940 B1 US6424940 B1 US 6424940B1
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gain
prediction error
digital sample
vector
calculating
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Meir Agassy
Amir Ilan
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ECI Telecom Ltd
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    • G—PHYSICS
    • G10—MUSICAL INSTRUMENTS; ACOUSTICS
    • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
    • G10L19/00—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis
    • G10L19/04—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis using predictive techniques
    • G10L19/08—Determination or coding of the excitation function; Determination or coding of the long-term prediction parameters
    • G10L19/083—Determination or coding of the excitation function; Determination or coding of the long-term prediction parameters the excitation function being an excitation gain

Definitions

  • the present invention relates to telecommunication systems in general, and in particular to the transmission of compressed signals in telecommunications systems.
  • VBR variable bit rate
  • VBR voice-band data
  • VBD voice-band data
  • the codec described in the 40 kbps algorithm basically uses a transmission rate of 40 kbit/s.
  • the algorithmic delay is 5-samples long, totaling 0.625 msec, and the codec can perform a mode-switch every “adaptation-cycle” (2.5 msec).
  • the suggested 40 kbps algorithm was intended mainly to solve problems in the transmission of compressed VBD for applications such as DCME, and was suggested to replace the 40 kbps ADPCM mode (ITU-T Rec.—G.726) in DCME systems where LD-CELP algorithm is incorporated.
  • this algorithm is the soft transition to and from the LD-CELP algorithm, and the maintaining of toll-quality or near toll-quality of speech.
  • the adaptation cycle used for the speech mode in the 40 kbps algorithm is essentially provided by G. 728 Recommendation. Therefore, when reverting to speech mode type of operation, the LD-CELP mode specified in Recommendation G.728 will be applied rather than the 40 kbps algorithm.
  • TCQ Trellis Coded Quantization
  • U.S. Pat. No. 4,677,423 recognizes a somewhat similar problem associated with another type of algorithm, the ADPCM algorithm, and discloses a solution to that problem.
  • the mechanism described in U.S. Pat. No. 4,677,423 is one for overcoming the problem associated with transitions in partial band energy signals, by locking and unlocking the adaptation speed.
  • the adaptation speed is locked in cases of very slow speed of adaptation, while the unlocked mode is used when high speed of adaptation is required.
  • LP Linear Prediction
  • a method for determining the compensated scaling of a quantizer in a process of encoding/decoding a VBD type transmission by using a vectorial linear non-adaptive predicting type algorithm is provided.
  • VBD is used to denote digital signals modulated for transmission in the voice band frequency (up to 4 KHz), e.g. modem signals, DTMF signals, or any other such narrow band type of signals.
  • the method provided by the present invention preferably comprises the steps of:
  • step (viii) in the case that the determination in step (vii) is that a gain compensation is required, determining the compensation required for the impulse in the prediction error of said digital sample vector;
  • step (v) combining the scaling of the quantizer as obtained by step (v) with the gain compensation determined in step (viii) to obtain the compensated scaling of the quantizer.
  • linear non-adaptive predicting algorithm is an algorithm of the type all poles modeling.
  • the determination whether a signal can be qualified as a steady signal is done by comparing the differences existing between the gains associated with a pre-defined number of preceding digital sample vectors and the average values associated therewith, with the second pre-defined threshold. If these differences do not exceed that second pre-defined threshold, the signal may be qualified as a steady signal.
  • the method described further comprises a step of calculating the value of a pre-defined function, which function is based on the calculated LP coefficients associated with the digital sample vector.
  • the value of the pre-defined function thus obtained may be used in determining the required gain compensation. According to this embodiment, this can be done for example, by setting a constrain that unless the calculated value is higher than that of a pre-defined value, no gain compensation will be carried out. Another possible example is by applying a factor on the gain compensation that depends on the difference existing between the calculated value and that of the pre-defined value.
  • gain compensation decision mechanisms can also be used and their results be incorporated in the final decision upon the actual compensation to be carried out.
  • a peak threshold value is pre-defined, and the calculated value of the difference as calculated in step (v) of the above method, is compared with that peak threshold.
  • This embodiment enables among others, extending a first pre-defined period of time during which the gain is compensated while its value does not exceed that peak threshold.
  • the gain compensation period can be extended for example until either the peak is reduced below the level of that peak threshold, or to a longer, pre-defined period of time.
  • the linear prediction error vector is derived by performing a Trellis code quantization on the prediction error vector, and selecting a preferred quantized linear prediction error vector from among a number of quantized linear prediction error vectors calculated. More preferably, such selection is made by choosing the linear prediction error vector that has the minimal prediction error.
  • the determination of the gain compensation required as set at step (viii) is subjected to a limiting threshold to prevent from reaching over-compensation of the gain.
  • digital telecommunication station operative in a digital telecommunication system, and comprises:
  • processing means adapted to calculate:
  • first determination means for determining whether a gain compensation for the impulse in the prediction error of said digital sample vector is required, based on:
  • second determination means adapted to determine the gain compensation required to compensate for the impulse in the prediction error of said digital sample vector if the determination made by the first determination means is affirmative;
  • output interface adapted to transmit a voiceband data signal.
  • the device described above may comprise further features that are known in the art per se, and should thus be understood as being encompassed by the present invention.
  • telecommunication network should be understood to encompass the various types of networks known in the art, such as TDM, synchronous and asynchronous transfer networks, IP networks, IP frame relaying networks and any other applicable communication networks.
  • telecommunication station is used herein to describe a combination of at least one pair of encoding/decoding devices, one of which is used for converting, when required, signals received to a new coded form, while the other is used as its corresponding decoder, converting signals received in this new coded form to essentially their pre-encoder form.
  • Such two devices may either be included within one apparatus or be separated from each other.
  • a telecommunication apparatus operative in a digital telecommunication system and adapted to produce temporal change in quantization gain in a process of encoding/decoding transmission of the VBD type, comprising the following:
  • the average calculator is operative to calculate the average of the gain estimation by using the most recent vector gain value, and the difference, G diff , between said most recent vector gain value and said average of the gain compensation. More preferably, the difference G diff is received and compared with a pre-determined first threshold, by the impulse detector which is operative to detect sudden changes in the gain after a predetermined period of time.
  • the signal classifier is adapted to detect pre-defined VBD transmissions, and more preferably, the decision means is adapted to receive the output of the impulse detector and the signal classifier, and to activate the gain compensator accordingly.
  • the gain compensator is operative to increase the gain for a pre-defined period of time.
  • a digital communication system for interconnecting a plurality of telecommunication trunks via a transmission path, comprising:
  • first transmission means at at least a first end of the transmission network for transmitting digital signals
  • FIG. 1 illustrates a schematic representation of a coder incorporating the method of handling VBD signals according to the present invention.
  • FIG. 2 describes schematically a typical state machine for generating a Trellis diagram.
  • FIG. 3 presents an example of a Trellis diagram generated by a state machine demonstrated in FIG. 2 .
  • FIG. 4 illustrates schematically a method of carrying out a temporal change in the quantization gain in accordance with the present invention.
  • FIG. 1 The schematic partial structure of a coder 1 of the present invention is presented in FIG. 1 .
  • Signal Sn is introduced into a summing device 3 together with the predicted value thereof S′n.
  • the difference is passed through a pre-amplifier 5 to a TCQ Search. & Viterbi decision block, 10 .
  • the information received by this block following the processing of the difference together with the relevant input derived from block 12 , a set of expanded super codebook, is passed through gain scaling device 15 and to predictor 16 . All the operations required by the TCQ (Trellis Coded Quantization) algorithm are carried out in the set up demonstrated in this Figure, by block 10 . Such operations may include for example, management of the Trellis survivors and the specified reproduction values, calculation and comparison of matrices, and determination of the Viterbi decisions.
  • Each node of a given set of nodes comprises a number of legitimate branches.
  • a limited number of these branches is selected, where the selected branches are those that will lead to a smaller error.
  • the path connecting the branches that would lead to the minimal overall error is selected.
  • block 10 also releases 5 channel indices designated in FIG. 1 as j, referencing the best survivor Yj for the 5 source samples by the Viterbi algorithm.
  • FIGS. 2 and 3 A typical state machine that generates the Trellis diagram and the Trellis diagram itself, are illustrated schematically in FIGS. 2 and 3.
  • Section 7.1 of the “40 kbps algorithm” provides the allowed path to the previous nodes through the Trellis lattice, for every node.
  • the allowed previous nodes for the first node are node 0 under branch 0 (b[ 0 ]) and node 2 under branch 1 (b[ 1 ]).
  • Section 7.2 of the “40 kbps algorithm” provides the allowed path to the next nodes through the Trellis lattice, for every node.
  • the allowed next nodes for the first node are node 0 under branch 0 (b[ 0 ]) and node 2 under branch 1 (b[ 1 ]).
  • Section 7.3 of the “40 kbps algorithm” provides the quantization subset ⁇ D 0 , D 1 , D 2 , D 3 ⁇ associated with every Trellis path. For example, the transition from s[ 0 ] to s[ 0 ] is associated with subset D 0 . Transition from s[ 0 ] to s[ 1 ] is associated with subset D 2 , and transitions to s[ 2 ] and s[ 3 ] are not allowed and are, therefore, marked with X.
  • Section 7.4 of the “40 kbps algorithm” provides the index bit that labels each transition, and identifies the two branches that emanate from each node. For example, transition from s[ 0 ] to s[ 0 ] is associated with 0. Transition from s[ 0 ] to s[ 1 ] is associated with 1 (note that bit 5 is used, and 0 ⁇ 10 is 10h in C), and transitions to s[ 2 ] and s[ 3 ] are not allowed, and are therefore marked with X.
  • block 12 is the Super Codebook which is a set-expanded scalar Lloyd-Max quantizer.
  • the 64 output levels are partitioned into four subsets, starting with the most negative point and proceeding towards the most positive point, labeling consecutive points as ⁇ D 0 , D 1 , D 2 , D 3 , . . . D 0 , D 1 , D 2 , D 3 ⁇ .
  • the quantization levels are given in section 7.6 of the “40 kbps algorithm” and the interval limits are given in section 7.5 of the “40 kbps algorithm”.
  • the levels that belong to subset D 0 are shown in the column marked s[ 0 ].
  • D 1 levels are shown below s[ 1 ], . . . , and D 3 are shown below s[ 3 ].
  • VBD signals are handled by the backward gain adapter 14 , there are several differences in accordance with the present invention in its operation as compared with the way speech signals are handled in accordance with G 728 ITU-T standard. The major differences are:
  • the RMS value of the codebook output values is calculated over a sequence of output levels (quantized residuals) that are specified by the survivor path.
  • the RMS is calculated over a sequence of 8 samples.
  • Eq. (1) provides the logarithmic approximation.
  • the coefficients d 0 , d 1 , d 2 , d 3 , d 4 are provided in section 8 of the “40 kbps algorithm” and the detailed description of the logarithmic calculator is provided in section 4.12 therein.
  • the log RMS value replaces the output of the shape and gain codebook, log-gain tables blocks #G.93 and #G.94 (the last two terms in equation G-14).
  • a smoothing filter may be introduced in the log gain loop, to reduce the steady-state oscillation for signals with stationary variance, such as voice-band data waveform.
  • a Dynamic Locking Quantizer (“DLQ”) algorithm generates a variable speed adaptation.
  • DLQ Dynamic Locking Quantizer
  • the input to the processor using the DLQ algorithm is the offset removed log-gain d(n). This input is averaged by the weighting filter (section 4.13 of the “40 kbps algorithm” block #J.14) to produce the locked gain G L .
  • a 1 is calculated by comparing the long-term and the short-term energy of the quantized residuals ET(n) (section 4.10, block #J.12 of the “40 kbps algorithm”). The comparison characterizes the constancy of the variance of quantized residuals.
  • Prediction error impulses might cause the saturation of the quantizer.
  • a temporal change in quantization gain is carried out in accordance with the method provided by the present invention.
  • a preferred way of performing the average calculation for carrying out the method of the present invention is by assigning more weight to the most recent gain values in the calculation.
  • FIG. 4 illustrates schematically a method of carrying out the temporal change in the quantization gain. In accordance with this method, the following steps are taken:
  • a smoothing filter 40 calculates the average of the gain estimation, Gave, using the most recent vector gain value, GSTATE [ 0 ].
  • the calculated average is a weighted average, giving higher weight to recent values than to past values. Equation 3 presents an optional way of calculating such an average.
  • the difference between GSTATE [ 0 ] and G ave , designated as G diff is then calculated and passed to an Impulse Detection block 42 .
  • Impulse Detection block 42 The function of this block is essentially the detection of sudden changes in the gain following a predetermined period of time wherein impulses were not detected. In order to accomplish that, G diff is compared with a second fixed pre-defined threshold. If the value of G diff were less than that of the second pre-defined threshold for a period exceeding a predefined period of time, then the signal would be treated as a “steady” signal. A linear prediction error impulse is detected when the value of G diff exceeds that of a first pre-defined threshold while the preceding signal was determined to be a “steady” signal. According to a preferred embodiment of the present invention, the first pre-defined threshold is equal to the second pre-defined threshold.
  • Signal Classifier During certain VBD transmission, error impulses are more likely to happen. Thus, upon their detection, the parameters of the gain compensation can be maximized. In signal classifier block 44 these transmissions are detected e.g. by using the LP coefficients, and the classification is forwarded to the decision block 46 .
  • Decision block 46 The decision block 46 receives both the output of the signal classifier block 44 and that of the impulse detection block 42 . Based on these outputs, a decision is taken whether a compensation is required, and how will the gain compensation parameters described in the following paragraph, be affected when activating the gain compensation block 48 .
  • Gain Compensation block 48 The major task carried by block 48 , is to define the gain compensation required, and allow the increase in the gain factor for a first pre-determined period of time. This first pre-defined period of time may, in accordance with another embodiment of the invention, be changed. According to this other embodiment, a third pre-defined threshold is set for the gain peak threshold. Once this third pre-defined threshold is reached, an extended period of time is used for the gain compensation, where this period can be re-defined as a second pre-defined period of time. The use of such an embodiment allows extending the period of gain compensation in case the impulse change is relatively very high.
  • the level of the gain compensation can be changed so as to achieve the required effect.
  • the value of that limiter may be adapted to provide a better way to carry out the required gain compensation.
  • Predictor 16 is a shorter version of the G. 728 synthesis filter (block #G.22).
  • the prediction is based on the survivor path (section 4.4, block #J.7 of the “40 kbps algorithm”), in the following manner: at time n, a prediction of the current sample is formed for each node (section 4.5, block #J.8 of the “40 kbps algorithm”), using the sequence of reproductions specified by the survivor selected at time n ⁇ 1.
  • this method only a one-step scalar prediction is performed, and the prediction does not have to be extended far into the future. This makes the prediction more “localized” than in many other predictive VQ schemes.
  • the backward prediction coefficient adapter, 18 is similar to the backward synthesis filter adapter (block #G.23). the major differences are the following:
  • the hybrid windowing module (block #G.49) constantly calculates 51 auto-correlation coefficients, enhancing the performance of data-to-voice transitions.
  • the bandwidth expansion factor of the synthesis filter is now 240/256.
  • the bandwidth expansion coefficients are provided in section 9 of the “40 kbps algorithm”.
  • a VBD transmission of the V.23 type, in character mode was evaluated using the G.728 40 kbps algorithm.
  • the transmitted characters were compared with those received, and the number of discrepancies found out of the total number of characters transmitted, was calculated. This ratio was defined as the average error.
  • the method provided by the present invention was evaluated.
  • the values of the first and second pre-defined thresholds were pre-set to be equal to 1800. Once an impulse in the prediction gain was found to exceed the value of 1800, the gain compensation mechanism was activated provided that the preceding 80 digital sample vectors each comprising 5 samples where every sample was 125 ⁇ sec long were determined as being signals of the “steady” type. A dramatic decrease in the average error defined above was observed, as it dropped to about 0.05%.

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CN1272728A (zh) 2000-11-08
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AU1848400A (en) 2000-11-09
AU767450B2 (en) 2003-11-13
IL129752A0 (en) 2000-02-29
IL129752A (en) 2003-01-12
EP1058237A2 (de) 2000-12-06
CN1218501C (zh) 2005-09-07
PL338826A1 (en) 2000-11-06
EP1058237A3 (de) 2004-01-28
AR023121A1 (es) 2002-09-04

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