US4868810A - Multi-stage transmitter aerial coupling device - Google Patents

Multi-stage transmitter aerial coupling device Download PDF

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Publication number
US4868810A
US4868810A US07/083,548 US8354887A US4868810A US 4868810 A US4868810 A US 4868810A US 8354887 A US8354887 A US 8354887A US 4868810 A US4868810 A US 4868810A
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gtfm
digital
signals
coupler
filter
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US07/083,548
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Peter Vary
Ulrich Wellens
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Nokia of America Corp
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US Philips Corp
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Assigned to U.S. PHILIPS CORPORATION, 100 EAST 42ND STREET, NEW YORK, N.Y. 10017, A CORP. OF DE. reassignment U.S. PHILIPS CORPORATION, 100 EAST 42ND STREET, NEW YORK, N.Y. 10017, A CORP. OF DE. ASSIGNMENT OF ASSIGNORS INTEREST. Assignors: VARY, PETER, WELLENS, ULRICH
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Assigned to LUCENT TECHNOLOGIES INC. reassignment LUCENT TECHNOLOGIES INC. ASSIGNMENT OF ASSIGNORS INTEREST (SEE DOCUMENT FOR DETAILS). Assignors: U.S. PHILIPS CORPORATION
Assigned to LUCENT TECHNOLOGIES INC. reassignment LUCENT TECHNOLOGIES INC. ASSIGNMENT OF ASSIGNORS INTEREST (SEE DOCUMENT FOR DETAILS). Assignors: U.S. PHILIPS CORPORATION
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J1/00—Frequency-division multiplex systems
    • H04J1/02—Details
    • H04J1/08—Arrangements for combining channels

Definitions

  • the invention relates to an arrangement for coupling a plurality of transmitters having different carrier frequencies to an aerial according to the preamble of claims 1 and 2.
  • a transmitter coupler comprising a plurality of branches each having a circulator network and a band-pass filter. Multiplexing the output signals of the various transmitters having different carrier frequencies is carried out by the band-pass filters, which have one end connected to a star network and from there to the aerial via an aerial transmission line, and the other end to the transmitters via circulators.
  • the individual channel frequencies at a transmitter coupler must be spaced by a multiple of a channel spacing of the system (coupler channel spacing).
  • the maximum number of radio transmission channels connected to a transmitter coupler depends on the permissible insertion loss and the available bandwidth of the radio transmission system. In practice, a power loss of approximately 3 dB can be found when multiplexing. This power loss cons of the loss in the Y circulator of 0.5 dB, the loss in the band-pass filter of approximately 1.5 dB and other losses such as reflexion losses of approximately 1 dB.
  • the band-pass filters are adjusted to a specific operating frequency and can only work at another operating frequency after they have been adjusted anew. If an increased system capacity can be achieved by dynamic channel assignment, further transmitter coupler inputs can be provided, which are then utilized only occasionally. When this measure is implemented, more space will be required, stiffer demands will be made on the coupler channel spacing and the cost of the transmitter coupler will go up, so that a dynamic channel assignment in radio transmission systems has not been utilized so far.
  • a hybrid circuit is connected behind the transmitter coupler for switching operations between the various frequency groups, this hybrid circuit operating as a power divider, more specifically, a branching circuit, thus achieving a spreading of the frequency groups over the respective directional aerials.
  • each individual radio transmission channel has to be amplified accordingly prior to the transmitter coupler.
  • the subject matter of DE PS 28 44 776 does not provide a multiple use of the power amplifiers and integration of the transmitter coupler is not possible either.
  • a suitable dielectric though, the outside dimensions of, for example, an air-filled band-pass filter can be reduced.
  • a plurality of directly adjacent channels preferably in the baseband are combined in the first stage and the groups of channels constituted thus are multiplexed by means of band-pass filters in the second stage.
  • the first stage of the transmitter coupler is arranged as a digital coupler in which in addition to multiplexing also the modulation of the useful signal takes place.
  • the cost of the ensuing circuit components of the transmitter coupler, for converting the signals to the output frequency and for power gain, is reduced by the baseband-multiplex factor M as a consequence of the combination of the output signal from each transmitter in the pre-group phase.
  • the cost of the second coupling stage for the frequently conversion of the pre-group phase to the group-phase can be reduced by the multiplex factor M.
  • a dynamic channel assignment can be utilized when combining the output signals of the transmitters in the pre-group phase.
  • the transmitter coupler can be manufactured with monolithically integrated components resulting in a considerable cost reduction.
  • a second arrangement according to the invention has the advantage of a dynamic channel assignment feature and a reduction of the power gain by the baseband-multiplex factor M.
  • the circuit arrangement for making the device for the first stage of the transmitter coupler requires little circuit complexity and can be integrated.
  • FIG. 1 shows an embodiment of the invention of a known transmitter coupler
  • FIG. 2 shows the structure of the transmitter coupler in accordance with the invention
  • FIG. 3a and 3b show the spectral arrangements for digital signal processing in the digital coupler implementing the GTFM method
  • FIG. 4 shows an embodiment of the invention of combining the output signals of each transmitter in the pregroup phase by implementing the GTFM method
  • FIG. 5 shows a further embodiment of the invention of multiplexing in the first stage whilst utilizing not more than one interpolator
  • FIG. 6 shows an embodiment of the invention of a periodic time-variable interpolator for the embodiment according to FIG. 5,
  • FIG. 7 shows a further embodiment of the invention of digital signal processing in the first stage without DFT and having enlarged Cos-Sin-stores and
  • FIG. 8 shows a further embodiment of the invention of the signal processing in the digital coupler whilst utilizing additive address increments.
  • FIG. 1 shows an embodiment of the invention of a known transmitter coupler.
  • multiplexing takes place by means of band-pass filters BP1, . . . , BPn.
  • the band-pass filters BP1, . . . , BPn have one end consisting of lines L1, . . . , Ln connected to a star network S, which leads to aerial A via transmission lines, and the other end to the transmitters S1, . . . , Sn via circulators Z1, . . . , Zn. If double circulators are used the return loss lies at approximately 50 dB.
  • each transmitter frequency f1 to fn one circulator and one band-pass filter are arranged in each branch.
  • the frequencies f1 to fn situated in the output frequency band cannot be transferred mutually uneffected to aerial A until (fi - fj) meets the requirements of the minimal frequency spacing.
  • FIG. 2 shows the structure of the transmitter coupler in accordance with the invention designed as a two-stage coupler.
  • the output signals (source signals) b 0 , . . . b M-1 of the sources or transmitters S1, . . . , SM, respectively, are combined by a digital coupler K in a pre-group phase.
  • the output signals of the parallel arranged digital couplers K are each time fed to a circulator ZI, ZII, . . .
  • n is the number of inputs without a digital coupler K of the first stage) subsequent to the digital-to-analog conversion (by means of DA), frequency conversion in the group phase (by means of FUS) and power gain (by means of LVS).
  • the signal produced by digital signal processing by means of the digital coupler K is available at its output in normal and quadrature components (N, Q). These signals can be used for the direct frequency conversion to the output frequency, that is to say the group phase, or for conversion to an intermediate frequency with a subsequent frequency conversion to the output frequency and subsequent band-pass filtering. Depending on the smaller number of inputs of the second stage, the requirements made on this second stage can be diminished considerably.
  • the second stage of the digital transmitter coupler can preferably be dimensioned such that the available frequency band is completely utilized to its full extent.
  • the known transmitter couplers cf DE-PS 28 44 7766 which combine analog signals
  • the band-pass filters BPI, BPII, . . . BPk can be designed to have a single or a double circuit, in order to guarantee a constant insertion loss in the available frequency band.
  • a dynamic channel assignment within the first stage of the transmitter coupler does not require a manual adjustment of the band-pass filters BPI, BPII, . . . , Bk in the second coupler stage. Consequently, the second coupler stage does not have to meet stricter requirements made on the frequency spacing.
  • Intermodulation products produced by the power amplifier stage LVS appear (when attenuated adequately) on all radio transmission channels, which can be used in the stationary radio station for the transmission of information, and are also radiated by the aerial A.
  • the levels of the intermodulation products can be reduced.
  • FIG. 3a shows by way of a diagram the spectrums of the individual GTFM baseband signals Wi, shifted and superposed in their respective pre-group phase by quadrature modulation and single sideband modulation, respectively (cf. FIG. 3b).
  • the multiplex signal x(k) generated by digital signal processing additively consists of the components xi(k) according to the following equation: ##EQU1##
  • k denotes a time index (sampling point) M the baseband-multiplex factor and i the index of the respective radio transmission channel.
  • the output signals b i of the transmitter Si are applied to a sampler ATi operating with a first sampling frequency fK.
  • the sampler ATi is connected tc a GTFM filter Gi.
  • the complex GTFM baseband signal Wi is generated, whose spectrum is shown in FIG. 3a.
  • the GTFM baseband signal Wi can be represented in normal and quadrature components in accordance with the equation:
  • the GTFM baseband signal Wi can be sampled with the sampling frequency fK.
  • the source signal b i is, subsequently, sampled with the sampling frequency fK and a GTFM baseband signal Wi is generated in a way and manner described in DE-AS 28 38 984.
  • the output signal b i sampled with the sampling frequency fK is filtered digitally by means of GTFM filter Gi, connected to a Cos-Sin store Si.
  • the output signal vi of the GTFM filter Gi serves as an address for the Cos-Sin store Si.
  • the Cos-Sin store Si is connected to an interpolator Ii.
  • the digital interpolation filtering utilizing the second sampling frequency fx increased by the factor L is carried out separately for normal and quadrature components Ni, Qi of the GTFM baseband signal Wi.
  • the interpolator Ii is connected to a multiplier Mi, which multiplicatively combines the filtered normal and quadrature components Ni, Qi with a Cos- and Sin-oscillation, respectively, leading to the component xi in the pre-group phase. This process corresponds with a complex quadrature modulation.
  • each radio transmission channel requires one interpolation filter Ii.
  • the interpolation is carried out simultaneously for M ⁇ L radio transmission channels having a common filter I.
  • the output signals b 0 , . . . , b M-1 from m transmitters S1, . . . , SM are each applied to a GTFM filter Gi.
  • the GTFM filters Gi are connected to a processor P for constituting an inverse discrete Fourier transform.
  • the output signals of the processor P are applied to a parallel-serial converter PS, connected to an interpolator I for filtering its output signal by way of interpolation.
  • Equation (3) With the impulse response H(k) of the interpolator I the component x(k) according to the equation (3) is: ##EQU2##
  • the first sum term denotes the superpositioning of M complex band-pass signals xi.
  • the second sum term denotes the effected filtering by using modulation, Aa the impulse response h(k) is independent of channel index i, the order of the summation in equation (3) can be changed.
  • equation (4) With the complex exponential function the perodicity holds according to equation (4) ##EQU3##
  • equation (3) can be transformed.
  • IDFT inverse discrete Fourier transform
  • the embodiment of the invention of the digital coupler K according to FIG. 5 has the advantages, that only a single interpolator I is required instead of the otherwise required number of M available interpolators Ii and that by simple product summation with few terms a simultaneous quadrature modulation can be realised.
  • FIG. 6 shows an embodiment of the invention of a periodic time-variable interpolator.
  • a non-recursive filter having two-channel delay networks is utilized as interpolator I.
  • the interpolator I has a number of p stores SP1 having the length L for storing the last p values of the transform and a number of p stores SP2. having the length L for storing the impulse response.
  • the interpolator I further includes a number of p two-channel multipliers MU for forming the products of the values of the transform and values of the impulse response.
  • the two-channel multipliers MU are connected to an adder A having a number of 2p inputs, which adder adds up the product terms divided into normal and quadrature components N(k), Q(k).
  • the p ⁇ L filter coefficients are filed in the stores SP2 each having L registers. For a calculation of the sampling values of the normal and quadrature components N(k), Q(k) of the multiplex signal x(k) only p multiplications and additions are required for each sampling. In view of a proper selection of an adjacent channel, values preferably ranging from 3 to 8 are chosen for p.
  • the product is unequivocally determined by the output value vi of the GTFM filter Gi as well as the time index k and the channel index i (by i ⁇ k) modL , respectively. Therefore, in the embodiment of the invention shown in FIG. 7 the actual product formation is dispensed with and an enlarged Cos-Sin-store Sj is connected to the GTFM filters Gi.
  • the information vi changes with a lower clock fb, whilst the index addresses are changed with the higher clock frequency fK. Under the valid addresses are the values: ##EQU8## These values are also read out from the Cos-Sin-store Sj and applied to an adder network AN and added there.
  • the interpolator I is connected to the adder network AN.
  • an adder Ai is connected to each GTFM filter Gi.
  • the sum is formed from the respective increment (2 ⁇ /L) ⁇ i ⁇ k according to equation (8) and argument vi.
  • a store Sk connected to each adder A is directly addressed with this sum. Consequently, in comparison with the embodiment of the invention shown in FIG. 7 an increase of the number of address inputs of the store Sk can be dispensed with.
  • the increments themselves required for the addition can be filed in further stores. All in all only a maximum number of L different increments can be made available on account of the periodicity of the exponential function.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Transmitters (AREA)
  • Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
US07/083,548 1986-08-08 1987-08-06 Multi-stage transmitter aerial coupling device Expired - Fee Related US4868810A (en)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
DE19863626862 DE3626862A1 (de) 1986-08-08 1986-08-08 Mehrstufige sender- antennenkoppeleinrichtung
DE3626862 1986-08-08

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US4868810A true US4868810A (en) 1989-09-19

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US07/083,548 Expired - Fee Related US4868810A (en) 1986-08-08 1987-08-06 Multi-stage transmitter aerial coupling device

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US (1) US4868810A (ja)
EP (1) EP0255972B1 (ja)
JP (1) JP2525420B2 (ja)
CN (1) CN1009986B (ja)
AU (1) AU603743B2 (ja)
CA (1) CA1271236A (ja)
DE (2) DE3626862A1 (ja)
DK (1) DK407487A (ja)
FI (1) FI873403A (ja)

Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0518835A1 (en) * 1991-06-12 1992-12-16 Telefonaktiebolaget L M Ericsson Method of compensating the dependence of the useful transmitter signal on the transfer function of a combiner filter
EP1039667A2 (en) * 1999-03-04 2000-09-27 Nippon Telegraph and Telephone Corporation Variable transmission rate digital modem with multi-rate filter bank
US20040185781A1 (en) * 1999-10-21 2004-09-23 Shervin Moloudi System and method for reducing phase noise
US8718563B2 (en) 1999-10-21 2014-05-06 Broadcom Corporation System and method for signal limiting

Families Citing this family (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
DE3929702A1 (de) * 1989-09-07 1991-03-14 Standard Elektrik Lorenz Ag Schaltungsanordnung zum gegeneinander entkoppelten verbinden mehrerer nach dem frequenzsprung-verfahren arbeitender sender an eine antenne
US5289464A (en) * 1992-09-21 1994-02-22 At&T Bell Laboratories Frequency-multiplexed cellular telephone cell site base station and method of operating the same
DE19747447A1 (de) * 1997-10-28 1999-04-29 Cit Alcatel Vorrichtung zum Zusammenführen und Verstärken von zwei breitbandigen Signalen
FR2784259B1 (fr) * 1998-10-01 2000-12-01 Telediffusion Fse Systeme aperiodique modulaire de reemission multicanaux numeriques
DE10003704A1 (de) * 2000-01-28 2001-08-09 Infineon Technologies Ag Schaltungsanordnung mit Bandpaßfiltern

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US3742149A (en) * 1970-05-06 1973-06-26 Nippon Electric Co A frequency division multiplex microwave communication system using polarization division multiplex technique
US3865990A (en) * 1972-03-22 1975-02-11 Siemens Ag Radio relay systems
US4161694A (en) * 1976-06-28 1979-07-17 Siemens Aktiengesellschaft Radio relay channel branch cascade exhibiting uniform transit-time-and-attenuation-characteristics of all channels
US4607362A (en) * 1983-08-18 1986-08-19 U.S. Philips Corporation Method of and circuit arrangement for establishing conference connections in a switching system
US4682361A (en) * 1982-11-23 1987-07-21 U.S. Philips Corporation Method of recognizing speech pauses
US4697260A (en) * 1984-12-22 1987-09-29 U.S. Philips Corporation Method of and arrangement for transmitting messages in a digital radio transmission system
US4700394A (en) * 1982-11-23 1987-10-13 U.S. Philips Corporation Method of recognizing speech pauses
US4766562A (en) * 1985-03-23 1988-08-23 U.S. Philips Corp. Digital analyzing and synthesizing filter bank with maximum sampling rate reduction

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JPS4823851B1 (ja) * 1968-12-19 1973-07-17
FR2191826A5 (ja) * 1972-07-03 1974-02-01 Cit Alcatel
NL180369C (nl) * 1977-04-04 1987-02-02 Philips Nv Inrichting voor het omzetten van discrete signalen in een discreet enkelzijband frequentie-multiplex-signaal en omgekeerd.
NL7709917A (nl) * 1977-09-09 1979-03-13 Philips Nv Systeem voor datatransmissie met behulp van een hoekgemoduleerde draaggolf van constante amplitude.
JPS581843B2 (ja) * 1977-10-14 1983-01-13 日本電信電話株式会社 陸上移動無線基地局送信アンテナ共用方式
US4581749A (en) * 1984-07-02 1986-04-08 Motorola, Inc. Data frequency modulator with deviation control

Patent Citations (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US3742149A (en) * 1970-05-06 1973-06-26 Nippon Electric Co A frequency division multiplex microwave communication system using polarization division multiplex technique
US3865990A (en) * 1972-03-22 1975-02-11 Siemens Ag Radio relay systems
US4161694A (en) * 1976-06-28 1979-07-17 Siemens Aktiengesellschaft Radio relay channel branch cascade exhibiting uniform transit-time-and-attenuation-characteristics of all channels
US4682361A (en) * 1982-11-23 1987-07-21 U.S. Philips Corporation Method of recognizing speech pauses
US4700394A (en) * 1982-11-23 1987-10-13 U.S. Philips Corporation Method of recognizing speech pauses
US4607362A (en) * 1983-08-18 1986-08-19 U.S. Philips Corporation Method of and circuit arrangement for establishing conference connections in a switching system
US4697260A (en) * 1984-12-22 1987-09-29 U.S. Philips Corporation Method of and arrangement for transmitting messages in a digital radio transmission system
US4766562A (en) * 1985-03-23 1988-08-23 U.S. Philips Corp. Digital analyzing and synthesizing filter bank with maximum sampling rate reduction

Cited By (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0518835A1 (en) * 1991-06-12 1992-12-16 Telefonaktiebolaget L M Ericsson Method of compensating the dependence of the useful transmitter signal on the transfer function of a combiner filter
US5499389A (en) * 1991-06-12 1996-03-12 Telefonaktiebolaget Lm Ericsson Method of compensating the dependence of the useful transmitter signal on the transfer function of a combiner filter
EP1039667A2 (en) * 1999-03-04 2000-09-27 Nippon Telegraph and Telephone Corporation Variable transmission rate digital modem with multi-rate filter bank
EP1039667A3 (en) * 1999-03-04 2005-08-17 Nippon Telegraph and Telephone Corporation Variable transmission rate digital modem with multi-rate filter bank
US20040185781A1 (en) * 1999-10-21 2004-09-23 Shervin Moloudi System and method for reducing phase noise
US7933555B2 (en) * 1999-10-21 2011-04-26 Broadcom Corporation System and method for reducing phase noise
US8718563B2 (en) 1999-10-21 2014-05-06 Broadcom Corporation System and method for signal limiting

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FI873403A (fi) 1988-02-09
CN87105897A (zh) 1988-05-18
CA1271236A (en) 1990-07-03
DE3788382D1 (de) 1994-01-20
DK407487A (da) 1988-02-09
EP0255972B1 (de) 1993-12-08
EP0255972A3 (en) 1990-03-21
CN1009986B (zh) 1990-10-10
JPS6346831A (ja) 1988-02-27
DE3626862A1 (de) 1988-02-11
JP2525420B2 (ja) 1996-08-21
EP0255972A2 (de) 1988-02-17
FI873403A0 (fi) 1987-08-05
AU603743B2 (en) 1990-11-22
DK407487D0 (da) 1987-08-05
AU7664487A (en) 1988-02-11

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