US3375434A - Constant current generator - Google Patents

Description

March 26, 1968 J. R. SHAPIRO CONSTANT CURRENT GENERATOR Filed May 6, 1965 ll IO RLI CONSTR NT cuRREM Q "AM/2 I C E sconce /ww*-- 3 TRANSDUCER .IT I? I8 RLZ Ext RL3 AMP.

l RL4 INVENTOR.

W 5mm ATTORNEYS United States Patent 3,375,434 CONSTANT CURRENT GENERATOR Jerome R. Shapiro, Palos Verdes Peninsula, Calif., as-

signor to Winsco Instruments & Controls Company, Inc., a corporation of California Filed May 6, 1965, Ser. No. 453,658 1 Claim. (Cl. 323-4) ABSTRACT OF THE DISCLOSURE sistor to render current passing through the load constant.

This invention relates to constant current generators and more particularly to an improved solid state circuit for providing a constant DC output current through an external load which is substantially independent of the resistance of the load within a given range.

A primary object of this invention is to provide a very high dynamic output impedance constant current generator capable of providing a relatively wide range of controlled output D-C currents as compared to prior art constant current generators.

Another important object is to provide a constant current generator which is temperature stabilized.

Still another object is to provide a constant current generator particularly well suited for monitoring a variable resistance, such as in a transducer to enable signal conditioning of the transducer.

Briefly, these and other objects and advantages of this invention are attained by providing a pre-regulated DC power supply which may be operated from a conventional 115 volt A-C electrical source. A variable resistance means and current regulating means are connected in series between a first line from the power supply and a first output terminal. A second line from the power supply is connected to a second output terminal. The first and second output terminals are adapted to be connected across a load through which constant current is to be passed. Also included is a circuit means responsive to a change in voltage across the variable resistance means, for controlling the current regulating means in series therewith in such a manner as to insure a constant current through any load connected across the output terminals. The adjustment of the output current to a desired constant value may be effected by changing the value of the variable resistance means.

A better understanding of the invention will be had by referring to a preferred embodiment thereof as illustrated in the acocmpanying drawings, in which:

FIGURE 1 is a circuit diagram of the constant current generating means; and,

FIGURE 2 illustrates a useful application of the constant current generating means of FIGURE 1 Referring to FIGURE 1, there is shown a circuit having input terminals 11 for connection to a conventional 115 volt A-C source. The terminals 11 feed into the circuit for passing constant current through a load 12 shown at the lower right of FIGURE 1. This circuit includes a transformer T having its secondary connected to a bridge network including rectifiers CR1, CR2, CR3 and CR4. The upper opposite terminal of the bridge connects through resistances R1 and R2 to a junction point J1 and the lower opposite terminal connects to a junction point J2. A condenser C1 is connected between the resistances R1 and R2 and the line to the junction point J2. The com ponents in the form of the transformer T, the bridge CR1, CR2, CR3, CR4, and R1 and C1 simply convert the 115 A-C line voltage to a suitably low D-C operating voltage for the circuitry. A Zener diode VR1 is connected between the junction points J1 and J2 and functions with the resistance R2 to further drop this voltage and regulate it. All of the components described thus far are hereinafter referred to as a pre-regulated D-C power supply which functions as a preliminary voltage regulator for changes in line or load voltage.

Connected to the terminal junctions J1 and J2 of the DC power supply are first and second output lines 13 and 14 shown at the upper and lower portion of the drawing. A temperature compensated Zener diode VR2, first NPN transistor Q1 and a first resistance R5 are all connected in series between the first and second lines 13 and 14 as shown. The temperature compensated Zener diode VR2 receives a current bias from the pre-regulated DC power supply through the resistance R5 and the collectoremitter circuit of the transistor Q1. This diode serves to provide a basic reference voltage ER for the circuit, A second PNP transistor Q2 has its collector terminal connected to the first line 13 and its emitter terminal connected through a second resistance R6 to the second line. The emitter terminal of Q2 is also connected to the base terminal of Q1 so that the transistors Q1 and Q2 form a complementary cascaded amplifier. The current gain of this combination is approximately equal to the product of the current gain of the individual transistors. Moreover, the base-emitter voltage of the transistor Q2 is made equal and opposite in polarity to the base-emitter voltage of the transistor Q1. The transistors Q1 and Q2 are selected so that any change in base-emitter voltage with temperature is the same for both so that the transistors are self-compensating with temperature.

A third NPN transistor Q3 has its base connected to the collector terminal of the first transistor Q1 at the junction point J3. A third resistance R7 in turn is connected between the emitter terminal of the transistor Q3 and the base terminal of the transistor Q2. R7 is used to provide a path for emitter current of Q3 so that it will not operate in a highly starved state. It is returned to the junction of R4 and a fourth NPN transistor Q4 so that all current flowing through the load also fiows through R3 and R4. The collector terminals of the transistors Q3 and Q4 in turn connect to a junction point J4 constituting a first output terminal of the circuit. The load 12, as shown, is connected across this first output terminal and a second output terminal J5 connecting to the second line 14.

The fourth NPN transistor Q4 has its collector and emitter terminals connected in series with a variable resistance means in the form of the resistances R3 and R4, the entire series circuit comprised of the fourth transistor Q4- and variable resistance means R3 and R4 being connected between the first line 13 and first output terminal J4 so as to be in series with the load as described. The base of the transistor Q4 connects to the emitter terminal of the transistor Q3 as shown and the base of the transistor Q2 connects to the emitter terminal of the transistor Q4 at its point of connection to the variable resistance means R3 and R4.

The third and fourth transistors Q3 and Q4 supply the regulated current to the output load. These transistors are in the form of cascaded amplifiers in what is commonly referred to as a Darlington connection. The transistor Q4 carries the bulk of the load current. The function of Q3 may be considered to divert the base current of the transistor Q4 back through the load. Practically all of the current that flows through the variable resistance means R3 and R4 also flows through the load 12. Essentially, this current is controlled by the base voltage, designated E2 and E3, on the transistor Q3 at the junction J 3 which in turn is determined by the reference voltage ER plus the collector-emitter voltage across the transistor Q1 designated E1 in FIGURE 1.

The first and second transistors Q1 and Q2 together with the reference diode VR2 function as a means responsive to any change in voltage across the variable resistance means R3 and R4 to control the base voltage of the transistor Q3 and thereby control the collectoremitter current through the transistor Q4 and thus the current through the load in such a manner as to maintain this current constant for a given setting of the variable resistance means R3 and R4. A condenser C2 is connected across the output terminals J4 and J5 to suppress high frequency noise across the load.

In the operation of the circuit of FIGURE 1, a load through which constant current is to be passed such as the load 12 is connected across the output terminals J4 and J5 as shown. Assuming that a specific desired constant current is to be passed through this load, the value of the variable resistance means R3 and R4 may be adjusted by the tap as shown to provide a given current I-L through the load. The bulk of this current flows through the transistor Q4 and the resistances R3 and R4 from the line 14 to the line 13 and results in a voltage drop, E3, across the resistances R3 and R4. If the resistance of the load changes, to thereby momentarily change the valve of current flowing therethrough, there will be a change in the voltage drop across the resistances R3 and R4. This voltage drop across R3 and R4 is detected and employed to adjust the base voltage of the transistor Q3 to control the current through the load in such a manner as to maintain it constant.

The foregoing is accomplished by sensing the voltage drop across R3 and R4 at the base of the transistor Q2. This voltage drop is compared to the reference voltage ER and any difference is employed to control the collectoremitter voltage, E1, across Q1. E1 is added to the reference voltage ER so as to maintain the required potential or voltage on the base of the transistor Q3 to maintain the voltage drop across R3 and R4 constant.

If the base current for the transistor Q2 is designated Ib2, the base current for the transistor Q3 is M3; the emitter current for the transistor Q4 is le4; and the current flowing through R3 and R4 is I1, the load current IL supplied by this circuit may be found as follows:

Since Ib2 and 1173 are very small in comparison with I1, IL substantially equals I1. 11 in turn may be expressed as follows:

where E3 is the voltage drop across the resistances R3 and R4.

The voltage drop E3 is made equal to ER, the zener reference voltage, by feedback around the circuitry. This may be shown by summing voltage drops around the loop including R3, R4, base to emitter of Q2, base to emitter of Q1, and ER. E3+Ebe2Ebe1-ER=0 where Ebe2 and Ebel represent the voltage between the base and emitter terminals of the transistors Q2 and Q1 respectively.

Since the base-emitter voltages of Q2 and Q1 are chosen to be equal in magnitude and opposite in polarity, they cancel from the above equation so that E3:ER. The condition that Ebe2 equals Ebel Will be assumed in all discussions to follow.

The load current IL which has been shown to be substantially equal to I1 may be expressed as Since ER is a carefully regulated reference volt-age, the load current is thus constant for any given setting of R3 and R4.

During normal operation of the circuit, the collectoremitter voltage drop of the transistor Q1 (E1 in FIG- URE 1) is very small. It will be shown that this voltage drop is equal to E2, the sum of the base to emitter voltage drops of Q3 and Q4. By summing voltage drops around the loop composed of ER, E1, E2, and E3, the following equation results: ER+E1=E2+E3, but ER=E3 from above, therefore, E1-E2=0, or E1=E2.

Thus, the collector-emitter voltage of the transistor Q1 automatically adjusts itself to always be equal to the voltage E2 no matter how it may vary. The circuit is therefore self-compensating for changes in the base-emitter voltages of Q1 and Q2 which may occur due to temperature variations.

For good temperature stability, the following require- .ments apply: resistors R3 and R4 must be highly temperature independent (wire wound resistors); zener diode VR2 must be highly temperature independent; and, the forward base to emitter voltage drops of Q1 and Q2 must have a matched temperature characteristic.

Since the operating collector to emitter voltage of Q1. is wholly dependent upon the base to emitter voltage drops of Q3 and Q4 transistors with large forward voltage drops, such as silicone transistors, should be chosen for use in this circuit. Also, transistor Q1 should be chosen to have a high dynamic grounded emitter current gain B, at low collector emitter operating voltages.

An alternate to the requirement that Q3 and Q4 have large voltage drops is with the addition of the zener diode VR4 in series with the emitter of Q4. With the addition of this component, Q3 and Q4 need not have any restrictions as to their forward base emitter drop, and transistor Q1 neednot be selected for its current gain at low operating voltage.

The addition of the zener diode VR4 will not materially affect the current regulating characteristics nor the temperature stability of the circuit.

One useful function for the circuit of FIGURE 1 is in the signal conditioning of transducers. In conventional signal conditioning processes, errors are introduced by the lead resistances to the transducer. The constant current generator of this invention when used to signal condition transducers will completely eliminate this error. For example, referring to FIGURE 2, the load 12 is shown as a variable resistance transducer including four lines having resistances designated RL1, RL2, RL3, and RL4. These resistances may represent resistance introduced by lead resistance, relay contacts, or other resistances'deli-berately inserted in the leads for other purposes, such as calibration.

The constant current source 10 is connected to supply a current through RLl, the resistance transducer 12 and RL4. The voltage drop across the resistance transducer is sensed by a high impedance external differential amplifier 15 through the leads RL4 and RL3 and an adjustablebias voltage supply 16. This latter element is inserted into the circuit for offsetting the DC transducer signal so that a zero signal reference level may be established for some fixed value of the transducer dependent variable.

If now the amplifier impedance is sufficiently high, for example of the order of 1 megohm, the current IA flowing through the amplifier circuit is too small to introduce a significant voltage drop across RL4 and RL3.

In the limiting case, the external amplifier is replaced by a passive nulling indicator and variable reference voltage circuit, This presents an infinite impedance at the external amplifier terminals, the loop current IA then becomes zero, and zero error is introduced by the existence of RL4 and RL3 in series with the amplifier.

Since the amplifier sensing terminals 17 and 18 are electrically at the variable resistance transducer, the signal output is equal to the product of the current generated by the constant current source and the transducer resistance, and no error is introduced by the existence of RL1 and RL2 in the transducer leads. The variable resistance transducer may thus be changed and since a constant current is always provided by the source through the variable resistance of the transducer, the voltage change across the transducer terminals will be directly proportional to the value of the resistance, the resistances RL1 and RL2 having no efiect.

From the foregoing description, it will thus be seen that the present invention has provided a greatly improved constant current generator. Not only is there provided a high dynamic output impedance, but the circuit is automatically temperature compensated and enables a wide range of output currents to be provided.

What is claimed is:

1. A constant current generator comprising, in combination: a pre-regulated D-C power supply having first and second output lines; a voltage reference zener diode, a first NPN transistor, and first resistance, connected in series across said output lines so that a given constant voltage is established across said diode between said first line and the emitter terminal of said first transistor; a second PNP transistor having its collector terminal connected to said first line and a second resistance connected between the emitter terminal of said second transistor and second line, the base terminal of said first transistor connecting to the emitter terminal of said second transistor so that said first and second transistors form a complementary cascaded amplifier in which the base-emitter voltage of said second transistor is made to be equal and opposite in polarity to the base-emitter voltage of said first transistor; a third NPN transistor having its base connected to the collector terminal of said first transistor; a third resistance connected between the emitter terminal of said third transistor and the base terminal of said second transistor; first and second output terminals for connection across a load through which constant current is to be passed, the collector terminal of said third transistor connecting to said first output terminal, said second line connecting to said second output terminal; a fourth NPN transistor; and variable resistance means connected between the emitter terminal of said fourth transistor and said first line, the collector terminal of said fourth transistor connecting to said first output terminal, the base terminal of said fourth transistor connecting to the emitter terminal of said third transistor, and the base terminal of said second transistor connecting to the emitter terminal of said fourth transistor whereby the emitter current of said fourth transistor is determined by said given voltage divided by the value of said variable resistance means said emitter current being controlled by the base voltage of said third transistor to be substantially constant for any value of load resistance within a given range connected across said output terminals.

References Cited UNITED STATES PATENTS 2,994,029 7/ 1961 Bixby. 3,005,147 10/ 1961 Thomas 323-9 3,031,608 4/ 1962 Von Eschen et al. 3,160,807 12/1964 Packard. 3,246,233 4/ 1966 Herz 323-4 3,311,814 3/1967 Cliifgard 3239 JOHN F. COUCH, Primary Examiner. A. D. PELLINEN, Assistant Examiner.

Images