US2347459A - Frequency modulation apparatus - Google Patents

Description

April 25, 1944. w. F. GO ETTER FREQUENCY MODULATION APPARATUS Filed May 28,- 1942 2 Sheets-Sheet 1 W Ym .m M .w cm W v. 1 Ki we eF M F vm M w B n AUDIO AMPLIFIER SOURCE OF HIGH FREQUENCY OSCILLATIONS April 25, 1944.

' FREQUENCY MODULATION APPARATUS- w. F: GOETTER 2,347,459

Filed ma 28', 1942 Fig.6.

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MODULATION POTENTIAL.

I nventor Wi l'l'iam F Goetter His Attorney application.

Patented Apr. 25, 1944 2,347,459 FREQUENCY MODULATION APPARATUS William-F. Goetter, Schenectady, N.

Y; assignmto General Electric Company, a corporation of New York Application May 28, 1942, Serial No. 444,843 9 Claims. (01. 179-1715) My invention relates to frequency modulation apparatus, and more particularly to that type of such apparatus commonly termed phase modulation apparatus.

In apparatus, which is utilized to modulate the phase of a carrier wave in accordance with a modulating potential, it is desired that such phase be modulated in linear relationship to the modulating potential. That type of apparatus which modulates the phase directly, as by phase splitting of the carrier' wave, and is arranged to modulate the phase of the carrier wave the same amount for the same modulating potential of any frequency, should modulate the phase of the carrier wave in linear relationship to the modulating potential intensity atany modulating frequency. If such linear relationship does not obtain, when the modulated carrier wave is demodulated in any desired fashion, harmonics and cross-modulation products of such har monies of the modulation potential are obtained. To avoid obtaining such harmonics and crossmodulation products of harmonics, it is necessary to preserve such linear relationship.

It is accordingly an object of my invention to provide a new and improved apparatus whereby the phase of a carrier wave is modulated in linear relationship to the intensity of a modulating potential. I

A similar apparatus for improving the linearity phase of a carrier wave and the modulating potential is disclosed and claimed in the copending application for Letters Patent of the United States, S. N. 440,172, filed April 23. 1942, of George M. Brown. Frequency modulation system, assigned to the same assignee as thepresent In this copending application a three-channel phase modulation system is utilized, in which three components of the carrier wave are generated. Two of these components are of opposite phase, and the third differs by a quarter cycle from each of the other two components. The two oppositely phased components are modulated oppositely in intensity in response to modulating potential, and the third component is reduced in intensity, in response to increases in intensity of either polarity of the modulating potential. an amount such that the phase of the resultant of the three components of the carrier wave is shifted by an amount substantially proportional to the intensity of the modulating potential.

It is a specific object of my invention to provide'a new and improved apparatus and method of the relationship between modulation of the for reducing the intensity of such third component in such a way as to increase the linearity of the relationship between phase shift of the carrier wave and intensity .of the modulating potential.

In such a multiple channel phase modulation system, the change in frequency of the carrier wave produced by its change in phase in response to modulating potentials is very small. It isnecessary to multiply the frequency of the carrier wave many times through suitable free quency multiplying, amplifiers which simultaneously increase the frequency change of the carrier wave, to obtain a change in frequency of the radiated carrier wave reasonably large with respect to the frequency of the modulating potential. Itis well known that the modulation of' the frequency of the radiated carrier waves should be large with respect to the frequency of the modulating potential to obtain noise-free signal transmission.

When a very large amount of frequency multiplication is thus used, very small random voltages incircuits preceding those utilized for such frequency multiplication produce substantial frequency changes in the radiated carrier wave. It is accordingly another object of my invention to reduce the amount of frequency multiplication necessary in a phase modulation system and thereby reduce the amount of random frequency shift of the carrier wave caused by small voltages, commonly called noise voltages, by providing a new and improved apparatus and method for modulating the phase of a carrier wave to a larger degree while retaining high linearity in the relationship between phase shift thereof may best be understood by reference to I the following description taken in connection with the accompanying drawings in which Fig. 1

illustrates one embodiment of my invention; I

Figs. 2 through 7 are vector diagrams and graphs representing certain characteristics of the apparatus of Fig. 1; and Figs. 8, 9 and 10 illustrate modified versions of a certain portion of the apparatus of Fig. 1.

In that form of my invention which-is illustrated in Fig. 1, a high frequency carrier wave from the source '10 istransmitted in different the terminals of the ing the electron discharge devices I I l2 and I3,

and after recombination through suitable fre-- quency multipliers and power amplifiers H to be radiated from an antenna l5. Those three components of the carrier wave from the source I transmitted through the devices H and I3 are suitably modulated in intensity in accordance with signals from a microphone it as described hereinafter.

The apparatus utilized to divide the high'frequency carrier wave from the source It into three components, two of which are in opposite phase and the third of which is difi'erent in phase by a quarter cycle from the first two, includes a pair of tuned circuits I! and I8. Tuned circuit I1 is connected across the output of source Ill, and one terminal is grounded. Tuned circuit I 8 is inductively coupled to tuned circuit l1, and

a center tap IS on the inductance of tuned circuit I8 is grounded. By reason of the inductive coupling between the tuned circuits Hand I8, the voltage across the terminals of the timed circuit l8 differs by a quarter cycle from that across circuit 1.. The voltages at the opposite terminals of the balanced with respect to ground byreason of the grounded center tap l9 of the inductance 20.

The first, or control, electrode 2| of device II is coupled through a condenser 22 tothe upper tenninal of the tuned circuit l8, and the first, or control, electrode 23 of device I3 is coupled through condenser 24 to the lower terminal of circuit I8. The oppositely phased voltages on the opposite terminals of tuned circuit l8 are accordingly impressed on the control electrodes 2| and 23. The first or control electrode 25 of device 2 is coupled through a condenser 26 to the ungrounded terminal of tuned circuit l1, and accordingly a voltage is impressed thereon which difl'ers in phase by a. quarter cycle from the oppositely phased v ltages on the control electrodes 2| and 23.

These voltages are illustrated as vectors in Fig. 2, in which the vector E21 represents the voltage on the control electrode 2|, the vector E23 represents the oppositely phased voltage on the control electrode 23, and the vector E25 represents the voltage on the control electrode 25 which differs by a quarter cycle from the voltages represented by vectors E21 and E23.

While I have shown only one arrangement for impressing such variously phased voltages on the devices II, and I3, any suitable arrangement .may be utilized, as,- for example, that shown in the above mentioned application of George M. Brown. ,In such application the voltage to be appliedto control electrode 25 is obtained by connecting across the tuned circuit I8 a path serially including a resistance and a condenser, the reactance of the condenser at the frequency of carrier wave from the source It being equal to the resistance. The condenser 28 couples the control electrode 25 to the point between the resistance and condenser and is energized with a voltage as represented by the vector E25 of Fig. 2.

Suitable operating potentials are supplied to the devices I, I2 and I3 as follows. The cathodes 21, 28 and 29-01 the devices l2 and I! are grounded. A resistance 30 is connected between the control electrode 2| and cathode 21. A resistance 3| is connected between control electrode 25 and cathode 28. A resistance 32 is connected between control electrode 23 a d cathode tuned circuit l8 are Fig. 3. The current vice I2 is connected to the cathode 29. The anodes 40, 4| and 42 of the devices l2 and I3 are connected together and through a tuned circuit including an inductance 43 and condenser 44 in shunt thereto, and an inductance 45, to the positive terminal of a suitable source 46 of operating potential. The negative terminal of the source 46 is grounded, and a point between the inductances 45 and 43 is bypassed to ground througha condenser 41 so that'the inductance 45 and condenser 41 act as a filter for continuous current flowing through the anodes of the devices ll, l2 and i3 from source 46.

The second, or screen, electrode 48 of the debypassing condenser 49, and to the positive terminal of the source 46 through resistance 50. This condenser 49 has low reactance at the frequency of waves from source II) but has high reactance at signal frequencies. The l3 are oi. the type having five control electrodes, of which the third may be used as a gain control electrode. Such third or gain control electrode 5| of the devices II is connected to one terminal of the secondary 52 of a transformer whose primary 53 is energized through an audio amplifier 54, through which signals from the microphone ll are amplified. Such third or gain control electrode 55 of the device I3 is connected to the other terminal of the secondary I52, and the center tap 56 of the secondary 52 is connected to ground. The second and fourth, or screen, electrodes of the device are connected to the cathode 21 through a condenser 6|, and to the second and fourth, or screen, electrodes 62 of the device l3 and to the positive terminal of the source 46. If desired, a condenser 63 may be rectly between the electrodes 62 and cathode 29.

With the operating circuits described, the operation is as follows. When there is no signal from the microphone l6, and the third control electrodes 5| and 55 of devices II and 13 are at ground potential, the three currents in the anodes 40, 4| and 42 are as illustrated by the vectors of flowing through the anode 40 is controlled in intensity by the voltage on the control electrode2l, represented by vector En in Fig. 2, such current being represented by vector I40 in Fig. 3. For the sake of simplicity, only the alternating component of current in the anodes 40, 4| and His considered in the vectors of Fig. 3. The current in the anode 4| of the device I2 is controlled by the voltage on the control electrode 25 and is represented Fig. 3. Similarly, the current in the anode 42 of the device I3 is represented by the vector In of Fig. 3 and is controlled by the voltage represented by vector E2: of Fig. 2 on control electrode While the phase of the three currents I40, I41 and I4: is completely controlled by the control not entirely dependent on the intensity of the control voltages E21, E25 and E23 respectively. Since the control voltages E21, E25 and E23 are constant, so far as these voltages are concerned. the currents I40, I41 and I42 are constant. However, each'of the devices II and I3. includes a When modulating signals from the microphone 28 through a devices II and connected diby the vector In of II; are transmitted through the amplifier M and appear in opposite phase on the gain control electrodes and 55, the gain of the devices I! and I3 varies oppositely, so that, when the current I40 increases, the current 14: decreases.- Conversely, when the current'Im decreases, the current I42 increases.

When there is no signal fromthe microphone Hi, the currents I40 and 142 are equal and in opposite phase and cancel, excitation of the tuned circuit 43, 44 being caused only by the current represented by vector I41. When the signals from microphone |6 cause one of the currents I40 and 142 to increase and the other to decrease, these currents do not cancel out, and a resultant current remains, which combines with the current 142 to produce a resultant current of phase difierent from current 142. This current flowing through the tuned circuit 43, 44 produces a'voltage thereacross which is different from the voltage thereacross when only current 141 flows through .the tuned circuit 43, 44.

This change in phase of the current In is illustrated in Fig. 4 for oneparticular instant at which signal potential from the microphone I6 is such that the gain control electrode 5| is positive with respect to ground and gain control electrode 55 is negative. At such instance the gain of the device H is increased, and the gain ofthe device l3 is decreased. Consequently, as

illustrated in Fig. 4, the vector 140 is much larger than the vector I42, and the three vectors I40, I41 and I42 combine to form a resultant I43, which represents the current flowing through the inductance 33. This current 14: leads the current I41 flowing through anode 4| of device |2 by an angle 6.

The vector difference, labelled 140-142, is represented by the dotted line extending between the separated ends of the vectors I41 and 14a, and

is proportional to the instantaneous intensity of the modulating signals from the microphone I6.

It is evident from an inspection of-Fig. 4 that incremental changes in the vector labelled 146- I42, which is proportional to the modulating potential, do not produce proportionate incremental changes in the angle 0. As set forth previously, such'proportionality is necessary to avoid distortion caused by the production of harmonics and cross-modulation products thereof. Actually, the greater the angle 0, the less is the changev in the angle 0 produced by a unit change in the modulating potential,

As set forth in the above mentioned application of George M. Brown for Frequency modulation systems-,a suitable reduction in the intensity of the current 141 made with every increase of the modulating potential is effective to increase the angle 0 just sufficiently during each incremental change thereof'to maintain substantial proportionality between changes of the angle 9 and changes of intensity of the modulating potential. Now ,it is evident that, without such corrective reduction of the current 141, the current I43 is greater than the current I41. If the current 141 be reduced upon increases in the resultant current-I43 by a suitable amount, a high degree of linearity may be obtained between the angle 0 and the modulating potential which produces it.

Apparatus for causing the current 141 to be reduced in response to increases in the resultant current 143' is provided as follows. The cathode 10 of a diode discharge device II is connected through a load resistance 12 in shunt to a bypassing condenser 13, whose reactance at the frequency of carrier waves from the source I0 is small but at the highest frequency of signals from the microphone I6 is relatively large. The

, device 1|, and is coupled through a coupling contial corresponding to that the cathode with respect to the control electrode but not Q in the vectors I40 and I42 is denser 16 to the anodes 40, 4|, and" of the devices l2 and i3. Consequently, a unidirectional voltage is produced across the resistance I2 which is proportional to the resultantv intensity of the carrier wave across the tuned circuit 43, 44,, and is accordingly proportional to the total current through that tuned circuit, represented by the vector 14:; in Fig. 4.

The cathode of a triode discharge device 8| is connected through a suitable vapor discharge device 82 to ground. The anode 83 of the device 8| is connected to the screen electrodes 48 of the device H. The control electrode 84 of the device 8| is connected'to a point between the cathode 10 of the, diode rectifier H and resistance I2.

The vapor discharge device 82 is of that type across which the voltage is substantially constant throughout large changes of current therethrough. The magnitude of resistance 12 is such that the rectified voltage thereacross is of. such magnitude asto cause the control electrode at of the device 8| to operate over a linear portion of its characteristics. That is, the voltage across the resistance 12 is less than the voltage across the vapor discharge device 82 by an amount such 80 is always maintained positive so positive that the control, electrode 84 approaches that voltage at which current flow through the device 8| is cut off.

The discharge device 8| together with the vapor discharge device 82 constitute a continuous potential amplifier which is effective to apply to the screen electrode 48 of device |2 amplified potential changes across'the resistance I2.

Referring to Fig. 5- the operability of the scheme of reducing the current 141 in response to increases in intensity of current 14: to produce linearity between the angle '0 and modulating potential is shown. In this figure, the current 141 is plotted as an abscissa on an arbitrary scale, and the vector difference -442, corresponding to modulating potential, is plotted as ordinates to the same scale. When a small modulating potential, suflicient to cause 0.1 unit difference present, the vector I43 is producedextending to point A, and making an angle of 0.1 radian with the vector 141. -At such small angles of phase shift, the relation between the angle and the modulating potential is substantially linear.

When modulating potentials of higher intensity produce larger phase shift, such, for example, as a modulating potential effective to produce a diiference between the currents I40 and 14: of 0.7 unit exists, the vector 143 extends to point B and is at an angle with vector 141 of 0.61 radian. It is thus evident that the angle between the vectors I43 and his substantially less than 0.7 radian, which is the phase shift that should be produced in response to a modulating poten- 0.7 unit of the vector difference 140-142. v

Now if the system including the rectifier II. and amplifier 8| were efiective to produce so much feedback that the intensity of the current proper angle 0.7 radian corresponding to a modulating potential which produces 0.7 unit difference between the currents I40 and I42, is produced by reducing the intensity of the vector I41 only enough to reduce the current I43 only partially toward the intensity of the current In. The vector I43 which extends to point D makes an angle of 0.7 radian with the vector I41, the vector I41 having been reduced by the amount BD to cause the vector I43 to lie at such angle.

It has thus been demonstrated graphically by the aid of Fig. 5 that, at a particular intensity of the modulating potential, the feedback through .the rectifier 1| and amplifier 8| may be so adjusted that that component of the carrier wave transmitted through device I2 may be reduced by an amount proportional to the increase in intensity of the resultant of the three components of the carrier wave transmitted through the devices IZ-and l3 to the tuned circuit 43, 44 so that the relation between the phase shift of the resultant carrier wave on the tuned circuit 43, 44 and the modulating potential producing it, is the same as the relationship between a certain small phas shift and the modulating potential producing it.

By reference to Fig. 6, it may be shown that the amount of feedback found to be desirable in Fig. 5 at one particular intensity of the modulating potential is effective to maintain a linear I relationship over a larg range of such intensity between modulating potential intensity and carrier wave phase shift. The abscissae and ordinates in Fig. 6 are similar to those in Fig. 5 and need not be further described. At twelve different intensities of the modulating potential, the largest being suflicient to cause a phase shift of the carrier wave in on direction of 1.2 radians, graphical solutions are found, utilizing the same proportionate amount of feedback shown to be necessary in Fig. 5, to determine the resultant phase angle and intensity of the current In in each of such twelve cases.

In each case the amount of reduction in the 0 vector I41 which has been plotted is proportional to the amount by which the resulting vector I43 exceeds the vector In in its unmodulated intensity.

A typical construction is illustrative. At an ordinate value of 1.1 unit, the dotted line 90 represents the shift in phas of voltage across the tuned circuit 43, 44 which would be produced if no reduction in intensity of the current through discharge device I! were caused. The solid line 9| is drawn at an angle with respect to. the vector I41 corresponding to the modulating potential unit 1.1, that is, at an angle of 1.1 radians. The line 92 represents the reduction in intensity of the vector In necessary to shift the phase of the resultant carrier wave the desired increased amount to render it proportional to modulating potential intensity. The line 93 represents the amount by which the resultant carrier wave intensity exceeds the unmodulated intensity thereof, and also is proportional to the amount of reduction of the component of the carrier wave transmitted through device I! and represented by vector I41.

The point 94 represents the end of the carrier wave vector which actually results when that amount of feedback is used which was determined as desirable by the graphical analysis of Fig. 5. It should be noted that point 94 does not correspond to the intersection of the line 9|, drawnat the desired phase angle, and the horizontal line at the unit intensity 1.1; The departure of point 94 from this intersection'represents an increase in phase angle larger than that desired, and consequently represents distortion.

In Fig. 6 there is no such distortion observable at modulating potential unit intensities less than 0.9, and the distortion is slight even mum intensity shown, which is 1.2.

In Fig. 7 there is illustrated a graph showing certain relations between modulation potential and the resultant phase shift in various types of phase modulation systems. Modulation potentials are plotted as abscissae, and the resultant phase shifts are plotted as ordinates. The dotted curve I00 is a straight line, and represents the desired ideal relation between modulation poten tial intensity and the resultant carrier wave phase shift. The downwardly drooping curve |0| represents the relation in a system in which the carrier wave intensity transmitted through device l2 of Fig. 1 is not modulated. It is evident that large departures from linearity are present in this system even at modulation potentials low enough to produce only smal1 amounts of carrier wave phase shift, of the order of ten or fifteen degrees.

The curve |02, which departs slightly from the ideal curve I00 at its upper end, corresponds to the relation between modulation potential intensity and carrier wave phase shift obtained with the system illustrated in Fig. 1. It is evident that a high degree of linearity exists in this relation even at relatively large amounts of carrier wave feedback may be obtained in the apparatus of' Fig. 1 by adjustment of the resistance 50. By adjustment of this resistance the gain of the amplifier 8| is adjusted to adjust the amount of gain in the feedback loop and consequently the amount of degenerative feedback. Of course, the gain of device 8| may be adjusted in any other convenient manner. This resistance 50 may be adjusted until it is observed by suitable means ,that the carrier wave phase shift is substantially proportional to modulating potentia1 intensity.

The non-linear relation between anode current and control grid voltage which occurs in any triode or pentode electron discharge device may be utilized to minimize the deviation of the curve I02 in Fig. 7 from the straight line I00. It is well known that, as the control electrode of an electron discharge device becomes less negative-or more positive, with respect to the cathode, the anode current tends to reach a maximum value. That is, with this type of electron discharge device, there is a certain region within which the control electrode voltage may be varied to produce proportionate changes in anode current, but when the control electrode voltage is made less negative or more positive than voltages within such region, the changes in anode current are less than those experienced within at the maxisuch regions for equal changes in control elec trode voltage.

By adjusting all the operating conditions of the electron discharge device 8| so that the relation between its control electrode potential and anode current is linear so long as the discharge device 8| operates along that portion of the curve I02 which coincides with the straight line I00, and by making these conditions such that the amplification of the device 8| is reduced when its control electrode 84 reaches a less negative or more positive potential with respect to thecathode 80, so that the device 8| is operating along the upper portion of the curve I02 which does not coincide with straight line I00. The

feedback may be reduced in any desired amount at times when the carrier wave is shifted a large amount in phase, so that the curve I02 of Fig. 7 is given any desired linearity.

In general, any discharge device may be operated under such conditions as to have the above described characteristics. However, it is desirable to use a discharge device whose cathode or filament is capable of emitting a relatively small discharge current. Uncoated tungsten filaments are ,of this type. The anode current and control electrode bias are adjusted so that saturation of the electron emission from the cathode 80 is approached for the apparatus of Fig. 1 where the apparatus of Fig. 1 begins to operate along that portion of the curve I02 of Fig. '7 which ,deviates from the straight line I00.

The same eifect'ma-y be achieved by utilizing that portion of the control electrode voltageanode current characteristic wher the control electrode becomes sufficiently negative to approach anode current cutofi. In this region of that characteristic, equal reductions in anode current require larger andlarger increases in negative control electrode potential. By the in sertion of an additional continuous potential amplifier between the control electrode 8 of device 8| and resistance I2, and by making such additional discharge device operate with its control electrode becoming increasingly negative in po-' tential as the carrier wave in the tuned circuit 43, 44 is increasingly shifted in phase the same result may be obtained. It is, of course, necessary to reverse the polarity of the diode rectifier II to obtain the correct polarity across resistance 12.

Where signal voltages from the microphone H5,

.or other source of signals, are transmitted through the amplifier 5 1 in equal intensity at all frequencies, the apparatus of Fig. 1 is very satisfactory, producing a phase shift of the carrier wave proportional to the intensity of signal voltage at any frequency. Consequently, the fre quency of the carrier wave is modulated over a range which is proportional both to signal voltage intensity and frequency. Such transmission is desirable in certainapplications since noise voltages are more objectionable at high frequencies, and the large frequency shift of the carrier wave obtained in response to high frequency signal voltages are effective in minimizing the effect of such noise voltages.

Where, as in the present standard frequency modulation broadcasting, it may be desired to obtain a frequency shift of the carrier wave which isv substantially proportional to si a V0117- age intensity at all frequencies, it is necessary to modify the transmission of signals from the microphone I6 through the amplifier 54. To obtain in the apparatus of Fig. 1 a frequency shift of the carrier wavewhich is proportional to signal intensity at any frequency, it is necessary to attenuate in the amplifier 5L signals from the microphone IS an amount proportional to signal frequency. Generally this attenuation is made something less than proportional at high signal frequencies, so that high frequency signals are transmitted with somewhat greater intensity than lower frequency signals, the process being commonly called preemphasis, in order to obtain some high frequency noise suppression.

In any case, where signals from the microphone I6 are so attenuated as to produce substantially proportional frequency shift of the carrier wave in apparatus such as thatshown in Fig. 1, the phase shift of the carrier wave produced by low frequency signals is much larger than the phase shift of the carrier waves produced by high frephone lt, even without the feedback through.

the discharge device 8i. Such feedback need not be used for. such high frequency components of the signals.

in Fig. 8 I have shown a low pass filter to be inserted between the diode rectifier device H and the amplifier discharge device 0! for the purpose of minimizing feedback for such. high frequency components. In this figure certain elements are identical with certain elements in the apparatus of Fig. 1, and are marked with like reference numerals. An inductance H0 is connected between the control electrode 84 of the device 8| and the upper terminal of the resistance I2 where it is connected to the cathode I0 of device II. A condenser III is connected between the control electrode 85 and the lower terminal of the resistancev I2, which is grounded. By its formation the filter including the inductance H0 and the condenser III may be recognized as a low pass filter, and its characteris-' tics may be suitably adjusted by adjusting the values of the inductance I I0 and condenser III so that it transmits voltages which appear across the resistance l2 and have frequencies corresponding to low frequency components of signals from the microphone I8. This filter, so adjusted, strongly attenuates higher frequency voltages which appear across the resistance I2, which voltages correspond to high frequency components of signals from the microphone I6.

By the use of'this low pass filter, possible tendencies of the apparatus to sing or oscillate at certain frequencies are minimized.

The apparatus of Fig. 1 may be somewhat simplifiedif the gain of the discharge device I2 is adjusted over a sufiiciently widefrange. by adjustment of the potential of the screen electrode 48. This simplification is illustrated by Fig. 9, in which only a portion of the apparatus of Fig. 1, thus simplified, is shown. In this figure certain elements are identical with certain elements of the apparatus of Fig. 1, and are marked with like reference numerals. The cathode I0 of discharge device II is connected to the upper terminal .of the inductance 48, that is, to the ungrounded or high frequency potential end of the tuned circuit 43, 44. The anode 74 of the device II is connected serially in order through the resistance 72 and condenser 78 to ground. The condenser I3 is connected in shunt to the resistance I2 to bypass high frequency currents therearound. A point between the resistance 12 and condenser 16 is connected to the positive terminal of source 46. An adjustable tap II2 on resistance '52 is connected through adjustable resistance 58 to the upper terminal of condenser 49 and thus to the screen electrode of device I 2 of Fig. l. 7

Carrier wave potentials appearing on the tuned circuit 43, 44 are impressed across the discharge device II and the load circuit therefor including resistance 12 and condenser I3, so that a continuous potential appears across the resistance I2, the intensity of this potential being proportional to the carrier wave intensity. A continuous current circuit for the diode II extends from the anode 14 through resistance 12, inductance 45 and inductance 43 back to the cathode I0.

Operating current for the screen electrode 48 is transmitted from the positive terminal of source 46 through resistance I2, adjustable tap H2, and resistance 58 to the screen electrode 48. The resistances 50 and 12, which are so connected that the operating current for the screen electrode 48 flows therethrough, must be 50 adjusted that, when no signal voltages are transmitted from the microphone I8, the screen electrode 48 is at the desired operating potential, so that discharge device I2 transmits the carrier wave to the tuned circuit 43, 44 in desired intensity.

Since the operating current for the screen electrode 48 flows through the resistance 12, it produces a unidirectional voltage thereacross, of such polarity as 'to prevent the discharge device "H from conducting and from rectifying carrier wave potentials, unless such potentials exceed in peak.

value the unidirectional potential across the resistance 12 caused by the flow'of operating current for screen electrode 48 therethrough. It is, therefore, necessary that the apparatus be arranged so that carrier wave potentials on the tuned circuit 43, 44 at all times exceed in peak intensity the unidirectional potential produced across resistance I2 by the flow of operating current for screen electrode 48 therethrough.

When the intensityof carrier wave potentials across the tuned circuit 43, 44 is varied as the phase of the carrier wave is varied in accordance with modulating signal potentials from microphone IS, the rectified unidirectional voltage across resistance 12 varies, so that the operating voltage of screen electrode 48 is reduced, there- 'by reducing the gain of discharge device I 2, when the carrier wave potentials on tuned circuit 48,44 increase. This result is the same as that obtained by the apparatus as shown in Fig. l, and, provided thatthe gain-of discharge device l2 may be sufficiently controlled by changes of operating potentialof the screen electrode 48, a suitable amount of rectified unidirectional voltage may be taken from the resistance 12 by adjustment of theadjustable tap II2 thereof "to produce the desired amount of feedback, in the same way as for the apparatus of Fig. 1. It must be remembered in connection with the apparatus of Fig. 9, that adjustment of the resistances I2 and 50 must be made simultaneously and oppositely in order to maintain the desired gain of the discharge device I2 wyenno signal is present.

In Fig. there is illustrated an additional arrangement which may be substituted in the apparatus illustrated in Fig. 1 to influence the control electrode 48 in response to the intensity of carrier Waves on the tuned circuit 43, 44. In this figure certain elements are identical with certain elements of the apparatus of Fig. l, and are marked with like reference numerals. The electron discharge amplifier device 8|, instead of being connected as a continuous potential amplifier as in Fig. 1, is capacity coupled so as to transmit alternating potentials down to very low frequencies. The cathode 80 of device 8| is connected to ground through a resistance I 2!] in shunt to which a bypassing condenser I2I is connected. The control electrode 84 is connected through a coupling condenser I22 to a point between the cathode 78 of diode rectifier II and resistance I2. The control electrode 84 is connected through a suitable resistance I23 to ground.

The device illustrated in Fig. 10 operates in much the same manner as that in Fig. 1, although it is preferred to use the device of Fig. l. The reason is that the correcting potentials transferred to the screen electrode 48 of device I2 through the amplifier device M are essentially.

unidirectional pulses. The first few pulses may be transferred properly through the amplifier of Fig. 10 when transmission of signals is initiated, but as the condenser I22 charges up, the amplifier device 8I tends alternately to increase and decrease the intensity of that carrier. wave com-. ponent transmitted through device I2 above and below the Value transmitted therethrough in the absence of a signal, instead of only reducing the value of that carrier wave component. However, reasonably good results may be attained with the apparatus shown in Fig. 10, even though a certain amount of amplified modulation of the carrier wave at syllable frequency results, and produces a slight dynamic modulation non-linearity.

Normally, this is of little consequence.

as 60 in either direction. Phaseshifts of the carrier wave which are substantially greater than this may be utilized if non-linearities no greater than those in previous systems be tolerated. With this high degree of linearity over such a large phase shift of the carrier wave, it is quite feasible to produce an excellent frequency modulated carrier wave without the enormous amounts of frequency multiplication heretofore necessary.

While I have shown and described a particular embodiment of my invention, it will be obvious to those skilled in the art that changes and modifications may be made without departing from my invention in its broader aspects and I, therefore, aim in the appended claims to cover all such changes and modifications as fall within the true spirit and scope of my invention.

What I claim as new and desire to Letters Patent of the United States is:

1. In a system for modulating the phase of a carrier wave in linear relationship in accordance with a desired signal, the combination of means for modulating the time phase of said wave in accordance with a function of said signal, said means being efl'eetive to change the intensity of said wave as the time phase thereof is modulated, and means responsive to said change in intensity secure by of said carrier wave to modulate the time phase of said carrier wave further by such an amount that said time phase varies substantially according to a linear function of the intensity of said signal.

2. In asystem for modulating the phase of a carrier-wave in linear relationship in accordance with the desired signal, the combination of m'eans for modulating the time phase of said wave in accordance with a function of said signal over a time approaching the time of a half cycle of said wave, said means being effective to increase the intensity of said wave as the time phase thereof is advanced or retarded and also being effective to modulate said time phase less than proportionately to the intensity of said signal, and means responsive to increased intensity of said carrier wave to increase said modulation of the time phase of said carrier wave to a linear relationship with the intensity of said signal.

-3. In a system for modulating the phase of a carrier wave in linear relationship in accordance with a desired signal, the combination of means for generating two components of such a carrier wave, the time phase between said two components being substantially greater than zero and substantially less than one-half cycle, means for modulating the intensity of one of said components in accordance with such desired signal, means for combining said components after such modulation of one to produce a carrier wave whose phase is shifted in response to such desired signal, and means responsive to increases in the amplitude of such combined components for reducing the intensity of the other of said com-'- ponenm.

4. In a system for modulating the phase of a carrier wave in linear relationship in accordance with the desired signal, the combination of means for generating two components of such a carrier wave, the time phase between said two components being substantially greater than zero and substantially less than a half cycle, mean for modulating the intensity of one of said components in accordance with such desired signal, means for combining said components after such modulation of one to produce a carrier wave whose phase is shifted in response to such desired signal, means for detecting variations in intensity of such combined components for generating a continuous potential varying in accordance with a function of said signal, and means for reducing the intensity of the other of said components in accordance with said continuous potential.

5. In a system for modulating the phase of a carrier wave in linear relationship in accordance with the desired signal, the combination of mean for modulating the time phase of said wave in accordance with a function -of said signal, said means being effective to change the intensity of said wave as the time phase thereof is modulated, means for detecting changes in intensity of said wave for generating a continuous potential varying in accordance with a function of said signal, and means comprising an amplifier for said continuous potentials for modulating further the time phase of said carrier wave in accordance with said continuous potential in such amount that said time phase varies substantially according to a linear function of the intensity of said $1811 8. a system for modulating the phase of a carrier wave in linear relationship in accordance with the desired signal, the combination of means for modulating the time phase of said wave in accordance with the function of said signal, said means being efl'ective to change the intensity of said wave as the time phase thereof is modulated, means responsive to changed intensity of said carrier wave to modulate the time phaseof said carrier wave further in such amount that said time phase varies substantially according to a linear function of the intensity of said signal, and means'for disabling said last means whenever the intensity and time phase of said wave are modulated by said first means in accordance with high frequency components of said signal.

7. In a system for modulating the phase of a carrier wave in linear relationship in accordance with a desired signal, the combination of means comprising a pair of electron discharge amplifier devices for generating separately two components ofsuch a carrier wave, the time phase between said two components being substantially greater than zero and substantially less than a half cycle, means for varying the gain of a first amplifier device to modulate the intensity of one of said components in accordance with such desired signal, means for combining said components after such modulation of one to produce a carrier wave whose phase is shifted in response to such desired signal, means for detecting increases in intensity of said combined components for generating a continuous potential varying in accordance with a function of said signal, and means for impressing said continuous potential upon an electrode of the other amplifier device to vary its gain in such sense as to reduce the intensity of the other of said components in accordance with said continuous potential in such amount as to cause modulation of the time phase of said carrier wave to vary substantially accordcordance with a function of said signal, and means for further modulating the time phase of said carrier wave in accordance with said continuous potential by such amounts and in such sense. that said time phase varies substantially according to a linear function. of the intensity of said signal.

9. In a system for modulating the phase of a carrier wave in linear relationship in accordance with a desired signal, the combination of means 'for generating three components of such a carrier wave, two of said components being oppositely phased and the third being different by a quarter cycle in time phase from the other two, means for modulating oppositely the intensities of said other two components, means for combining said modulated components with said third component to produce a carrier wave whose phase is shifted less than proportionately to said signal, and means responsive to variations in intensity of said combined components for modulating the intensity of said third component in such sense and such amount as to produce a-resultant phase shift. of said combined component substantially proportional to the intensity of said signal,

. -WILLIAM F. GOETIER.

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