US2315442A - Negative feedback detector - Google Patents

Description

J. W. MCRAE NEGATIVE FEEDBACK DETECTORS March 3o, 1943.

Filed Feb. 11, 1942 3 Sheets-Sheet 1 3% FIG.l 3

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/NVENTOR J. W MC RAE ATTORNEY March-30,1943. J. w. MORA: Y2,315,442

NEGATIVE FEEDBACK DETEcTons ATrQRA/Ey March 30, 1943. J. w, McRAE 2,315,442

NEGATIVE FEEDBACK DETECTORS Filed Feb. 11. 1942 s sheets-sheet s SIGNAL OUTPUT SOURCE 0F DOUBLE B. NODULTED MODULATED CARRIER SIGNAL OUTPUT /NVENTQR J W MCR/1E Arrow/Ey l Patented Mar. 30, 1943 UNITED STATES Para orrics l Telephone Laboratories,

Incorporated, New

York, N. Y., a corporation of New York Application February 11, 1942, Serial No. 430,383

(Cl. Z50-27) 8 Claims.

This invention relates to modulation, especially detection. Y

An object of the invention is to increase fidelity of detection or demodulation.

In one specific aspect of the invention an amplifier operating Vat envelope frequencies and having a large amountof negative feedback is so associated with a self-biasing detector that the feedback not only reduces distortion and phase shift between the detectorv bias voltage and the amplifier output but so modifies the apparent de-r tector load impedance as to increase fidelity of detection. Y

Other objectsand aspects of the invention will be apparent'from the following description and claims. A

Fig.` 1 is a detector circuit diagram for facilitating explanation of the invention; y

Fig. 1A is the envelopeA frequency equivalent circuit corresponding to Figfl;

Fig. 2 shows a detector circuit embodying a form of the invention; Y

FigfSshows a modified formV of circuit with means for feeding backenvelope frequency currents to the exclusion of directl current;

Fig. 4 shows a generalized form of detector circuit from which circuits such as those of Figs. 2 and 3 canbe derived;

Figs. 5, 5A and 5B show three modified forms of detector circuits employing detector tubes having in addition to the anode and cathode at least one control element; Fig. 5C shows a modification of a portion of the circuit of Fig. 5B;

i Fig. 6 showsa circuit adapted for modulation of a carrier wave orjdemodulation of a modulated wave; and Y Y Fig. 7 shows an envelope frequency feedback system with the feedback path including a detector circuit generally similar to the detector circuit of Fig. 3.

A fundamental circuit for a self-biasing diode detector is shown in Fig. l, wherein I is a tuned radio frequency input circuit or circuit to which the signal modulated carrier waves to be demodulated are applied, 2 is a diode having inherent plate-cathode capacity Cd, R is a load resistance across which the detected signal voltage is desired, and Cc is a blocking condenser for the detected signal but a lowv or negligible impedance for the input waves. The signal to distortionl ratioVv of such detectors is limited in two ways.

Firs t there is the non-linearity inherent in theV diode characteristic and, second, there is the effect of the frequency characteristic of the load 4impedance Z, which is shown in its simplest form in Fig. 1A wherein E is the diode bias voltage and C=Cc-l-Cd. In order to reduce the effect of diode non-linearity it is desirable to increase the absolute value of Z. However, because of the presence of capacities Ce and Cs, such an increase in absolute value of Z tends to result in a decrease in what may be called the critical frequency of Z, i. e. the frequency at which Z has its shunt reactance equal to its shunt resistance. Where deep modulation must be handled at relatively high `envelope frequencies, excessive reduction of this critical frequency tends to result in increased distortion. For thel detection of broadcast signals, where the modulation at high audio frequencies is very small, no difficulty is ordinarily encountered in securing sufficiently high values of Z. In other applications it may be found that Z cannot be made sufficiently high to overcome diode non-linearity without making its critical frequency too low. An example of this kind of application .is the beta-circuit demodulator V(i. e. the demodulator in the envelope feedback circuit) in multi-plex radio transmitter circuits employing envelope feedback, (for instance circuits of the general type of the circuit of Fig. 7 described hereinafter). Additional examples may be found in some forms of. measuring equipment and also possibly in television receivers.

vThis difficulty can be overcome in many cases with the help of a negative feedbackamplier responding tofrequencies in the envelope of the incoming signal, as for example, amplifier A in Fig. 2, wherein the impedance Z1 across the grid and cathode of the amplifier comprises resistance R and radio frequency by-pass capacity C'. In discussing operation of this circuit, only the direct and envelope frequency components of the diode current, and the voltages set up by these currents, need be considered. The diode current id (constituted by these current components) owing through the impedance Z1 across the grid and cathode of the amplifier A, sets up across Z1 a voltage E1=z`dZ1 and this voltage sets upa voltage Eo across the output terminals of the amplifier, tending to bias the diode to cutoff'. The resultant diode biasvoltage vis E=E1+Eo. Designating the amplification of the amplifier,

andthe apparent diode load impedance, which is defined as the ratio of the diode bias voltage to diode current (averaged over a number of radio frequency cycles), is

Z=f=z1 f1+o 2 If A is large the apparent diode load impedance is therefore approximately equal to the impedance Z1 multiplied by the amplification of the amplifier and the output voltage Ec is almost exactly in phase with and slightly less than the diode bias voltage. If, further, A is real and constant over the required band of envelope frequencies, Z will have the same frequency characteristics as Z1, and since Z1 may be made much smaller than Z, it may be made to have much more desirable characteristics than a physical impedance of the magnitude of Z. By taking ad- Vantage of this fact, distortion in the diode may be kept small, even for signals containing a wide band of envelope frequencies.

The negative bias on the diode, tending to cut it off, is obtained from the output of the amplifier rather than directly from the fiow of current through a physical impedance of magnitude Z. Thus the diode current, which is only required to control the input voltage of the amplifier, may be considerably reduced from the same diode bias voltage, and since the apparent diode load impedance is the ratio of diode bias to diode current this reduction is equivalent to an increase in the apparent diode load impedance Z. More important, the fact that this increase in impedance is obtained by the action of a feedback amplifier means that the variation of Z with frequency may be made more satisfactory than in the case of a purely passive physical impedance. In this way, distortion generated in the diode in many cases may be made considerably smaller than in the simple diode circuits (i. e., the diode circuits without the feedback amplifier). The circuit configuration results in a large amount of feedback around the amplifier. In addition to its effect upon the apparent diode load impedance, this feedback tends to reduce distortion generated in the amplifier itself. Thus, a relatively large detected output, with relativelyvlow distortion, may be obtained with only a small input. Besides the resultant high detector input impedance, this means that sufciently low distortion may often be obtained with smaller diodes than Would normally be required. At ultra-high. carrier frequencies this is sometimes an important advantage.

Because of the presence of feedback, the envelope frequency circuits should be designed to pass a frequency band somewhat wider than the required envelope band, in order to facilitate proper shaping of the Inu-beta loop transmission characteristic as disclosed in II. W. Bode Patent 2,123,178, July 12, 1938. Therefore, to avoid difficulties in keeping radio frequencies out of the amplifier, it is desirable to have the highest envelope frequency low compared to the carrier frequency.

The amplifier may have any desired number of stages', provided the phase of the feedback is maintained correct for the desired negative feedback action. In the envelope frequency band the phase-shift through the amplifier is 180 degrees. Although the amplifier may be capable of responding to direct as well as alternating currents and voltages, still more desirable operation, (in particular, even more desirable impedance characteristics), may be obtained by a modified circuit that makes use of an amplifier which does not transmit direct current but is responsive only to alternating voltage, as for example, the circuit of Fig. 3.

In this circuit of Fig. 3, condensers CB are blocking condensers for direct current. A radiofrequency choke 3 isolates the plate of the diode 2 from the envelope-frequency amplifier A at radio frequencies, to prevent overloading the amplifier. Re is a grid leak resistor, Rp a plate resistor, R2 a coupling resistor and RL, a load resistor for the amplifier. There is a phase reversal in transmission at envelope frequencies through the amplifier. C1 is a radio-frequency by-pass condenser. The average diode bias corresponding to the carrier is set up'across the resistors Ri and R2 in series, the amplifier being isolated from this voltage by the blocking condensers CB. On the other hand, the apparent impedance presented to the alternating components of envelope frequency in the diode current is still given by Equation 2. The effect of the blocking condensers must of course be considered in computing the amplification A of the amplifier A and, except at the lowest en velope frequencies, Zi now includes the effect of the grid resistor Rc at the grid of the amplifier tube, (or at the grid of the first amplifier tube Where the amplifier is a multistage amplifier).

This arrangement makes it possible to select the direct and alternating components of the apparent diode load impedance independently. In this way the occurrence of the most serious type of distortion in the diode can be avoided. This type of distortion is often known as nontracking distortion since it occurs when the envelope of the signal is decreasing at too great a rate for the bias voltage to follow or track When the diode load impedance consists of a simple resistance-capacitance parallel combination, the criterion for the occurrence of this kind of distortion may be simply related to the time constant of the load impedance. However, in more complicated cases, such as those under consideration here, another Vapproach seems desirable.

Non-tracking distortion occurs when the al ternating component of diode current becomes greater than the direct, or carrier component. In that case, if there were to be no distortion, the diode current would have to become negative during a portion of the envelope cycle. Since this is impossible, the result follows that the negative peak of the diode alternating current is cut off and during this interval the load is isolated from the incoming signal. Thus the output voltage no longer follows the envelope but simply decreases in transient fashion.

To avoid this kind of distortion, the peak magnitude of the ratio between envelope frequency components and the detected carrier component of diode current must be less than unity. For single tone modulation this criterion may be written where 7c is the modulation factor for the incoming signal voltage, R is the resistance presented to the direct (i. e., carrier) component of diode current, and Z is the apparent diode load impedance at the frequency of the envelope of the signal. Now Z usually falls off with increasing frequency. VThus a usually suflicientV condition for the avoidance of non-tracking distortion, even for complex modulation, is obtained by setting up the above criterion, using the maximum total modulation factor and the maximum frequency in the envelope band. In many cases, this condition is much too severe and may be relaxed considerably. However it may be taken as the basis for the following discussion of the design procedure.

The first step in this procedure is to set the value for R (equal to Ri-l-Rz in Fig. 3) sufficiently high to give the required low distortion due to diode non-linearities. Then the impedance Z1 is decided upon, the capacitance being large enough to give the required radio frequency by-passing and the resistance then being adjusted to give a reasonably at characteristic over the required band of envelope frequencies. The amplifier gain necessary to raise the apparent diode load impedance to a value approximating Rrnay then be found. Since the amplification will, in general, not be flat over the entire band, it will -be necessary to use Relation 2 in order to find the characteristics of Z. At the highest frequency in the band, Criterion 3 will give a minimum permissible value of Z, in terms of R and the characteristics of the expected signal. .The aim of the design should be to make Z exceed this minimum as little as posible, consistent with necessary allowances for loss of gain in the amplifier as the tubes age. In addition, Z should be as fiat as posible, so as not to exceed the value of R greatly at the low vfrequency end of the required band of envelope frequencies. Such excessive valuesv of Z would introduce a type of distortion not as yet mentioned. However, experience indicates that even with ratios of Z to R as great as 3 this type of distortion is not important. Y

If more than a single stage is required in the amplifier, precautions must be taken to avoid singing around the feedback loop. The usual design methods may of course be employed for this purpose, and it will usually be found satisfactory to consider the diode as -a zero impedance generator in making such calculations.

Means should also be provided to keep excessive radio frequency voltages out of the amplifier, for

'example filtering circuits such as shown in Fig. 7

described hereinafter. This is particularly true if the amplifier tubes are operating nearly to capacity under the given envelope frequency conditions. Then a relatively Vsmall radio frequency voltage in the amplifier may considerably increase the distortion. The effect of the necessary filter.- ing .circuits on the amplifier vfeedback loop must, of course, be considered in the stability calculations.

v 'A specific design example' of a detector of the type shown in Fig. 3 was one for use in a l2-channel multiplex radio telephone transmitter employing envelope-l feedback and operating on a carrier frequency of 141 megacycles. A single 6L6 tube was used as the amplifier. This detector was to be used (in the fashion indicated in Fig. 7 described hereinafter) to obtain a sample of the output for feeding back through the beta circuit to the input of the signal amplifier, and therefore it was important that it should introduce negligible distortion throughout the envelope frequency band and that its overall phase shift should be negligibly small up to frequencies of the order of a mega-cycle. The design was made to permit operation over an envelope frequency band from l2 to 108 kilocycles. v

The value of R was set at 550,000 ohms, and values computed for curves (not shown) of A,

A A l Z1, and Z. At 108 kilocycles, the absolute value of Z was about 4 decibels higher than R (that is, about 870,000 ohms). There was consequently a large safety factor against non-tracking, even for per cent modulation at 108 kilocycles. In fact, this safety factor was excessive in this application and could have been reduced by increasing R to 870,000 ohms or more. At the lower frequencies, Z increased to a maximum value of about 1.5 megohms. On the other hand, at very high frequencies, Z fell on" rather rapidly, the slope being nearly 12 decibels per octave over a portion of the range. In this region the real part of Z was negative. Nevertheless, because of the low internal impedance of the diode, the circuit was found to be stable. In addition, despite the somewhat low value of R, the distortion was found to be sufficiently low.

Fig. 4 shows a general form of detector circuit of which the circuits of Figs. 2 and 3 can be considered specic types. In Fig. 4 the amplifier is shown as having a plurality of sections A1 and A2 in tandem, (each of any desired number of stages), with a pair of output terminals 4 between two of the sections. An attenuating or transmission control network c may be provided between the amplifier output terminals 5 and the diode. Equalization (or control of the attenuation versus frequency and phaseshift versus frequency characteristics of the feedback path) to prevent singing may be provided, at least partially, in this attenuating network, and this network may include condensers (indicated in dotted lines) preventing feedback of direct current. The presence of this network provides a choice as to the outputfi. load) connection for the syst-em. For example, the output of the system may be taken off at the diode (from terminals S, across resistor R2', or at the amplifier output (terminals 5), or at some intermediate point on the attenuating network or at terminals For each of these conditions analysis of the operation of the system would take the same general forni as that presented in connection with the preceding figures of the drawings.

Figs. 5, 5A and 5B show examples of ways in which multielement tubes, that is, tubes having in addition to an anode and cathode at least one discharge control element, may be used as the detector elements cooperating with the amplifier devices. As in the case of the amplier elements of Figs. 2 to 4, the amplifiers A of Figs. 5, 5A and 5B may be linear devices, the detector elements being the only necessarily non-linear elements of the systems. The multielement detector tubes are shown at 2 and may be, for example, screen grid tubes or suppressor grid pentodes. The ampliiiers A may be assisted to some extent in the amplication function by the gain (to envelope frequency voltages) existing in the detector tube. However since this tube is operated around its cut-off, its gain will probably be very small or may even be negative. In addition the cut-off in multielement tubes ordinarily is not as sharp as in a diode, and because of this greater nonlinearity it may be necessary to present a higher equivalent load impedance to the detector than in the case of a diode.

'Ihe operation of these circuits is similar in general to those already described. In each case,

the principal function of the feedback at envelope frequencies is to facilitate the maintenance of a high load impedance for the detector and to reduce the possibility of non-tracking distortion. In addition, much greater output power may be obtained from the amplifier-detector combination for a given amount of distortion than could be obtained from a simple detector. Finally, the feedback tends to reduce distortion in the amplier in addition to its effect on the apparent detector load impedance.

One feature of these circuits which is not characteristic of the circuits using diodes is the existence of what may be an appreciable Zero signal current in the detector tube. This component. of current could -be set up in the diode of any of the preceding circuits by the application of a positive direct current bias to the diode anode. Such a bias is known to reduce the possibility of non-tracking in a simple diode detector but it has the disadvantage of reducing the diode input impedance for small signals. The circuits of Figs. 5. 5A and 5B do not have this disadvantage and the input impedance presented to the signal will be considerably higher than that for any type of detector using a diode. This is a result of the use of a multielement tube as the detector, and of the fact that the detector may be operated with no grid current. At the same time, as will be shown later, the no-signal current in the platecathode circuit of the detector tube is similar to a positive bias on a diode in reducing the probability of non-tracking distortion.

Consider first the circuit of Fig. 5, which is drawn for the case in which amplifier A is a direct-current amplier. With no signal, an equilibrium condition is established between the plate-cathode current in the detector tube 2' and the direct current bias voltage developed across cathode impedance Ze, which may be, for example, a resistance 'i and a carrier or radio frequency by-pass condenser 8 connected parallel and having its impedance value Z@ and its resistance value Re.

The detector plate current flowing through plate circuit impedance Zp, which may be a resistor or other suitable impedance of impedance value Zp and resistance value Rp, sets up a voltage between amplifier input terminals II and I2 which, after amplicatio-n, appears across terminals I3 and I4 and thus across cathode impedance Ze. The sense of this amplified voltage is such as to drive tube 2 toward cut-off and thus only a small plate current is required to set up the 11o-signal bias. Let impedance Z0' be defined as the ratio Ec/Ic, where E@ is the bias developed across ZC and Ic is the detector tube plate-cathode current. Then this apparent impedance will be greater than the actual value of Zc and its magnitude will be a measure of the effectiveness of the feedback in reducing the necessary detector current. The magnitude of Ze may be calculated in terms of Zp, Ze, A, i and an additional impedance Z0. Impedance Z is the impedance measured between terminals I3 and I4, with the connections to Ze broken at XX. During this measurement the output load I must be left connected. The apparent impedance Ze' is then:

where Ai is the voltage gain of the ampliiierattenuator combination measured between input terminals Il and I2 and output terminals I3 and I4, with the load connected and with the leads to Ze broken at XX. The factor in parentheses evidently multiplies the Value of Ze up to its apparent value Ze. This expression is evidently valid at any frequency and reduces to the form at zero frequency.

Having established the value of apparent im-` pedance in the cathode lead of the detector tube, the analysis of the system becomes relatively simple. When a steady carrier is applied through input circuit I an additional direct current component of current flows through tube 2, of sufcient magnitude to develop across Ze a voltage E very nearly equal to the radio frequency voltage between grid and cathode of tube 2. (Ze is arranged to include a path, as for example the path through condenser 8, of low impedance at the carrier frequency so that practically the entire applied voltage appears between the grid and cathode of the tube.) Thus there are now two components of current through the detector tube, the no-signal component and the component due to the signal carrier. These two components may be written as E lod-F where Eb is the volta-ge developed across Ze, or

between grid and cathode of tube 2', in the aby sence of signal.

When the carrier is modulated sinusoidally with a modulation factor 7c, the voltage across Ze must vary proportionally and a third component of current therefore must flow in tube 2. This component is of envelope frequency and has an amplitude comparing this criterion with that for the diode, it will be observed that the existence of the factor E/Eb-l-E tends to reduce the diiculty of preventing non-tracking distortion.

The circuit shown could be modified in several ways without changing its method of operation. For example, attenuator i might be omitted, or the output obtained from terminals I3 and I4 instead of as shown.

More important modications would be required in order to make the amplifier responsive only to alternating voltages. In that case, blocking condensers (not shown) could be inserted at the points marked II and I3. Then the direct current path through Ze should be made to have a high resistance Re, comparable in magnitude to Zo', as computed for envelope frequencies in the required range. This computation should be made from the same formulas as for the direct current amplifier. The non-tracking criterion for this case is rn. E Z E11-l E- From this expression, it will be evident that Re should be about equal to lZc'l. It should not exceed IZcI by more than, say, a factor of 2.

Radio frequency chokes IS and I1 prevent the incoming radio frequency waves from bein-g fed back or from reaching the amplifier A.

In Fig. 5A, the amplifier is effective in reducing the current through tube 2 in much the same way as in Fig. 5. The most important difference between the two circuits is that the envelope frequency bias voltage set up by the signal appears on the grid in Fig. 5A and on the cathode in Fig. 5. -Since Fig. 5A is drawnl for the case in which the amplifier is isolated from direct voltages, the direct current biases due to the rio-signal and carrier components of current are developed on the cathode, across resistance R2.

Blocking condensers CB isolate the amplifier from direct currents, and condenser C2 by-passes resistance R2. These three condensers all have low reactanc-e for envelope frequencies and `C2 must also have a low reactance at radio frequencies. Condenser CR must have a high reactance for envelope frequencies and a low reactance for radio frequencies. The resistance Rg is -a high resistance grid leak used to maintain the -grid of tube 2 at a definite average voltage C-. .The zero-signal current may be adjusted by Varying the Voltage C-. Since resistance R2 is large, the necessary zero signal current tube 2' is small.

When a steady carrier is applied to tube 2 through tuned circuit I, the current through R2 increases an amount sufficient to give an increase in voltage across R2 nearly equal to the magnitude E of thecarrier voltage. The two direct current components of detector current are therefore E E reife where Eb is the zero signal bias across R2.

The envelope frequency component of detector current must be of such magnitude as to develop a voltage of magnitude almost equal to lcE on the grid. This requires a current Ip through Zp and tube 2 of magnitude From this expression, it is evident that an apparent transfer impedance Z may be defined such that Z=l.cE/Ip where lsE is the magnitude of the input modulation envelope and Ip is the envelope frequency component of detector current. In that case Z'=ZpA1 and the criterion for no non-tracking is sistance R. With these definitions made, the criterion for no non-tracking becomes @gil [Z Eb-i-E The circuit of Fig. 5B differs from both the preceding circuits in that the amplifier input voltage is obtained from an impedance in the cathode lead, rather than the plate lead, of tube 2'. Otherwise, it most nearly resembles Fig. 5A. However, the amplifier in Fig. 5B must have essentially degree phase shift throughout the envelope frequency band, as opposed to Zero degrees in Fig. 5A.

In the form shown, the circuit is adapted for a direct current amplifier. In that case, the zero-signal current through tube 2 depends on the characteristics of the tube 2 and of amplifier A and attenuator i. However, the entire detector tube current flows through impedance Zc and the bias voltage Eb which will appear in the no non-tracking criterion is the voltage appearing across Ze in the absence of signal.

The apparent impedance Ze' which limits the fiow of detector tube plate-cathode current is 'Ilhis reduces to Re' at zero frequency. Following the same type of reasoning as outlined above, the criterion for no non-tracking is where al1 the symbols have already been defined. r

The circuit of Fig. 5B may be converted for the use of an amplifier isolated from direct voltages by following the general procedure already described. Blocking condensers, as shown at CB in Fig. 5A should be inserted at the points marked B in Fig. 5B and a grid leak resistor and C-supply arranged to hold the grid of tube 2 in Fig. 5B-at Vthe desired steady bias Voltage as resistor Rg and the C-supply in Fig. 5A bias the grid of tube 2' in Fig. 5A. The Vdirect current path in Ze should be made a high resistance of magnitude approximately equal to Ze. Apparent impedance Ze should be calculated in the same way as before, and the criterion for no non-tracking becomes i cRc E' WIEN-E51 where Re is the resistance of the direct current path in Ze.

Fig. 5C shows a form of cathode-lead network suitable as la substitute for that shown between point P andY ground in Fig. 5B when the amplier is to be an alternating current amplifier t(i. e., when the amplifier is to be isolated for direct current). In Fig. 5C the cathode lead impedance comprises resistances Y 9 and Ill in series, with a radio-frequency by-pass condenser I8 `(of high reactance at the envelope frequencies) connected across both 9 and l0, and with The magnitude of the age Eo is given by Ec=(Is-I1)Z (2) where I1 is the carrier frequency component of the diode current with switch 22 closed and switch 2| open. Thus,

and under these conditions the average value of diode current will be carrier frequency voltwhere En is the voltage of bias source 40 and R is the resistance of resistance 28, the factor MA having already been defined. This relation expresses the fact that there are two direct current components of diode current, the rst being that sufficient to build up a negative bias on the diode approximately equal to the magnitude of voltage Ee, while the second is the component of current due to the bias source 40. From these four equations, inserting the requirement that Eo must be at least twice Ee, the following criterion for the adjustment of resistance R and impedance Z may be established: i

MR l-ZTE l (5) Observance of this criterion will insure that proper operating conditions are obtained for the case of an unmodulated carrier. In order to determine the conditions necessary for operation with switch 2| closed and a sinusoidal modulating voltage supplied at the terminals Vof the bias source, it will be necessary to consider the magnitude of the envelope frequency component of current through the diode. Using double primes to represent quantities evaluated at envelope or side band frequencies, this current becomes 121%? 4 6) where lc is the modulation factor for the resultant modulated frequency voltage appearing across the diode. The condition necessary for no nontracking distortion may therefore be written 5- IZ//I EblZ//l 1;;- MkR nMkR l (7) which reduces to MICR E@ f wrm-1 y (8)..

I-t will be observed that this criterion is very similar to those developed for the preceding circuits. The factor M will ble approximately one-half and its actual value may be determined by measurement of the fundamental component of current to the diode compared with the resultant average current through resistor 2|3.l In making this measurement, it will be advisable to reduce the voltage Et from the bias source 40 to zero.

If `switch 22 is open while the system lis operating in accordance with the above description, the signal frequency voltage appearing across resistor 23 may be used to monitor the operation of the system. Resistor 23 should then have a magnitude as small as is consistent with the desired sensitivity of the monitor.

With switch 2| open, and switch 24 operated to replace source 20 by the double sideband modulation generator 29 or source of double sideband modulated signal, the system may be operated as a detector. In that case a condenser M3 having a low reactance at signal frequencies should be connected in parallel with resistance 28 as by switch 47. A steady bias will then appear across this condenser and resistance 28, approximately equal to the carrier value of the voltage appearing across the diode. When this carrier is modulated at signal frequency, an additional current, varying in magnitude at signal frequency, will flow through the diode and impedance 32. This current, consisting of a series of pulses at carrier frequency, will tend to hold the diode radiofrequency voltage constant through the combined action of impedance 32, amplifier A and attenuator ,6. Thus the current through the diode will contain a component whose magnitude is proportional to the modulation on the input signal. If switch 22 is opened, a voltage corresponding to the signal will therefore be developed across resistance 23.

Under these conditions the device will also op'- erate as a limiter. In other words, the radiofrequency voltage delivered at the output of the radio-frequency amplifier will be held nearly constant, despite variations in the voltage delivered from the source.

Fig. 7 shows a detector circuit 50, generally similar to that of Fig. 3, in an envelope frequency negative feedback system comprising radio transmitter 5| (including carrier oscillation generating and modulating means) and signal input amplifier 52 for amplifying signals supplied to its input transformer 53 by signal input circuit 54 and transmitting the amplified signals through connection 5l to the modulator in the radio transmitter to modulate the carrier oscillations generated in the radio transmitter. The modulated carrier wave generated by the combination of the carrier oscillations and the signals in the modulator is transmitted from the radio transmitter to a radio-frequency load circuit 55, for example, an antenna circuit.

The detector circuit 5i] is shown with its input connected across the circuit 55 and its output connected across resistor 56 which is in series with-the secondary winding of the signal input transformer 53 in the input circuit of the signal amplifier 52. Modulated waves, from the modulator in the transmitter 5|, are demodulated in detector 50 and the resulting signal waves are fed back through connection 58 to the input of amplifier'V 52 in phase opposition to signal waves supplied to the amplier from circuit 54. Thus, there is negative feedback around the feedback loop including the modulator in the radio transmitter 5|, the detector 50, and the amplier 52. In other words there is feedback around this loop in the proper phase and amplitude to obtain the improvements in distortion, noise, etc., which accrue from the application of negative feedback.

The amount of this negative feedback around this envelope feedback loop may be large, as for example several times 10 decibels. However, inA envelope-,frequency feedback operation, where a frequency change occurs at the modulator in the Inu-circuit of the feedback loop, a limitation on the amount of improvement obtainable (in envelope distortion, etc.) by large amounts of feedback around the loop is encountered due to the presence of a detector or demodulator; because inasmuch as the demodulator is in the beta-circuit of this feedback loop, any distortion generated in the demodulator is present in the transmitter output. Thus the problem of reducing non-linear distortion at the transmitter is not merely a. matter of obtaining sufficient envelope feedback around this loop, but involves obtaining a demodulator whose distortion is sufficiently low to realize the improvement of transmitter distortion possible with the increased amount of feedback. Moreover, if a usual type of high level linear rectifier were used, it would need to have a large filament emission with small transmit time and Work into a high impedance load, and these requirements would not be compatible with those for the maximum amount of feedback, which are low tube capacitance and low output impedance. However, all these requirements are satisfied by the detector t in the system of Fig. 7. This detector is a circuit which includes a linear rectifier, the diode I, and effectively applies local negative feedback to the diode through linear amplifier A in the general fashion described above in connection With Fig. 3, for example. This circuit allows the use of a small rectifier tube with small transit time, low capacitance and only moderate lament emission working into a low output impedance. This feedback makes the rectifier operate as if it had a high output impedance (load impedance) so that large filament emission i's not required for small distortion. Though Without this feedback a very high physical resistance might be used, it would so increase the time constant that non-tracking in the diode circuit might result. The feedback in the detector may be, for example, several times 10 decibels, and reduces the time constant by a large factor. Thus the fidelity of the detector is improved by local feedback in the demodulator which so modifies the impedance presented to the diode as to maintain its fidelity even when the transmitter is deeply modulated at high envelope frequencies.

In Fig. '7, condensers CB are blocking condensers. Co is a blocking condenser for envelope frequencies. Circuit l comprising parallel connected inductance, capacity and resistance is tuned to the radio carrier frequency, fo, which may be for example of the order of 162 megacycles, the signal being for example a group of 12 carrier telephone messages occupying the frequency band extending from 60 kilocycles to 108 kilocycles. This circuit i acts as a shunt to the signal frequency that is present along with the modulated radio Wave on the modulator plate. It helps to insure that the output will be fed back by means of modulation and demodulation and not directly at the signal frequency, its inductance presenting a very low impedance at the envelope frequency so as to minimize any distortion pick-up of envelope voltage from the modulating amplifier. Also, it increases the input impedance of the detector by anti-resonating the tube and the stray capacities.

R1 and C1 are the resistance and condenser across which the envelope voltage is developed that is to be amplified by A'. The phase of the envelope voltage developed in the plate circuit of A is such as to produce the desired local feedback for the diode, the plate of A being coupled to the plate of the diode by connection which includes a blocking condenser CB and a, circuit 3 tuned to fu. The circuit 3 isolates the plate of the diode from the envelope frequency amplifier at radio frequencies. It prevents the radio frequency from getting on the plate of amplifier A and helps prevent it from getting through to the grid of the first tube of the signal-frequency amplifier 52. Radio-frequency choke 6I and series-tuned circuit 62 which is tuned to fo serve as a filter for the grid of amplifier A', minimizing the radio-frequency carrier voltage which otherwise might appear on the grid of A and cause overloading. Radio-frequency choke 10 further reduces the radio-frequency voltage on the grid of A. Choke B3, anti-resonant to fo, further suppresses the radio-frequency voltage to a value which will not cause distortion on the first signal-frequency amplifier grid. Resistor B4 is a grid bias and local negative feedback resistor for A. Condenser 65 cooperates With resistance Rp to filter the plate current for tube A'. Resistances 66 and 61 form a potential -divider for the screen grid circuit of the tube.

Elements 65, 65 and 68 filter the screen grid current. Resistance R2 corresponds to the resistance R2 in Fig. 3. Network 12 is part of the equalizing network needed in the main feedback loop. It has no significant effect on the operation of the demodulator circuit.

In the case of the detector 5i) as well as others of the detectors described above, an advantageous feature is that it is not necessary that the radio-frequency feeding circuit have low impedance for the modulation frequencies.

What is claimed is:

1. A wave translating system comprising a source of signal modulated carrier Waves, a detector for demodulating said waves, an amplifier responsive to signal energy from said detector, means for transmitting signal energy from said detector to said amplifier, means connecting said source across a portion of the output circuit of said amplifier, and means for reducing the signal current in said detector comprising means for transmitting signal energy from the output circuit of said amplifier to said detector.

2. A demodulating system comprising a source of signal modulated carrier waves, a detectoramplifier circuit comprising a detector and a signal amplifier, said detector-amplifier circuit having an input circuit and an amplifier output circuit, means for connecting said source across said input circuit, and means for applying signal waves from a portion of said output circuit across said input circuit, the phase shift through said amplifier having such value as to produce negative feedback of signal waves from said output circuit across said input circuit.

3. A demodulating system comprising a source of signal modulated carrier waves, a detector, a signal amplifier having an input circuit and an output circuit, means for connecting said source across said detector and said input circuit in series, and signal transmitting means for connecting a portion of said output circuit across said detector and said input circuit in series, the phase shift through said amplier having such value as to produce negative feedback of signal waves from its output circuit to its input circuit through said detector.

4. A demodulating system comprising a source of signal modulated carrier waves, a detector, a detector load circuit including an amplifier for signal Waves having an input circuit and an output circuit, a circuit for transmitting signal Waves including said detector, said input circuit and a portion of said output circuit in series, and means connecting said source across said portion, said amplifier having its phase shift such that it produces negative feedback of signal Waves from its output circuit to its input circuit through said detector and negative biasing voltage for the detector varying in accordance with the signal and tending to reduce signal current flow through the detector resulting from detection of modulated Waves supplied to said detector from said source.

5. A Wave translating system comprising a source of Waves, a rectifier, an amplifier, means transmitting energy from said rectifier to said amplifier, negative feedback means for said amplifier connecting in serial relation the amplier input circuit, a path through said rectifier and a path forming a portion of the amplifier output circuit, and means connecting said source across one of said paths.

6. A demodulating circuit comprising a source of signal modulated carrier waves, a diode, an amplifier responsive to signal Waves, and means for feeding said diode from said source and said amplier from said diode and producing negative feedback of signal Waves in said amplifier, comprising means connecting in serial relation said diode, the input circuit of said amplifier and a branched circuit including in one branch said source and in another branch a portion of the output circuit of said amplifier.

7. A Wave translating system comprising a source of signal modulated carrier waves to be demodulated, a diode detector, an amplifier, means for transmitting signal Waves from said detector to the input circuit of said amplifier, negative feedback means for said amplifier connecting in serial relation the input circuit of said amplifier, a path through said rectifier and a portion of the output circuit of -said amplifier, and means connecting said source across said portion.

8. A Wave translating system comprising a source of signal modulated carrier Waves, a detector comprising an anode, a cathode and a control electrode connected to said source for demodulating said Waves, an amplifier responsive to signal energy from said detector, means transmitting signal energy from said detector to said amplier, and means for producing negative feedback of signal energy from the output circuit of said amplifier to said control electrode and cathode of said detector.

JAMES W. MCRAE.

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