US20230052807A1 - Three-phase inverter control system and three-phase inverter control method - Google Patents

Three-phase inverter control system and three-phase inverter control method Download PDF

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US20230052807A1
US20230052807A1 US17/818,544 US202217818544A US2023052807A1 US 20230052807 A1 US20230052807 A1 US 20230052807A1 US 202217818544 A US202217818544 A US 202217818544A US 2023052807 A1 US2023052807 A1 US 2023052807A1
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voltage
inverter
value
controller
axis
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Huahong Ning
Dengke ZHANG
Hualiang Li
Dawei Zheng
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Santak Electronic Shenzhen Co Ltd
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Santak Electronic Shenzhen Co Ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/493—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode the static converters being arranged for operation in parallel
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53873—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with digital control

Definitions

  • the present invention belongs to the field of inverters, and in particular relates to a control system and a control method for a three-phase inverter.
  • An inverter is a converter that converts DC electric energy of battery, storage battery, and the like into constant-frequency and constant-voltage AC or variable-frequency and variable-voltage AC.
  • An inverter is a core part of power supply equipment such as an uninterruptible power supply (UPS), an AC power frequency converter, or a new energy supply system. Therefore, research on inverters is of great significance to the development of modern industry.
  • UPS uninterruptible power supply
  • AC power frequency converter or a new energy supply system. Therefore, research on inverters is of great significance to the development of modern industry.
  • the quality of an inverter output waveform may deteriorate or even lead to system instability due to the presence of nonlinear factors.
  • feedback control is performed directly by current or voltage, which has slow response speed and poor stability.
  • an objective of the present invention is to overcome the foregoing deficiencies in the prior art, and provide a control system for a three-phase inverter, which comprises an instantaneous value voltage controller and an equivalent effective value voltage controller, wherein the instantaneous value voltage controller is configured to feed back and control an instantaneous value of an inverter output voltage, the equivalent effective value voltage controller is configured to perform an orthogonal decomposition feedback control on an effective value of the inverter output voltage, and wherein the equivalent effective value voltage controller is configured to perform integral compensation respectively on a real-axis voltage and an imaginary-axis voltage of a two-phase rotating coordinate system of the three-phase inverter, and an output of the instantaneous value voltage controller and an output of the equivalent effective value voltage controller are used to obtain the inverter output voltage through a delay stage transfer function and a controlled object transfer function.
  • the equivalent effective value voltage controller is an integral compensator
  • B BB ( B ) B BBB ⁇ B ⁇ 1 B ,
  • B BBBB B is a gain coefficient
  • s is a frequency domain operator
  • the gain coefficient is less than 1.
  • the real-axis voltage is represented as B B
  • the imaginary-axis voltage is represented as B B
  • B B B BBB_BEB ⁇ B BBB
  • B B B BBB_BBB ⁇ B BBB ,
  • B BBB_BBB represents an inverter sine voltage given value
  • B BBB_BBB represents an inverter cosine voltage given value
  • B BBB represents a difference between the inverter sine voltage given value and the inverter output voltage
  • B B B B + ⁇ circumflex over (B) ⁇ B
  • B B B B + ⁇ circumflex over (B) ⁇ B
  • B B B B + ⁇ circumflex over (B) ⁇ B
  • B B is a real-axis voltage average value
  • ⁇ circumflex over (B) ⁇ B is an imaginary-axis voltage average value
  • ⁇ circumflex over (B) ⁇ B is a real-axis voltage disturbance value
  • ⁇ circumflex over (B) ⁇ B is an imaginary-axis voltage disturbance value.
  • BB BB B is a difference between a given inverter voltage effective value and a feedback voltage effective value
  • B is an angular frequency of the inverter output voltage
  • t is time
  • the integral compensator is configured to convert the real-axis voltage B B into a controller real-axis output value B BB , and convert the imaginary-axis voltage B B into a controller imaginary-axis output value B BB , wherein
  • a duty cycle B B of an output of an equivalent real-axis voltage loop is obtained based on the controller real-axis output value B BB
  • a duty cycle B B of an output of an equivalent imaginary-axis voltage loop is obtained based on the controller imaginary-axis output value B BB
  • a duty cycle of an output of the instantaneous value voltage controller B B (B) is B BB
  • the total duty cycle d is used to obtain the inverter output voltage through the delay stage transfer function and the controlled object transfer function.
  • the instantaneous value voltage controller is a hysteresis controller and represented as
  • B B ( B ) B BB ⁇ B + B B + B ,
  • B BB is a gain coefficient of the instantaneous value voltage controller
  • B is a zero point of an instantaneous voltage loop controller
  • B is a pole of the instantaneous voltage loop controller
  • s is the frequency domain operator
  • control system further comprises a coordinate system conversion part which is configured to implement conversion between an BBB coordinate system and an ⁇ coordinate system.
  • the present invention further provides a control method for a three-phase inverter, comprising:
  • a step of performing orthogonal decomposition feedback control on an effective value of the inverter output voltage which comprises performing integral compensation on a real-axis voltage and an imaginary-axis voltage of a two-phase rotating coordinate system of the three-phase inverter;
  • an output of the step of feeding back and controlling an instantaneous value of an inverter output voltage and an output of the step of performing orthogonal decomposition feedback control on an effective value of the inverter output voltage are used to obtain the inverter output voltage through delay stage transfer and controlled object transfer.
  • a gain coefficient of the integral compensation is less than 1.
  • the real-axis voltage is represented as B B
  • the imaginary-axis voltage is represented as B B
  • B B B BBB_BEB ⁇ B BBB
  • B B B BBB_BBB ⁇ B BBB ,
  • B BBB_BBB represents an inverter sine voltage given value
  • B BBB_BBB represents an inverter cosine voltage given value
  • B BBB represents a difference between the inverter sine voltage given value and the inverter output voltage
  • the method further includes converting an BBB coordinate system into an ⁇ coordinate system in a feedback process and converting the ⁇ coordinate system into the BBB coordinate system in an output process.
  • the method further includes a step of adjusting an initial phase angle of the inverter output voltage to zero.
  • the advantages of the present invention lie in that compared with a conventional control system, in an equivalent effective value voltage control loop, a relatively large crossover frequency can be designed, so that sufficiently fast regulating speed is ensured, and there are still sufficient phase and gain margins.
  • an integral controller in the system ensures stability and almost has no net difference, thereby providing excellent performance in dynamic status/stability.
  • FIG. 1 and FIG. 2 respectively show a three-phase four-wire inverter topology and a three-phase three-wire inverter topology
  • FIG. 3 shows a simplified inverter control system used for a three-phase four-wire inverter topology according to a first embodiment of the present invention
  • FIG. 4 shows an equivalent inverter control system according to the first embodiment of the present invention
  • FIG. 5 is an open-loop Bode diagram of an equivalent effective value voltage loop according to the first embodiment of the present invention.
  • FIG. 6 shows an instantaneous value voltage loop in FIG. 4 ;
  • FIG. 7 is an open-loop Bode diagram of the instantaneous value voltage loop in FIG. 4 ;
  • FIG. 8 shows a simplified inverter control system used for a three-phase three-wire inverter topology according to a second embodiment of the present invention
  • FIG. 9 a and FIG. 9 b show simulation results
  • FIG. 10 a and FIG. 10 b show results of test under hardware-in-the-loop (HIL).
  • HIL hardware-in-the-loop
  • FIG. 1 and FIG. 2 respectively show a three-phase four-wire inverter topology and a three-phase three-wire inverter topology.
  • a power supply voltage supplies power to a load through an inverter topology.
  • B BBB represents a bus capacitor of a single side busbar.
  • B BBB represents a voltage of the bus capacitor.
  • Brg_A_X/Brg_A_Y, Brg_B_X/Brg_B_Y, and Brg_C_X/Brg_C_Y respectively represent inverter X and Y bridge arms of phases A, B, and C.
  • B B and B B respectively represent filter inductors of the inverter X bridge arm and Y bridge arm;
  • B B_B and B B_B respectively represent currents of the filter inductors of the inverter X bridge arm and Y bridge arm;
  • B BBB represents an inverter filter capacitor;
  • BBBB and BBBB respectively represent an X bridge arm and a Y bridge arm of a three-phase inverter.
  • a control system and a control method for an inverter of the present invention are provided based on the three-phase four-wire inverter topology shown in FIG. 1 .
  • a control object of the control system and control method in this embodiment is the three-phase four-wire inverter topology shown in FIG. 1 .
  • a phase A is only used as an example for discussion.
  • Those skilled in the art can understand that the cases of phases B and C are similar to the case of the phase A.
  • ABC and abc herein are both used for representing three phases of the three-phase inverter.
  • an inverter sine voltage given value B BBB_BBB and an inverter cosine voltage given value B BBB_BBB are obtained based on the given inverter voltage effective value B BB B_BBB ,
  • B BBB_BBB B BB B_BBB BBB ( BB ),
  • B BBB_BBB B BB B_BBB BBB ( BB )
  • B is an angular frequency of the inverter output voltage
  • t is time.
  • a given value is also referred to as a reference value, that is, an expected value output by an inverter.
  • B BBB is a difference between the inverter sine voltage given value and the feedback voltage
  • B BB B is a difference between the given inverter voltage effective value and the feedback voltage effective value.
  • the feedback voltage is a sine value. Therefore, in the discussion of the present invention, a sine symbol is no longer labeled on a feedback signal.
  • B BBB including the feedback voltage B B is input into a multiplier respectively with the inverter sine voltage given value and the inverter cosine voltage given value to obtain a real-axis voltage B B and an imaginary-axis voltage B B
  • B B B BBB_BBB ⁇ B BBB
  • B B B BBB_BBB ⁇ B BBB (1).
  • pu represents a relative unit system, and is a common term in the field of engineering.
  • an inverter voltage standard value is 230 V, represented by 1 pu. In this case, (1+15%)*230 V is 1.15 pu.
  • Formula (2) may be simplified as:
  • the real-axis voltage and the imaginary-axis voltage are further respectively represented as including a constant part (a direct current amount) and a disturbance part (an alternating current amount):
  • B B is a real-axis voltage average value
  • B B is an imaginary-axis voltage average value
  • ⁇ circumflex over (B) ⁇ B is a real-axis voltage disturbance value
  • ⁇ circumflex over (B) ⁇ B is an imaginary-axis voltage disturbance value
  • a real-axis voltage and an imaginary-axis voltage of a two-phase rotating coordinate system of a three-phase inverter are obtained, and the real-axis voltage and the imaginary-axis voltage are respectively written in a form including a direct current amount and an alternating current amount.
  • Compensation control is further performed on the real-axis voltage and the imaginary-axis voltage.
  • an integral compensator also referred to as an integral controller
  • B BB ( B ) B BBB ⁇ B ⁇ 1 B
  • B BB (B) is an effective value voltage controller configured to control a real-axis voltage and an imaginary-axis voltage, and is referred to as an “equivalent effective value voltage controller” in the present invention to differentiate from a conventional effective value voltage controller.
  • B BBB B is a gain coefficient of the controller
  • s is a frequency domain operator.
  • B BB is a controller real-axis output value
  • B BB is a controller imaginary-axis output value
  • B BB ( B BBB ⁇ B ⁇ 1 B ) ⁇ ( B B + B B )
  • B BB ( B BBB ⁇ B ⁇ 1 B ) ⁇ ( B B + B B ) .
  • an inverter voltage effective value after orthogonal decomposition includes a real part B 3B and an imaginary part B BB equal to zero, which is equivalent to an instantaneous effective value voltage control system. Therefore, the foregoing derivation process is proved to be accurate and reasonable.
  • B B is a duty cycle of an output of the instantaneous value voltage controller
  • B B is a duty cycle of an output of an equivalent real-axis voltage loop
  • B B is a duty cycle of an output of an equivalent imaginary-axis voltage loop
  • B BBBBB (B) is a digitally controlled delay stage transfer function
  • B BB (B) is a controlled object transfer function of the voltage loop, that is, a mathematical control formula obtained through a Laplace transform of hardware of the three-phase inverter
  • B BBB is a voltage of a bus capacitor
  • B is an inverter filter inductor
  • B is an inverter filter capacitor
  • B is an equivalent resistance value of a load.
  • B BBB_BBB 1BB ⁇ BBB(BB)
  • B BBB_BBB 1BB ⁇ BBB(BB).
  • the duty cycle of the equivalent real-axis voltage loop output and the duty cycle of the equivalent imaginary-axis voltage loop output may be further obtained.
  • B B ⁇ E ( B ) B B ⁇ B ⁇ B ⁇ B ⁇ 1 B ,
  • An effective value error BB BB B may be quickly calculated.
  • a calculation period is a control period B B , for example,
  • an output B BB of the controller may quickly reach a reference value B BB B_BBB that is, B BB ⁇ B BB B_BBB
  • a conventional calculation period of the effective value error is a utility power period B B , for example, for utility power 50 Hz/20 ms,
  • a bandwidth (that is, a crossover frequency B B_BBB ) of the equivalent effective value voltage controller in the present invention may be configured to be much greater than a crossover frequency B B_BBB of a conventional effective value voltage controller.
  • the crossover frequency of the equivalent effective value voltage controller in the present invention is
  • the entire inverter control system in this embodiment is equivalent to a voltage inner-outer loop system.
  • the inner loop is a voltage instantaneous value open-loop system
  • the outer loop is a voltage effective value closed-loop system.
  • the instantaneous value voltage controller B B (B) can ensure the convergence of an initial state of the system and improve the dynamic state of the system.
  • An imaginary-axis branch in the voltage effective value closed-loop system can ensure that an initial phase angle ⁇ of the inverter voltage is zero.
  • a total duty cycle B B BB +B B +B B ⁇ B B , that is, B BB ⁇ 0. Therefore, the instantaneous value voltage controller B B (B) accounts for a very small weight ratio in the steady state.
  • FIG. 4 which clearly shows the voltage inner-outer loop system.
  • the inner loop is an equivalent instantaneous value voltage open loop 3 which is equivalent to an instantaneous value voltage loop 1
  • the outer loop is an equivalent voltage effective value closed loop 2 .
  • B BB B_B is an effective value of the inverter output voltage
  • B ⁇ BB is a delay stage transfer function
  • B B_BB (B) is a transfer function for making the instantaneous value voltage loop 1 equivalent to the instantaneous value voltage open loop 3 .
  • the equivalent effective value voltage loop is specifically analyzed below with reference to FIG. 4 .
  • a controller is
  • B BB ( B ) B B ⁇ B ⁇ B ⁇ B ⁇ 1 B ,
  • B BB B_BB (B) B BB (B)B BBBBB (B), and
  • B BBB ⁇ _ ⁇ BB ( B ) B BB ( B ) ⁇ B BBBBB ( B ) 1 + B BB ( B ) ⁇ B BBBBB ( B ) .
  • An appropriate PM/GM is set to obtain a parameter B BBB B .
  • PM stands for a phase margin
  • GM stands for a gain margin.
  • B BB must be an integral controller.
  • GM>6 and PM>30 it is usually required that GM>6 and PM>30. Therefore, the design in this embodiment can meet the performance requirement.
  • FIG. 6 specially shows an instantaneous value voltage loop 1 in FIG. 4 .
  • B B ( B ) B B ⁇ B ⁇ B + B B + B
  • B BB is a gain coefficient of the instantaneous value voltage loop controller
  • B is a zero point of an instantaneous value voltage loop controller
  • B is a pole of the instantaneous voltage loop controller
  • s is the frequency domain operator.
  • the other types of controllers well known in the field may be selected as the instantaneous value voltage loop controller, as long as they can ensure the convergence of an initial state of the system and account for a very small weight ratio in a steady state.
  • the instantaneous value voltage loop accounts for a relatively small weight ratio, and is approximately 0 in a steady state. Therefore, the crossover frequency may be configured to be relatively small.
  • a crossover frequency of a conventional instantaneous value voltage controller is usually greater than 100 Hz. Therefore, the crossover frequency of the instantaneous value voltage loop in the present invention is clearly lower than that in conventional standards.
  • FIG. 8 is a simplified schematic diagram of a three-phase control system of a three-phase three-wire inverter.
  • a coordinate system conversion part is added based on the control system shown in FIG. 3 , wherein, BBB/ ⁇ represents a conversion part from an BBB coordinate system into an ⁇ coordinate system, and BBB/BBB represents an inverse conversion part from the ⁇ coordinate system into the BBB coordinate system.
  • a three-phase system has a three-phase BBB static coordinate system and a three-phase ⁇ static coordinate system and further has a two-phase rotating coordinate system BBB.
  • inverter voltages B BB B_B and B BB B_B of inverter controlled objects are still controlled by sine and cosine fundamental waves.
  • an instantaneous value voltage controller B B (B) and an equivalent effective value voltage controller B BB (B) are still used.
  • the inverter voltage B BB B_B is controlled by a 3 rd harmonic.
  • the 3 rd harmonic is injected to improve the utilization of a DC voltage.
  • the inventors prove the effects of the present invention through the results of simulation and test under hardware-in-the loop (HIL), respectively.
  • HIL hardware-in-the loop
  • FIG. 9 a shows an inverter phase abc voltage V inv_abc , an phase abc inductor current V inv_abc_xy of inverter bridge arms x and y, and an output phase abc current I o_abc in a full-load steady state sequentially from top to bottom.
  • the signals are normal, and there is almost no net difference.
  • phases b and c have similar harmonic data.
  • FIG. 10 a shows an instantaneous process of the load from 0% to 100%, an inverter phase abc voltage V inv_abc , and a phase a inductor current I inv_a_x of an inverter bridge arm x.
  • the signals are stable.
  • a dynamic state period is approximately less than one utility power period of 20 ms.
  • FIG. 10 b shows an unloading process of the load from 100% to 0%. The results show that the inverter voltage and the current are both stable, and the period of the dynamic state process is less than that of a utility power. Therefore, the inverter is stable with dynamic load, and the dynamic state performance and the steady state net difference satisfy requirements.
  • the present invention provides a novel orthogonal decomposition method for an inverter voltage, which comprises: obtaining a real-axis voltage and an imaginary-axis voltage, decomposing the real-axis voltage to obtain a direct current amount and a disturbance amount by means of a small signal analysis method, and similarly obtaining a direct current amount and a disturbance amount of the imaginary-axis voltage.
  • a novel inverter voltage controller is designed, which includes an equivalent effective value voltage controller and an instantaneous value voltage controller.
  • a complete novel three-phase inverter control system is designed.
  • the novel system Compared with a conventional inverter control system, the novel system has much better dynamic state performance and steady state performance than the conventional system in terms of effective value voltage control.
  • the instantaneous value voltage control loop is equivalent to an open-loop system, so as to ensure that the application to an inverter parallel system has comparable performance.
  • One of the effects of the inverter control system of the present invention lies in that compared with a conventional control system, in an equivalent effective value voltage control loop, a relatively large crossover frequency can be designed, so that sufficiently fast regulating speed is ensured, and there are still sufficient phase and amplitude margins.
  • an integral controller part in the system ensures stability and almost has no net difference, thereby providing excellent performance in dynamic status/stability.

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Abstract

The present invention provides a control system for a three-phase inverter, which comprises an instantaneous value voltage controller and an equivalent effective value voltage controller, wherein the instantaneous value voltage controller is configured to feed back and control an instantaneous value of an inverter output voltage, the equivalent effective value voltage controller is configured to perform an orthogonal decomposition feedback control on an effective value of the inverter output voltage, and wherein the equivalent effective value voltage controller is configured to perform integral compensation respectively on a real-axis voltage and an imaginary-axis voltage of a two-phase rotating coordinate system of the three-phase inverter, and an output of the instantaneous value voltage controller and an output of the equivalent effective value voltage controller are used to obtain the inverter output voltage through a delay stage transfer function and a controlled object transfer function. The control system of the present invention has fast regulating speed and high stability.

Description

    TECHNICAL FIELD
  • The present invention belongs to the field of inverters, and in particular relates to a control system and a control method for a three-phase inverter.
    • BACKGROUND
  • An inverter is a converter that converts DC electric energy of battery, storage battery, and the like into constant-frequency and constant-voltage AC or variable-frequency and variable-voltage AC. With the rapid development of science and technology, people's life and production have higher and higher requirements for power supply quality. An inverter is a core part of power supply equipment such as an uninterruptible power supply (UPS), an AC power frequency converter, or a new energy supply system. Therefore, research on inverters is of great significance to the development of modern industry. In an existing inverter control system, the quality of an inverter output waveform may deteriorate or even lead to system instability due to the presence of nonlinear factors. In an existing inverter control method, feedback control is performed directly by current or voltage, which has slow response speed and poor stability.
    • SUMMARY
  • Therefore, an objective of the present invention is to overcome the foregoing deficiencies in the prior art, and provide a control system for a three-phase inverter, which comprises an instantaneous value voltage controller and an equivalent effective value voltage controller, wherein the instantaneous value voltage controller is configured to feed back and control an instantaneous value of an inverter output voltage, the equivalent effective value voltage controller is configured to perform an orthogonal decomposition feedback control on an effective value of the inverter output voltage, and wherein the equivalent effective value voltage controller is configured to perform integral compensation respectively on a real-axis voltage and an imaginary-axis voltage of a two-phase rotating coordinate system of the three-phase inverter, and an output of the instantaneous value voltage controller and an output of the equivalent effective value voltage controller are used to obtain the inverter output voltage through a delay stage transfer function and a controlled object transfer function.
  • Preferably, the equivalent effective value voltage controller is an integral compensator
  • B BB ( B ) = B BBB ⁢ B ⁢ 1 B ,
  • where BBBBB B is a gain coefficient, and s is a frequency domain operator.
  • Preferably, the gain coefficient is less than 1.
  • Preferably, the real-axis voltage is represented as BB, and the imaginary-axis voltage is represented as BB, wherein

  • B B =B BBB_BEB ×B BBB

  • B B =B BBB_BBB ×B BBB,
  • where BBBB_BBB represents an inverter sine voltage given value, BBBB_BBB represents an inverter cosine voltage given value, and BBBB represents a difference between the inverter sine voltage given value and the inverter output voltage.
  • Preferably, BB=BB+{circumflex over (B)}B, BB=BB+{circumflex over (B)}B where BB is a real-axis voltage average value, {circumflex over (B)}B is an imaginary-axis voltage average value, {circumflex over (B)}B is a real-axis voltage disturbance value, and {circumflex over (B)}B is an imaginary-axis voltage disturbance value.
  • Preferably,
  • B B = 1 2 ⁢ BB BB ⁢ B B ^ B = - 1 2 ⁢ B ⁢ B BB ⁢ B ⁢ BBB ( 2 ⁢ B ⁢ B ) , and ⁢ B B = 0 B ^ B = 1 2 ⁢ BB BB ⁢ B ⁢ BBB ( 2 ⁢ B ⁢ B ) ,
  • where BBBB B is a difference between a given inverter voltage effective value and a feedback voltage effective value, B is an angular frequency of the inverter output voltage, and t is time.
  • Preferably, the integral compensator is configured to convert the real-axis voltage BB into a controller real-axis output value BBB, and convert the imaginary-axis voltage BB into a controller imaginary-axis output value BBB, wherein
  • B BB ≈ ( B BBB ⁢ B ⁢ 1 B ) ⁢ B B = ( B BBB ⁢ B ⁢ 1 B ) ⁢ 1 2 ⁢ B ⁢ B BB ⁢ B B BB ≈ ( B BBB ⁢ B ⁢ 1 B ) ⁢ B B = 0 .
  • Preferably, a duty cycle BB of an output of an equivalent real-axis voltage loop is obtained based on the controller real-axis output value BBB, a duty cycle BB of an output of an equivalent imaginary-axis voltage loop is obtained based on the controller imaginary-axis output value BBB, a duty cycle of an output of the instantaneous value voltage controller BB(B) is BBB, and a total duty cycle is d, where B=BBB+BB+BB.
  • Preferably, the total duty cycle d is used to obtain the inverter output voltage through the delay stage transfer function and the controlled object transfer function.
  • Preferably, the instantaneous value voltage controller is a hysteresis controller and represented as
  • B B ( B ) = B BB ⁢ B + B B + B ,
  • where BBB is a gain coefficient of the instantaneous value voltage controller, B is a zero point of an instantaneous voltage loop controller, B is a pole of the instantaneous voltage loop controller, and s is the frequency domain operator.
  • Preferably, the control system further comprises a coordinate system conversion part which is configured to implement conversion between an BBB coordinate system and an αβγ coordinate system.
  • The present invention further provides a control method for a three-phase inverter, comprising:
  • a step of feeding back and controlling an instantaneous value of an inverter output voltage;
  • a step of performing orthogonal decomposition feedback control on an effective value of the inverter output voltage, which comprises performing integral compensation on a real-axis voltage and an imaginary-axis voltage of a two-phase rotating coordinate system of the three-phase inverter; and
  • an output of the step of feeding back and controlling an instantaneous value of an inverter output voltage and an output of the step of performing orthogonal decomposition feedback control on an effective value of the inverter output voltage are used to obtain the inverter output voltage through delay stage transfer and controlled object transfer.
  • Preferably, a gain coefficient of the integral compensation is less than 1.
  • Preferably, the real-axis voltage is represented as BB, and the imaginary-axis voltage is represented as BB, where

  • B B =B BBB_BEB ×B BBB

  • B B =B BBB_BBB ×B BBB,
  • where BBBB_BBB represents an inverter sine voltage given value, BBBB_BBB represents an inverter cosine voltage given value, and BBBB represents a difference between the inverter sine voltage given value and the inverter output voltage.
  • Preferably, the method further includes converting an BBB coordinate system into an αβγ coordinate system in a feedback process and converting the αβγ coordinate system into the BBB coordinate system in an output process.
  • Preferably, the method further includes a step of adjusting an initial phase angle of the inverter output voltage to zero.
  • Compared with the prior art, the advantages of the present invention lie in that compared with a conventional control system, in an equivalent effective value voltage control loop, a relatively large crossover frequency can be designed, so that sufficiently fast regulating speed is ensured, and there are still sufficient phase and gain margins. In addition, an integral controller in the system ensures stability and almost has no net difference, thereby providing excellent performance in dynamic status/stability.
    • BRIEF DESCRIPTION OF DRAWINGS
  • Embodiments of the present invention are further described below with reference to the accompanying drawings, in which:
  • FIG. 1 and FIG. 2 respectively show a three-phase four-wire inverter topology and a three-phase three-wire inverter topology;
  • FIG. 3 shows a simplified inverter control system used for a three-phase four-wire inverter topology according to a first embodiment of the present invention;
  • FIG. 4 shows an equivalent inverter control system according to the first embodiment of the present invention;
  • FIG. 5 is an open-loop Bode diagram of an equivalent effective value voltage loop according to the first embodiment of the present invention;
  • FIG. 6 shows an instantaneous value voltage loop in FIG. 4 ;
  • FIG. 7 is an open-loop Bode diagram of the instantaneous value voltage loop in FIG. 4 ;
  • FIG. 8 shows a simplified inverter control system used for a three-phase three-wire inverter topology according to a second embodiment of the present invention;
  • FIG. 9 a and FIG. 9 b show simulation results; and
  • FIG. 10 a and FIG. 10 b show results of test under hardware-in-the-loop (HIL).
    •  DETAILED DESCRIPTION
  • In order to make the objectives, technical solutions, and advantages of the present invention more clear, the present invention is further described below in detail with reference to the accompanying drawings and specific embodiments. It should be understood that the specific embodiments described herein are only used to explain the present invention, but not to limit the present invention.
  • FIG. 1 and FIG. 2 respectively show a three-phase four-wire inverter topology and a three-phase three-wire inverter topology. A power supply voltage supplies power to a load through an inverter topology. BBBB represents a bus capacitor of a single side busbar. BBBB represents a voltage of the bus capacitor. Brg_A_X/Brg_A_Y, Brg_B_X/Brg_B_Y, and Brg_C_X/Brg_C_Y respectively represent inverter X and Y bridge arms of phases A, B, and C. BB and BB respectively represent filter inductors of the inverter X bridge arm and Y bridge arm; BB_B and BB_B respectively represent currents of the filter inductors of the inverter X bridge arm and Y bridge arm; BBBB represents an inverter filter capacitor; and BBBB and BBBB respectively represent an X bridge arm and a Y bridge arm of a three-phase inverter.
    • First Embodiment
  • In this embodiment, a control system and a control method for an inverter of the present invention are provided based on the three-phase four-wire inverter topology shown in FIG. 1 . A control object of the control system and control method in this embodiment is the three-phase four-wire inverter topology shown in FIG. 1 . For simplicity, a phase A is only used as an example for discussion. Those skilled in the art can understand that the cases of phases B and C are similar to the case of the phase A. ABC and abc herein are both used for representing three phases of the three-phase inverter.
  • Refer to a simplified inverter control system shown in FIG. 3 . First, an inverter sine voltage given value BBBB_BBB and an inverter cosine voltage given value BBBB_BBB are obtained based on the given inverter voltage effective value BBB B_BBB,

  • B BBB_BBB =B BB B_BBB BBB(BB),

  • B BBB_BBB =B BB B_BBB BBB(BB)
  • where B is an angular frequency of the inverter output voltage, and t is time. Those skilled in the art know that a given value is also referred to as a reference value, that is, an expected value output by an inverter.
  • It is assumed that BB=BBB B_BBBB(BB+B), where BB is the inverter output voltage (that is, a feedback voltage), BBB B_B is an effective value of the inverter output voltage, and B is an initial phase angle of the inverter voltage. In the present invention, for ease of control, it is required that B=0.
  • In this case,

  • B BBB =B ref_sin −B B =B BB B_BBB BBB(BB)−B BB B_B BBB(BB)—BB BB B BBB(BB),
  • where BBBB is a difference between the inverter sine voltage given value and the feedback voltage, and BBB B is a difference between the given inverter voltage effective value and the feedback voltage effective value. In the field of circuit control, the feedback voltage is a sine value. Therefore, in the discussion of the present invention, a sine symbol is no longer labeled on a feedback signal.
  • BBBB including the feedback voltage BB is input into a multiplier respectively with the inverter sine voltage given value and the inverter cosine voltage given value to obtain a real-axis voltage BB and an imaginary-axis voltage BB

  • B B =B BBB_BBB ×B BBB

  • B B =B BBB_BBB ×B BBB  (1).
  • Therefore,
  • B B = B BB ⁢ B ⁢ _ ⁢ BBB ⁢ B ⁢ B BB ⁢ B ⁢ BBB 2 ( B ⁢ B ) = B BB ⁢ B ⁢ _ ⁢ BBB ⁢ BB BB ⁢ B ⁢ 1 2 ⁢ ( 1 - BBB ⁡ ( 2 ⁢ B ⁢ B ) ) ( 2 ) B B = B BB ⁢ B ⁢ _ ⁢ BBB ⁢ B ⁢ B BB ⁢ B ⁢ BBB ⁡ ( B ⁢ B ) ⁢ BBB ⁡ ( B ⁢ B ) = B BB ⁢ B ⁢ _ ⁢ BBB ⁢ B ⁢ B BB ⁢ B ⁢ 1 2 ⁢ BBB ⁡ ( 2 ⁢ B ⁢ B ) .
  • The given inverter voltage effective value is normalized into a per-unit pu, that is, BBB B_BBB=1. Where pu represents a relative unit system, and is a common term in the field of engineering. For example, an inverter voltage standard value is 230 V, represented by 1 pu. In this case, (1+15%)*230 V is 1.15 pu.
  • In this case, Formula (2) may be simplified as:
  • B B = B ⁢ B BB ⁢ B ⁢ BBB 2 = ( B ⁢ B ) = B ⁢ B BB ⁢ B ⁢ 1 2 ⁢ ( 1 - BBB ( 2 ⁢ B ⁢ B ) ) B B = B ⁢ B BB ⁢ B ⁢ BBB ⁢ ( B ⁢ B ) ⁢ BBB ( B ⁢ B ) = B ⁢ B BB ⁢ B ⁢ 1 2 ⁢ BBB ( 2 ⁢ B ⁢ B ) .
  • The real-axis voltage and the imaginary-axis voltage are further respectively represented as including a constant part (a direct current amount) and a disturbance part (an alternating current amount):

  • B B =B B +{circumflex over (B)} B

  • B B =B B +{circumflex over (B)} B  (3),
  • where BB is a real-axis voltage average value, BB is an imaginary-axis voltage average value, {circumflex over (B)}B is a real-axis voltage disturbance value, and {circumflex over (B)}B is an imaginary-axis voltage disturbance value.
  • Real-axis part:
  • B B = 1 2 ⁢ B ⁢ B BB ⁢ B B B = - 1 2 ⁢ B ⁢ B BB ⁢ B ⁢ BBB ( 2 ⁢ B ⁢ B ) ; ( 4 )
  • and
  • Imaginary-axis part:
  • B B = 0 B ^ B = 1 2 ⁢ BB BB ⁢ B ⁢ BBB ( 2 ⁢ B ⁢ B ) . ( 5 )
  • According to the foregoing derivation, a real-axis voltage and an imaginary-axis voltage of a two-phase rotating coordinate system of a three-phase inverter are obtained, and the real-axis voltage and the imaginary-axis voltage are respectively written in a form including a direct current amount and an alternating current amount.
  • Compensation control is further performed on the real-axis voltage and the imaginary-axis voltage. Starting from decompositions (3), (4), and (5), an integral compensator (also referred to as an integral controller)
  • B BB ( B ) = B BBB ⁢ B ⁢ 1 B
  • is added. In the present invention, BBB(B) is an effective value voltage controller configured to control a real-axis voltage and an imaginary-axis voltage, and is referred to as an “equivalent effective value voltage controller” in the present invention to differentiate from a conventional effective value voltage controller. BBBB B is a gain coefficient of the controller,
  • 1 B
  • is an integral pan or the controller, and s is a frequency domain operator.
  • In this case,
  • B BB = ( B BBB ⁢ B ⁢ 1 B ) × B B B BB = ( B BBB ⁢ B ⁢ 1 B ) × B B , ( 6 )
  • where BBB is a controller real-axis output value, and BBB is a controller imaginary-axis output value.
  • Formula (3) is substituted into Formula (6),
  • B BB = ( B BBB ⁢ B ⁢ 1 B ) × ( B B + B B ) B BB = ( B BBB ⁢ B ⁢ 1 B ) × ( B B + B B ) .
  • It is known that
  • ❘ "\[LeftBracketingBar]" B B ^ × ( B BBB ⁢ B ⁢ 1 B ) ❘ "\[RightBracketingBar]" = ❘ "\[LeftBracketingBar]" B BBB ⁢ B 2 ⁢ B ⁢ B BB ⁢ B ⁢ 1 2 ⁢ B ⁢ BBB ⁢ ( 2 ⁢ B ⁢ B ) ❘ "\[RightBracketingBar]" ≪ 1 ; ❘ "\[LeftBracketingBar]" B B ^ × ( B BBB ⁢ B ⁢ 1 B ) ❘ "\[RightBracketingBar]" = ❘ "\[LeftBracketingBar]" B BBB ⁢ B 2 ⁢ B ⁢ B BB ⁢ B ⁢ 1 2 ⁢ B ⁢ BBB ⁢ ( 2 ⁢ B ⁢ B ) ❘ "\[RightBracketingBar]" ≪ 1 ; ⁢ ( B BBB ⁢ B < 1 ) . ( 7 )
  • Therefore, the disturbance part is omitted to obtain
  • B BB ≈ ( B BBB ⁢ B ⁢ 1 B ) ⁢ B B = ( B BBB ⁢ B ⁢ 1 B ) ⁢ 1 2 ⁢ B ⁢ B BB ⁢ B B BB ≈ ( B BBB ⁢ B ⁢ 1 B ) ⁢ B B = 0 . ( 8 )
  • As can be seen from Formula (8), in this embodiment, an inverter voltage effective value after orthogonal decomposition includes a real part B3B and an imaginary part BBB equal to zero, which is equivalent to an instantaneous effective value voltage control system. Therefore, the foregoing derivation process is proved to be accurate and reasonable.
  • Continue to refer to the simplified inverter control system shown in FIG. 3 . An instantaneous value of the inverter output voltage is controlled by the instantaneous value voltage controller BB(B). BBB is a duty cycle of an output of the instantaneous value voltage controller, BB is a duty cycle of an output of an equivalent real-axis voltage loop, BB is a duty cycle of an output of an equivalent imaginary-axis voltage loop, BBBBBB(B) is a digitally controlled delay stage transfer function, BBB (B) is a controlled object transfer function of the voltage loop, that is, a mathematical control formula obtained through a Laplace transform of hardware of the three-phase inverter, BBBB is a voltage of a bus capacitor, B is an inverter filter inductor, B is an inverter filter capacitor, B is an equivalent resistance value of a load. When a controlled object (an inverter topology) has different parameters, transfer functions are different. A total duty cycle d is used to obtain the inverter output voltage through the delay stage transfer function and the controlled object transfer function.
  • In the case of normalization, BBBB_BBB=1BB×BBB(BB) BBBB_BBB=1BB×BBB(BB).
  • Based on this, the duty cycle of the equivalent real-axis voltage loop output and the duty cycle of the equivalent imaginary-axis voltage loop output may be further obtained.

  • B B =B BB ×B BBB_BBB =B BB sin(BB)

  • B B =B BB ×B BBB_BBB =B BB cos(BB)≈0.
  • It is known that
  • B B ⁢ E ( B ) = B B ⁢ B ⁢ B ⁢ B ⁢ 1 B ,
  • which represents an equivalent effective value voltage controller in the present invention. An effective value error BBBB B may be quickly calculated. A calculation period is a control period BB, for example,
  • B B = 1 B B = 1 1 ⁢ 8 ⁢ 0 ⁢ 0 ⁢ 0
  • here.
  • Therefore, an output BBB of the controller may quickly reach a reference value BBB B_BBB that is, BBB≈BBB B_BBB A conventional calculation period of the effective value error is a utility power period BB, for example, for utility power 50 Hz/20 ms,
  • B B = 1 B B = 1 5 ⁢ 0 .
  • a bandwidth (that is, a crossover frequency BB_BBB) of the equivalent effective value voltage controller in the present invention may be configured to be much greater than a crossover frequency BB_BBB of a conventional effective value voltage controller. For example, the crossover frequency of the equivalent effective value voltage controller in the present invention is
  • B B ⁢ _ ⁢ BBB ≈ 1 1 ⁢ 0 ⁢ B B = 1800 ⁢ Hz ,
  • and the crossover frequency of the conventional effective value voltage controller is
  • R B ⁢ _ ⁢ BBB ≈ ( 1 1 ⁢ 0 ∼ 1 5 ) ⁢ B B = ( 5 ∼ 10 ⁢ BB ) .
  • This means that a regulating speed of the equivalent effective value voltage controller in the present invention is greater than that of a conventional control method. In addition, with the presence of an integral stage, there is almost no steady state error in the control method of the present invention.
  • The entire inverter control system in this embodiment is equivalent to a voltage inner-outer loop system. The inner loop is a voltage instantaneous value open-loop system, and the outer loop is a voltage effective value closed-loop system. The instantaneous value voltage controller BB(B) can ensure the convergence of an initial state of the system and improve the dynamic state of the system. An imaginary-axis branch in the voltage effective value closed-loop system can ensure that an initial phase angle θ of the inverter voltage is zero. In a steady state of the system, BBBB B≈0, BB≈BBB B_BBB BBB(BB), and BB≈0. A total duty cycle B=BBB+BB+BB≈BB, that is, BBB≈0. Therefore, the instantaneous value voltage controller BB(B) accounts for a very small weight ratio in the steady state.
  • Refer to an equivalent inverter control system shown in FIG. 4 , which clearly shows the voltage inner-outer loop system. The inner loop is an equivalent instantaneous value voltage open loop 3 which is equivalent to an instantaneous value voltage loop 1, and the outer loop is an equivalent voltage effective value closed loop 2. BBB B_B is an effective value of the inverter output voltage, B−BB is a delay stage transfer function, and BB_BB(B) is a transfer function for making the instantaneous value voltage loop 1 equivalent to the instantaneous value voltage open loop 3.
  • The equivalent effective value voltage loop is specifically analyzed below with reference to FIG. 4 . A controller is
  • B BB ( B ) = B B ⁢ B ⁢ B ⁢ B ⁢ 1 B ,
  • and a controlled object is BBBBBB(B)=B−BB, where
  • B = 1 1 ⁢ 7 ⁢ B ⁢ B ⁢ B .
  • Pade equivalent linearization is performed, and a controller is designed. In the open loop and closed loop of the system:
  • an equivalent voltage effective value open-loop transfer function is:
  • BBB B_BB(B)=BBB(B)BBBBBB(B), and
  • an equivalent voltage effective value closed-loop transfer function is:
  • B BBB ⁢ _ ⁢ BB ( B ) = B BB ( B ) ⁢ B BBBBB ( B ) 1 + B BB ( B ) ⁢ B BBBBB ( B ) .
  • An appropriate PM/GM is set to obtain a parameter BBBB B. PM stands for a phase margin, and GM stands for a gain margin. It should be noted that BBB must be an integral controller.
  • Refer to an open-loop Bode diagram of the equivalent effective value voltage loop in FIG. 5 . A loop crossover frequency is BB_BB=16.6BB, a phase margin is BBBB=89.6BBB, and a gain margin is BBBB=48.2BB. In the field of inverter control, it is usually required that GM>6 and PM>30. Therefore, the design in this embodiment can meet the performance requirement.
  • The instantaneous value voltage loop in FIG. 4 is then discussed below. For clarity, FIG. 6 specially shows an instantaneous value voltage loop 1 in FIG. 4 .
  • B B ( B ) = B B ⁢ B ⁢ B + B B + B
  • is selected exemplarily but not restrictively, representing a hysteresis controller. BBB is a gain coefficient of the instantaneous value voltage loop controller, B is a zero point of an instantaneous value voltage loop controller, B is a pole of the instantaneous voltage loop controller, and s is the frequency domain operator. In the present invention, the other types of controllers well known in the field may be selected as the instantaneous value voltage loop controller, as long as they can ensure the convergence of an initial state of the system and account for a very small weight ratio in a steady state.
  • Refer to an open-loop Bode diagram of the instantaneous value voltage loop in FIG. 7 . A crossover frequency is BB_B=15.9BB, a phase margin is BBB=116BBB, and a gain margin is BBB=37.1BB. The instantaneous value voltage loop accounts for a relatively small weight ratio, and is approximately 0 in a steady state. Therefore, the crossover frequency may be configured to be relatively small. A crossover frequency of a conventional instantaneous value voltage controller is usually greater than 100 Hz. Therefore, the crossover frequency of the instantaneous value voltage loop in the present invention is clearly lower than that in conventional standards.
    • Second Embodiment
  • In this embodiment, an inverter control system for the three-phase three-wire inverter topology shown in FIG. 2 is provided. FIG. 8 is a simplified schematic diagram of a three-phase control system of a three-phase three-wire inverter. A coordinate system conversion part is added based on the control system shown in FIG. 3 , wherein, BBB/αβγ represents a conversion part from an BBB coordinate system into an αβγ coordinate system, and BBB/BBB represents an inverse conversion part from the αβγ coordinate system into the BBB coordinate system. In the electrical field, a three-phase system has a three-phase BBB static coordinate system and a three-phase αβγ static coordinate system and further has a two-phase rotating coordinate system BBB. After coordinate system conversion, inverter voltages BBB B_B and BBB B_B of inverter controlled objects are still controlled by sine and cosine fundamental waves. For controllers of the control system, an instantaneous value voltage controller BB(B) and an equivalent effective value voltage controller BBB(B) are still used. The inverter voltage BBB B_B is controlled by a 3rd harmonic. The 3rd harmonic is injected to improve the utilization of a DC voltage.
  • The inventors prove the effects of the present invention through the results of simulation and test under hardware-in-the loop (HIL), respectively.
  • Refer to the results of simulation shown in FIG. 9 a and FIG. 9 b . FIG. 9 a shows an inverter phase abc voltage Vinv_abc, an phase abc inductor current Vinv_abc_xy of inverter bridge arms x and y, and an output phase abc current Io_abc in a full-load steady state sequentially from top to bottom. The signals are normal, and there is almost no net difference. FIG. 9 b shows harmonic data of phase a of an inverter voltage in the full-load steady state, wherein a total distortion rate of a voltage harmonic is THDv=0.88%, satisfying the requirement of steady-state performance of products. Those skilled in the art can understand that phases b and c have similar harmonic data.
  • Refer to the results of test under HILg shown in FIG. 10 a and FIG. 10 b . FIG. 10 a shows an instantaneous process of the load from 0% to 100%, an inverter phase abc voltage Vinv_abc, and a phase a inductor current Iinv_a_x of an inverter bridge arm x. In a loading process, the signals are stable. A dynamic state period is approximately less than one utility power period of 20 ms. FIG. 10 b shows an unloading process of the load from 100% to 0%. The results show that the inverter voltage and the current are both stable, and the period of the dynamic state process is less than that of a utility power. Therefore, the inverter is stable with dynamic load, and the dynamic state performance and the steady state net difference satisfy requirements.
  • In summary, the present invention provides a novel orthogonal decomposition method for an inverter voltage, which comprises: obtaining a real-axis voltage and an imaginary-axis voltage, decomposing the real-axis voltage to obtain a direct current amount and a disturbance amount by means of a small signal analysis method, and similarly obtaining a direct current amount and a disturbance amount of the imaginary-axis voltage. Based on the novel orthogonal decomposition method for a voltage, a novel inverter voltage controller is designed, which includes an equivalent effective value voltage controller and an instantaneous value voltage controller. In a three-phase inverter, a complete novel three-phase inverter control system is designed. Compared with a conventional inverter control system, the novel system has much better dynamic state performance and steady state performance than the conventional system in terms of effective value voltage control. In addition, the instantaneous value voltage control loop is equivalent to an open-loop system, so as to ensure that the application to an inverter parallel system has comparable performance.
  • One of the effects of the inverter control system of the present invention lies in that compared with a conventional control system, in an equivalent effective value voltage control loop, a relatively large crossover frequency can be designed, so that sufficiently fast regulating speed is ensured, and there are still sufficient phase and amplitude margins. In addition, an integral controller part in the system ensures stability and almost has no net difference, thereby providing excellent performance in dynamic status/stability.
  • Although the present invention has been described by way of preferred embodiments, the present invention is not limited to the embodiments described herein, but includes various changes as well as variations made without departing from the scope of the present invention.

Claims (16)

1. A control system for a three-phase inverter, comprising an instantaneous value voltage controller and an equivalent effective value voltage controller, wherein the instantaneous value voltage controller is configured to feed back and control an instantaneous value of an inverter output voltage, the equivalent effective value voltage controller is configured to perform an orthogonal decomposition feedback control on an effective value of the inverter output voltage, and wherein the equivalent effective value voltage controller is configured to perform integral compensation respectively on a real-axis voltage and an imaginary-axis voltage of a two-phase rotating coordinate system of the three-phase inverter, and an output of the instantaneous value voltage controller and an output of the equivalent effective value voltage controller are used to obtain the inverter output voltage through a delay stage transfer function and a controlled object transfer function.
2. The control system of claim 1, wherein the equivalent effective value voltage controller is an integral compensator Dri(s)=Kcomp1/s, where Kcomp is a gain coefficient, and s is a frequency domain operator.
3. The control system of claim 2, wherein the gain coefficient is less than 1.
4. The control system of claim 3, wherein the real-axis voltage is represented as vr, and the imaginary-axis voltage is represented as vi, wherein
{ v r = v ref ⁢ _ ⁢ sin × v err v i = v ref ⁢ _ ⁢ cos × v err ,
where vref sin represents an inverter sine voltage given value, vref cos represents an inverter cosine voltage given value, and verr represents a difference between the inverter sine voltage given value and the inverter output voltage.
5. The control system of claim 4, wherein
{ v r = V r + v ^ r v i = V i + v ^ i ,
where Vr is a real-axis voltage average value, Vi is an imaginary-axis voltage average value, {circumflex over (v)}r is a real-axis voltage disturbance value, and {circumflex over (v)}i is an imaginary-axis voltage disturbance value.
6. The control system of claim 5, wherein
{ V r = 1 2 ⁢ Δ ⁢ V rms v ^ r = - 1 2 ⁢ Δ ⁢ V rms ⁢ cos ⁢ ( 2 ⁢ wt ) , and ⁢ { V i = 0 v ^ i = 1 2 ⁢ Δ ⁢ V rms ⁢ sin ⁢ ( 2 ⁢ wt ) ,
where ΔVrms is a difference between a given inverter voltage effective value and a feedback voltage effective value, w is an angular frequency of the inverter output voltage, and t is time.
7. The control system of claim 6, wherein the integral compensator is configured to convert the real-axis voltage vr into a controller real-axis output value Vrs, and convert the imaginary-axis voltage v, into a controller imaginary-axis output value Vim, wherein
{ V re ≈ ( K comp ⁢ 1 s ) ⁢ V r = ( K comp ⁢ 1 s ) ⁢ 1 2 ⁢ Δ ⁢ V rms V im ≈ ( K comp ⁢ 1 s ) ⁢ V i - 0 .
8. The control system of claim 7, wherein a duty cycle dd of an output of an equivalent real-axis voltage loop is obtained based on the controller real-axis output value Vre, a duty cycle do of an output of an equivalent imaginary-axis voltage loop is obtained based on the controller imaginary-axis output value Vim, a duty cycle of an output of the instantaneous value voltage controller Dv(
Figure US20230052807A1-20230216-P00999
) is ddr, and a total duty cycle is d, where d=dv+dd+dα.
9. The control system of claim 8, wherein the total duty cycle d is used to obtain the inverter output voltage through the delay stage transfer function and the controlled object transfer function.
10. The control system of claim 1, wherein the instantaneous value voltage controller is a hysteresis controller and represented as
D v ( s ) = ? s + z s + p , ? indicates text missing or illegible when filed
where kjw is a gain coefficient of the instantaneous value voltage controller, z is a zero point of an instantaneous voltage loop controller, p is a pole of the instantaneous voltage loop controller, and s is the frequency domain operator.
11. The control system of claim 1, further comprising a coordinate system conversion part which is configured to implement conversion between an abc coordinate system and an αβγ coordinate system.
12. A control method for a three-phase inverter, comprising:
a step of feeding back and controlling an instantaneous value of an inverter output voltage;
a step of performing orthogonal decomposition feedback control on an effective value of the inverter output voltage, which comprises performing integral compensation on a real-axis voltage and an imaginary-axis voltage of a two-phase rotating coordinate system of the three-phase inverter; and
an output of the step of feeding back and controlling an instantaneous value of an inverter output voltage and an output of the step of performing orthogonal decomposition feedback control on an effective value of the inverter output voltage are used to obtain the inverter output voltage through delay stage transfer and controlled object transfer.
13. The control method of claim 12, wherein a gain coefficient of the integral compensation is less than 1.
14. The control method of claim 12, wherein the real-axis voltage is represented as vr, and the imaginary-axis voltage is represented as vi, wherein
{ v r = v ref ⁢ _ ⁢ sin × v err v i = v ref ⁢ _ ⁢ cos × v err ,
where vref sin represents an inverter sine voltage given value, vref cos represents an inverter cosine voltage given value, and verr, represents a difference between the inverter sine voltage given value and the inverter output voltage.
15. The control method of claim 12, further comprising converting an abc coordinate system into an αβγ coordinate system in a feedback process and converting the αβγ coordinate system into the abc coordinate system in an output process.
16. The control method of claim 12, further comprising a step of adjusting an initial phase angle of the inverter output voltage to zero.
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