US20110026578A1 - Method for reception of long range signals in bluetooth - Google Patents

Abstract

The invention of a method for reception of long transmission range Bluetooth signals impaired by multipath are disclosed. The new reception method proposed allows to increase the transmission range for data transmission in Bluetooth. The invention proposes the use of a new FDE adapted to SC transmission without a GI or CP. The proposed FDE very successfully mitigates ISI while being very implemention-friendly.

Description

CROSS REFERENCE TO RELATED APPLICATIONS
• This application is a Divisional of co-pending application Ser. No. 11/435,948, filed on May 18, 2006, and for which priority is claimed under 35 U.S.C. 120, the entire contents of all of which are hereby incorporated by reference.
BACKGROUND OF THE INVENTION
• 1. Field of the Invention
• The present invention relates to a method for reception of Bluetooth signals, and more especially, to a method for reception of long-range signals in Bluetooth.
• 2. Background of the Related Art
• The Bluetooth standard distinguishes devices by their so-called power class {[IEEE 802.15.1], [BT SIG 1.2], [BT SIG EDR]}. For each power class, a maximum output power (Pmax), a nominal output power and a minimum output power is specified as shown in Table 1.

• TABLE 1
  	Maximum	Nominal	Minimum
  Power	Output	Output	Output
  Class	Power (Pmax)	Power	Power	Power Control

  1	100	mW	N/A	1 mW (0 dB)	Pmin <+4 dBm
  	(20	dBm)			to Pmax
  				Optional:
  				Pmin2 to Pmax
  2	2.5	mW	1 mW	0.25 mW (−6	Optional:
  	(4	dBm)	(0 dBm)	dBm)	Pmin2 to Pmax
  3	1	mW	N/A	N/A	Optional:
  	(0	dBm)			Pmin2 to Pmax

• The Bluetooth technology is intended to implement wireless personal area networks (WPAN). Therefore, the typical range of Bluetooth devices is expected to be limited to about 10 meters. Bluetooth devices according to power class 1, however, are capable to transmit over a range significantly larger than the so-called personal operating space (POS) of about 10 meters.
• [IEEE 802.15.1]: the IEEE Std 802.15.1-IEEE Standard for Information technology Telecommunications and information exchange between systems Local and metropolitan area networks Specific requirements-Part 15.1: Wireless Medium Access Control (MAC) and Physical Layer (PHY) Specifications for Wireless Personal Area Networks (WPANs), 14 Jun. 2002.
• [BT SIG 1.2]: Bluetooth SIG Specification of the Bluetooth System, Version 1.2, 5 Nov. 2003.
• [BT SIG EDR]: Bluetooth SIG Specification of the Bluetooth System with EDR, Version 2.0, 4 Nov. 2004.
Sensitivity Performance in Bluetooth
• In [BT SIG EDR], a reference sensitivity level of −70 dBm is given for an uncoded bit error rate (BER) of 0.0001 (0.01%). In FIG. 1 and FIG. 2, the uncoded BER versus SNR is shown for PI/4-DQPSK and D8PSK, respectively. For PI/4-DQPSK, about 14 dB SNR are needed to achieve an uncoded BER of 0.01%. For D8PSK, about 20 dB SNR are needed to achieve an uncoded BER of 0.01%. Additional SNR margin is needed to accommodate fixed-point implementation losses as well as losses introduced by radio front end impairments and non-ideal time and frequency synchronization. Therefore, about 25 dB SNR are assumed to achieve an uncoded BER of 0.01%.
Path Loss in Bluetooth
• The signal power received by a Bluetooth device depending on the signal power transmitted by another Bluetooth device is given by Equation 1:
• 
  P _RX =P _TX −L _Path −L _Fade +G _TX +G _RX  (1)
• with
• • P_RX: received signal power
  • P_TX: transmitted signal power
  • L_Path: path loss
  • L_Fade: fade margin
  • G_TX: received antenna gain
  • G_RX: transmit antenna gain
    The following assumptions are applied in Equation 2 and Equation 3:
• 
  G_TX=G_RX=0 dBi  (2)
• 
  L_Fade=8 dB  (3)
• Therefore, based on Equation 1 and Equation 2, the path loss is given by Equation 4:
• 
  L _Path =P _TX −P _RX−8 dB  (4)
• The transmitted signal power under consideration (maximum signal power) is in Equation 5:
• 
  P _TX

  

  P _TX,max=20 dBm (Power class 1 device)  (5)
• The received signal power under consideration (minimum signal power) is given by Equation 6:
• 
  P _RX

  

  P _RX,min =N _Floor +W+SNR _RX +NF _RX  (6)
• with
• • P_Tx,max: maximum transmit power
  • P_RX,min: minimum received power
  • N_Floor: noise floor due to thermal noise
  • W: noise bandwidth
  • SNR_RX: signal-to-noise-ratio required for BER=0.0001 for D8PSK
  • NF_Rx: receiver noise figure
• The noise floor due to thermal noise amounts to −174 dBm per Hz signal bandwidth. The signal bandwidth for Bluetooth technology equals 1 MHz. The receiver noise figure is assumed to be 20 dB.
• The minimum signal power can now be computed by Equation 7:
• $\begin{array}{cc}\begin{array}{c}{P}_{\mathrm{RX},\min }=-174\mathrm{dBm}\text{/}\mathrm{Hz}+1\mathrm{MHz}+25\mathrm{dB}+20\mathrm{dB}\\ =-114\mathrm{dBm}+45\mathrm{dB}\\ =-69\mathrm{dBm}\end{array}& \left(7\right)\end{array}$
• The maximum path loss based on maximum transmit signal power and minimum received signal power and fade margin based on Equation 4 is now given by Equation 8:
• $\begin{array}{cc}\begin{array}{c}{L}_{\mathrm{Path},\max }=P_{\mathrm{TX},\max }-P_{\mathrm{RX},\min }-8\mathrm{dB}\\ =20\mathrm{dB}+69\mathrm{dB}-8\mathrm{dB}\\ =81\mathrm{dB}\end{array}& \left(8\right)\end{array}$
• It follows that the maximum path loss for a Bluetooth device of power class 1 equals 81 dB. For power class 2 and power class 3, the maximum path loss amounts to 73 dB and 69 dB, respectively.
On Transmission Range in Bluetooth
• The path loss depending on the transmission range for line-of-sight (LOS) conditions in a Bluetooth network is given by Equation 9:
• $\begin{array}{cc}{L}_{\mathrm{Path}}=20\log \left(\frac{4\Pi }{\lambda }R\right)& \left(9\right)\end{array}$
• or by Equation 10 approximately
• 
  L _Path=40+20 log(R)  (10)
• with
• • R: transmission range in [meters]
  • λ: wavelength of transmission signal
• The path loss depending on the transmission range for non-line-of-sight (NLOS) conditions in a Bluetooth network is given by Equation 11.
• $\begin{array}{cc}{L}_{\mathrm{Path}}=36\log \left(\frac{4\Pi }{\lambda }R\right)-46.7\mathrm{dB}& \left(11\right)\end{array}$
• or Equation 12 approximately
• 
  L _Path=25.3+36 log(R)  (12)
• Equation 9, by Equation 10, Equation 11 and Equation 12 are visualized in FIG. 3. It follows that for a maximum pathloss of 81 dB, a maximum transmission range R_max of 113 meters (in large office) is achieved (LOS conditions) using a power class 1 device while ensuring reliable communication. For power class 2 and power class 3, 18 meters (in small office) and 11 meters (POS) are achieved, respectively.
On Multipath Propagation in Bluetooth
• In Bluetooth, the symbol rate equals 1 Msps while the symbol duration T_symbol equals 1 μs (1000 ns). According the radio propagation theory, a radio frequency signal propagates 300 m in 1 μs (3e8 meters per second). The maximum echo delay (1st versus 2nd echo) based on the maximum transmission range is given by Equation 13:
• $\begin{array}{cc}\begin{array}{c}{D}_{\max }=\frac{T_{\mathrm{Symbol}}}{300m}·R_{\max }\\ =\frac{1\mathrm{\mu s}}{300m}113m\\ =377\mathrm{ns}\end{array}& \left(13\right)\end{array}$
• It follows that for a maximum transmission range R_max of 113 meters a maximum echo delay of 377 ns is obtained.
• For power class 2 and power class 3, 60 ns and 37 ns are obtained, respectively.
• Multipath propagation results in inter-symbol interference (ISI). The amount of ISI introduced depends on the number and power of all echo paths following the first arriving path.
• Using the result from Equation 13, one gets a maximum ISI percentage shown in Equation 14:
• $\begin{array}{cc}\begin{array}{c}{\mathrm{ISI}}_{\max }=\frac{D_{\max }}{T_{\mathrm{Symbol}}}\\ =\frac{377\mathrm{ns}}{1\mathrm{\mu s}}\\ =37.7\%\end{array}& \left(14\right)\end{array}$
• For power class 2 and power class 3, 6% and 3.7% are obtained, respectively.
• The ISI is modelled as an echo path having a relative power (with regards to the first arriving path) equal to ISI_max. With that assumption, a worst-case multipath channel profile with a 1st (obstructed) path @ 0 dB w/ delay of 0 samples and a 2nd path (echo) @10*log₁₀(0.377)=−4.24 dB w/ delay of 1 sample (1 μs).
• The 2-path multipath propagation model for Bluetooth long transmission range applications is shown in FIG. 4. A large office scenario for Bluetooth long transmission range applications is shown in FIG. 5.
Impact of Multipath Propagation on Bluetooth Demodulation Performance
• In FIG. 6, FIG. 7, FIG. 8, and FIG. 9, the simulated impact of multipath propagation on Bluetooth EDR demodulation performance is shown.
• For the 2-path multipath propagation model, the power of the second path is varied relative to the first arriving path. For the exponential multipath propagation model, the RMS delay spread is varied.
• In FIG. 6, one can see that even for Pi/4-DQPSK and a very small second path such as −15 dB (10 log₁₀(0.0313)), there is a degradation exceeding 3 dB already. For path larger than −9 dB (10 log₁₀(0.125)), successful demodulation is no longer possible independent of the SNR.
• In FIG. 7, one can see that even for D8PSK and a very small second path such as −15 dB (10 log₁₀(0.0313)), there is a degradation exceeding 12 dB (!) already. For path larger than −15 dB, successful demodulation is no longer possible independent of the SNR.
• In FIG. 8, one can see that for Pi/4-DQPSK and an RMS delay spread >250 ns, there is a degradation exceeding 3 dB.
• In FIG. 9, one can see that for D8PSK and an RMS delay spread >200 ns, there is a degradation exceeding 3 dB.
• It was also shown that even for very moderate multipath propagation, no reliable data transmission using Bluetooth technology is possible. That is due to the inter-symbol interference (ISI) introduced by multipath propagation. Current (state-of-the-art) Bluetooth receivers are not capable of mitigating the unfavorable impact of ISI on the data demodulation in Bluetooth.
• It is concluded that with current (state-of-the-art) Bluetooth receivers, no reliable data transmission is possible with regards to transmission ranges provided the transmission power of power class 1 devices.
SUMMARY OF THE INVENTION
• In order to solve the problems mentioned above, the present invention provides a method for reception of long-range signals in Bluetooth. The present invention processes Bluetooth signals with linear minimum mean square error (MMSE) frequency-domain equalization (FDE) in single carrier (SC) system using a Fourier Transform and provides long transmission range Bluetooth service with reliable data transmission.
• The present invention improves the performance of Bluetooth service based on power class 2 and 3 devices in multipath environment.
• The present invention provides FFT/IFFT-based MMSE SC FDE receiving mode for all Bluetooth transmission modes for low-complexity and high-performance
• The present invention is used in multi-standard devices in efficient implementation by reuse of the FFT/IFFT circuitry
• To achieve the purpose mentioned above, one embodiment of the present invention provides a method for reception of long-range signals in Bluetooth is for all transmission modes on power class 1, power class 2 and power class 3, the method comprising: receiving Bluetooth signals; and processing signals with linear frequency-domain equalization (FDE) in single carrier (SC) system using a Fourier Transform.
BRIEF DESCRIPTION OF THE DRAWINGS
• The file of this patent contains at least one drawing executed in color. Copies of this patent with color drawing(s) will be provided by the Patent and Trademark Office upon request.
• The foregoing aspects and many of the accompanying advantages of this invention will becomes more readily appreciated as the same becomes better understood by reference to the following detailed description, when taken in conjunction with the accompanying drawings, wherein:
• FIG. 1 illustrating BER versus SNR for PI/4-DQPSK in AWGN;
• FIG. 2 illustrating BER versus SNR for D8PSK in AWGN;
• FIG. 3 illustrating the pathloss versus transmission range in a Bluetooth network;
• FIG. 4 illustrating 2-path multipath propagation model for Bluetooth long transmission range applications;
• FIG. 5 illustrating the scenario of multipath propagation in large office;
• FIG. 6 illustrating the BER performance for Pi/4-DQPSK in 2-Path multipath;
• FIG. 7 illustrating the BER performance for D8PSK in 2-path multipath;
• FIG. 8 illustrating the BER performance for Pi/4-DQPSK in exponential multipath;
• FIG. 9 illustrating the BER performance for D8PSK in exponential multipath;
• FIG. 10 illustrating the processing flow of h yielding H_inv in SC linear MMSE BLE according to one embodiment of the present invention;
• FIG. 11 illustrating h of exponential channel (left) and corresponding h_inv using M=64 (right);
• FIG. 12 illustrating the Pole-Zero Diagrams of h of exponential channel (left) and corresponding h_inv using M=64 (right);
• FIG. 13 illustrating the constellation diagrams for noiseless QPSK before equalization (left) and after equalization (right);
• FIG. 14 illustrating the constellation diagrams for noiseless QPSK before equalization (left) and after equalization (right);
• FIG. 15 illustrating the performance of Pi/4-DQPSK and FDE in 2-path multipath (varying N);
• FIG. 16 illustrating the performance of D8PSK and FDE in 2-path multipath (N=64);
• FIG. 17 illustrating the performance of D8PSK and FDE in 2-path multipath (varying N);
• FIGS. 18A, 18B and 18C illustrating the functional block diagram of Bluetooth transmitter;
• FIG. 19 illustrating the functional block diagram of radio channel model for Bluetooth transmission: introduction of multipath fading and additive White Gaussian Noise (AWGN); and
• FIGS. 20A and 20B illustrating the functional Block Diagram of Bluetooth Receiver according to one embodiment of the present invention.
DETAILED DESCRIPTION OF THE INVENTION
• As an invention, the use of equalization is proposed for Bluetooth data communication.
• While ISI mitigation by equalization is well-known in state-of-the-art digital wireless communications engineering, this invention proposes the use of a new FDE adapted to SC transmission without a GI or CP. The proposed FDE very successfully mitigates ISI while being very implemention-friendly.
• The resulting performance of long transmission range Bluetooth service based on power class 1 devices using BlueWARP technology is beyond state-of-the-art Bluetooth service based on power class 1 devices.
• In the following, a generalized system model is introduced. The model is similar to the one proposed in [Klein].
• At the transmit side, a block (vector) d of data of length N is formed in Equation 15:
• 
  d=(d ₁ ,d ₂ . . . d _N)^T  (15)
• Any coding, modulation or spreading is assumed to be included in d already. The data block is transmit through a channel characterized by its impulse response h in Equation 16:
• 
  h=(h ₁ ,h ₂ . . . h _w)^T  (16)
• The convolution of d and h is expressed in matrix notation using the matrix H in Equation 17:
• 
  H=(H _i,v), i=1 . . . N+W−1, v=1 . . . N  (17)
• with Equation 18:
• $\begin{array}{cc}{H}_{i,v}=\{\begin{array}{cc}{h}_{i-\mathrm{<figure-callout\; id="1"\; label="v\; +"\; filenames="US20110026578A1-20110203-D00002.png,US20110026578A1-20110203-D00006.png"\; state="\{\{state\}\}">v</figure-callout>}+1}& 1\le i-\mathrm{<figure-callout\; id="1"\; label="v\; +"\; filenames="US20110026578A1-20110203-D00002.png,US20110026578A1-20110203-D00006.png"\; state="\{\{state\}\}">v</figure-callout>}+1\le \mathrm{<figure-callout\; id="0"\; label="W"\; filenames="US20110026578A1-20110203-D00003.png,US20110026578A1-20110203-D00011.png"\; state="\{\{state\}\}">W</figure-callout>}\\ 0& \mathrm{else}\end{array}& \left(18\right)\end{array}$
• The received signal r is given by Equation 19:
• $\begin{array}{cc}\begin{array}{c}r={\left({\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="US20110026578A1-20110203-D00002.png,US20110026578A1-20110203-D00006.png"\; state="\{\{state\}\}">r</figure-callout>}}_1,{\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="US20110026578A1-20110203-D00002.png,US20110026578A1-20110203-D00006.png"\; state="\{\{state\}\}">r</figure-callout>}}_2\dots r_{N+W-1}\right)}^T\\ =H·d+n\end{array}& \left(19\right)\end{array}$
• where n denotes an additive white Gaussian noise sequence with zero mean and covariance matrix R_nn.
• Using (block) linear equalization technique for SC systems, an estimate of the transmit data is obtained using one of the following criteria. Equation 20: Matched Filter (MF) Criterion
• 
  {circumflex over (d)} _MF =H ^H ·r  (20)
Equation 21: Zeros Forcing (ZF) Criterion
• 
  {circumflex over (d)} _ZF=(H ^H ·H)⁻¹ ·H ^H ·r  (21)
Equation 22: Minimum Mean Square Error (MMSE) Criterion
• 
  {circumflex over (d)} _MMSE=(H ^H ·H+σ ²)⁻¹ ·H ^H ·r  (22)
• Typically, the MMSE criterion yields superior results. Therefore, only the MMSE criterion is pursued. Nevertheless, all newly proposed receiver architectures are applicable as well to MF or ZF equalization.
• Single Carrier Linear MMSE Frequency-Domain Equalization using FFT without Guard Period
• In order to avoid complex receiver processing tasks such as Cholesky decomposition for solving Equation 22, the (block) linear MMSE equalization for SC systems can be performed efficiently in frequency domain expressed in Equation 23 and Equation 24:
• $\begin{array}{cc}{\hat{d}}_{\mathrm{MMSE}}=F^{-1}\left\{H_{\mathrm{inv}}·F\left(r\right)\right\}& \left(23\right)\\ H_{\mathrm{inv}}=\frac{{\left(F\left\{\hat{h}\right\}\right)}^{*}}{{\left(F\left\{\hat{h}\right\}\right)}^{*}·F\left\{\hat{h}\right\}+{\sigma }^2}& \left(24\right)\end{array}$
• where F denotes the Discrete Fourier Transform (DFT), F⁻¹ denotes the Inverse Discrete Fourier Transform (IDFT).
  ĥ refers to an estimated channel impulse response. The ĥ is obtained by a separate processing step typically called channel estimation.
  H_inv represents the frequency response of the propagation channel being inverted using the MMSE criterion. Its time-domain equivalent is given by h_inv.
• For actual implementations, DFT and IDFT are realized by Fast Fourier Transform (FFT) and Inverse Fast Fourier Transform (IFFT), respectively.
• H_inv in Equation 23 can be interpreted as the frequency response of a transversal linear equalizer filter which time (impulse) response has to be convolved with the received signal r. The filter can be categorized as an IIR filter. Therefore, the filter length is infinite. However, a length can be defined which contains most of the large coefficients and neglects small coefficients. The length of this approximated equalizer filter is denoted by L_eq.
• The length of the received data blocks varies significantly depending on packet type and service. A fixed-length FFT/IFFT implementation based on the maximum data block length N of all packet types and services to be integrated in the receiver architecture would not be efficient. Also, N can be rather large (>2̂13) which would require to implement a very large FFT (>8 k). However, it is well-known that convolution (e.g. filtering) operations for continuous data streams or long blocks of data can be implemented efficiently using overlap-add-technique (OAT) FFT or overlap-save-technique (OST) FFT algorithms. The further description focuses on OAT FFT.
• As suggested in [Falconer], FDE for SC systems requires a GI to be inserted at the transmitter. The following method, however, allows to apply FFT-based FDE as well for systems without GI.
• An M-point FFT (M=8, 16, 32, 64) is assumed. For every M-point FFT/IFFT based convolution operation, a length M-L_eq output data block is generated. The start index within the data block is advanced by M-L_eq samples per FFT-IFFT operation.
• The single FFTs/IFFTs overlap by L_eq samples. Therefore, L_eq<M/2 must hold. For such short FFT/IFFT sizes, the approximated equalizer filter must be limited which can be accomplished either by circular convolution with a rectangular window transformed into frequency domain RW (see equation 26) or by multiplication with a rectangular window rw in time domain (see equation 25). The latter approach requires one additional frequency-time and time-frequency conversion.
• The extended versions of Equation 23 are given below:
• $\begin{array}{cc}{\hat{d}}_{\mathrm{MMSE}}=F^{-1}\left\{F\left\{F^{-1}\left\{\frac{{\left(F\left\{\hat{h}\right\}\right)}^{*}}{{\left(F\left\{\hat{h}\right\}\right)}^{*}·F\left\{\hat{h}\right\}+{\sigma }^2}\right\}·\mathrm{rw}\right\}·F\left(r\right)\right\}& \left(25\right)\\ {\hat{d}}_{\mathrm{MMSE}}=F^{-1}\left\{\left(\frac{{\left(F\left\{\hat{h}\right\}\right)}^{*}}{{\left(F\left\{\hat{h}\right\}\right)}^{*}·F\left\{\hat{h}\right\}+{\mathrm{<figure-callout\; id="2"\; label="\sigma "\; filenames="US20110026578A1-20110203-D00002.png,US20110026578A1-20110203-D00006.png"\; state="\{\{state\}\}">\sigma </figure-callout>}}^2}\otimes \mathrm{RW}\right)·F\left(r\right)\right\}& \left(26\right)\end{array}$
• 

  denotes circular convolution.
• Also, h_inv must be shifted into the correct position. Performing this operation in frequency domain corresponds to rotating H_inv with phasors having an angle increasing with every sample of H_inv
• In FIG. 10, the entire processing flow of ĥ yielding H_inv is depicted:
• S01: FFT on estimated channel impulse response
• S02: conjugate complex operation on Ĥ
• S03: Multiplication of Ĥ with conj(Ĥ)
• SO4: Addition of Ĥ·H* with σ²
• S05: Division of Ĥ* by Ĥ·H*+σ²
• S06: Multiplication with phasors
• $1·{}^{j\frac{k}{M}}$
• S07: Circular convolution with
• $\frac{\sin x}{x}$
SC Linear MMSE FDE
• In FIG. 11, h of a noiseless exponential multipath channel without fading (upper subplot) and with fading (lower subplot) is shown. The channel parameter RMS delay spread s was set to s=150 ns. In addition, one can see the corresponding h_inv which was obtained by converting H_inv from Equation 24 back to time domain.
• In order to apply OAT for equalization, h_inv has to be shortened to the overlap length M/2. This shortened h_inv is constructed using the last quarter of samples of h_inv and appending the first quarter of samples of h_inv to it.
• In FIG. 12, one can see the impact of the minimum-phase/non-minimum phase character of the non-fading/fading exponential channel on the pole-zero diagrams of h and h_inv.
• FIG. 13, visualizes the successful outcome of the described equalization process by comparing noiseless QPSK constellations before and after SC linear MMSE FDE. In non-faded multipath conditions, the equalized constellation appears again to be perfect. In multipath fading conditions, however, some noise-like interference remains.
• Bluetooth Demodulation Performance in Multipath Propagation using SC Linear MMSE FDE
• In FIG. 14, FIG. 15, FIG. 16 and FIG. 17, the Bluetooth EDR demodulation performance using SC linear MMSE FDE is demonstrated. In addition, the simulated impact of multipath propagation on Bluetooth EDR demodulation performance is shown.
• For the 2-path multipath propagation model, the power of the second path is varied relative to the first arriving path.
• In FIG. 14, one can see that for Pi/4-DQPSK and severe multipath conditions (second path as high as −3 dB (10 log₁₀(0.5)), the degradation in demodulation performance is limited to 3 dB. The FFT size Mapplied equals 64.
• In FIG. 15, one can see that for Pi/4-DQPSK and a strong second path (−6 dB), M=128 and M=32 perform as well as M=64. Even M=16 performs within 1 dB of the optimum performance (M=128). Very small FFT sizes (M<16) cause servere degradation in equalization (and therefore demodulation) performance.
• In FIG. 16, one can see that for D8PSK and severe multipath conditions (second path as high as −3 dB (10 log₁₀(0.5)), the degradation in demodulation performance is limited to 3 dB. The FFT size Mapplied equals 64.
• In FIG. 17, one can see that for D8PSK and a strong second path (−6 dB), M=128 and M=32 perform as well as M=64. Even M=16 performs within 2 dB of the optimum performance (M=128). Very small FFT sizes (M<16) cause severe degradation in equalization (and therefore demodulation) performance.
• It was shown that for Bluetooth SC linear MMSE FDE can be used to efficiently mitigate severe ISI introduced by multipath propagation (second path as high as −3 dB (10 log₁₀(0.5)). In addition, it was demonstrated that using FFT sizes as small as M=16 still allow for equalization (and therefore demodulation) performance within 2 dB of the optimum performance using M=128.
• Integration with a Bluetooth Receiver
• In this section, it is described how to integrate the SC linear MMSE FDE into a Bluetooth receiver. The integration is described on a conceptual system level.
• The SC linear MMSE FDE is assumed to be used for EDR only. However, it can also be used for Basic Rate without modifications.
• In FIG. 18A, FIG. 18B and FIG. 18C, a (simplified) functional block diagram of a Bluetooth transmitter according to [BT SIG EDR] is shown:
• • FIG. 18A
    • The packet header and the packet data (payload) undergo bitstream processing as described in Volume 2 Core System Package, Part B Baseband Specification, Chapter 7: Bitstream Processing (encryption is not shown).
    • Bitstream-processed packet header and access code are multiplexed as described in Volume 2 Core System Package, Part B Baseband Specification, Chapter 6: Packets.
    • Bitstream-processed and multiplexed packet header and access code are GFSK modulated as described in Volume 2 Core System Package, Part A Radio Specification, Chapter 3: Transmitter Characteristics.
  • FIG. 18B
    • Bitstream-processed packet data (payload) is switched between either Basic Rate or EDR processing:
      • Basic Rate: Bitstream-processed packet data (payload) is GFSK modulated
      • EDR: Sync sequence and trailer are multiplexed with bitstream-processed packet data (payload)
      • EDR: Multiplexed sync sequence/trailer//bitstream-processed packet data (payload) is switched between Pi/4-DQPSK modulation or D8PSK modulation
      • EDR: Multiplexed and modulated sync sequence/trailer//bitstream-processed packet data (payload) is multiplexed with guard
      • EDR: guard and multiplexed and modulated sync sequence/trailer//bitstream-processed packet data (payload) is filtered upsampled (US) with root-raised-cosine (RRC) filter as described in Volume 2 Core System Package, Part A Radio Specification, Chapter 3: Transmitter Characteristics
  • FIG. 18C
    • Processed access code, packet header and packet data (payload) is multiplexed forming a complete transmit packet
• In FIG. 19, a functional block diagram of the radio channel model used for Bluetooth transmission is depicted. It introduces multipath fading according to the model described in prior art. Also, it introduces Additive White Gaussian Noise (AWGN).
• In FIG. 20A and FIG. 20B, a (simplified) functional block diagram of a Bluetooth receiver is shown:
• • FIG. 20A
    • A de-multiplxer 101 demultiplexing packet header and packet data (payload) (access code-related processing is not shown)
    • Packet data (payload) is switched by a switch 102 between either Basic Rate or EDR processing:
      • Basic Rate: packet data (payload) is processed by a FFT-based equalizer 105, and then demodulated by a GFSK demodulator 108.
      • Basic Rate: optional equalization of packet header using SC linear MMSE FDE (estimation of channel impulse response not shown).
      • Basic Rate: GFSK demodulation of packet header by a GFSK demodulator 107.
      • EDR: Packet data (payload) is filtered downsampled (DS) with root-raised-cosine (RRC) filter 103 (guard, sync sequence and trailer processing is not shown).
      • EDR: filtered packet data (payload) is switched by a switch 104 between Pi/4-DQPSK demodulation or D8PSK demodulation.
      • EDR: equalization of packet data (payload) using SC linear MMSE FDE (estimation of channel impulse response not shown)
      • EDR: Pi/4-DQPSK demodulation or 8-DPSK demodulation of packet data (payload) by a Pi/4-DQPSK demodulator 109 or an 8-DPSK demodulator 110.
  • FIG. 20B
    • EDR: reverse bitstream-processing on packet data (payload)
• The key in the description of the (simplified) processing flow in a Bluetooth receiver applying BlueWARP technology is the positioning of SC linear MMSE FDE directly before Pi/4-DQPSK demodulation or D8PSK demodulation.
• Accordingly, one of features of the present invention is to provide a method for reception of long-range signals in Bluetooth. The method has outstanding performance of long transmission range Bluetooth service based on power class 1 devices and performance improvement of Bluetooth service based on power class 2 and 3 devices in multipath environment.
• Accordingly, the Low-complexity/high-performance FFT/IFFT-based MMSE SC FDE receiver architecture is used for all Bluetooth transmission modes and has highly efficient implementation by reuse of the FFT/IFFT circuitry in the context of multi-standard devices,
• Although the present invention has been explained in relation to its preferred embodiment, it is to be understood that other modifications and variation can be made without departing the spirit and scope of the invention as hereafter claimed.

Claims (13)

1. A method for reception of long-range signals in Bluetooth for all transmission modes on power class 1, power class 2 and power class 3, the method comprising:

receiving Bluetooth signals; and

processing said Bluetooth signals with linear frequency-domain equalization (FDE) in single carrier (SC) system using a Fourier Transform.

2. The method for reception of long-range signals in Bluetooth according to claim 1, wherein said Fourier Transform is a Discrete Fourier Transform (DFT) deduced from a Fast Fourier Transform (FFT).

3. The method for reception of long-range signals in Bluetooth according to claim 2, wherein said Fast Fourier Transform (FFT) is overlap-add-technique FFT or overlap-save-technique FFT.

4. The method for reception of long-range signals in Bluetooth according to claim 1, wherein said Fourier Transform is an Inverse Discrete Fourier Transform deduced from an Inverse Fast Fourier Transform (IFFT).

5. The method for reception of long-range signals in Bluetooth according to claim 4, wherein said Inverse Fast Fourier Transform (IFFT) is overlap-add-technique IFFT or overlap-save-technique IFFT.

6. The method for reception of long-range signals in Bluetooth according to claim 1, further comprising executing an estimate of the transmit data after receiving said Bluetooth signals.

7. The method for reception of long-range signals in Bluetooth according to claim 6, wherein said estimate of the transmit data is obtained using Matched Filter (MF) criterion.

8. The method for reception of long-range signals in Bluetooth according to claim 6, wherein said estimate of the transmit data is obtained using Zero Forcing (ZF) criterion.

9. The method for reception of long-range signals in Bluetooth according to claim 6, wherein said estimate of the transmit data is obtained using Minimum Mean Square Error (MMSE) criterion.

10. The method for reception of long-range signals in Bluetooth according to claim 9, wherein said minimum mean square error (MMSE) is obtained from

{circumflex over (d)} _MMSE =F ⁻¹ {H _inv ·F(r)}

and H_inv is obtained from

$H_{\mathrm{inv}}=\frac{{\left(F\left\{\hat{h}\right\}\right)}^{*}}{{\left(F\left\{\hat{h}\right\}\right)}^{*}·F\left\{\hat{h}\right\}+{\sigma }^2}$

11. The method for reception of long-range signals in Bluetooth according to claim 10, wherein said F and F⁻¹ size is 8 or 16.

12. The method for reception of long-range signals in Bluetooth according to claim 9, wherein said minimum mean square error (MMSE) comprises:

using FFT on estimated channel impulse response ĥ yielding Ĥ;

conjugating complex operation on Ĥ;

multiplying of Ĥ with conj(Ĥ);

adding of Ĥ·H* with σ²;

dividing of Ĥ* by Ĥ·H*+σ²;

multiplying with phasors

$1·{}^{j\frac{k}{M}};$

and

circulating circular convolution with

$\frac{\sin x}{x}.$

13. The method for reception of long-range signals in Bluetooth according to claim 1, wherein said transmission modes comprises 1 Mbps transmission mode (GFSK), 2 Mbps transmission mode (DPSK), and 3 Mbps transmission mode (DPSK).

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