US20100135117A1

US20100135117A1 - Downhole Timing Recovery and Signal Detection

Info

Publication number: US20100135117A1
Application number: US12/618,906
Authority: US
Inventors: John G McRory
Original assignee: Xact Downhole Telemetry Inc
Current assignee: Baker Hughes Oilfield Operations LLC
Priority date: 2008-11-28
Filing date: 2009-11-16
Publication date: 2010-06-03
Also published as: CA2685688A1

Abstract

The present invention relates to telemetry apparatus and methods, and more particularly to acoustic telemetry apparatus and methods used in the oil and gas industry. More specifically, the invention relates to a method for enhancing a received signal transmitted by acoustic telemetry through a drill string by modifying the received signal by a multiplication of the received signal with a second waveform.

Description

RELATED APPLICATIONS

This application is related to and claims priority from U.S. Provisional Patent Application No. 61/118,501, filed Nov. 28, 2008, the entire contents of which are incorporated herein by reference for all purposes.

FIELD OF THE INVENTION

BACKGROUND OF THE INVENTION

As is known, acoustic telemetry is a method of communication in the well drilling and production industry. In a typical drilling environment, acoustic carrier waves from a downhole acoustic telemetry device are modulated in order to carry information via the drillpipe to the surface. Upon arrival at the surface, the waves are detected, decoded and displayed in order that drillers, geologists and others helping steer or control the well are provided with drilling and formation data.
In a typical acoustic telemetry drilling or production environment acoustic waves are produced and travel predominantly along the metal wall of the tubing(s) associated with the downhole section required to drill the well. The acoustic energy is usually detected by sensitive accelerometers and sometimes by relatively less sensitive strain gauges. Care needs to be taken about the positioning and coupling of such devices to the tubing in order that the maximum signal energy can be obtained in order to optimize the detection system's signal to noise ratio (SNR).
The theory of acoustic telemetry as applied to communication along drillstrings has a long history, and a comprehensive theoretical understanding has been achieved and backed up by accurate measurements. It is now generally recognized that the nearly regular periodic structure of drillpipe sections and the inter-section connections imposes a passband/stopband structure on the frequency response, similar to that of a comb filter. A typical passband/stopband frequency response for drill pipe sections is shown in FIG. 1 where multiple ranges of frequencies cannot be transmitted along the drill pipe sections.
Further still, other factors including significant surface and downhole noise from drilling, dispersion, phase non-linearity and frequency dependent attenuation as well as passband effects all contribute to the challenge of effectively enabling acoustic telemetry through the drillpipe.
Thus, as a result of the passband/stopband structure, the acoustic signal transmitted up the drillpipe to the surface must be limited in frequency bandwidth such that its frequencies fall within one or simultaneously two or more of the frequency passbands within the passband/stopband structure.
However, the main challenge in effective transmission up the drillpipe is the frequency dependant destructive interference of the signals within these passbands. This interference arises from both the effect of the regular structure of the drill string (Drumheller), and signal reflections local to the acoustic transmitter. These reflections arise from the different acoustic impedances offered to the signal by the various parts of the bottom hole assembly. The combination of the fine structure of the passband with the destructive interference of the transmitted signal by local reflections can result in narrow notches within the passband with attenuations of 10 dB or greater.
Several different digital modulation schemes have been proposed for this transmission, including Binary Phase Shift Keying (BPSK) and Phase Shift Keying (PSK) however one of the most effective is the use of a sinusoidal linear frequency chirp falling within the passband.
A linear chirp is a constant amplitude signal that begins at one frequency and then over a period of time sweeps to another in a linear fashion. From its starting point, the frequency can either increase forming an “up-chirp”, or decrease forming a “down-chirp”. The use of linear chirps in both radar and sonar applications is well established. In these applications the chirp offers both an improvement in signal to noise ratio and improved ranging resolution through pulse compression resulting from the use of the cross-correlation of the received signal with a reference chirp waveform in the receiver. A detailed explanation of the cross-correlation and its application in digital receivers can be found in Digital Communications, third edition by John G. Proakis (p. 65, pp. 235-238).
In the cases of sonar and radar applications, the cross-correlation operation is used to detect a short waveform in a longer data record by sliding the reference waveform through the received data one point at a time and carrying out a correlation calculation at each data point. The resulting output data record contains the autocorrelation waveform of the transmitted chirp in which most of the transmitted energy has been concentrated into a main lobe that has a much higher peak value and shorter duration than the transmitted chirp, leading to better range resolution and an improved SNR.
These aspects of the chirp's auto-correlation also work to advantage in downhole acoustic telemetry. The increase in the signal to noise ratio improves the system performance, while the narrower waveform peak aids in timing recovery and signal detection. An additional advantage is that the chirp effectively fills the available frequency band while sweeping through the notches in the passband.
However, these aspects are also an inverse function of the time-bandwidth product, or the product of the duration and the frequency span of the chirp. If the duration of the chirp is decreased by increasing the baud rate, or if the frequency span of the chirp is reduced, then the main lobe of the auto-correlation function widens, reducing the improvement in SNR and hence the improvement in signal detection.
The bandpass filter properties of the downhole acoustic channel limit the frequency span available for the chirp placing a limit on the time bandwidth product. Also, the market need for increasing baud rates requires a reduction in the chirp duration for each increment in baud rate, placing a further constraint on the time bandwidth product. This combination of constraints forms a boundary on the available performance of the acoustic telemetry system by limiting the shape of the auto-correlation waveform, making it more difficult to demodulate and detect the telemetry signal at higher data rates.
As a result, there has been a need for a system and method in which the above problems are addressed. In particular, there has been a need for a system and method in which detection of the acoustic signal at the surface and system synchronization of the acoustic signal is improved for improved timing recovery.

SUMMARY OF THE INVENTION

In accordance with the invention, a method that improves the detection and synchronization of an acoustic signal and that increases the operating baud rate by modifying the time-bandwidth product of the transmitted linear acoustic chirp at the receiver is provided.
More specifically, the invention provides a method for enhancing a received signal transmitted by acoustic telemetry through a drill string by modifying the received signal by a multiplication of the received signal with a second waveform.
In various embodiments, the received acoustic signal is a linear frequency chirp and/or the received acoustic signal is a linear frequency upchirp and the second waveform is a linear frequency downchirp.
In a more specific embodiment, the received acoustic signal is a linear frequency downchirp and the second waveform is a linear frequency upchirp.
In another embodiment, an autocorrelation function of the received signal is calculated and the autocorrelation function is optimized to compensate for limited chirp bandwidth. In another embodiment, the autocorrelation function is optimized to remove dispersion effects.
In another aspect, the invention provides a method of enhancing a received chirp signal within a receiver wherein the received chirp signal has been transmitted by acoustic telemetry through a drill string, comprising the step of: a) applying a non-constant frequency local oscillator signal to the received chirp signal to selectively shift component frequencies of the received chirp signal by spreading the received chirp in the frequency domain while maintaining baud rate in order to create a processed signal having an increased time-bandwidth.
In another embodiment, the non-constant frequency local oscillator is a linear chirp waveform.
In yet another embodiment, step a) includes adjusting an autocorrelation waveform to a desired form during down-conversion in the receiver.
In another embodiment, the local oscillator signal is a down chirp opposite in frequency span to the received chirp signal.
In a still further embodiment, the frequency span of the down chirp is chosen to obtain a desired correlation waveform.
In yet another embodiment, the processed signal is subjected to multiple frequency sweeps and the time-bandwidth is increased with each frequency sweep and wherein the frequency sweeps are limited to twice the received chirp's frequency sweep.

BRIEF DESCRIPTION OF THE FIGURES

The invention is described with reference to the accompanying figures in which:

FIG. 1 is a graph showing frequency response (signal strength vs. frequency) in a typical drill pipe section;

FIG. 2 shows a typical heterodyne receiver structure that can be used to shift the frequency of a bandpass linear chirp signal to base band through the use of a local oscillator (LO);

FIG. 3 is a graph showing a typical linear chirp autocorrelation;

FIG. 4 is a graph showing autocorrelation as a function of baud rate; and

FIG. 5 is a graph showing autocorrelation for a 20 baud rate with different local oscillator.

DETAILED DESCRIPTION

Overview

With reference to the figures, systems and methods of improved acoustic telemetry are described.
As noted above, the unique transmission characteristics of the down-hole environment within the available acoustic channels of a typical drill pipe string create an extremely difficult digital communications environment given that the frequency channel is a narrow, highly dispersive band-pass system with substantial levels of echoes, reverberation and attenuation.
As noted above, one digital communication scheme that has been used with success in this environment is a BPSK or PSK signal modulated on a linear chirp that is centered on one of the passbands of the channel. A working system using this approach is described in detail by Camwell and Neff in “Field Test Results of an Acoustic MWD System”.
The receiver in this system employs a correlation demodulator as described by Proakis pp: 234-238. In this case the demodulator correlates the received data stream with a reference wave form that has been derived from an ideal linear chirp. In the simplest case for a binary phase shift keying a single reference waveform is used, and the output of the correlation operation in the presence of a received chirp is either a positive or negative correlation peak. If a chirp is not present, then the output of the correlation operation is simply the correlation between the reference waveform and the in-band channel noise. Since the reference waveform is derived from an ideal linear chirp then the correlation between the reference and the received signal is effectively the autocorrelation waveform of the chirp.
However, the transmission channel reduces the effectiveness of the correlation receiver by limiting and distorting the autocorrelation waveform. In accordance with the invention, a method is described by which the autocorrelation function of the received signal at surface may be modified to compensate for the limited chirp bandwidth as well as to remove the effects of dispersion, and thereby optimize the performance of the receiver.
A common heterodyne receiver structure such as shown in FIG. 2 can be used to shift the frequency of the bandpass linear chirp signal to base band through the use of a local oscillator (LO). The LO waveform may comprise a simple sinusoid, a complex sinusoid or perhaps a more complex signal. The receiver as shown in FIG. 2 may be implemented in hardware, software or a combination thereof.
More specifically, a finite linear chirp modulating a carrier frequency f_o, of a duration T, and a chirp frequency rate a can be represented by
f(t)=exp(j2π(f ₀ t+0.5αt ²)), 0≦t≦T (1)
The instantaneous frequency of the chirp is the first time derivative of the phase in (1), 2π(f_o+αt). Thus, the frequency bandwidth of the chirp is αT over the chirp duration T, and that the time-bandwidth product of the chirp is αT².
The autocorrelation function for the linear chirp can be written as
$\begin{matrix} acf (τ) = Φ (τ) (T - \langle τ \rangle) \frac{\sin (π ατ (T - \langle τ \rangle))}{(πατ (t - \langle τ \rangle))} & (2) \end{matrix}$
Where in (2) Φ is the carrier harmonic that is modulated by both a triangle pulse (T−|τ|), and a sinc function of time. Examination of (2) shows that the autocorrelation function can be completely characterized by the frequency chirp rate α and the chirp duration T, and that the first root of the sinc function occurs at approximately at τ=1/αT so the width of the main lobe of the waveform is an inverse function of the time-bandwidth product as determined by the baud rate and frequency sweep.
In most linear chirp applications, the time/bandwidth product of the chirp is chosen such that its autocorrelation waveform displays a narrow peak with well controlled sidelobes. As an example, the autocorrelation waveform for αT²=40 is shown in FIG. 3. The well defined peak combined with the high peak to average ratio make the chirp an ideal waveform for ranging applications.
However, the downhole acoustic channel limits the frequency span of the chirp to tens of Hz. In the case of the third passband of the down-hole channel as shown in FIG. 1, the frequency passband encompasses 550 to 720 Hz. Assuming a symbol rate of T=0.2, 0.1 or 0.05 sec for 5, 10 or 20 baud, with a chirp frequency span of 40 Hz, results in a time/bandwidth product of 1.6, 0.4 and 0.1 respectively.
FIG. 4 shows the evolution of the baseband autocorrelation waveform for a linear chirp with a frequency span of 640 to 680 Hz at different baud rates at a normalized sample rate (equal number of samples per chirp). In this case a fixed frequency downconversion and lowpass filtering is used to generate the baseband chirp. The figure shows the increase in the width of the main lobe of the autocorrelation as the time-frequency product reduces with increasing baud rate until the limiting triangle is reached. As the time/bandwidth product drops, the peak to average ratio of the waveform is reduced, thereby also reducing the receiver's immunity to noise in the timing recovery and symbol detection. The difficulty arising in the downhole channel is that since the frequency span is limited by the channel, the autocorrelation waveform is determined by the baud rate. However as the baud rate increases, the autocorrelation waveform will reduce to a simple triangle pulse, thereby losing the advantages of the linear chirp carrier.
In accordance with the invention, the autocorrelation waveform is restored to its desired form is by using a non-constant frequency local oscillator in the downconversion stage of the receiver. More specifically, the linear chirp waveform is used for the local oscillator in order to selectively shift the component frequencies of the received chirp, thereby increasing the time-bandwidth product by spreading the received chirp in the frequency domain while maintaining the baud rate.
In order to achieve the frequency spreading operation, the local oscillator chirp must be opposite in its frequency span to received chirp. For example, for a received linear up-chirp of 640 to 680 Hz, the corresponding LO chirp must be a down-chirp with the same time period as the received chirp. The frequency span of the downchirp is chosen to obtain the desired correlation waveform.
As an example, FIG. 5 shows the effect of four different local oscillator signals on a 20 baud received signal using a 640-680 Hz linear chirp. Examination of the figure shows that that the time bandwidth product is increased with each increase in the frequency sweep of the local oscillator down chirp, and that diminishing returns limit the maximum LO sweep to twice the chirp's frequency sweep.
Although the present invention has been described and illustrated with respect to preferred embodiments and preferred uses thereof, it is not to be so limited since modifications and changes can be made therein which are within the full, intended scope of the invention as understood by those skilled in the art.

REFERENCES

Thomas G. Barnes and Bill R Kirkwood, “Passbands for Acoustic Transmission in an Idealized Drill String”, The Journal of the Acoustical Society of America, Vol. 51, Number 5, 1972, pp. 1606-1608.
Douglas S Drumheller, “Acoustical properties of drill strings”, The Journal of the Acoustical Society of America, Vol. 85, Number 3, March 1989, pp. 1048-1064.
John C. Burgess, “Chirp design for acoustical system identification”, The Journal of the Acoustical Society of America, Vol. 91, Number 3, March, 1992, pp. 1525-1530.
Coert Olmsted, Alaska SAR Facility, Scientific SAR User's Guide, July, 1993, pp. 5-6.
Douglas S Drumheller, “Attenuation of sound waves in drill strings”, The Journal of the Acoustical Society of America, Vol. 94, Number 4, October 1993, pp. 2387-2396.
Douglas S. Drumheller, “The propagation of sound waves in drill strings”, The Journal of the Acoustical Society of America, Vol. 97, Number 4, October 1995, pp. 2116-2125.
J. M. Neff and P. L. Camwell, “Field-Test Results of an Acoustic MWD System”, Proceedings of the SPE/IADC Drilling Conference, February, 2007.
John G. Proakis, Digital Communications, Third Edition, McGraw-Hill Inc. 1995.

Claims

1. A method for enhancing a received signal transmitted by acoustic telemetry through a drill string comprising the step of modifying the received signal by a multiplication of the received signal with a second waveform.

2. The method of claim 1 where the received acoustic signal is a linear frequency chirp.

3. The method of claim 2, where the received acoustic signal is a linear frequency upchirp and the second waveform is a linear frequency downchirp.

4. The method of claim 2, where the received acoustic signal is a linear frequency downchirp and the second waveform is a linear frequency upchirp.

5. The method of claim 2 wherein an autocorrelation function of the received signal is calculated and the autocorrelation function is optimized to compensate for limited chirp bandwidth.

6. The method of claim 5 wherein the autocorrelation function is optimized to remove dispersion effects.

7. A method of enhancing a received chirp signal within a receiver wherein the received chirp signal has been transmitted by acoustic telemetry through a drill string, comprising the step of:

applying a non-constant frequency local oscillator signal to the received chirp signal to selectively shift component frequencies of the received chirp signal by spreading the received chirp in the frequency domain while maintaining baud rate in order to create a processed signal having an increased time-bandwidth.

8. The method as in claim 7 wherein the non-constant frequency local oscillator is a linear chirp waveform.

9. The method as in claim 7 wherein step a) includes adjusting an autocorrelation waveform to a desired form during down-conversion in the receiver.

10. The method as in claim 7 wherein the local oscillator signal is a down chirp opposite in frequency span to the received chirp signal.

11. The method as in claim 8 wherein the local oscillator signal is a down chirp opposite in frequency span to the received chirp signal.

12. The method as in claim 9 wherein the local oscillator signal is a down chirp opposite in frequency span to the received chirp signal.

13. The method as in claim 10 wherein the frequency span of the down chirp is chosen to obtain a desired correlation waveform.

14. The method as in claim 7 wherein the processed signal is subjected to multiple frequency sweeps and the time-bandwidth is increased with each frequency sweep and wherein the frequency sweeps are limited to twice the received chirp's frequency sweep.

15. The method as in claim 8 wherein the processed signal is subjected to multiple frequency sweeps and the time-bandwidth is increased with each frequency sweep and wherein the frequency sweeps are limited to twice the received chirp's frequency sweep.

16. The method as in claim 9 wherein the processed signal is subjected to multiple frequency sweeps and the time-bandwidth is increased with each frequency sweep and wherein the frequency sweeps are limited to twice the received chirp's frequency sweep.

17. The method as in claim 13 wherein the processed signal is subjected to multiple frequency sweeps and the time-bandwidth is increased with each frequency sweep and wherein the frequency sweeps are limited to twice the received chirp's frequency sweep.