US10551469B2 - Calibration of inverting amplifier based impedance analyzers - Google Patents
Calibration of inverting amplifier based impedance analyzers Download PDFInfo
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- US10551469B2 US10551469B2 US15/445,762 US201715445762A US10551469B2 US 10551469 B2 US10551469 B2 US 10551469B2 US 201715445762 A US201715445762 A US 201715445762A US 10551469 B2 US10551469 B2 US 10551469B2
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- G—PHYSICS
- G01—MEASURING; TESTING
- G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
- G01R35/00—Testing or calibrating of apparatus covered by the other groups of this subclass
- G01R35/005—Calibrating; Standards or reference devices, e.g. voltage or resistance standards, "golden" references
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- G—PHYSICS
- G01—MEASURING; TESTING
- G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
- G01R27/00—Arrangements for measuring resistance, reactance, impedance, or electric characteristics derived therefrom
- G01R27/02—Measuring real or complex resistance, reactance, impedance, or other two-pole characteristics derived therefrom, e.g. time constant
- G01R27/14—Measuring resistance by measuring current or voltage obtained from a reference source
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- the preferred embodiments relate to electronic measurement and are more specifically directed to devices and methods of calibrating to compensate for non-idealities so as to better measure impedance of a circuit element.
- impedance As fundamental in the art, the electrical impedance of an electrical circuit or circuit component is the opposition to current that the circuit or component presents to an applied voltage.
- impedance is a complex quantity, namely the sum of a resistance and a reactance, and varies with the frequency of the applied voltage. Impedance is of course an important factor in the manufacture of electronic circuits and systems, especially in determining the efficiency with which energy is delivered to the load of a circuit.
- impedance measurement and analysis can be used in electronic sensors, for example in determining the properties of a material or workpiece, or conditions of the surrounding environment.
- Conventional impedance analyzers operate by applying a sinusoidal stimulus to the object under measurement (referred to herein as the “device under test,” or “DUT”), and measuring the electrical response of the DUT to that sinusoid waveform.
- the response is measured at more than one frequency of the sinusoidal stimulus, for example over a “sweep” of input frequencies.
- the use of a single frequency sinusoid as the measurement stimulus at each of the frequencies of interest greatly simplifies the measurements, as harmonic interference in the response of the DUT is largely avoided.
- CMOS complementary metal-oxide-semiconductor
- SoC system on a chip
- Typical modern microcontroller architectures include one or more processor cores that carry out the digital computer functions of retrieving executable instructions from memory, performing arithmetic and logical operations on digital data retrieved from memory, and storing the results of those operations in memory.
- Other digital, analog, mixed-signal, or even RF functions may also be integrated into the same integrated circuit for acquiring and outputting the data processed by the processor cores.
- microcontrollers and SoCs have reduced the cost of implementing complex measurement and computational functions in applications for which such functionality had been cost-prohibitive.
- sensors and controllers are now being deployed in a wide range of applications and environments, including in the widely-distributed networks of such sensors and controllers often referred to as the “Internet of Things” (IoT).
- IoT Internet of Things
- microcontroller-based sensors for the measurement and analysis of electrical impedance is attractive.
- such sensors may be vulnerable to inaccurate measures, for example at higher frequencies and/or with the inclusion of a lower cost operational amplifier as part of the sensor, where such attributes provide non-idealities which therefore can affect measurement performance.
- FIG. 1 illustrates a conventional microcontroller-based impedance analyzer 10 .
- analyzer 10 includes a microcontroller 12 , which includes a digital frequency synthesizer 14 .
- Synthesizer 14 generates a sample stream corresponding to a desired signal waveform indicated by signals from a processor 16 .
- this sample stream corresponds to a sinusoidal waveform of a selected frequency.
- the sample stream generated by digital frequency synthesizer 14 is applied to a digital-to-analog converter (DAC) 18 , which is also realized within microcontroller 12 , and that generates the output sinusoidal stimulus V in that will be applied to a device under test (DUT) 22 for measurement of its impedance.
- DAC digital-to-analog converter
- DUT 22 is a two-terminal device, having one terminal receiving stimulus voltage V in (after additional filtering, if desired), and its other terminal coupled, via a switch 30 S1 discussed below, to the inverting input ( ⁇ ) of an operational amplifier (op amp) 24 .
- Op amp 24 receives a reference voltage 26 , for example at 1 ⁇ 2 the peak-to-peak amplitude of stimulus voltage V in , at its non-inverting input (+).
- a reference impedance 28 is connected in negative feedback fashion between the output of op amp 24 and its inverting input.
- the output voltage V out from op amp 24 is received by microcontroller 12 , and converted to the digital domain by an analog-to-digital converter (ADC) 20 .
- ADC analog-to-digital converter
- the ratio of output voltage V out to stimulus voltage V in reflects the impedance of DUT 22 , relative to the impedance Z REF of reference impedance 28 .
- op amp 24 maintains a virtual ground at its inverting input, and with the ideal expectation that the voltage drop across DUT 22 will equal the input voltage V in .
- the input of op amp 24 exhibits a significantly higher impedance than Z REF of reference impedance 28 , effectively all of the current conducted through DUT 22 will pass through reference impedance 28 .
- Output voltage V out will thus be proportional to this DUT current conducted through reference impedance 28 .
- the impedance of DUT 22 can be determined from the output voltage V out presented by op amp 24 , based on the ratio of voltage V out relative to voltage V in . As mentioned above, this measurement is performed over frequency by the conventional architecture of FIG. 1 , typically by processor 16 controlling digital frequency synthesizer 14 to sweep the frequency of the stimulus voltage V in applied to DUT 22 .
- ADC 20 samples and digitizes output voltage V out representing the response of DUT 22 to the stimulus at each frequency, and processor 16 analyzes that sample stream, for example via a discrete Fourier transform (DFT), to determine the impedance of DUT 22 at each frequency in the sweep. Both the amplitude and phase of output voltage V out relative to stimulus voltage V in are considered in quantifying the inductive and capacitive components of the impedance of DUT 22 .
- DFT discrete Fourier transform
- DUT 22 is connected, via a first switch 30 S1 of a switching block 30 , to the inverting input of op amp 24 .
- Switching block 30 includes a second switch 30 S2 , which is operable to connect a calibration impedance 32 between the input signal V in and the non-inverting input of op amp 24 .
- switches 30 are operated to select one of two loads at a time as an input to the non-inverting input of op amp 24 , whereby, therefore, in one instance calibration impedance 32 may be so selected, so as to perform a calibration operation given that calibration impedance 32 is a known precision impedance that is useful in calibrating the impedance measurement given a lack of precision of reference impedance 28 .
- calibration impedance 32 may be a variable impedance device (e.g., a bank of selectable precision resistors) to provide accurate calibration over a wide range of impedances.
- switch 30 S2 selecting the known impedance of Z CAL into the analyzer loop (while DUT 22 is switched out of the loop by switch 30 S1 ), then a ratio is determinable of V out relative to voltage V in , which because Z CAL is known can provide a corresponding value of Z REF .
- DUT 22 is instead selected as the circuit load (i.e., switch 30 S1 is closed, while switch 30 S2 is opened), and the result of the calibration can be used to adjust the impedance estimation of DUT 22 .
- op amp 24 can change, such as the infinite or very high input impedance can drop, and the relatively low output impedance can rise.
- One manner of attempting to address these factors is to implement a more expensive op amp, but such an approach may be cost prohibitive and also may still be somewhat vulnerable to these changes, again therefore diminishing the accuracy of DUT impedance measure that arises from the analyzer 10 which is relying on the op amp.
- a circuit for measuring an impedance of a device under test comprises: (i) circuitry for generating a stimulus wave at a stimulus frequency; (ii) an amplifier circuit coupled to the DUT to present a response signal from the DUT in response to the stimulus wave; (iii) switching circuitry for selectively coupling, between the stimulus wave and an input to the amplifier, either the DUT, a first calibration impedance, or a second calibration impedance; and (iv) processor circuitry programmed to sample a signal responsive to the response signal.
- the processor is programmed to: (A) in a first iteration, with the switching circuitry selectively coupling the first calibration impedance between the stimulus wave and an input to the amplifier, sampling a first signal responsive to the response signal; (B) in a second iteration, with the switching circuitry selectively coupling the second calibration impedance between the stimulus wave and an input to the amplifier, sampling a second signal responsive to the response signal; and (C) in a third iteration, with the switching circuitry selectively coupling the DUT between the stimulus wave and an input to the amplifier, sampling a third signal responsive to the response signal.
- the processor circuitry is further programmed to provide a measure of impedance of the DUT in response to the first signal responsive to the response signal, the second signal responsive to the response signal, and the third signal responsive to the response signal.
- FIG. 1 illustrates an electrical diagram, in block form, of a prior art microcontroller-based impedance analyzer.
- FIG. 2 illustrates, in part, a microcontroller-based impedance analyzer 200 constructed according to a preferred embodiment, where as detailed later additional calibration aspects are added thereto.
- FIG. 3 illustrates a schematic of a proposed circuit model for establishing a relationship, so as to include and consider circuit non-idealities, between the current I D through a DUT and the voltage across an impedance analyzer load Z Load .
- FIG. 4 illustrates a schematic of a proposed circuit model for establishing a relationship, so as to include and consider circuit non-idealities, between an input voltage to an impedance analyzer and the voltage dropped across a DUT.
- FIG. 5 illustrates a schematic of a preferred embodiment impedance analyzer, which is preferably a microcontroller-based impedance analyzer constructed according to a preferred embodiment.
- FIG. 1 was discussed in the Background Of The Invention section of this document and the reader is assumed familiar with the aspects of that discussion.
- FIG. 2 illustrates, in part, a microcontroller-based impedance analyzer 200 constructed according to a preferred embodiment, where as detailed later additional calibration aspects are added thereto, but for purposes of introducing various aspects recognized by the present inventor, it is shown first without such calibration aspects—moreover, FIG. 2 in many respects is the same as FIG. 2 in the above-incorporated U.S. patent application Ser. No. 15/344,565, entitled “Impedance Analyzer Using Square Wave Stimuli,” which has the same inventor as the present application, claims priority to the same date as the present application, and is commonly-assigned. Thus, various aspects are stated here, with additional detail available in the incorporated U.S. patent application Ser. No. 15/344,565.
- Impedance analyzer 200 may be implemented into a stand-alone sensor (e.g., in the IoT context) or within a larger-scale system or equipment.
- analyzer 200 includes a microcontroller 202 , which includes the appropriate functional circuitry for generating a stimulus waveform to be applied to a device under test (DUT) 204 , and for analyzing the response of that device to the stimulus in order to determine its electrical impedance.
- microcontroller 202 includes one or more processors 206 (also referred to as “processor cores”) that are capable of executing program instructions for carrying out the operations described in this specification.
- microcontroller 202 represents the memory capacity of microcontroller 202 , and as such may include memory blocks of various types, including non-volatile memory (e.g., “flash” or other electrically programmable memory) storing program instructions and configuration data for processor 206 and other functions in microcontroller 202 , and also volatile (e.g., dynamic or static RAM) memory for storing data involved in those operations. Some of memory resource 208 may be embedded within processor(s) 206 . Examples of microcontroller devices that are suitable for implementation as microcontroller 202 according to these embodiments include the MSP and C2000x families of microcontrollers available from Texas Instruments Incorporated.
- microcontroller 202 includes a general purpose input/output (GPIO) function 210 , which is coupled to a terminal SW of microcontroller 202 .
- GPIO 210 includes both input circuitry for receiving and forwarding a digital logic level to terminal SW, and driver circuitry for driving a digital voltage level at terminal SW.
- GPIO 210 is configured and operates under program control, as executed by processor 206 .
- the digital logic levels driven at terminal SW by GPIO 210 in its form as an output are constituted by a power supply voltage V pp and ground (V ss , or 0 volts).
- GPIO 210 is so configured and operates to drive a square wave signal V sq at these two levels (V pp , V ss ) that will serve as the stimulus applied to DUT 204 , so as to facilitate a measure of its electrical impedance.
- Processor 206 is also coupled to analog-to-digital converter (ADC) 212 , which is in turn coupled (via conventional “analog front end” circuitry, not shown) to a terminal RS of microcontroller 202 .
- ADC 212 operates to periodically sample and digitize the voltage V out at its terminal RS, producing a sample stream V adc that is forwarded to processor 206 .
- attention is also directed to a transfer function that can arise from any effects (e.g., gain, impedance) imposed by ADC 212 (and any related front end circuitry, not shown) on its input, relative to its output.
- the voltage sampled by ADC 212 represents the response of DUT 204 to the stimulus of square wave signal V sq applied from GPIO 210 .
- Processor 206 executes the appropriate program instructions, for example as stored in memory resource 208 , to determine an impedance measurement for DUT 204 from those sampled voltages.
- processor 206 will determine that impedance measurement by performing a discrete Fourier transform (DFT) on the V adc sample stream acquired by ADC 212 from the response of DUT 204 to the applied stimulus.
- DFT discrete Fourier transform
- the stimulus applied to DUT 204 for the impedance measurement is not a sinusoid as in the conventional architecture of FIG. 1 , but rather is a square wave signal V sq as generated by GPIO 210 .
- V sq square wave signal
- the use of a square wave will contain frequency components other than the single frequency of a sinusoid, which in this context will complicate the measurement of the electrical impedance of DUT 204 .
- the generation of the square wave stimulus and the timing of the sampling of the response are based on the same clock signal at a relationship that accounts for lower harmonics of the fundamental square wave stimulus frequency.
- a clock generator circuitry 214 of microcontroller 202 generates a relatively high-speed base clock signal CLK, at frequency f CLK , on which both the square wave stimulus V sq and the sampling frequency f ADC applied by ADC 212 are based.
- base clock frequency f CLK will be at a higher frequency than either the square wave stimulus frequency f sq or the sampling frequency f ADC .
- the stimulus and sampling frequencies can be generated within microcontroller 202 by relatively simple frequency divider functions, without requiring expensive and complex circuitry such as fractional phase-locked loops and the like as conventionally used to generate sinusoids at specific frequencies.
- base clock signal CLK is applied to digital timers 216 associated with GPIO 210 , which divide down the frequency f CLK by an integer divisor to derive the timing of the square wave stimulus.
- digital timers 216 may include a digital counter that issues a control signal to GPIO 210 to begin a cycle of the square wave (e.g., issue a rising edge) upon the elapsing of a specified number of cycles of clock signal CLK.
- a second digital counter also may be included within digital timers 216 to define the duty cycle of the square wave stimulus, for example by controlling GPIO 210 to end a pulse (e.g., issue a rising edge) upon the elapsing of a specified number of cycles of clock signal CLK.
- a pulse e.g., issue a rising edge
- CLK clock signal
- a digital timer 218 is provided in microcontroller 202 to control the sampling frequency f ADC at which ADC 212 samples the response voltage at its corresponding terminal.
- digital timer 218 controls ADC 212 to sample and digitize the response voltage upon the elapsing of a specified number of cycles of base clock signal CLK.
- sampling frequency f ADC is divided down, by a selected integer divisor value, from the frequency f CLK of base clock signal CLK. The relationship of this integer value that defines sampling frequency f ADC and the integer value that defines square wave stimulus frequency f sq according to these embodiments is described in the above-incorporated U.S. patent application Ser. No. 15/344,565.
- terminal SW driven by GPIO 210 is coupled to an anti-aliasing filter 220 , which is constructed to attenuate higher harmonics of the fundamental frequency of square wave stimulus V sq .
- Filter 220 may be a conventional off-chip (i.e., outside of microcontroller 202 ) analog low-pass filter of the desired frequency response.
- filter 220 may be constructed as a conventional 4 th order multiple feedback low-pass filter, or alternatively as any one of a number of filter architectures and topologies to attain the desired characteristic.
- Anti-aliasing filter 220 alternatively may be constructed as a band-pass frequency selective frequency filter, rather than a low-pass filter, if desired.
- anti-aliasing filter 220 may have a gain less than 1 in order to reduce the peak-to-peak voltage swing of the square wave stimulus as applied to DUT 204 , to prevent signal saturation.
- anti-aliasing filter 220 is provided to minimize the effect of higher harmonics of the square wave stimulus, so that these harmonics do not significantly contaminate the measured response of DUT 204 at the fundamental frequency of that stimulus waveform.
- attention is also directed to a transfer function that can arise from any effects (e.g., gain, impedance) imposed by filter 220 on its input, relative to its output.
- V in is the voltage connected to a first terminal of DUT 204 , so that DUT 204 is connected to receive V in , which is the filtered square wave stimulus V sq .
- DUT 204 is switchably connected in and out of the loop shown in FIG. 2 , so as to achieve preferred embodiment calibration techniques, detailed later.
- An inverting amplifier circuit receives and amplifies the response of DUT 204 to the stimulus from GPIO 210 according to a preferred embodiment.
- DUT 204 (or calibration impedances, as detailed later) is connected to an inverting input of differential operational amplifier (“op amp”) 222 .
- the non-inverting input of op amp 222 receives a DC voltage equal to the expected DC voltage of the square wave signal; in this example, a voltage source 224 applies a voltage of one-half the peak-to-peak amplitude of the square wave stimulus V sq , for example one-half the supply voltage (V pp /2).
- the output of op amp 222 is coupled to terminal RS of microcontroller 202 , and thus to ADC 212 (via front end circuitry within microcontroller 202 , not shown).
- a reference impedance 226 is connected between the output and the inverting input of op amp 222 , in the well-known negative feedback manner.
- Reference impedance 226 is preferably a precision resistor or variable impedance (e.g., a bank of precision resistors in combination with switches for selectably switching one or more of the resistors into the circuit), and thus has a known impedance for purposes of this impedance measurement.
- preferred embodiments improve upon the prior art, as well as the partial diagram shown in FIG. 2 , particularly at higher frequencies (e.g., above 100 kHz), by proposing and implementing additional calibration structure and methodology so as to overcome the non-idealities that arise at such frequencies, and that also may arise from other impedances in an impedance analyzing op-amp based circuit.
- the preferred embodiments implement an architecture based on proposed modeling, wherein such architecture is based on the ideally surprising proposition that certain elements are reducible to a voltage division based model, wherein the voltage division accounts for various changes in impedance and op amp gain (i.e., op am impairments) over a wide range of high frequencies.
- FIG. 3 illustrates a schematic of a proposed circuit model 300 for establishing a relationship between the current I D through an impedance analyzer connected DUT and the voltage across a load Z Load , which may represent the input impedance of an ADC (and related circuitry) of that analyzer.
- the model therefore, proposes a transfer function as between that current and voltage, where as demonstrated later this transfer function in view of additional observations lends to a preferred embodiment that provides improved performance given the non-ideal performance of the prior art in high frequencies.
- Model 300 includes a current source 302 representing the current I D through the DUT, which is a result of the input voltage V in coupled to the DUT.
- Node 304 connects through an input impedance Z in to ground, through a reference impedance Z REF (e.g., akin to reference impedance 226 in FIG. 2 ) to a node 308 , and through a gain coupling, which is shown with conventional representations and depicting the gain relationship such that the output of op amp model 306 is its gain, G, times its input (hence the voltage at the terminal of Z out connected to the voltage-controlled voltage source in FIG.
- Z REF e.g., akin to reference impedance 226 in FIG. 2
- the gain coupling of op amp model 306 includes an output impedance Z out , also connected to node 308 .
- Node 308 is connected through load impedance Z Load , to ground.
- I in is the current through input impedance Z in ;
- I REF is the current through impedance Z REF .
- Equation 3 Each of the currents in Equation 3 may be re-written as the voltage drop across the respective impedance divided by that respective impedance, yielding the following Equation 4:
- Equation 5 V - V in + V - - V out Z REF Equation ⁇ ⁇ 4
- Equation 5 may be re-written in terms of a factored out value of V out , and given the parallel nature of the added reciprocal impedances in Equation 5, as in the following Equation 6:
- I D V out ⁇ ( ( V - ⁇ / ⁇ V out ) ( Z in
- I Load is the current through the load impedance Z Load .
- Equation 7 Each of the currents in Equation 7 may be re-written as the voltage drop across the respective impedance divided by that respective impedance, yielding following Equation 8:
- V - - V out Z REF + - GV - - V out Z out V out Z Load Equation ⁇ ⁇ 8
- Equation 9 Equation 9:
- Equation 10 Grouping common term, V ⁇ and V out , from the left of Equation 9 yields the following Equation 10:
- Equation 10 simplifies as shown in the following Equation 11:
- V - ( Z out - ( GZ REF ) Z REF ⁇ Z out - V out ⁇ ( Z out + Z REF ) Z REF ⁇ Z out V out Z Load Equation ⁇ ⁇ 11
- V - ⁇ ( 1 Z REF - G Z out ) V out Z Load + V out ⁇ Z out Z REF ⁇ Z OUT + V out ⁇ Z REF Z REF ⁇ Z OUT Equation ⁇ ⁇ 12
- Equation 13
- V - ⁇ ( 1 Z REF - G Z out ) V out ⁇ ( 1 Z Load + 1 Z out + 1 Z REF ) Equation ⁇ ⁇ 13
- V - ⁇ / ⁇ V out ( Z REF
- Equation 14 can be substituted into the numerator term of (V ⁇ /V out ) in Equation 6, providing the following Equation 15:
- Equation 15 the factors other than the impedance Z REF are combined into a single factor, Q, representing a transfer function that accounts for the scaling factor on current that arises from the other impedances shown in Equation 15, as well as from any change in the gain ⁇ G of op amp 306 .
- Equation 16 Equation 16:
- V out I D ⁇ ( Z REF - Q ) Equation ⁇ ⁇ 16
- Equation 16 readily demonstrates how I D is scaled by the factor of
- FIG. 4 illustrates a schematic of a proposed circuit model 400 for establishing a relationship, so as to include and consider circuit non-idealities, between an input voltage to an impedance analyzer and the voltage dropped across a DUT. More particularly, therefore, FIG.
- model 400 which is model 300 of FIG. 3 , with the addition of the input voltage, V in , such as from an anti-aliasing filter (see, e.g., FIG. 2 , filter 220 ), and to include the impedance Z DUT of the DUT or, more appropriately, what will be the measured value of that impedance, hereafter referred to as Z DUT Meas .
- Z DUT Meas the impedance analyzer is typically on a circuit board, and the DUT may be connected to that impedance analyzer circuit board by a cable or some other connector.
- the measure of Z DUT Meas therefore, includes the gain and phase shift caused by this connection. Indeed, in some instances, after certain measures are taken, such as those in accordance with the preferred embodiments described below (i.e., compensation for op amp impairments), there may be an additional calibration procedure to correct for the effect of these cables and connectors between the circuit board and the DUT.
- the second calibration procedure going from Z DUT Meas to an actual estimated impedance value for the DUT (i.e., Z DUT ), is well known.
- the reader is referred to Appendix C of the reference, “ Impedance Measurement Handbook, A guide to measurement technology and techniques,” 6 th Edition, by Keysight Technologies, which is hereby incorporated fully herein by reference.
- Equation 17 Each of the currents in Equation 17 may be re-written as the voltage drop across the respective impedance divided by that respective impedance, and here by representing current I D based on the measured impedance Z DUT Meas , yielding following Equation 18:
- V in - V - Z DUT Meas V - - V out Z REF + V - Z in Equation ⁇ ⁇ 18
- Equation 19 Separating the left side term into two addends and moving the one relating to V ⁇ to the right of Equation 18 yields the following Equation 19:
- V in Z DUT Meas V - Z REF + - V out Z REF + V - Z in + V - Z DUT Meas Equation ⁇ ⁇ 19
- Equation 20 Factoring out the common term V ⁇ from the right of Equation 19 yields the following Equation 20:
- V in Z DUT Meas V - ⁇ ( 1 Z REF + 1 Z in + 1 Z DUT Meas ) + - V out Z REF Equation ⁇ ⁇ 20
- V in Z DUT Meas V - ⁇ ( 1 Z REF ⁇ ( 1 - V out V - ) + 1 Z in + 1 Z DUT Meas ) Equation ⁇ ⁇ 21
- Equation 14 the reciprocal of Equation 14 can be substituted into the term
- Equation 22 V out V - of Equation 21, and the parallel impedances expanded, which yields the following Equation 22:
- V in Z DUT Meas V - ⁇ [ ( 1 Z DUT Meas + 1 Z REF + 1 Z in ) - ( ( 1 Z REF - G Z out ) ⁇ / ⁇ ( 1 + Z REF Z Load + Z REF Z out ) ) ] Equation ⁇ ⁇ 22
- Equation 24 can be observed from rearranging the terms in the equation to collect the voltage variables on the left hand side of the equals sign, and to collect the impedance variables on the right hand side:
- V - V in ( 1 Z DUT Meas ) ⁇ / ⁇ [ ( 1 Z DUT Meas + 1 Z REF + 1 Z in ) - ( 1 Z REF - G Z out ) ⁇ / ⁇ ( 1 + Z REF Z out + Z REF Z Load ) ] Equation ⁇ ⁇ 23
- Equation 24 can be observed from multiplying all the terms of the right hand side denominator by Z DUT Meas :
- V - V in 1 ⁇ / ⁇ [ ( 1 + Z DUT Meas Z REF + Z DUT Meas Z in ) - ( Z DUT Meas Z REF - GZ DUT Meas Z out ) ⁇ / ⁇ ( 1 + Z REF Z out + Z REF Z Load ) ] Equation ⁇ ⁇ 24
- Equation 24 the equivalent of all factors in Equation 24, other than Z DUT Meas , can be represented by a collective admittance value P, in which case Equation 24 reduces to the following Equation 25:
- V - V in 1 1 + PZ DUT Meas Equation ⁇ ⁇ 25
- Equation 26 the relationship of the voltage drop across the DUT to the variables V in and V ⁇ also may be mathematically stated as in the following Equation 26:
- V in - V - V in ⁇ ( 1 - V - V in ) Equation ⁇ ⁇ 26
- Equation ⁇ ⁇ 28 the transfer function arising from the anti-aliasing filter from Equation 2 may be added to Equation 27, thereby providing the following Equation 28:
- Equation 29 demonstrates that with circuit non-idealities including op am impairments, the measured op amp output voltage V adc relates to the excitation voltage V sq in attempting to measure the DUT impedance by an impedance analyzer, by two factors, namely, an added offset
- the multiplier represents the transconductance of the amplifier 222 and reference impedance 226 , corrected for the non-idealities of the circuit elements.
- the offset models the degradation of the op amp virtual ground by impedance in series with the DUT. The full voltage swing of V sq drops across the voltage divider formed by the DUT and the offset impedance. Hence, there are two unknowns in this relationship, giving rise to a preferred embodiment impedance analyzer 500 , which is now explored.
- FIG. 5 illustrates a schematic of a preferred embodiment impedance analyzer 500 , which is preferably the microcontroller-based impedance analyzer 200 of FIG. 2 , but now constructed according to a preferred embodiment to include various calibration aspects, as introduced earlier and as now should be better understood with reference to the preceding teachings from the present inventor.
- analyzer 500 includes all the aspects illustrated above in connection with FIG. 2 , and the reader is assumed to be familiar with that illustration and the accompanying details provided earlier. Thus, the following discussion attends to the additional aspects.
- analyzer 500 includes a switching block 530 with three switches, shown as 530 S1 , 530 S2 , and 530 S3 , each connected for selecting a respective load at a time and including it in the analyzer 500 loop.
- switches 530 S1 , 530 S2 , and 530 S3 may be under control of microcontroller 202 or some intermediary control block so as to close one switch at a time, leaving the other two open, and in a desired order so as to accomplish functionality and unknown parameter determination as detailed later.
- DUT 204 is connected to a first terminal of a first switch 530 S1 , a first calibration impedance Z CAL1 is connected to a first terminal of a second switch 530 S2 , and a second calibration impedance Z CAL2 is connected to a first terminal of a third switch 530 S3 ; the second terminals of each of switches 530 S1 , 530 S2 , and 530 S3 is connected to the inverting input of op amp 222 .
- switches 530 are operated to select one of three loads at a time as an input to the inverting input of op amp 222 , whereby, therefore, in one instance calibration impedance Z CAL1 may be so selected, so as to perform one calibration operation given that calibration impedance Z CAL1 is a known precision impedance, and in another instance calibration impedance Z CAL2 may be so selected, so as to perform another calibration operation given that calibration impedance Z CAL2 is also a known precision impedance, while lastly DUT 204 may be selected and, as discussed below, the results from the two calibrations, involving calibration impedances Z CAL1 and Z CAL2 , may be used to provide values for adjusting the measured voltage V out so as to arrive at an estimated value of the impedance of DUT 204 (i.e., finding Z DUT Meas ).
- Equation 29 may be simplified as shown in the following Equation 30:
- V sq V adc H TOT ⁇ ( Z DUT Meas + 1 P ) Equation ⁇ ⁇ 30
- Equation 30 ( - Q Z REF ⁇ H TX ⁇ H RX ) .
- analyzer 500 includes two different selectable, and known, calibration impedances, then in a preferred embodiment analyzer 500 is operated in one iteration wherein switch 530 S2 is exclusively closed, for a given V sq at a given frequency, and since Z CAL1 is known, then Equation 31 becomes as shown in the following Equation 30.1.
- Equation 31 H TOT ⁇ ( Z CAL ⁇ ⁇ 1 + 1 P ) Equation ⁇ ⁇ 30.1
- switch 530 S3 is exclusively closed, for a given V sq at a given frequency, and since Z CAL2 is known, then Equation 31 becomes as shown in the following Equation 30.2.
- V sq V adc ⁇ ⁇ 2 H TOT ⁇ ( Z CAL ⁇ ⁇ 2 + 1 P ) Equation ⁇ ⁇ 30.2
- Equations 30.1 and 30.2 will be known (either at the outset (i.e., V sq and Z CAL1 or Z CAL2 ) or as measured (i.e., V adc1 for the measure when switching in Z CAL1 and V adc2 for the measure when switching in Z CAL2 )). With two equations and two unknowns, one skilled in the art may readily solve for the unknowns.
- analyzer 500 is readily operable to determine the impedance of DUT 204 , at any of the frequencies for which those values were stored. More particularly, when such an impedance measure is desired, switch 530 S1 is exclusively closed and V adc is measured at a particular frequency. From that measure, and from the already-determined values of H TOT and
- Equation 30 may be re-arranged to show the ready application of those values to the rest of the Equation, so as to arrive at the impedance of DUT 204 , as shown in the following Equation 31:
- microcontroller 202 can control switch 530 S1 to close and issue, via GPIO 210 a stimulus signal V sq , while then measuring the response V adc with the additional scaling of H TOT and offset of
- microcontroller 202 can thereby apply those various values, per Equation 31, so as to provide a measure of impedance for the DUT.
- FIG. 5 illustrates the two switchable calibration elements in an analyzer that uses a non-sinusoid stimulus
- an alternative contemplated preferred embodiment would apply such two switchable calibration elements in an analyzer that resembles that of FIG. 1 , that is, where the sinusoid is a driving stimulus signal to the DUT.
Abstract
Description
Vadc=HRXVout Equation 1
Thus, according to a preferred embodiment, the voltage sampled by
Vin=HTXVsq Equation 2
I D =I in +I REF Equation 3
where,
Re-arranging Equation 4 gives the following Equation 5:
Equation 5 may be re-written in terms of a factored out value of Vout, and given the parallel nature of the added reciprocal impedances in Equation 5, as in the following Equation 6:
I REF +I out =I Load Equation 7
where,
Thus,
so as to influence the op amp output voltage Vout, and independent of the DUT impedance.
I D =I REF +I in Equation 17
of Equation 21, and the parallel impedances expanded, which yields the following Equation 22:
into the last term of
Still further, the transfer function arising from the anti-aliasing filter from
Lastly in view of the above, and further given the receive transfer function of HRX of
and a multiplier
The multiplier represents the transconductance of the
-
- HTOT is a total transform function, thereby singularly representing the product of four unknowns,
Thus, by collecting the various factors into HTOT, it may now be appreciated that Equation 30 (and Equation 29), represents for a given measurement cycle by analyzer 500 a total of two unknowns, one being the multiplier HTOT, and the other being the offset
Inasmuch as
Similarly,
Hence, from the two iterations, and for the given frequency, then the results will be two sets of results for two different unknowns, those being the multiplier HTOT and
as the other items shown in Equations 30.1 and 30.2 will be known (either at the outset (i.e., Vsq and ZCAL1 or ZCAL2) or as measured (i.e., Vadc1 for the measure when switching in ZCAL1 and Vadc2 for the measure when switching in ZCAL2)). With two equations and two unknowns, one skilled in the art may readily solve for the unknowns.
are repeated for respective other frequencies. Thus, an entire set of values may be stored (e.g., table in memory) wherein, for each frequency, a value of multiplier HTOT and offset
determined from a paired iteration at the respective frequency, are stored. Thereafter,
then
From Equation 31, therefore,
Claims (18)
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Citations (14)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4041382A (en) * | 1976-08-16 | 1977-08-09 | The Sippican Corporation | Calibrating a measurement system including bridge circuit |
US5297425A (en) | 1992-03-23 | 1994-03-29 | Tennessee Valley Authority | Electromagnetic borehole flowmeter system |
US20010010467A1 (en) | 2000-01-31 | 2001-08-02 | Tanita Corporation | Bioelectrical impedance measuring apparatus constructed by one-chip integrated circuit |
US20050072874A1 (en) | 2001-02-09 | 2005-04-07 | Georgia-Pacific Corporation | Paper dispenser with proximity detector |
US20060182231A1 (en) | 2005-01-27 | 2006-08-17 | Kan Tan | Apparatus and method for processing acquired signals for measuring the impedance of a device under test |
US20070268012A1 (en) | 2005-03-04 | 2007-11-22 | Masayuki Kawabata | Waveform input circuit, waveform observation unit and semiconductor test apparatus |
US20080303538A1 (en) * | 2005-12-16 | 2008-12-11 | Timothy Orr | Measuring Electrical Impedance at Various Frequencies |
US20110074392A1 (en) | 2009-09-30 | 2011-03-31 | Tektronix, Inc. | Signal Acquisition System Having Reduced Probe Loading of a Device Under Test |
US20110115509A1 (en) | 2009-11-18 | 2011-05-19 | Samsung Electronics Co., Ltd. | Semiconductor Devices Including Design for Test Capabilities and Semiconductor Modules and Test Systems Including Such Devices |
US20110227587A1 (en) | 2010-03-16 | 2011-09-22 | Yokogawa Electric Corporation | Ac impedance measuring device |
US8604809B2 (en) | 2008-11-02 | 2013-12-10 | Siemens Aktiengesellschaft | Current sensor capacity measuring system |
US20140145729A1 (en) | 2011-08-22 | 2014-05-29 | Keithley Instruments, Inc. | Low frequency impedance measurement with source measure units |
US20160077140A1 (en) | 2014-09-12 | 2016-03-17 | Dialog Semiconductor B.V. | Low Power On-Chip Impedance Detector |
US20160274060A1 (en) * | 2013-10-22 | 2016-09-22 | Jentek Sensors, Inc. | Method and Apparatus for Measurement of Material Condition |
-
2017
- 2017-02-28 US US15/445,762 patent/US10551469B2/en active Active
Patent Citations (14)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4041382A (en) * | 1976-08-16 | 1977-08-09 | The Sippican Corporation | Calibrating a measurement system including bridge circuit |
US5297425A (en) | 1992-03-23 | 1994-03-29 | Tennessee Valley Authority | Electromagnetic borehole flowmeter system |
US20010010467A1 (en) | 2000-01-31 | 2001-08-02 | Tanita Corporation | Bioelectrical impedance measuring apparatus constructed by one-chip integrated circuit |
US20050072874A1 (en) | 2001-02-09 | 2005-04-07 | Georgia-Pacific Corporation | Paper dispenser with proximity detector |
US20060182231A1 (en) | 2005-01-27 | 2006-08-17 | Kan Tan | Apparatus and method for processing acquired signals for measuring the impedance of a device under test |
US20070268012A1 (en) | 2005-03-04 | 2007-11-22 | Masayuki Kawabata | Waveform input circuit, waveform observation unit and semiconductor test apparatus |
US20080303538A1 (en) * | 2005-12-16 | 2008-12-11 | Timothy Orr | Measuring Electrical Impedance at Various Frequencies |
US8604809B2 (en) | 2008-11-02 | 2013-12-10 | Siemens Aktiengesellschaft | Current sensor capacity measuring system |
US20110074392A1 (en) | 2009-09-30 | 2011-03-31 | Tektronix, Inc. | Signal Acquisition System Having Reduced Probe Loading of a Device Under Test |
US20110115509A1 (en) | 2009-11-18 | 2011-05-19 | Samsung Electronics Co., Ltd. | Semiconductor Devices Including Design for Test Capabilities and Semiconductor Modules and Test Systems Including Such Devices |
US20110227587A1 (en) | 2010-03-16 | 2011-09-22 | Yokogawa Electric Corporation | Ac impedance measuring device |
US20140145729A1 (en) | 2011-08-22 | 2014-05-29 | Keithley Instruments, Inc. | Low frequency impedance measurement with source measure units |
US20160274060A1 (en) * | 2013-10-22 | 2016-09-22 | Jentek Sensors, Inc. | Method and Apparatus for Measurement of Material Condition |
US20160077140A1 (en) | 2014-09-12 | 2016-03-17 | Dialog Semiconductor B.V. | Low Power On-Chip Impedance Detector |
Non-Patent Citations (3)
Title |
---|
CDCE(L)913: Flexible Low Power LVCMOS Clock Generator with SSC Support for EMI Reduction, Texas Instruments datasheet dated Jun. 2007, revised Oct. 2016; located at http://www.ti.com/lit.ds.symlink/cdce913.pdf (35 pages). |
Chen, Chiouguey J., "Modified Goertzel Algorithm in DTMF Detection Using the TMS320C80," Texas Instruments Application Report, SPRA066, Jun. 1996 (19 pages). |
Mock, Pat, "Add DTMF Generation and Decoding to DSP-uP Designs," Texas Instruments Application Report, SPRA168, copyright 1997 (19 pages). |
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