TWI816719B - Bidirectional DC-AC converter and control method thereof - Google Patents

Abstract

The present invention provides a bidirectional DC-AC converter and a control method thereof. The bidirectional DC-AC converter includes: a full-bridge or half-bridge inverter; and a transformer. One of the primary sides of the transformer is connected to the full-bridge or half-bridge. The AC terminal of the bridge inverter; the AC-AC converter, which has a first AC terminal and a second AC terminal, the first AC terminal is connected to the secondary side of the transformer, and the second AC terminal is assembled The state is connected to the load or AC power supply; and the inductor is connected between the primary side of the transformer and the AC end of the full-bridge or half-bridge inverter, or between the secondary side of the transformer and between the first AC terminals of the AC-AC converter. The bidirectional DC-AC converter of the present invention improves power density and conversion efficiency, and reduces costs.

Description

Bidirectional DC-AC converter and control method thereof

The invention relates to the field of electronic circuits, and specifically relates to a bidirectional DC-AC converter and a control method thereof.

Uninterruptible power supplies can continuously supply power to loads and have been widely used in various fields.

Existing uninterruptible power supplies usually include DC-DC converters, inverters and chargers. In the battery mode, the DC power of the rechargeable battery is converted into the required AC power by the DC-DC converter and inverter; in the charging mode, the charger is used to charge the rechargeable battery.

However, the existing DC-DC converter, inverter and charger are three independent devices. As a result, the uninterruptible power supply has lower power density, lower efficiency and higher cost.

In view of the above technical problems existing in the prior art, embodiments of the present invention provide a bidirectional DC-AC converter, which includes: a full-bridge or half-bridge inverter; a transformer, with one primary side of the transformer connected to the full-bridge or half-bridge inverter. The AC terminal of the half-bridge inverter; the AC-AC converter, which has a first AC terminal and a second AC terminal, the first AC terminal is connected to the secondary side of the transformer, and the second AC terminal is configured to be connected to a load or an AC power source; and an inductor connected between the secondary side of the transformer and the first AC terminal of the AC-AC converter.

Preferably, the AC-AC converter is a half-bridge AC-AC converter composed of two bidirectionally controllable switches connected in series and two capacitors connected in series. The two bidirectionally controllable switches connected in series are The node formed by the tube and the node formed by the two series-connected capacitors serve as the first AC terminal.

Preferably, each of the two bidirectionally controllable switching transistors connected in series includes two switching transistors connected in reverse series.

Preferably, the half-bridge AC-AC converter includes: a fifth switch tube and a sixth switch tube connected in reverse series, and a seventh switch tube and an eighth switch tube connected in reverse series; wherein when the fifth switch tube When the switch tube and the seventh switch tube are controlled to be turned on, the sixth switch tube, the eighth switch tube and the two capacitors constitute the first half-bridge inverter, and when the sixth switch tube and the eighth switch tube When controlled to be turned on, the fifth switch tube, the seventh switch tube and the two capacitors form a second half-bridge inverter.

Preferably, the AC-AC converter is a full-bridge AC-AC converter composed of four bidirectional controllable switching tubes, and the nodes of the two bridge arms of the full-bridge AC-AC converter serve as the first communication side.

Preferably, each of the four bidirectionally controllable switching transistors includes two switching transistors connected in reverse series.

Preferably, the full-bridge AC-AC converter includes: a fifth switch tube and a sixth switch tube connected in reverse series, a seventh switch tube and an eighth switch tube connected in reverse series, and a ninth switch connected in reverse series. tube and the tenth switching tube, the eleventh switching tube and the twelfth switching tube connected in reverse series; wherein when the fifth, seventh, ninth and eleventh switching tubes are controlled to be turned on, the The sixth, eighth, tenth and twelfth switching tubes constitute the first full-bridge inverter, and when the sixth, eighth, tenth and twelfth switching tubes are controlled to be turned on, the fifth , the seventh, ninth and eleventh switch tubes constitute the second full-bridge inverter.

Preferably, the bidirectional DC-AC converter further includes a filter capacitor connected between the DC terminals of the full-bridge or half-bridge inverter.

Preferably, the bidirectional DC-AC converter further includes a control device, which is used to: when the AC power supply fails, control the full-bridge or half-bridge inverter to convert the DC power at its DC end into the first AC power. square wave, and control the AC-AC converter to convert the second AC square wave at its first AC end into industrial frequency AC power, wherein the periods of the first AC square wave and the second AC square wave are provided to the The period of the pulse width modulation signal of the full-bridge or half-bridge inverter; when the AC power supply is normal, the AC-AC converter is controlled to convert the power frequency alternating current at its second AC end into a third AC square wave , and control the full-bridge or half-bridge inverter to convert the fourth AC square wave at its AC end into direct current. The periods of the third AC square wave and the fourth AC square wave are provided to the full bridge or half-bridge inverter. The period of the pulse width modulation signal of the half-bridge inverter.

The present invention also provides a control method for the bidirectional DC-AC converter as described above. The full-bridge inverter includes a first switch tube and a second switch tube connected to its DC end in sequence, and The third switch tube and the fourth switch tube at the DC end, the half-bridge AC-AC converter includes: the fifth switch tube and the sixth switch tube connected in reverse series, and the seventh switch tube and the third switch tube connected in reverse series. Eight switching tubes; wherein when the fifth switching tube and the seventh switching tube are controlled to be turned on, the sixth switching tube, the eighth switching tube and the two capacitors constitute a first half-bridge inverter, and when the When the sixth switch tube and the eighth switch tube are controlled to be turned on, the fifth switch tube, the seventh switch tube and the two capacitors form a second half-bridge inverter. The control method includes: Within the frequency cycle, the first switch tube and the second switch tube are controlled to be turned on alternately, the third switch tube and the fourth switch tube are controlled to be turned on alternately, the fifth and seventh switch tubes are controlled to be turned on, and the third switch tube is controlled to be turned on alternately. The sixth and eighth switch tubes are alternately turned on; during the negative half power frequency cycle, the first switch tube and the second switch tube are controlled to be turned on alternately, the third switch tube and the fourth switch tube are controlled to be turned on alternately, and the The fifth and seventh switch tubes are alternately turned on to control the sixth and eighth switch tubes to be turned on.

Preferably, in the positive half power frequency cycle, the pulse width modulation signal provided to the sixth switching tube is delayed by a first period of time than the pulse width modulation signal provided to the fourth switching tube. The pulse width modulation signal provided by the fourth switching tube is delayed by a second period of time than the pulse width modulation signal provided by the first switching tube; and the pulse width modulation signal provided by the eighth switching tube is delayed by a second time period. The pulse width modulation signal provided by the third switching tube is delayed by a first period of time, and the pulse width modulation signal provided to the third switching tube is delayed by a second period than the pulse width modulation signal provided to the second switching tube. time period; in the negative half power frequency cycle, the pulse width modulation signal provided to the seventh switching tube is delayed by a first time period than the pulse width modulation signal provided to the fourth switching tube, and the pulse width modulation signal provided to the fourth switching tube is The pulse width modulation signal provided by the four switching tubes is delayed by a second period of time than the pulse width modulation signal provided by the first switching tube; and the pulse width modulation signal provided by the fifth switching tube is delayed by a second period. The pulse width modulation signal provided by the third switching tube is delayed by a first period of time, and the pulse width modulation signal provided to the third switching tube is delayed by a second time period than the pulse width modulation signal provided to the second switching tube. part.

Preferably, in the positive half power frequency cycle, the pulse width modulation signal provided to the second switching tube is delayed by a third period of time than the pulse width modulation signal provided to the eighth switching tube. The pulse width modulation signal provided by the eighth switching tube is delayed by a fourth period of time than the pulse width modulation signal provided by the fourth switching tube; and the pulse width modulation signal provided by the first switching tube is delayed by a fourth time period. The pulse width modulation signal provided by the sixth switching tube is delayed by a third time period, and the pulse width modulation signal provided by the sixth switching tube is delayed by a fourth period than the pulse width modulation signal provided by the third switching tube. time period; during the negative half power frequency cycle, the pulse width modulation signal provided to the second switching tube is delayed by a third time period than the pulse width modulation signal provided to the fifth switching tube, and the pulse width modulation signal provided to the third switching tube is The pulse width modulation signal provided by the fifth switch tube is delayed by a fourth period of time than the pulse width modulation signal provided by the fourth switch tube; and the pulse width modulation signal provided by the first switch tube is delayed by a fourth time period; The pulse width modulation signal provided by the seventh switching tube is delayed by a third time period, and the pulse width modulation signal provided by the seventh switching tube is delayed by a fourth time period than the pulse width modulation signal provided by the third switching tube. part.

Preferably, a pulse width modulation signal with a delay difference of zero is provided to the first switch tube and the fourth switch tube, and a pulse width modulation signal with a delay difference of zero is provided to the second switch tube and the third switch tube. signal; and in the positive half power frequency cycle, the pulse width modulation signal provided to the sixth switching tube is delayed by a fifth period of time compared to the pulse width modulation signal provided to the first switching tube, and the pulse width modulation signal provided to the third switching tube is The pulse width modulation signal provided by the eighth switch tube is delayed by a fifth period of time than the pulse width modulation signal provided by the second switch tube; in the negative half power frequency cycle, the pulse width modulation signal provided by the seventh switch tube is The modulation signal is delayed by a fifth period of time compared to the pulse width modulation signal provided to the first switching tube, and the pulse width modulation signal provided to the fifth switching tube is greater than the pulse width provided to the second switching tube. The modulation signal is delayed for a fifth time period.

Preferably, a pulse width modulation signal with a delay difference of zero is provided to the first switch tube and the fourth switch tube, and a pulse width modulation signal with a delay difference of zero is provided to the second switch tube and the third switch tube. signal; and in the positive half power frequency cycle, the pulse width modulation signal provided to the first switching tube is delayed by a sixth period of time than the pulse width modulation signal provided to the sixth switching tube, and the pulse width modulation signal provided to the sixth switching tube is delayed by a sixth time period. The pulse width modulation signal provided by the six switching tubes is delayed by a seventh period of time than the pulse width modulation signal provided by the second switching tube; the pulse width modulation signal provided to the second switching tube is delayed by a seventh time period; The pulse width modulation signal provided by the eight switching tubes is delayed for a sixth time period, and the pulse width modulation signal provided to the eighth switching tube is delayed by a seventh time period than the pulse width modulation signal provided to the first switching tube. ; In the negative half power frequency cycle, the pulse width modulation signal provided to the first switch tube is delayed by a sixth period of time than the pulse width modulation signal provided to the seventh switch tube, and the pulse width modulation signal provided to the seventh switch tube is The pulse width modulation signal provided by the tube is delayed by a seventh period of time than the pulse width modulation signal provided to the second switch tube; the pulse width modulation signal provided to the second switch tube is delayed by a seventh period of time than the pulse width modulation signal provided to the fifth switch. The pulse width modulation signal provided by the switch is delayed by a sixth period of time, and the pulse width modulation signal provided to the fifth switch tube is delayed by a seventh period of time than the pulse width modulation signal provided to the first switch tube.

Preferably, the half-bridge inverter includes a first switch and a second switch connected to its DC end in sequence, and the half-bridge AC-AC converter includes: a fifth switch and a third switch connected in reverse series. Six switching tubes, and a seventh switching tube and an eighth switching tube connected in reverse series; wherein when the fifth switching tube and the seventh switching tube are controlled to be conductive, the sixth switching tube, the eighth switching tube and The two capacitors constitute the first half-bridge inverter, and when the sixth switch tube and the eighth switch tube are controlled to be turned on, the fifth switch tube, the seventh switch tube and the two capacitors constitute the second half-bridge inverter. Bridge inverter, the control method includes: controlling the first switch tube and the second switch tube to conduct alternately, controlling the fifth and seventh switch tubes to conduct, and controlling the third switch tube to conduct during the positive half power frequency cycle. The sixth and eighth switch tubes are alternately turned on; during the negative half power frequency cycle, the first switch tube and the second switch tube are controlled to be turned on alternately, the fifth and seventh switch tubes are controlled to be turned on alternately, and the sixth switch tube is controlled to be turned on alternately. and the eighth switch tube is turned on.

Preferably, in the positive half power frequency cycle, the pulse width modulation signal provided to the sixth switching tube is delayed by an eighth period of time than the pulse width modulation signal provided to the first switching tube. The pulse width modulation signal provided by the eighth switching tube is delayed by an eighth period of time than the pulse width modulation signal provided by the second switching tube; during the negative half power frequency cycle, the pulse width modulation signal provided by the seventh switching tube is The wide modulation signal is delayed by an eighth period of time compared to the pulse width modulation signal provided to the first switching tube, and the pulse width modulation signal provided to the fifth switching tube is delayed from the pulse width modulation signal provided to the second switching tube. The wide modulation signal is delayed for an eighth time period.

Preferably, in the positive half power frequency cycle, the pulse width modulation signal provided to the first switching tube is delayed by a ninth period of time than the pulse width modulation signal provided to the sixth switching tube. The pulse width modulation signal provided by the sixth switching tube is delayed by a tenth period of time than the pulse width modulation signal provided by the second switching tube; the pulse width modulation signal provided by the second switching tube is delayed by a tenth period; The pulse width modulation signal provided by the eighth switch tube is delayed for a ninth time period, and the pulse width modulation signal provided to the eighth switch tube is delayed by a tenth time period than the pulse width modulation signal provided to the first switch tube. segment; in the negative half power frequency cycle, the pulse width modulation signal provided to the first switching tube is delayed by a ninth period of time than the pulse width modulation signal provided to the seventh switching tube, and the pulse width modulation signal provided to the seventh switching tube is The pulse width modulation signal provided by the switching tube is delayed by a tenth time period than the pulse width modulation signal provided by the second switching tube; the pulse width modulation signal provided to the second switching tube is delayed by a tenth time period; the pulse width modulation signal provided to the second switching tube is delayed by a tenth time period. The pulse width modulation signal provided by the switching tube is delayed for a ninth time period, and the pulse width modulation signal provided to the fifth switching tube is delayed by a tenth time period than the pulse width modulation signal provided to the first switching tube.

The bidirectional DC-AC converter of the present invention can convert the DC power of the rechargeable battery into the required AC power in the battery mode, and can charge the rechargeable battery in the charging mode. The charging power and charging current are large, and it can realize Power factor correction. The bidirectional DC-AC converter of the present invention improves power density and conversion efficiency, and reduces costs.

In order to make the purpose, technical solutions and advantages of the present invention more clear, the present invention will be further described in detail below with reference to the accompanying drawings through specific embodiments.

FIG. 1 is a circuit diagram of a bidirectional DC-AC converter according to the first embodiment of the present invention. As shown in FIG. 1 , the bidirectional DC-AC converter 1 includes a full-bridge inverter 11 , a transformer Tr, an inductor 14 , a half-bridge AC-AC converter 12 and a control device 17 .

The full-bridge inverter 11 includes metal-oxide semi-field effect transistors (MOSFETs) S ₁ , MOSFET S ₂ , MOSFET S ₃ and MOSFET S ₄ . MOSFET S ₁ and MOSFET S ₂ are connected in series between the positive and negative electrodes of the rechargeable battery 13 and form node A, and MOSFET S ₃ and MOSFET S ₄ are connected in series between the positive and negative electrodes of the rechargeable battery 13 and form Node B.

The half-bridge AC-AC converter 12 includes a capacitor C ₁ and a capacitor C ₂ , and a bidirectional controllable switch tube 121 and a bidirectional control switch tube 122 . The capacitor C ₁ and the capacitor C ₂ are connected in series to either the load 15 or the AC power supply. end, and forms node D. The bidirectional switch 121 and the bidirectional switch 122 are connected in series at both ends of the series capacitor C ₁ and the capacitor C ₂ and form a node C, where the nodes C and D form the first node of the half-bridge AC-AC converter 12 An AC terminal, the two ends of the capacitor C ₁ and the capacitor C ₂ connected in series form the second AC terminal of the half-bridge AC-AC converter 12 . The bidirectional switch 121 further includes MOSFET S ₅ and MOSFET S ₆ connected in reverse series, and the bidirectional switch 122 further includes MOSFET S ₇ and MOSFET S ₈ connected in reverse series. When MOSFET S ₅ and MOSFET S ₇ are controlled to be turned on, MOSFET S ₆ , MOSFET S ₈ , capacitor C ₁ and capacitor C ₂ form a half-bridge inverter; similarly, when MOSFET S ₆ and MOSFET S ₈ are controlled to be When turned on, MOSFET S ₅ , MOSFET S ₇ capacitor C ₁ and capacitor C ₂ form the other half-bridge inverter.

The primary side of the transformer Tr is connected between the node A and the node B, and the secondary side of the transformer Tr is connected between the node C and the node D. The inductor 14 is connected between one end of the secondary side of the transformer Tr and the node C.

The working principle of the battery discharge mode of the bidirectional DC-AC converter 1 will be described below in conjunction with the equivalent circuit diagram.

Figure 2 is a waveform diagram of the pulse width modulation signal provided by the control device to the switching tube in the bidirectional DC-AC converter shown in Figure 1. As shown in Figure 2, V _o is the voltage across the load 15, MOSFET S ₁ and MOSFET S ₂ are controlled to alternately conduct, and MOSFET S ₃ and MOSFET S ₄ are also controlled to alternately conduct. During the positive half power frequency cycle, MOSFET S ₅ and MOSFET S ₇ are controlled to be continuously conductive, and MOSFET S ₆ and MOSFET S ₈ are controlled to be alternately conductive; during the negative half power frequency cycle, MOSFET S ₅ and MOSFET S ₇ are controlled to be continuously conductive. The control is to alternate conduction, and MOSFET S ₆ and MOSFET S ₈ are controlled to be continuously conductive.

Figure 3 is a partial enlarged view of the pulse width modulation signal in Figure 2 during the positive half power frequency cycle. Figure 3 further shows the voltage between nodes A and B, the voltage between nodes C and D and the inductance. The waveform diagram of the current. Among them: the pulse width modulation signal provided to MOSFET S ₄ is delayed by d ₂ T (T is the period of the pulse width modulation signal) relative to the pulse width modulation signal of MOSFET S _1. The pulse width modulation signal provided to MOSFET S ₃ The signal is delayed d ₂ T with respect to the pulse width modulation signal of MOSFET S ₂ ; in addition, the pulse width modulation signal provided to MOSFET S ₆ is delayed d ₁ T with respect to the pulse width modulation signal of MOSFET S ₄ , which is provided to MOSFET S The pulse width modulation signal of ₈ is delayed by d ₁ T relative to the pulse width modulation signal of MOSFET S _3.

FIG. 4 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₀ -t ₁ shown in FIG. 3 . At time t ₀ -t ₁ , MOSFET S ₁ and MOSFET S ₄ are turned on, and MOSFET S ₅ , MOSFET S ₆ and MOSFET S ₇ are turned on, and the resulting current direction is shown by the dotted arrow in Figure 4 . On the primary side of the transformer Tr, the current flows from the positive electrode of the rechargeable battery 13, MOSFET S ₁ , node A, the primary side of the transformer Tr, node B, MOSFET S ₄ to the negative electrode of the rechargeable battery 13. At this time, the rechargeable battery 13 Discharge, and the voltage V _AB between nodes A and B is |V _b |. On the secondary side of transformer Tr, the current flows from the secondary side of transformer Tr, inductor 14, node C, MOSFET S ₆ , MOSFET S ₅ in sequence. Part of the current passes through capacitor C ₁ to node D, and the other part of the current passes through load 15 and Capacitor C ₂ reaches node D. At this time, inductor 14 stores energy, and the voltage V _CD between nodes C and D is 0.5 | V _O |.

FIG. 5 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₁ -t ₂ shown in FIG. 3 . At time t ₁ -t ₂ , MOSFET S ₂ and MOSFET S ₄ are turned on, and MOSFET S ₅ , MOSFET S ₆ and MOSFET S ₇ are turned on, and the resulting current direction is shown by the dotted arrow in Figure 5 . On the secondary side of transformer Tr, the current direction is the same as shown in Figure 4. At this time, the voltage V _CD between nodes C and D is 0.5 | V _O |. On the primary side of transformer Tr, the current flows from node A, the primary side of transformer Tr, node B, MOSFET S ₄ to MOSFET S ₂ in sequence. At this time, the voltage V _AB between nodes A and B is 0.

FIG. 6 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₂ -t ₃ shown in FIG. 3 . At time t ₂ -t ₃ , MOSFET S ₂ and MOSFET S ₃ are turned on, and MOSFET S ₅ , MOSFET S ₆ and MOSFET S ₇ are turned on. The direction of the current formed is as shown by the dotted arrow in Figure 6 . On the secondary side of transformer Tr, the current direction is the same as shown in Figure 4. At this time, the voltage V _CD between nodes C and D is 0.5 | V _O |. On the primary side of transformer Tr, the current flows from the negative electrode of rechargeable battery 13, MOSFET S ₂ , node A, the primary side of transformer Tr, node B, MOSFET S ₃ to the positive electrode of rechargeable battery 13. At this time, nodes A and B The voltage between V _AB is -|V _b |.

FIG. 7 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₃ -t ₄ shown in FIG. 3 . At time t ₃ -t ₄ , MOSFET S ₂ and MOSFET S ₃ are turned on, and MOSFET S ₅ , MOSFET S ₇ and MOSFET S ₈ are turned on. The direction of the current formed is as shown by the dotted arrow in Figure 7 . On the primary side of the transformer, the current flows sequentially from the positive electrode of the rechargeable battery 13, MOSFET S ₃ , node B, the primary side of the transformer Tr, node A, MOSFET S ₂ to the negative electrode of the rechargeable battery 13. At this time, the rechargeable battery 13 is discharged , and the voltage V _AB between nodes A and B is -|V _b |. On the secondary side of the transformer Tr, the current flows from the secondary side of the transformer Tr to node D. Part of the current passes through the capacitor C ₁ and the load 15. The other part of the current passes through the capacitor C ₂ and then passes through MOSFET S ₈ and MOSFET S _{7 in} sequence. , node C to inductor 14. At this time, inductor 14 stores energy, and the voltage V _CD between nodes C and D is -0.5 | V _O |.

FIG. 8 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₄ -t ₅ shown in FIG. 3 . At time t ₄ -t ₅ , MOSFET S ₁ and MOSFET S ₃ are turned on, and MOSFET S ₅ , MOSFET S ₇ and MOSFET S ₈ are turned on. The resulting current direction is shown by the dotted arrow in Figure 8 . On the secondary side of the transformer Tr, the current direction is the same as shown in Figure 7. At this time, the inductor 14 releases energy and provides it to the load 15, and the voltage V _CD between nodes C and D is -0.5 | V _O |. On the primary side of the transformer Tr, the current flows from the primary side of the transformer Tr, node A, MOSFET S ₁ , MOSFET S ₃ to node B. At this time, the voltage V _AB between nodes A and B is 0.

FIG. 9 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₅ -t ₆ shown in FIG. 3 . At time t ₅ -t ₆ , MOSFET S ₁ and MOSFET S ₄ are turned on, and MOSFET S ₅ , MOSFET S ₇ and MOSFET S ₈ are turned on, and the resulting current direction is shown by the dotted arrow in Figure 9 . On the secondary side of the transformer Tr, the current direction is the same as shown in Figure 7. At this time, the inductor 14 releases energy and provides it to the load 15, and the voltage V _CD between nodes C and D is -0.5 | V _O |. On the primary side of the transformer Tr, the current flows sequentially from the primary side of the transformer Tr, node A, MOSFET S ₁ , the positive and negative poles of the rechargeable battery 13 , MOSFET S ₄ to the contact B. At this time, the voltage between nodes A and B V _AB is |V _b |.

Figure 10 is a partial enlarged view of the pulse width modulation signal in Figure 2 during the negative half power frequency cycle. Figure 10 further shows the voltage between nodes A and B, the voltage between nodes C and D and the inductance. The waveform diagram of the current. Among them, the pulse width modulation signal provided to MOSFET S ₇ is delayed by d ₁ T relative to the pulse width modulation signal of MOSFET S ₄ ; the pulse width modulation signal provided to MOSFET S ₅ is delayed relative to the pulse width modulation of MOSFET S ₃ . The signal delay is d ₁ T.

FIG. 11 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₀ -t ₁ shown in FIG. 10 . At time t ₀ -t ₁ , MOSFET S ₁ and MOSFET S ₄ are turned on, and MOSFET S ₆ , MOSFET S ₇ and MOSFET S ₈ are turned on, forming a current direction as shown by the dotted arrow in Figure 11 . On the primary side of the transformer Tr, the current direction is the same as shown in Figure 4, which will not be described again. At this time, the voltage V _AB between nodes A and B is |V _b |. On the secondary side of transformer Tr, the current flows from the secondary side of transformer Tr, inductor 14, node C, MOSFET S ₇ , MOSFET S ₈ in sequence. Part of the current passes through capacitor C ₂ to node D, and the other part of the current passes through load 15 and Capacitor C ₁ to node D. At this time, inductor 14 stores energy, and the voltage V _CD between nodes C and D is 0.5 | V _O |.

FIG. 12 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₁ -t ₂ shown in FIG. 10 . At time t ₁ -t ₂ , MOSFET S ₂ and MOSFET S ₄ are turned on, and MOSFET S ₆ , MOSFET S ₇ and MOSFET S ₈ are turned on. The resulting current direction is shown by the dotted arrow in Figure 12 . On the primary side of the transformer Tr, the current direction is the same as that shown in Figure 5. On the secondary side of the transformer Tr, the current direction is the same as that shown in Figure 11, which will not be described again.

FIG. 13 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₂ -t ₃ shown in FIG. 10 . At time t ₂ -t ₃ , MOSFET S ₂ and MOSFET S ₃ are turned on, and MOSFET S ₆ , MOSFET S ₇ and MOSFET S ₈ are turned on. The direction of the current formed is as shown by the dotted arrow in Figure 13 . On the primary side of the transformer Tr, the current direction is the same as shown in Figure 7. On the secondary side of the transformer Tr, the current flows sequentially from the inductor 14, the secondary side of the transformer Tr, node D, capacitor C ₂ , MOSFET S ₈ , MOSFET S ₇ to node C. There is also reactive current in capacitor C ₂ , load 15 and capacitor C ₁ .

FIG. 14 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₃ -t ₄ shown in FIG. 10 . At time t ₃ -t ₄ , MOSFET S ₂ and MOSFET S ₃ are turned on, and MOSFET S ₅ , MOSFET S ₆ and MOSFET S ₈ are turned on. The direction of the current formed is shown by the dotted arrow in Figure 14 . On the primary side of the transformer Tr, the current direction is the same as shown in Figure 7. On the secondary side of the transformer Tr, the current flows from the secondary side of the transformer Tr to the node D. Part of the current passes through the capacitor C 2 and the load 15, and the other part of the current flows through the capacitor C ₂ and the load 15. After passing through capacitor C ₁ , it passes through MOSFET S ₅ , MOSFET S ₆ , and node C to inductor 14 in sequence. At this time, the inductor 14 stores energy, and the voltage V _CD between nodes C and D is -0.5 | V _O |.

FIG. 15 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₄ -t ₅ shown in FIG. 10 . At time t ₄ -t ₅ , MOSFET S ₁ and MOSFET S ₃ are turned on, and MOSFET S ₅ , MOSFET S ₆ and MOSFET S ₈ are turned on. The resulting current direction is shown by the dotted arrow in Figure 15 . On the secondary side of the transformer Tr, the current direction is the same as that shown in Figure 14. On the primary side of the transformer Tr, the current direction is the same as that shown in Figure 8, which will not be described again.

FIG. 16 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₅ -t ₆ shown in FIG. 10 . At time t ₅ -t ₆ , MOSFET S ₁ and MOSFET S ₄ are turned on, and MOSFET S ₅ , MOSFET S ₆ and MOSFET S ₈ are turned on. The resulting current direction is shown by the dotted arrow in Figure 16 . On the secondary side of transformer Tr, the current direction is from the secondary side of transformer Tr, inductor 14, node C, MOSFET S ₆ , MOSFET S ₅ , capacitor C ₁ to node D, among which capacitor C ₂ , load 15 and capacitor C There are also reactive currents in ₁ . On the primary side of the transformer Tr, the current direction is the same as shown in Figure 9 and will not be described again here.

Since the waveforms of i _L shown in Figure 3 and Figure 10 are the same, that is, the current i _L in the inductor 14 is at the time t ₀ -t ₆ of the positive half power frequency cycle and at the time t ₀ -t ₆ of the negative half power frequency cycle. same. The following does not distinguish between the positive half power frequency cycle and the negative half power frequency cycle. The current i _L in the inductor 14 is expressed by the following equation: Among them, n is the turns ratio between the primary side and the secondary side of the transformer Tr, and L is the inductance value of the inductor 14 . The relationship between t ₀ ~ t ₆ is expressed by the following equation:

According to the above equation, it can be concluded that the constraints of d ₁ and d ₂ are:

In addition, the waveform of the current i _L in the inductor 14 at time t ₀ -t ₃ is symmetrical with the waveform at time t ₃ -t ₆ , that is, it satisfies: i _L (t ₀ )=-i _L (t ₃ )=i _L ( _t6 ).

It can be concluded that the current i _L in the inductor 14 is expressed by the following equation:

The calculated output power P is expressed by the following equation: where f _s is the frequency of the pulse width modulation signal. If the resistance of the load 15 is R, when (-4d ₁ ² -2d ₂ ² -4d ₁ d ₂ +2d ₁ +d ₂ )>4f _s L/R is satisfied, the rechargeable battery 13 realizes boost discharge. When (-4d ₁ ² -2d ₂ ² -4d ₁ d ₂ +2d ₁ +d ₂ )＜4f _s L/R is satisfied, the rechargeable battery 13 realizes step-down discharge. It can be seen from this that the bidirectional DC-AC converter 1 of this embodiment can realize voltage step-up or step-down discharge.

The working principle of the battery charging mode of the bidirectional DC-AC converter 1 will be described below in conjunction with the equivalent circuit diagram.

Figure 17 is a waveform diagram of the pulse width modulation signal provided by the control device to the switching tube in the bidirectional DC-AC converter shown in Figure 1. As shown in Figure 17, MOSFET S ₁ and MOSFET S ₂ are controlled to alternately conduct, and MOSFET S ₃ and MOSFET S ₄ are also controlled to alternately conduct. During the positive half power frequency cycle, MOSFET S ₅ and MOSFET S ₇ are controlled to be continuously conductive, and MOSFET S ₆ and MOSFET S ₈ are controlled to be alternately conductive; during the negative half power frequency cycle, MOSFET S ₅ and MOSFET S ₇ are controlled to be continuously conductive. The control is to alternate conduction, and MOSFET S ₆ and MOSFET S ₈ are controlled to be continuously conductive.

Figure 18 is a partial enlarged view of the pulse width modulation signal in Figure 17 during the positive half power frequency cycle. Figure 18 further shows the voltage between nodes A and B, the voltage between nodes C and D and the inductance. The waveform diagram of the current. Among them, the pulse width modulation signal provided to MOSFET S ₈ is delayed by d ₂ 'T relative to the pulse width modulation signal of MOSFET S ₄ , and the pulse width modulation signal provided to MOSFET S ₂ is delayed relative to the pulse width modulation signal of MOSFET S ₈ . The variable signal delay is d ₁ 'T; similarly, the pulse width modulation signal provided to MOSFET S ₆ is delayed d ₂ 'T relative to the pulse width modulation signal of MOSFET S ₃ , and the pulse width modulation signal provided to MOSFET S ₁ is delayed relative to The pulse width modulation signal of MOSFET S ₆ is delayed by d ₁ 'T.

FIG. 19 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₀ -t ₁ shown in FIG. 18 . At time t ₀ -t ₁ , MOSFET S ₁ and MOSFET S ₄ are turned on, and MOSFET S ₅ , MOSFET S ₆ and MOSFET S ₇ are turned on, and the resulting current direction is shown by the dotted arrow in Figure 19 . On the secondary side of the transformer Tr, part of the current in the AC power supply 16 passes through the MOSFET S ₅ , MOSFET S ₆ , node C, the inductor 14 , the secondary side of the transformer Tr to the node D, and the other part of the current passes through the capacitor C ₁ to the node D, and finally to the AC power supply 16 through the capacitor C _2. At this time, the voltage V _CD between nodes C and D is 0.5 | V _O |. On the primary side of transformer Tr, the current flows from the negative electrode of rechargeable battery 13, MOSFET S ₄ , node B, the primary side of transformer Tr, node A, MOSFET S ₁ to the positive electrode of rechargeable battery 13. At this time, nodes A and B The voltage between V _AB is |V _b |.

FIG. 20 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₁ -t ₂ shown in FIG. 18 . At time t ₁ -t ₂ , MOSFET S ₁ and MOSFET S ₄ are turned on, and MOSFET S ₅ , MOSFET S ₇ and MOSFET S ₈ are turned on. The direction of the current formed is as shown by the dotted arrow in Figure 20 . On the secondary side of the transformer Tr, the current flows from the AC power supply 16, the capacitor C ₁ to the node D, and part of the current flows through the secondary side of the transformer Tr, the inductor 14, the node C, MOSFET S ₇ , and MOSFET S ₈ to the AC power supply. 16. The other part of the current directly returns to the AC power supply 16 through the capacitor C _2. At this time, the voltage V _CD between nodes C and D is -0.5 | V _O |. On the primary side of the transformer Tr, the current flows from the positive electrode of the rechargeable battery 13, MOSFET S ₁ , node A, the primary side of the transformer Tr, node B, MOSFET S ₄ to the negative electrode of the rechargeable battery 13. At this time, nodes A and B The voltage between V _AB is |V _b |.

FIG. 21 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₂ -t ₃ shown in FIG. 18 . At time t ₂ -t ₃ , MOSFET S ₂ and MOSFET S ₄ are turned on, and MOSFET S ₅ , MOSFET S ₇ and MOSFET S ₈ are turned on. The direction of the current formed is as shown by the dotted arrow in Figure 21 . On the secondary side of transformer Tr, the current direction is the same as shown in Figure 20. At this time, the voltage V _CD between nodes C and D is -0.5 | V _O |. On the primary side of the transformer Tr, the current flows from the primary side of the transformer Tr, node B, MOSFET S ₄ , MOSFET S ₂ to node A. At this time, the voltage V _AB between nodes A and B is 0.

FIG. 22 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₃ -t ₄ shown in FIG. 18 . At time t ₃ -t ₄ , MOSFET S ₂ and MOSFET S ₃ are turned on, and MOSFET S ₅ , MOSFET S ₇ and MOSFET S ₈ are turned on. The direction of the current formed is as shown by the dotted arrow in Figure 22 . On the secondary side of transformer Tr, the current direction is the same as in Figure 20. At this time, the voltage V _CD between nodes C and D is -0.5 | V _O |. On the primary side of transformer Tr, the current flows from the negative electrode of rechargeable battery 13, MOSFET S ₂ , node A, the primary side of transformer Tr, node B, MOSFET S ₃ to the positive electrode of rechargeable battery 13. At this time, nodes A and B The voltage between V _AB is -|V _b |.

FIG. 23 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₄ -t ₅ shown in FIG. 18 . At time t ₄ -t ₅ , MOSFET S ₂ and MOSFET S ₃ are turned on, and MOSFET S ₅ , MOSFET S ₆ and MOSFET S ₇ are turned on. The direction of the current formed is as shown by the dotted arrow in Figure 23 . On the secondary side of transformer Tr, the current direction is the same as in Figure 19. At this time, the voltage V _CD between nodes C and D is 0.5 | V _O |. On the primary side of transformer Tr, the current flows from the positive electrode of rechargeable battery 13, MOSFET S ₃ , node B, the primary side of transformer Tr, node A, MOSFET S ₂ to the negative electrode of rechargeable battery 13. At this time, nodes A and B The voltage between V _AB is -|V _b |.

FIG. 24 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₅ -t ₆ shown in FIG. 18 . At time t ₅ -t ₆ , MOSFET S ₁ and MOSFET S ₃ are turned on, and MOSFET S ₅ , MOSFET S ₆ and MOSFET S ₇ are turned on. The direction of the current formed is as shown by the dotted arrow in Figure 24 . On the secondary side of transformer Tr, the current direction is the same as in Figure 19. At this time, the voltage V _CD between nodes C and D is 0.5 | V _O |. On the primary side of the transformer Tr, the current flows from the primary side of the transformer Tr, node A, MOSFET S ₁ , MOSFET S ₃ to node B. At this time, the voltage V _AB between nodes A and B is 0.

Figure 25 is a partial enlarged view of the pulse width modulation signal in Figure 17 during the negative half power frequency cycle. Figure 25 further shows the voltage between nodes A and B, the voltage between nodes C and D and the inductance. The waveform diagram of the current. Among them, the pulse width modulation signal provided to MOSFET S ₅ is delayed by d ₂ 'T relative to the pulse width modulation signal of MOSFET S ₄ , and the pulse width modulation signal provided to MOSFET S ₂ is delayed relative to the pulse width modulation signal of MOSFET S ₅ . The variable signal delay is d ₁ 'T; similarly, the pulse width modulation signal provided to MOSFET S ₇ is delayed d ₂ 'T relative to the pulse width modulation signal of MOSFET S ₃ , and the pulse width modulation signal provided to MOSFET S ₁ is delayed relative to The pulse width modulation signal of MOSFET S ₇ is delayed by d ₁ 'T.

FIG. 26 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₀ -t ₁ shown in FIG. 25 . At time t ₀ -t ₁ , MOSFET S ₁ and MOSFET S ₄ are turned on, and MOSFET S ₆ , MOSFET S ₇ and MOSFET S ₈ are turned on. The direction of the current formed is as shown by the dotted arrow in Figure 26 . On the secondary side of the transformer Tr, the current flows from the secondary side of the transformer Tr to the node D. Part of the current returns to the AC power supply 16 through the capacitor C ₁ , and the other part of the current passes through the capacitor C ₂ and then passes through the MOSFET S ₈ , MOSFET S ₇ , node C to inductor 14. At this time, the voltage V _CD between nodes C and D is 0.5 | V _O |. On the primary side of the transformer Tr, the current direction is the same as shown in Figure 19. At this time, the voltage V _AB between nodes A and B is |V _b |.

FIG. 27 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₁ -t ₂ shown in FIG. 25 . At time t ₁ -t ₂ , MOSFET S ₁ and MOSFET S ₄ are turned on, and MOSFET S ₅ , MOSFET S ₆ and MOSFET S ₈ are turned on, and the resulting current direction is shown by the dotted arrow in Figure 27 . On the secondary side of transformer Tr, the current flows from the secondary side of transformer Tr, inductor 14, node C, MOSFET S ₆ to MOSFET S ₅ , part of the current passes through capacitor C ₁ to node D, and the other part passes through AC power supply 16 and capacitor C ₂ to node D. At this time, the voltage V _CD between nodes C and D is -0.5 | V _O |. On the primary side of the transformer Tr, the current direction is the same as shown in Figure 20. At this time, the voltage V _AB between nodes A and B is |V _b |.

FIG. 28 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₂ -t ₃ shown in FIG. 25 . At time t ₂ -t ₃ , MOSFET S ₂ and MOSFET S ₄ are turned on, and MOSFET S ₅ , MOSFET S ₆ and MOSFET S ₈ are turned on. The direction of the current formed is as shown by the dotted arrow in Figure 28 . On the secondary side of transformer Tr, the current direction is the same as shown in Figure 27. At this time, the voltage V _CD between nodes C and D is -0.5 | V _O |. On the primary side of transformer Tr, the current flows from MOSFET S ₂ , node A, the primary side of transformer Tr, node B to MOSFET S ₄ in sequence. At this time, the voltage V _AB between nodes A and B is 0.

FIG. 29 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₃ -t ₄ shown in FIG. 25 . At time t ₃ -t ₄ , MOSFET S ₂ and MOSFET S ₃ are turned on, and MOSFET S ₅ , MOSFET S ₆ and MOSFET S ₈ are turned on. The direction of the current formed is as shown by the dotted arrow in Figure 29 . On the secondary side of transformer Tr, the current direction is the same as in Figure 27. At this time, the voltage V _CD between nodes C and D is -0.5 | V _O |. On the primary side of transformer Tr, the current flows from the negative electrode of rechargeable battery 13, MOSFET S ₂ , node A, the primary side of transformer Tr, node B, MOSFET S ₃ to the positive electrode of rechargeable battery 13. At this time, nodes A and B The voltage between V _AB is -|V _b |.

FIG. 30 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₄ -t ₅ shown in FIG. 25 . At time t ₄ -t ₅ , MOSFET S ₂ and MOSFET S ₃ are turned on, and MOSFET S ₆ , MOSFET S ₇ and MOSFET S ₈ are turned on, and the direction of the current formed is as shown by the dotted arrow in Figure 30 . On the secondary side of transformer Tr, the current direction is the same as in Figure 26. At this time, the voltage V _CD between nodes C and D is 0.5 | V _O |. On the primary side of transformer Tr, the current flows from the positive electrode of rechargeable battery 13, MOSFET S ₃ , node B, the primary side of transformer Tr, node A, MOSFET S ₂ to the negative electrode of rechargeable battery 13. At this time, nodes A and B The voltage between V _AB is -|V _b |.

FIG. 31 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₅ -t ₆ shown in FIG. 25 . At time t ₅ -t ₆ , MOSFET S ₁ and MOSFET S ₃ are turned on, and MOSFET S ₆ , MOSFET S ₇ and MOSFET S ₈ are turned on. The direction of the current formed is as shown by the dotted arrow in Figure 31 . On the secondary side of transformer Tr, the current direction is the same as in Figure 26. At this time, the voltage V _CD between nodes C and D is 0.5 | V _O |. On the primary side of the transformer Tr, the current flows from the primary side of the transformer Tr, node A, MOSFET S ₁ , MOSFET S ₃ to node B. At this time, the voltage V _AB between nodes A and B is 0.

In battery charging mode, the expression of the output power P can also be calculated as follows:

According to the above conclusion, it can be seen that in the battery charging mode, the bidirectional DC-AC converter 1 of the above embodiment can also realize voltage step-up operation or step-down operation. The expressions of the charging power in the charging mode and the discharging power in the discharging mode are similar. It can be seen that the bidirectional DC-AC converter 1 has larger charging power and charging current. However, the charger in the UPS of the prior art is a separate flyback circuit, and its charging power is much smaller than the rated output power of the UPS. Therefore, compared with the charger in the prior art UPS, the charging power and charging current are significantly increased.

Figure 32 is a waveform diagram of the voltage and current of the AC power supply in battery charging mode. Figure 32 further shows the waveform diagram of the current i _L in the inductor 14 and the current i _C in the capacitor C1 or C2, and the current i _O in the AC power supply is equal to i _L -i _C , so the current in the AC power supply i _O is in the same phase as the voltage V _O of the AC power supply, thus realizing the power factor correction function.

No matter in the battery discharge mode or the battery charging mode, the voltages of MOSFET S ₁ ~ MOSFET S ₄ are all clamped below V _b , and the voltages of MOSFET S ₅ ~ MOSFET S ₈ are all clamped at 0.5|V _O | below, so there is no overshoot voltage, avoiding the failure of components in the circuit.

The bidirectional DC-AC converter of the present invention can realize discharge mode or charging mode through one conversion, has high conversion efficiency, omits a charger, has a small number of components, low cost, and high power density.

Embodiments of the present invention also provide another battery discharging method for the bidirectional DC-AC converter 1 . The working principle of the battery discharge mode will be described below with reference to FIGS. 33 to 35 .

Figure 33 is a waveform diagram of the pulse width modulation signal provided by the control device to the switching tube in the bidirectional DC-AC converter shown in Figure 1. As shown in Figure 33, MOSFET S ₁ and MOSFET S ₂ are controlled to alternately conduct, and MOSFET S ₃ and MOSFET S ₄ are also controlled to alternately conduct. When the pulse width modulation signal is provided to MOSFET S ₁ and MOSFET S ₄ The delay difference is zero. Similarly, the delay difference of the pulse width modulation signals provided to MOSFET S ₂ and MOSFET S ₃ is zero. During the positive half power frequency cycle, MOSFET S ₅ and MOSFET S ₇ are controlled to be continuously conductive, and MOSFET S ₆ and MOSFET S ₈ are controlled to be alternately conductive; during the negative half power frequency cycle, MOSFET S ₅ and MOSFET S ₇ are controlled to be continuously conductive. The control is to alternate conduction, and MOSFET S ₆ and MOSFET S ₈ are controlled to be continuously conductive.

Figure 34 is a partially enlarged view of the pulse width modulation signal in Figure 33 during the positive half power frequency cycle. Figure 34 further shows the voltage between nodes A and B, the voltage between nodes C and D and the inductance. Current waveform diagram. The pulse width modulation signal provided to MOSFET S ₆ is delayed by d ₁ T relative to the pulse width modulation signal of MOSFET S _4. Similarly, the pulse width modulation signal provided to MOSFET S ₈ is delayed relative to the pulse width modulation signal of MOSFET S ₃ . The variable signal delay is d ₁ T.

Among them, the operating mode of the bidirectional DC-AC converter 1 at time t ₀ -t ₁ in Figure 34 is the same as the operating mode at time t ₀ -t ₁ in Figure 3 , and the operation mode at time t ₁ -t ₃ in Figure 34 The working mode is the same as the working mode at time t ₁ -t ₃ in Figure 3. The working mode at time t ₃ -t ₄ in Figure 34 is the same as the working mode at time t 3 -t 4 in Figure 3. The working mode at time t ₃ -t ₄ in Figure 34 is the same. The working mode at time t ₄ -t ₆ is the same as the working mode at time t ₄ -t ₆ in Figure 3 and will not be described again here.

Figure 35 is a partial enlarged view of the pulse width modulation signal in Figure 33 during the negative half power frequency cycle. Figure 35 further shows the voltage between nodes A and B, the voltage between nodes C and D and the inductance. Current waveform diagram. Among them, the pulse width modulation signal provided to MOSFET S ₇ is delayed by d ₁ T relative to the pulse width modulation signal of MOSFET S ₄ ; the pulse width modulation signal provided to MOSFET S ₅ is delayed relative to the pulse width modulation of MOSFET S ₃ . The signal delay is d ₁ T.

Among them, the operating mode of the bidirectional DC-AC converter 1 at time t ₀ -t ₁ in Figure 35 is the same as the operating mode at time t ₀ -t ₁ in Figure 10 , and the operation mode at time t ₁ -t ₃ in Figure 35 The working mode is the same as the working mode at time t ₁ -t ₃ in Figure 10. The working mode at time t ₃ -t ₄ in Figure 35 is the same as the working mode at time t ₃ -t ₄ in Figure 10. In Figure 35 The working mode at time t ₄ -t ₆ is the same as the working mode at time t ₄ -t ₆ in Figure 10 and will not be described again here.

In the battery discharge mode, since d ₂ T=0, the expression of the output power P can also be calculated as follows:

The working principle of the battery charging mode will be described below with reference to Figures 36 to 38.

Figure 36 is a waveform diagram of the pulse width modulation signal provided by the control device to the switching tube in the bidirectional DC-AC converter shown in Figure 1. As shown in Figure 36, MOSFET S ₁ and MOSFET S ₂ are controlled to alternately conduct, and MOSFET S ₃ and MOSFET S ₄ are also controlled to alternately conduct. When the pulse width modulation signal is provided to MOSFET S ₁ and MOSFET S ₄ The delay difference is zero. Similarly, the delay difference of the pulse width modulation signals provided to MOSFET S ₂ and MOSFET S ₃ is zero. During the positive half power frequency cycle, MOSFET S ₅ and MOSFET S ₇ are controlled to be continuously conductive, and MOSFET S ₆ and MOSFET S ₈ are controlled to be alternately conductive; during the negative half power frequency cycle, MOSFET S ₅ and MOSFET S ₇ are controlled to be continuously conductive. The control is to alternate conduction, and MOSFET S ₆ and MOSFET S ₈ are controlled to be continuously conductive.

Figure 37 is a partially enlarged view of the pulse width modulation signal in Figure 36 during the positive half power frequency cycle. Figure 37 further shows the voltage between nodes A and B, the voltage between nodes C and D and the inductance. Current waveform diagram. Among them, the pulse width modulation signal provided to MOSFET S ₂ is delayed d ₁ 'T compared to the pulse width modulation signal provided to MOSFET S ₈ ; the pulse width modulation signal provided to MOSFET S ₁ is delayed compared to the pulse width modulation signal provided to MOSFET S ₆ . The variable signal delay is d ₁ 'T; the pulse width modulation signal provided to MOSFET S ₈ is compared to the pulse width modulation signal provided to MOSFET S ₁ with a delay d ₂ 'T, and the pulse width modulation signal ratio provided to MOSFET S ₆ is provided Give the pulse width modulation signal of MOSFET S ₂ a delay of d ₂ 'T.

Among them, the operating mode of the bidirectional DC-AC converter 1 at time t ₀ -t ₁ in Figure 37 is the same as the operating mode at time t ₄ -t ₅ in Figure 18 , and the operation mode at time t ₁ -t ₂ in Figure 37 The working mode is the same as the working mode at time t ₀ -t ₁ in Figure 18. The working mode at time t ₂ -t ₃ in Figure 37 is the same as the working mode at time t ₁ -t ₂ in Figure 18. The working mode at time t in Figure 37 The working mode of ₃ - t ₄ is the same as the working mode of time t ₃ - t ₄ in Figure 18 and will not be described again here.

Figure 38 is a partially enlarged view of the pulse width modulation signal in Figure 36 during the negative half power frequency cycle. Figure 38 further shows the voltage between nodes A and B, the voltage between nodes C and D and the inductance. Current waveform diagram. Among them, the pulse width modulation signal provided to MOSFET S ₅ is delayed by d ₂ 'T relative to the pulse width modulation signal of MOSFET S _4. Similarly, the pulse width modulation signal provided to MOSFET S ₇ is delayed relative to the pulse width modulation signal of MOSFET S ₃ . The wide modulation signal is delayed by d ₂ 'T. The pulse width modulation signal provided to MOSFET S ₃ is delayed by d ₁ 'T relative to the pulse width modulation signal of MOSFET S ₅ , and the pulse width modulation signal provided to MOSFET S ₄ is delayed relative to the pulse width modulation signal of MOSFET S ₇ . Delay d ₁ 'T.

Among them, the operating mode of the bidirectional DC-AC converter 1 at time t ₀ -t ₁ in Figure 38 is the same as the operating mode at time t ₄ -t ₅ in Figure 25 , and the operation mode at time t ₁ -t ₂ in Figure 38 The working mode is the same as the working mode at time t ₀ -t ₁ in Figure 25. The working mode at time t2-t3 in Figure 38 is the same as the working mode at time t ₁ -t ₂ in Figure 25. The working mode at time t ₃ - in Figure 38 The working mode at t ₄ is the same as the working mode at time t ₃ - _{t 4} in Figure 25 and will not be described again.

Figure 39 is a circuit diagram of a bidirectional DC-AC converter according to the second embodiment of the present invention. Figure 39 is basically the same as Figure 1, except that a half-bridge inverter 21 is used instead of the full-bridge inverter 11 in Figure 1. That is, the capacitor C ₃ and the capacitor C ₄ in the bidirectional DC-AC converter 2 are connected in sequence between the positive electrode and the negative electrode of the rechargeable battery 13, and the node B formed by the connection of the capacitors C ₃ and C ₄ is connected to the transformer Tr. One end of the primary side (i.e. the end with the same name).

The working principle of the battery discharge mode of the bidirectional DC-AC converter 2 will be described below with reference to FIGS. 40 to 42 .

Figure 40 is a waveform diagram of the pulse width modulation signal provided by the control device to the switching tube in the bidirectional DC-AC converter shown in Figure 39. As shown in Figure 40, the pulse width modulation signals provided to MOSFET S ₁ , MOSFET S ₂ , and MOSFET S ₅ ~ S ₈ in Figure 39 are exactly the same as the pulse width modulation signals provided to the corresponding MOSFETs in Figure 33 .

Figure 41 is a partially enlarged view of the pulse width modulation signal in Figure 40 during the positive half power frequency cycle. Figure 41 further shows the voltage between nodes A and B, the voltage between nodes C and D and the inductance. Current waveform diagram. Among them, the working modes of the bidirectional DC-AC converter 2 at times t ₀ -t ₁ , t ₁ -t ₂ , t ₂ -t ₃ , and t ₃ -t ₄ in Figure 41 are respectively the same as those of the bidirectional DC-AC converter 1 at The working modes at time t ₀ -t ₁ , t ₁ -t ₃ , t ₃ -t ₄ and t ₄ -t ₆ in Figure 34 are the same and will not be described again.

Figure 42 is a partial enlarged view of the pulse width modulation signal in Figure 40 during the negative half power frequency cycle. Figure 42 further shows the voltage between nodes A and B, the voltage between nodes C and D and the inductance. Waveform diagram of current. Among them, the working modes of the bidirectional DC-AC converter 2 at times t ₀ -t ₁ , t ₁ -t ₂ , t ₂ -t ₃ , and t ₃ -t ₄ in Figure 42 are respectively the same as those of the bidirectional DC-AC converter 1 at The working modes at times t ₀ -t ₁ , t ₁ -t ₃ , t ₃ -t ₄ and t ₄ -t ₆ in Figure 35 are the same and will not be described again.

The working principle of the battery charging mode of the bidirectional DC-AC converter 2 will be described below with reference to FIGS. 43 to 45 .

Figure 43 is a waveform diagram of the pulse width modulation signal provided by the control device to the switching tube in the bidirectional DC-AC converter shown in Figure 39. As shown in Figure 43, the pulse width modulation signals provided to MOSFET S ₁ , MOSFET S ₂ , and MOSFET S ₅ ~ S ₈ in Figure 39 are exactly the same as the pulse width modulation signals provided to the corresponding MOSFETs in Figure 36 .

Figure 44 is a partially enlarged view of the pulse width modulation signal in Figure 43 during the positive half power frequency cycle. Figure 44 further shows the voltage between nodes A and B, the voltage between nodes C and D and the inductance. Current waveform diagram. Among them, the operating modes of the bidirectional DC-AC converter 2 at times t ₀ -t ₁ , t ₁ -t ₂ , t ₂ -t ₃ , and t ₃ -t ₄ in Figure 44 are respectively the same as those of the bidirectional DC-AC converter 1 at The working modes at time t ₀ -t ₁ , t ₁ -t ₂ , t ₂ -t ₃ , and t ₃ -t ₄ in Figure 37 are the same and will not be described again.

Figure 45 is a partial enlarged view of the pulse width modulation signal in Figure 43 during the negative half power frequency cycle. Figure 45 further shows the voltage between nodes A and B, the voltage between nodes C and D and the inductance. Current waveform diagram. Among them, the working modes of the bidirectional DC-AC converter 2 at times t ₀ -t ₁ , t ₁ -t ₂ , t ₂ -t ₃ , and t ₃ -t ₄ in Figure 45 are respectively the same as those of the bidirectional DC-AC converter 1 at The working modes at time t ₀ -t ₁ , t ₁ -t ₂ , t ₂ -t ₃ , and t ₃ -t ₄ in Figure 38 are the same and will not be described again.

Figure 46 is a circuit diagram of a bidirectional DC-AC converter according to the third embodiment of the present invention. It is basically the same as Figure 1. The difference is that a full-bridge AC-AC converter 32 is used instead of the half-bridge AC-AC converter 12 in Figure 1, that is, the capacitor C is replaced by MOSFET S ₉ and MOSFET S ₁₀ connected in reverse series. ₁ , and the capacitor C ₂ is replaced by MOSFET S ₁₁ and MOSFET S ₁₂ connected in reverse series. When MOSFET S ₅ , MOSFET S ₇ , MOSFET S ₉ and MOSFET S ₁₁ are controlled to be turned on, MOSFET S ₆ , MOSFET S ₈ , MOSFET S ₁₀ and MOSFET S ₁₂ form a full-bridge inverter; and when MOSFET S _6. When MOSFET S ₈ , MOSFET S ₁₀ and MOSFET S ₁₂ are controlled to be turned on, MOSFET S ₅ , MOSFET S ₇ , MOSFET S ₉ and MOSFET S ₁₁ form another full-bridge inverter.

Among them, in the battery discharge mode, the control device provides the pulse width modulation signals shown in Figure 2 to MOSFET S ₁ ~ MOSFET S ₈ , and provides MOSFET S ₁₁ and MOSFET S ₁₂ with the same signal as MOSFET S ₅ and MOSFET S ₆ . The same pulse width modulation signal provides MOSFET S ₉ and MOSFET S ₁₀ with identical pulse width modulation signals to MOSFET S ₇ and MOSFET S ₈ .

In the battery charging mode, the control device provides pulse width modulation signals as shown in Figure 17 to MOSFET S ₁ ~ MOSFET S ₈ , and provides MOSFET S ₁₁ and MOSFET S ₁₂ with the same pulse as MOSFET S ₅ and MOSFET S ₆ . The wide modulation signal provides MOSFET S ₉ and MOSFET S ₁₀ with the same pulse width modulation signal as MOSFET S ₇ and MOSFET S ₈ .

In another bidirectional DC-AC converter of the present invention, capacitors are used to replace MOSFET S ₃ and MOSFET S ₄ in FIG. 46 .

In other embodiments of the present invention, the inductor 14 is connected between the secondary side of the transformer Tr and the node D, or between the primary side of the transformer Tr and the node A or B.

In other embodiments of the present invention, the bidirectional DC-AC converter further includes a filter capacitor connected in parallel with the rechargeable battery for high-frequency filtering in the battery charging mode, thereby effectively protecting the rechargeable battery.

In other embodiments of the present invention, an insulated gate bipolar transistor is used to replace the MOSFET in the above embodiment.

Although the present invention has been described through preferred embodiments, the present invention is not limited to the embodiments described herein, and further includes various changes and changes made without departing from the scope of the present invention.

1‧‧‧Bidirectional DC-AC converter 2‧‧‧Bidirectional DC-AC converter 11‧‧‧Full-bridge inverter 12‧‧‧Half-bridge AC-AC converter 13‧‧‧Rechargeable battery 14‧‧ ‧Inductor 15‧‧‧Load 16‧‧‧AC power supply 17‧‧‧Control device 21‧‧‧Half-bridge inverter 32‧‧‧Full-bridge AC-AC converter 121‧‧‧Bidirectional controllable switch tube 122‧ ‧‧Bidirectional controllable switch A‧‧‧Node B‧‧‧Node C‧‧‧Node C ₁ ‧‧‧Capacitor C ₂ ‧‧‧Capacitor C ₃ ‧‧‧Capacitor C ₄ ‧‧‧Capacitor D‧‧‧ Node S ₁ ‧‧‧Metal Oxygen Semi Field Effect Transistor S ₂ ‧‧‧Metal Oxygen Semi Field Effect Transistor S ₃ ‧‧‧Metal Oxygen Semi Field Effect Transistor S ₄ ‧‧‧Metal Oxygen Semi Field Effect Transistor S ₅ ‧‧‧ Metal Oxygen Semi Field Effect Transistor S ₆ ‧‧‧Metal Oxygen Semi Field Effect Transistor S ₇ ‧‧‧Metal Oxygen Semi Field Effect Transistor S ₈ ‧‧‧Metal Oxygen Semi Field Effect Transistor S ₉ ‧‧‧Metal Oxygen Semi Field Effect Transistor S ₁₀ ‧‧‧Metal Oxygen Semi Field Effect Transistor S ₁₁ ‧‧‧Metal Oxygen Semi Field Effect Transistor S ₁₂ ‧‧‧Metal Oxygen Semi Field Effect Transistor Tr‧‧‧Transformer V _o ‧‧‧Voltage

The embodiments of the present invention will be further described below with reference to the accompanying drawings, in which: Figure 1 is a circuit diagram of a bidirectional DC-AC converter according to the first embodiment of the present invention. Figure 2 is a waveform diagram of the pulse width modulation signal provided by the control device to the switching tube in the bidirectional DC-AC converter shown in Figure 1. Figure 3 is a partial enlarged view of the pulse width modulation signal in Figure 2 during the positive half power frequency cycle. FIG. 4 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₀ -t ₁ shown in FIG. 3 . FIG. 5 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₁ -t ₂ shown in FIG. 3 . FIG. 6 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₂ -t ₃ shown in FIG. 3 . FIG. 7 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₃ -t ₄ shown in FIG. 3 . FIG. 8 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₄ -t ₅ shown in FIG. 3 . FIG. 9 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₅ -t ₆ shown in FIG. 3 . Figure 10 is a partial enlarged view of the pulse width modulation signal in Figure 2 during the negative half power frequency cycle. FIG. 11 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₀ -t ₁ shown in FIG. 10 . FIG. 12 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₁ -t ₂ shown in FIG. 10 . FIG. 13 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₂ -t ₃ shown in FIG. 10 . FIG. 14 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₃ -t ₄ shown in FIG. 10 . FIG. 15 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t4-t5 shown in FIG. 10 . FIG. 16 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₅ -t ₆ shown in FIG. 10 . Figure 17 is a waveform diagram of the pulse width modulation signal provided by the control device to the switching tube in the bidirectional DC-AC converter shown in Figure 1. Figure 18 is a partial enlarged view of the pulse width modulation signal in Figure 17 during the positive half power frequency cycle. FIG. 19 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₀ -t ₁ shown in FIG. 18 . FIG. 20 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₁ -t ₂ shown in FIG. 18 . FIG. 21 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₂ -t ₃ shown in FIG. 18 . FIG. 22 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₃ -t ₄ shown in FIG. 18 . FIG. 23 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₄ -t ₅ shown in FIG. 18 . FIG. 24 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₅ -t ₆ shown in FIG. 18 . Figure 25 is a partial enlarged view of the pulse width modulation signal in Figure 17 during the negative half power frequency cycle. FIG. 26 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₀ -t ₁ shown in FIG. 25 . FIG. 27 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₁ -t ₂ shown in FIG. 25 . FIG. 28 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₂ -t ₃ shown in FIG. 25 . FIG. 29 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₃ -t ₄ shown in FIG. 25 . FIG. 30 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₄ -t ₅ shown in FIG. 25 . FIG. 31 is an equivalent circuit diagram of the bidirectional DC-AC converter shown in FIG. 1 at time t ₅ -t ₆ shown in FIG. 25 . Figure 32 is a waveform diagram of the voltage and current of the AC power supply in battery charging mode. Figure 33 is a waveform diagram of the pulse width modulation signal provided by the control device to the switching tube in the bidirectional DC-AC converter shown in Figure 1. Figure 34 is a partial enlarged view of the pulse width modulation signal in Figure 33 during the positive half power frequency cycle. Figure 35 is a partial enlarged view of the pulse width modulation signal in Figure 33 during the negative half power frequency cycle. Figure 36 is a waveform diagram of the pulse width modulation signal provided by the control device to the switching tube in the bidirectional DC-AC converter shown in Figure 1. Figure 37 is a partial enlarged view of the pulse width modulation signal in Figure 36 during the positive half power frequency cycle. Figure 38 is a partial enlarged view of the pulse width modulation signal in Figure 36 during the negative half power frequency cycle. FIG. 39 is a circuit diagram of a bidirectional DC-AC converter according to the second embodiment of the present invention. Figure 40 is a waveform diagram of the pulse width modulation signal provided by the control device to the switching tube in the bidirectional DC-AC converter shown in Figure 39. Figure 41 is a partial enlarged view of the pulse width modulation signal in Figure 40 during the positive half power frequency cycle. Figure 42 is a partial enlarged view of the pulse width modulation signal in Figure 40 during the negative half power frequency cycle. Figure 43 is a waveform diagram of the pulse width modulation signal provided by the control device to the switching tube in the bidirectional DC-AC converter shown in Figure 39. Figure 44 is a partial enlarged view of the pulse width modulation signal in Figure 43 during the positive half power frequency cycle. Figure 45 is a partial enlarged view of the pulse width modulation signal in Figure 43 during the negative half power frequency cycle. Figure 46 is a circuit diagram of a bidirectional DC-AC converter according to the third embodiment of the present invention.

1‧‧‧Bidirectional DC-AC converter

11‧‧‧Full bridge inverter

12‧‧‧Half-bridge AC-AC converter

13‧‧‧Rechargeable battery

14‧‧‧Inductor

15‧‧‧Load

17‧‧‧Control device

121‧‧‧Bidirectional controllable switch tube

122‧‧‧Bidirectional controllable switch tube

A‧‧‧Node

B‧‧‧Node

C‧‧‧Node

C ₁ ‧‧‧Capacitor

C ₂ ‧‧‧Capacitor

D‧‧‧Node

S ₁ ‧‧‧Metal Oxygen Semi Field Effect Transistor

S ₂ ‧‧‧Metal Oxygen Semi Field Effect Transistor

S ₃ ‧‧‧Metal Oxygen Semi Field Effect Transistor

S ₄ ‧‧‧Metal Oxygen Semi Field Effect Transistor

S ₅ ‧‧‧Metal Oxygen Semi Field Effect Transistor

S ₆ ‧‧‧Metal Oxygen Semi Field Effect Transistor

S ₇ ‧‧‧Metal Oxygen Semi Field Effect Transistor

S ₈ ‧‧‧Metal Oxygen Semi Field Effect Transistor

Claims (17)

A bidirectional DC-AC converter, characterized in that it includes: a full-bridge or half-bridge inverter; a transformer, the primary side of the transformer is connected to the AC end of the full-bridge or half-bridge inverter; AC-AC A converter having a first AC terminal connected to the secondary side of the transformer and a second AC terminal configured to be connected to a load or an AC power source; and An inductor connected between the primary side of the transformer and the AC terminal of the full-bridge or half-bridge inverter, or between the secondary side of the transformer and the first AC terminal of the AC-AC converter between ends. The bidirectional DC-AC converter described in item 1 of the patent application, wherein the AC-AC converter is a half-bridge AC-AC converter composed of two bidirectional controllable switches connected in series and two capacitors connected in series. In the AC converter, the node formed by the two bidirectionally controllable switches connected in series and the node formed by the two capacitors connected in series serve as the first AC terminal. The bidirectional DC-AC converter as described in item 2 of the patent application, wherein each of the two bidirectionally controllable switch transistors connected in series includes two switch transistors connected in reverse series. The bidirectional DC-AC converter described in item 3 of the patent application, wherein the half-bridge AC-AC converter includes: a fifth switch tube and a sixth switch tube connected in reverse series, and a third switch tube connected in reverse series. Seven switching tubes and an eighth switching tube; when the fifth switching tube and the seventh switching tube are controlled to be turned on, the sixth switching tube, the eighth switching tube and the two capacitors form a first half-bridge inverter. and when the sixth switch tube and the eighth switch tube are controlled to be turned on, the fifth switch tube, the seventh switch tube and the two capacitors form a second half-bridge inverter. The bidirectional DC-AC converter described in item 1 of the patent application, wherein the AC-AC converter is a full-bridge AC-AC converter composed of four bidirectional controllable switching tubes, and the full-bridge AC-AC converter is The nodes of the two bridge arms of the AC converter serve as the first AC terminal. The bidirectional DC-AC converter as described in item 5 of the patent application, wherein each of the four bidirectional controllable switch tubes includes two switch tubes connected in reverse series. The bidirectional DC-AC converter described in item 6 of the patent application, wherein the full-bridge AC-AC converter includes: a fifth switch tube and a sixth switch tube connected in reverse series, and a seventh switch tube connected in reverse series. The switching tube and the eighth switching tube, the ninth switching tube and the tenth switching tube connected in reverse series, the eleventh switching tube and the twelfth switching tube connected in reverse series; wherein the fifth, seventh and ninth switching tubes are And when the eleventh switching tube is controlled to be turned on, the sixth, eighth, tenth and twelfth switching tubes constitute the first full-bridge inverter, and when the sixth, eighth, tenth and twelfth switching tubes When the twelfth switching tube is controlled to be turned on, the fifth, seventh, ninth and eleventh switching tubes constitute a second full-bridge inverter. The bidirectional DC-AC converter as described in item 1 of the patent application, wherein the bidirectional DC-AC converter further includes a filter capacitor connected between the DC terminals of the full-bridge or half-bridge inverter. The bidirectional DC-AC converter as described in item 1 of the patent application, wherein the bidirectional DC-AC converter further includes a control device, the control device is used to control the full bridge when the AC power supply fails. Or a half-bridge inverter to convert the DC power at its DC end into a first AC square wave, and control the AC-AC converter to convert the second AC square wave at its first AC end into power frequency AC power, wherein The period of the first AC square wave and the second AC square wave is the period of the pulse width modulation signal provided to the full-bridge or half-bridge inverter; when the AC power supply is normal, the AC-AC converter is controlled To convert the power frequency alternating current at its second AC end into a third AC square wave, and control the full-bridge or half-bridge inverter to convert the fourth AC square wave at its AC end into direct current, and the third The period of the AC square wave and the fourth AC square wave is the period of the pulse width modulation signal provided to the full-bridge or half-bridge inverter. A control method for a bidirectional DC-AC converter as described in item 1 of the patent application. The full-bridge inverter includes a first switch tube and a second switch connected to its DC end in sequence. tube, and a third switching tube and a fourth switching tube connected in sequence to its DC end. The half-bridge AC-AC converter includes: a fifth switching tube and a sixth switching tube connected in reverse series, and a fifth switching tube and a sixth switching tube connected in reverse series. The seventh switch tube and the eighth switch tube; when the fifth switch tube and the seventh switch tube are controlled to be turned on, the sixth switch tube, the eighth switch tube and the two capacitors form a first half-bridge inverter. inverter, and when the sixth switch tube and the eighth switch tube are controlled to be turned on, the fifth switch tube, the seventh switch tube and the two capacitors constitute a second half-bridge inverter, which is characterized in that, The control method includes: during the positive half power frequency cycle, controlling the first switch tube and the second switch tube to alternately conduct, controlling the third switch tube and the fourth switch tube to alternately conduct, controlling the fifth and fourth switch tubes to alternately conduct. The seventh switch tube is turned on, and the sixth and eighth switch tubes are controlled to be turned on alternately; during the negative half power frequency cycle, the first switch tube and the second switch tube are controlled to be turned on alternately, and the third switch tube and The fourth switch tube is alternately turned on, and the fifth and seventh switch tubes are controlled to be turned on alternately, and the sixth and eighth switch tubes are controlled to be turned on. The control method as described in item 10 of the patent application, wherein in the positive half power frequency cycle, the pulse width modulation signal provided to the sixth switching tube is greater than the pulse width modulation signal provided to the fourth switching tube. The variable signal is delayed for a first period of time, and the pulse width modulation signal provided to the fourth switch is delayed by a second period of time than the pulse width modulation signal provided to the first switch; and to the eighth switch The pulse width modulation signal provided by the tube is delayed by a first period of time than the pulse width modulation signal provided to the third switch tube, and the pulse width modulation signal provided to the third switch tube is delayed by a first period of time than the pulse width modulation signal provided to the second switch. The pulse width modulation signal provided by the tube is delayed for a second period of time; in the negative half power frequency cycle, the pulse width modulation signal provided to the seventh switching tube is greater than the pulse width modulation signal provided to the fourth switching tube. The signal is delayed for a first period of time, and the pulse width modulation signal provided to the fourth switching tube is delayed by a second period of time than the pulse width modulation signal provided to the first switching tube; and to the fifth switching tube The pulse width modulation signal provided is delayed by a first period of time compared with the pulse width modulation signal provided to the third switching tube, and the pulse width modulation signal provided to the third switching tube is delayed by a first period of time than the pulse width modulation signal provided to the second switching tube. provided The pulse width modulated signal is delayed for a second period of time. The control method as described in item 10 of the patent application, wherein in the positive half power frequency cycle, the pulse width modulation signal provided to the second switching tube is greater than the pulse width modulation signal provided to the eighth switching tube. The variable signal is delayed for a third time period, and the pulse width modulation signal provided to the eighth switch tube is delayed by a fourth time period than the pulse width modulation signal provided to the fourth switch tube; and to the first switch The pulse width modulation signal provided by the tube is delayed by a third period of time than the pulse width modulation signal provided to the sixth switching tube, and the pulse width modulation signal provided to the sixth switching tube is delayed by a third period of time than the pulse width modulation signal provided to the third switch. The pulse width modulation signal provided by the tube is delayed for a fourth time period; in the negative half power frequency cycle, the pulse width modulation signal provided to the second switching tube is greater than the pulse width modulation signal provided to the fifth switching tube. The signal is delayed for a third time period, and the pulse width modulation signal provided to the fifth switching tube is delayed by a fourth time period than the pulse width modulation signal provided to the fourth switching tube; and to the first switching tube The pulse width modulation signal provided is delayed by a third period of time than the pulse width modulation signal provided to the seventh switching tube, and the pulse width modulation signal provided to the seventh switching tube is delayed by a third period of time. The provided pulse width modulation signal is delayed for a fourth time period. The control method as described in item 10 of the patent application, wherein a pulse width modulation signal with a delay difference of zero is provided to the first switch tube and the fourth switch tube, and a pulse width modulation signal with a delay difference of zero is provided to the second switch tube and the third switch tube. The tube provides a pulse width modulation signal with a delay difference of zero; and in the positive half power frequency cycle, the pulse width modulation signal provided to the sixth switching tube is greater than the pulse width modulation signal provided to the first switching tube. The signal is delayed for a fifth time period, and the pulse width modulation signal provided to the eighth switching tube is delayed by a fifth time period than the pulse width modulation signal provided to the second switching tube; during the negative half power frequency cycle, The pulse width modulation signal provided to the seventh switching tube is delayed by a fifth period of time than the pulse width modulation signal provided to the first switching tube, and the pulse width modulation signal provided to the fifth switching tube is delayed by a fifth period. The pulse width modulation signal provided to the second switching tube is delayed for a fifth time period. The control method as described in item 10 of the patent application, wherein the first opening The switching tube and the fourth switching tube provide a pulse width modulation signal with a delay difference of zero, and the second switching tube and the third switching tube provide a pulse width modulation signal with a delay difference of zero; and in the positive half power frequency cycle Within, the pulse width modulation signal provided to the first switching tube is delayed by a sixth period of time than the pulse width modulation signal provided to the sixth switching tube, and the pulse width modulation signal provided to the sixth switching tube is The signal is delayed by a seventh period of time than the pulse width modulation signal provided to the second switching tube; the pulse width modulation signal provided to the second switching tube is slower than the pulse width modulation signal provided to the eighth switching tube. The signal is delayed for a sixth time period, and the pulse width modulation signal provided to the eighth switching tube is delayed for a seventh time period than the pulse width modulation signal provided to the first switching tube; during the negative half power frequency cycle, The pulse width modulation signal provided to the first switching tube is delayed by a sixth period of time than the pulse width modulation signal provided to the seventh switching tube. The pulse width modulation signal provided to the seventh switching tube is delayed by a sixth period. The pulse width modulation signal provided to the second switching tube is delayed by a seventh time period; the pulse width modulation signal provided to the second switching tube is delayed than the pulse width modulation signal provided to the fifth switching tube. In the sixth time period, the pulse width modulation signal provided to the fifth switching tube is delayed by the seventh time period than the pulse width modulation signal provided to the first switching tube. A control method for a bidirectional DC-AC converter as described in item 1 of the patent application. The half-bridge inverter includes a first switch tube and a second switch tube connected to its DC end in sequence. The half-bridge AC-AC converter includes: a fifth switch tube and a sixth switch tube connected in reverse series, and a seventh switch tube and an eighth switch tube connected in reverse series; wherein the fifth switch tube and the seventh switch tube When the tube is controlled to be turned on, the sixth switching tube, the eighth switching tube and the two capacitors form a first half-bridge inverter, and when the sixth switching tube and the eighth switching tube are controlled to be turned on, The fifth switch tube, the seventh switch tube and the two capacitors constitute a second half-bridge inverter, wherein the control method includes: controlling the first switch tube and The second switch tube is alternately turned on, controlling the fifth and seventh switch tubes to be turned on, and controlling the sixth and eighth switch tubes to be turned on alternately; during the negative half power frequency cycle, the first switch tube and the second switch tube are controlled to be turned on. The switch tubes are turned on alternately, controlling the The fifth and seventh switching tubes are alternately turned on, and the sixth and eighth switching tubes are controlled to be turned on. The control method as described in item 15 of the patent application, wherein in the positive half power frequency cycle, the pulse width modulation signal provided to the sixth switching tube is greater than the pulse width modulation signal provided to the first switching tube. The variable signal is delayed for an eighth time period, and the pulse width modulation signal provided to the eighth switching tube is delayed by the eighth time period than the pulse width modulation signal provided to the second switching tube; in the negative half power frequency cycle , the pulse width modulation signal provided to the seventh switching tube is delayed by an eighth period of time than the pulse width modulation signal provided to the first switching tube, and the pulse width modulation signal provided to the fifth switching tube is The pulse width modulation signal provided to the second switching tube is delayed by an eighth period of time. The control method as described in item 15 of the patent application, wherein in the positive half power frequency cycle, the pulse width modulation signal provided to the first switching tube is greater than the pulse width modulation signal provided to the sixth switching tube. The variable signal is delayed for a ninth time period, and the pulse width modulation signal provided to the sixth switching tube is delayed by a tenth time period than the pulse width modulation signal provided to the second switching tube; to the second switching tube The pulse width modulation signal provided is delayed by a ninth time period than the pulse width modulation signal provided to the eighth switch tube, and the pulse width modulation signal provided to the eighth switch tube is delayed by a ninth period of time than the pulse width modulation signal provided to the first switch tube. The provided pulse width modulation signal is delayed for a tenth time period; in the negative half power frequency cycle, the pulse width modulation signal provided to the first switching tube is greater than the pulse width modulation signal provided to the seventh switching tube. Delay the ninth time period, and the pulse width modulation signal provided to the seventh switch tube is delayed by the tenth time period than the pulse width modulation signal provided to the second switch tube; The pulse width modulation signal is delayed by a ninth period of time than the pulse width modulation signal provided to the fifth switching tube, and the pulse width modulation signal provided to the fifth switching tube is delayed by a ninth time period than the pulse width modulation signal provided to the first switching tube. The pulse width modulation signal is delayed for a tenth time period.