TWI741050B - Sending device, sending method, receiving device, and receiving method - Google Patents

Abstract

The present invention adopts the following configuration: a transmitting device including: a precoding unit that performs precoding processing on a first baseband signal and a second baseband signal to generate a first precoding signal and a second precoding signal; the order is reversed The turning part inverts the order of the symbols constituting the second precoding signal to generate a second inversion signal; the phase changing part generates a second phase change signal, and the second phase change signal is the second inversion signal The signal with the phase change implemented; and the transmitting part, which respectively transmits the first precoding signal and the second phase change signal from different antennas.

Description

Sending device, sending method, receiving device, and receiving method

FIELD OF THE INVENTION The present disclosure relates to a transmitting device, a transmitting method, a receiving device, and a receiving method for communication using multi-antenna.

BACKGROUND OF THE INVENTION One of the standards related to wireless LAN, the IEEE802.11ad standard, is a standard for wireless communication using millimeter waves in the 60 GHz band (Non-Patent Document 1). In the IEEE802.11ad specification, transmission by a single carrier (Single Carrier) is specified.

In addition, one type of communication technology using multiple antennas is MIMO (Multiple-Input Multiple-Output) (Non-Patent Document 2). By using MIMO, the space diversity effect can be improved, and the reception quality can be improved. Previous technical literature Non-patent literature

Non-Patent Document 1: IEEE802.11ad ^TM -2012, December 28, 2012 Non-Patent Document 2: "MIMO for DVB-NGH, the next generation mobile TV broadcasting," IEEE Commun. Mag., vol.57, no. 7, pp.130-137, July 2013 Non-Patent Document 3: IEEE802.11-16/0631r0, May 15, 2016 Non-Patent Document 4: IEEE802.11-16/0632r0, May 15, 2016

Summary of the Invention Problems to be Solved by the Invention However, in MIMO communication using a single carrier, there may be cases where the frequency diversity effect cannot be sufficiently obtained.

The non-limiting embodiments of the present disclosure help to provide a transmitting device, a transmitting method, a receiving device, and a receiving method that can improve the frequency diversity effect in MIMO communication using a single carrier. Means to solve the problem

The transmitting device of one aspect of the present disclosure includes: a precoding unit that performs precoding processing on the first baseband signal and the second baseband signal to generate the first precoding signal and the second precoding signal; and the sequence reversal unit , The sequence of the symbol sequence constituting the second precoding signal is reversed to generate a second inversion signal; the phase changing section generates a second phase change signal, and the second phase change signal is the second inversion signal The signal undergoes a phase change signal; and the transmitting unit transmits the first precoding signal and the second phase change signal from different antennas.

A receiving device in one aspect of the present disclosure includes: a receiving unit that uses different antennas to receive the first received signal and the second received signal, respectively; and a demodulation unit that generates from the first received signal and the second received signal The first baseband signal and the second baseband signal. The first received signal and the second received signal include a first precoding signal and a second phase change signal. The first precoding signal is a transmitting device. The first baseband signal and the second baseband signal are pre-encoded to generate a signal, and the second phase change signal is the following signal: The frequency signal is pre-encoded to generate a second pre-encoded signal, and the sequence of the symbol sequence constituting the second pre-encoded signal is reversed to generate a second inverted signal, and the generated second inverted signal The signal generated by the implementation of the phase change.

Furthermore, these comprehensive or specific aspects can be implemented using systems, methods, integrated circuits, computer programs, or recording media, or any of systems, devices, methods, integrated circuits, computer programs, and recording media. The combination to achieve. Invention effect

According to an aspect of the present disclosure, the frequency diversity effect in MIMO communication using a single carrier can be improved.

The further advantages and effects of the one aspect of the present disclosure can be clearly understood from the description and the drawings. Although the advantages and/or effects are separately provided by some embodiments and the features described in the specification and drawings, in order to obtain one or more of the same features, it is not necessary to provide all of them.

Modes for Carrying Out the Invention Hereinafter, embodiments of the present disclosure will be described in detail with reference to the drawings.

(Embodiment 1) Fig. 1 is a diagram showing an example of the configuration of a MIMO communication system. The transmitting device includes a plurality of transmitting antennas. The receiving device includes a plurality of receiving antennas.

The wireless transmission path between each transmitting antenna and each receiving antenna is called a channel. In Figure 1, between the first transmitting antenna and the first receiving antenna, between the first transmitting antenna and the second receiving antenna, between the second transmitting antenna and the first receiving antenna, and between the second transmitting antenna and the second transmitting antenna Among the receiving antennas, there are channel H ₁₁ (k), channel H ₁₂ (k), channel H ₂₁ (k), and channel H ₂₂ (k). In each channel, for example, direct waves, reflected waves, diffracted waves, and/or scattered waves can be synthesized. The values of channels H ₁₁ (k), H ₁₂ (k), H ₂₁ (k), and H ₂₂ (k) are the frequency response of each channel. The frequency response is the complex number in the frequency index k.

The transmitting device simultaneously transmits different transmission data from each transmitting antenna, that is, at the same sampling timing in the D/A converter. The receiving device includes a plurality of receiving antennas. In the receiving device, each receiving antenna simultaneously receives the received data, that is, receives the same sampling timing in the A/D converter. However, due to the different delays of each channel, the transmission data that has been sent by the sending device at the same time may not be received by the receiving device at the same time.

Fig. 2 is a diagram showing an example of the amplitude component of the frequency response. In Figure 2, what is shown is an example of a state where the frequency response of each channel is different, and the correlation between the channels is low.

When the receiving device receives the transmission data x ₁ (b, n) from the first transmission antenna, for example, the following processing is performed. In other words, the receiving device multiplies the received data from the first receiving antenna and the received data from the second receiving antenna by a complex number weighting coefficient, and adds the data to enhance the signal from the channel H ₁₁ (k) And channel H ₁₂ (k) reception signal, and suppress the reception signal from channel H ₂₁ (k) and channel H ₂₂ (k). The weighting coefficient is calculated using, for example, MMSE (Minimum Mean Square Error) described later.

FIG. 3 is a diagram showing an example of the configuration of the transmission device 100. In FIG. 3, the transmitting device 100 includes a MAC unit 101, a stream generation unit 102, coding units 103a, 103b, data modulation units 104a, 104b, a precoding unit 105, and GI (Guard Interval) addition Units 106a, 106b, symbol order inversion unit 107, data symbol buffers 108a, 108b, phase rotation unit 109, transmission F/E circuits (filtered D/A conversion RF circuits) 110a, 110b, and transmission antennas 111a, 111b.

The transmitting device 100 performs π/2-BPSK modulation in the data modulation sections 104a and 104b, and transmits different data from the transmitting antennas 111a and 111b, respectively.

The MAC unit 101 generates transmission data, and outputs the generated transmission data to the stream generation unit 102.

The stream generation unit 102 divides the transmission data into two streams, the first stream data and the second stream data. For example, the stream generating unit 102 allocates the odd-numbered bits of the transmission data to the first stream data, and allocates the even-numbered bits of the transmission data to the second stream data. In addition, the stream generating unit 102 outputs the first stream data to the encoding unit 103a, and outputs the second stream data to the encoding unit 103b. The stream generation unit 102 may calculate a CRC (Cyclic Redundancy Check) of the transmission data, and add the CRC to the end of the transmission data to generate the stream data.

The processing of the first stream data output from the stream generating unit 102 is referred to as first transmission stream processing. The first transmission stream processing is performed by the encoding unit 103a and the data modulation unit 104a.

The processing of the second stream data output from the stream generating unit 102 is referred to as second transmission stream processing. The second transmission stream processing is performed by the encoding unit 103b and the data modulation unit 104b.

The encoding units 103a and 103b perform error correction encoding processing on each stream data. The encoding units 103a and 103b can also use, for example, LDPC (Low Density Parity Check) codes for error correction encoding methods.

The data modulation units 104a and 104b perform modulation processing on each stream data that has been subjected to error correction coding by the coding units 103a and 103b. The data modulation units 104a and 104b use, for example, π/2-BPSK in the data modulation method.

FIG. 4A is an example of a constellation of π/2-BPSK (Binary Phase Shift Keying) with an odd symbol index m. Fig. 4B is an example showing a π/2-BPSK horoscope with an even symbol index m. The data output by the data modulation unit 104a (also referred to as a "modulation signal") is represented as a modulation symbol s ₁ (m). In addition, the data output from the data modulation unit 104b is expressed as a modulation symbol s ₂ (m). Here, m represents a symbolic exponent and is a positive integer.

When the data modulation unit 104a performs π/2-BPSK modulation, the modulation symbols s ₁ (m) and s ₂ (m) will have the following values. When m is an odd number, s ₁ (m) and s ₂ (m) are arranged on the I axis and have any value of +1 or -1. When m is an even number, s ₁ (m) and s ₂ (m) are arranged on the Q axis, and have any value of +j or -j. Here, j is an imaginary unit.

(算式1)As shown in Equation 1, the precoding unit 105 _{multiplies the modulation symbols s 1} (m) and s ₂ (m) of the data modulation units 104a and 104b by a matrix of 2 rows and 2 columns to calculate the precoding symbol x ₁ (m), x ₂ (m).

(Equation 1)

(算式2)In Equation 1, the matrix of 2 rows and 2 columns that is multiplied by s ₁ (m) and s ₂ (m) is called a precoding matrix (hereinafter denoted as "G"). That is, the precoding matrix G is expressed by Equation 2.

(Equation 2)

(算式2-1)However, the precoding matrix of Equation 2 is only an example, and other matrices may be used for the precoding matrix G. For example, other unitary matrices can also be used for the precoding matrix G. Here, the so-called monopositive matrix is a matrix that satisfies Equation 2-1. In Equation 2-1, G ^H represents the complex conjugate transpose of the matrix G, and I represents the identity matrix.

(Equation 2-1)

Since the precoding matrix G of Equation 2 satisfies Equation 2-1, it is an example of a unitary matrix.

(算式2-2)When the precoding matrix G of Equation 2 is used, x ₁ (m) and x ₂ (m) satisfy the relationship of Equation 2-2. Furthermore, the symbol "*" means complex conjugate.

(Equation 2-2)

(算式2-3)Next, another example of the precoding matrix G is shown in Equation 2-3.

(Equation 2-3)

(算式2-4)When the precoding matrix G of Equation 2-3 is used, x ₁ (m) and x ₂ (m) satisfy the relationship of Equation 2-4.

(Equation 2-4)

(算式2-5)Next, another example of the precoding matrix G is shown in Equation 2-5. In formula 2-5, a is a real number and b is a constant of a complex number. In addition, ρ is a constant representing the amount of phase shift.

(Equation 2-5)

(算式2-6)When the precoding matrix G of Equation 2-5 is used, x ₁ (m) and x ₂ (m) satisfy the relationship of Equation 2-6.

(Equation 2-6)

In Equation 2-5, when both a and b are set to 1, and ρ is set to -π/4, Equation 2-5 becomes equal to Equation 2.

FIG. 4C is a diagram showing an example of a star map of _{the output data x 1} (m) and x _{2 (m) of the precoding unit 105.} The astrological diagrams of Fig. 4C and QPSK modulation are the same. In other words, the precoding unit 105 uses Equation 1 to _{convert the two modulation symbols s 1} (m) and s ₂ (m) modulated by π/2-BPSK into two QPSK symbols. Precoding symbols x ₁ (m), x ₂ (m).

_{The processing of the precoding symbol x 1} (m) output from the precoding unit 105 is called a first transmission RF chain processing. The first transmission RF chain processing is performed by the GI adding unit 106a, the data symbol buffer 108a, the transmission F/E (Front End) circuit 110a, and the transmission antenna 111a.

_{The processing of the precoding symbol x 2} (m) output from the precoding unit 105 is called a second transmission RF chain processing. The second transmission RF chain processing is performed by the complex conjugate GI adding unit 106b, the symbol order inversion unit 107, the data symbol buffer 108b, the phase rotation unit 109, the transmission F/E circuit 110b, and the transmission antenna 111b.

FIG. 5A is a diagram showing an example of a GI adding method in the GI adding unit 106a and the complex conjugate GI adding unit 106b.

The GI adding unit 106a divides the precoding symbol x ₁ (m) into data blocks of 448 symbols each. For example, _{the first 448 symbols of x 1} (m) are divided into the first data block (x ₁ (1,n)), and the following 448 symbols are divided into the second data block (x ₁ ( 2,n)),..., and divide the b-th 448 symbols into b-th data blocks (x ₁ (b,n)). Here, in the case of this embodiment, n is an integer of 1 or more and 448 or less, and b is a positive integer. In other words, x ₁ (b, n) represents the n-th precoding symbol in the b-th data block. In addition, the number of these symbols is only an example, and the number of symbols other than these may be used in this embodiment.

The GI adding unit 106a adds 64-symbol GI to the top of each data block. GI is a symbol sequence modulated by π/2-BPSK on a conventional sequence. In addition, the GI adding unit 106a adds 64 symbols of GI to the end of the last data block. In this way, the transmission symbol u1 as shown in FIG. 5A can be generated.

Similarly, the complex conjugate GI appending unit 106b also divides the precoding symbol x ₂ (m) into data blocks of 448 symbols each, and appends 64 symbols of GI to the front of each data block, and at the end 64-symbol GI is appended after the data block of. However, the GI added by the complex conjugate GI adding unit 106b is the complex conjugate of the GI added by the GI adding unit 106a. In this way, the transmission symbol u2 as shown in FIG. 5A can be generated.

(算式3)Here, the p-th symbol of the GI added by the GI adding unit 106a is represented as GI ₁ (p). In addition, the p-th symbol of the GI added by the complex conjugate GI adding unit 106b is represented as GI ₂ (p). In the case of this embodiment, p is an integer of 1 or more and 64 or less. In this case, GI ₁ (p) and GI ₂ (p) have the relationship shown in Equation 3. Furthermore, the symbol "*" means complex conjugate.

(Equation 3)

_{Fig. 5B is an example showing the DFT signal X 1} (b,k) after DFT (Discrete Fourier Transform) is performed on the symbol block (refer to the transmission symbol u1 in Fig. 5A), where the symbol block is in the pre- A symbol block of GI(p) is added to the coded symbol x _{1 (b, n). FIG. 5C is an example showing the DFT signal X 2} (b,k) after DFT is performed on the symbol block (refer to the transmission symbol u2 in FIG. 5A), where the symbol block is in the precoding symbol x ₂ (b,n) A symbol block ^{with GI *} (p) appended to it. Next, the DFT signal X ₁ (b, k) is used to illustrate the frequency characteristics of the signal output from the GI adding unit 106a. In addition, the DFT signal X ₂ (b, k) is used to illustrate the frequency characteristics of the signal output from the complex conjugate GI adding section 106b.

(算式3-1)In the case of using the precoding matrix G of Equation 2, since x ₂ (b,n) and GI ^* (p) are _{the complex conjugates of x 1} (b,n) and GI(p), the DFT signal X ₂ (b,k) will be a signal that _{reverses the frequency of the complex conjugate of the DFT signal X 1} (b,k) and applies phase rotation to the frequency region. In other words, X ₂ (b,k) can be expressed by Equation 3-1.

(Equation 3-1)

(算式3-2)In addition, the amount of phase rotation (exp(j×2πk/N)) in Equation 3-1 is expressed as W as described below.

(Equation 3-2)

Through the precoding process, two modulation symbols s ₁ (m) and s ₂ (m) can be mixed and transmitted using two different transmitting antennas. In this way, the space diversity effect can be obtained. In addition, through the precoding process, two modulation symbols s ₁ (m) and s ₂ (m) can be mixed and transmitted using two different frequency indices k and -k. In this way, the frequency diversity effect can be obtained.

Furthermore, in Fig. 5B and Fig. 5C, when the absolute value of two different frequency indices k and -k |k| is small, since the two frequencies are close, the frequency diversity effect will be achieved. Reduce. In the following, a technique for suppressing such a case where two frequencies are close to each other and reducing the frequency diversity effect will be described.

FIG. 6A shows an example of the sign order reversal processing in the sign order reversal unit 107.

(算式4)6A, the symbol sequence is reversed for each symbol portion 107 blocks, the sequence of precoded symbols x ₂ (b, n) is inverted, and in addition to the pre-coded symbol x ₂ (b, n The order of GI(p) of) is reversed. In order to make the description easier to understand, the precoding symbol x ₂ ^{(time reversal)} (b, n) whose order has been reversed is expressed as Equation 4. In other words, use "-n" to indicate a sequence of symbols whose order has been reversed.

(Equation 4)

(算式5) _{Also, GI 2} ^{(time reversal)} (p) whose order has been reversed is expressed as Equation 5. In other words, use "-p" to indicate a sequence of symbols whose order has been reversed.

(Equation 5)

_{6C is an example of the DFT signal X 1} (b,k) after DFT is performed on the symbol block (refer to the transmission symbol u1 in FIG. 5A), where the symbol block is in the precoding symbol x ₁ (b,n) A symbol block with GI(p) attached. Fig. 6C is the same as Fig. 5B. Moreover, FIG. 6D is an example of the inverted DFT signal X _2r (b,k) _{after performing DFT on the inverted symbol x 2 (-m).} Here, the reversal sign x ₂ (-m) includes the pre-encoded sign signal x ₂ (b, -n) after the sign order is reversed, and the GI ^* (p ). Next, the inverted DFT signal X _2r (b, k) is used to illustrate the frequency characteristics of the signal output from the sign order inverting unit 107.

(算式5-2)In the case of using the precoding matrix G of Equation 2, since x ₂ (b,-n) and GI ^* (-p) are obtained _{by reversing the order of x 1} (b,n) and GI(p) The complex conjugate of the symbol block, so X _2r (b,k) is represented by equation 5-2.

(Equation 5-2)

The inverted DFT signal X _2r (b,k) is a signal for applying phase rotation to the complex conjugate _{of the DFT signal X 1 (b,k).} In addition, in Equation 5-2, N included in W is the DFT size (for example, the length of the symbol block is "512").

In the examples shown in Figs. 6C and 6D, different from the cases of Figs. 5B and 5C, the DFT signal X ₁ (b, k) processed by the first transmission RF chain and the reverse of the processing of the second transmission RF chain are combined. The converted DFT signal X _2r (b,k)=X ₁ ^* (b,k)×W, which is sent with the same frequency index k. Thus, the space diversity effect can be obtained.

FIG. 6B shows another example of the sign order reversal processing in the sign order reversal unit 107.

As shown in FIG. 6B, the symbol sequence reversing unit 107 reverses the sequence of the symbol sequence (arrangement of the symbol sequence) of the entire symbol block for each symbol block. At this time, the symbol sequence reversing unit 107 removes the last data area in order to equalize the position of the GI between the symbol block before the symbol sequence reversal and the symbol block after the symbol sequence reversal. The GI attached to the stage after the block is attached, and the GI whose sign order has been reversed is attached before the first data block. Furthermore, as already proposed, the symbol block is, for example, a 512-symbol block formed by combining a 64-symbol GI and a 448-symbol data block.

The symbol order reversing unit 107 may also sequentially save 448 symbols of data symbols in the data symbol buffer 108b among the transmission symbols u2 output by the complex conjugate GI adding unit 106b, and in a different order from the time of saving. (Reverse order) to read the data symbols from the data symbol buffer 108b, thereby realizing the reversal of the symbol sequence. In other words, the data symbol buffer 108b is also a buffer equivalent to a LIFO (Last In, First Out) buffer. Furthermore, the data symbol buffer 108b can also be a memory, RAM, or register, etc.

Since the sign order reversal unit 107 performs the process of reversing the sign order of the transmission symbol u2, a delay of the output data with respect to the input data is caused. Therefore, using the data symbol buffer 108a, among the transmission symbols u2 output by the GI adding unit 106a, the data symbol (for example, x ₂ (b, n)) can be given the same delay as the symbol sequence reversing unit 107 The same time delay. Thereby, the transmission symbol u1 output by the GI addition unit 106a and the transmission symbol u2 output by the complex conjugate GI addition unit 106b are transmitted at the same timing. In addition, in the following description, the symbol block in which the transmission symbol u2 has been inverted by the symbol order reversing unit 107 may be expressed as the inverted symbol u2r.

(算式6)

(算式7) _{The phase rotation unit 109 applies a different phase rotation to the data symbol (for example, x 2} (b, n)) for each symbol among the inverted symbols u2r output by the symbol order inverting unit 107. That is, the phase rotation unit 109 performs a different phase change for each symbol. The phase rotation unit 109 uses Equation 6 to _{apply phase rotation to the data symbol (for example, x 2} (b, n)), and uses Equation 7 to apply phase rotation to GI (for example, GI ₂ (p)). In addition, in Equation 6 and Equation 7, θ represents the amount of phase rotation.

(Equation 6)

(Equation 7)

(算式8)Among the transmission symbols output by the precoding unit 105, the transmitting device 100 does not apply phase rotation to x1(b,n), but applies phase rotation to x2(b,n). The transmitted symbol after the phase rotation is represented by Equation 8.

(Equation 8)

(算式9)Furthermore, in FIG. 3, although the phase rotation unit 109 is arranged in the second transmission RF chain processing, it is also possible to arrange the phase rotation unit in both the first transmission RF chain processing and the second transmission RF chain processing. Can. In the case of this configuration, the phase rotation matrix shown in Equation 9 can be used.

(Equation 9)

Furthermore, in Equation 8, the case where n is 1 or more and 448 or less can also be regarded as an expression related to the data symbol (for example, Equation 6), and the case where n is 449 or more and 512 or less can also be considered as having Regarding the formula for GI (for example, formula 7 when the value obtained by subtracting 448 from n in formula 8 is set to p). In this case, the formula 8 is that n is 1 or more and 512 or less, and x ₁ (b,n) and x ₂ (b,-n) will include both the data symbol and the GI.

_{FIG. 6E is a diagram showing the DFT signal T 1} (b, k) after the phase rotation of the symbol t ₁ (b, n) is DFT performed for each symbol block. _{FIG. 6F is a diagram showing the DFT signal T 2} (b, k) after the phase rotation of the symbol t ₂ (b, n) is DFT performed for each symbol block. Next, use T ₁ (b,k) and T ₂ (b,k) to illustrate the frequency characteristics of the phase-rotated signal.

According to Equation 8, X ₁ (b,k) and T ₁ (b,k) are equal. That is, FIG. 6C and FIG. 6E are the same except that the symbol is _{replaced from X 1} to T ₁ .

(算式9-1) _{T 2} (b,k) shown in FIG. 6F _{is a signal that X 2r} (b,k) is phase-rotated in the time region. When the phase rotation is applied to the time zone using Equation 8, the frequency index will be shifted in the frequency zone by the amount of the frequency bin d calculated in Equation 9-1. N is the DFT size (for example, the length of the symbol block is "512").

(Equation 9-1)

(算式9-2)Therefore, X ₁ (b,k) is calculated by Equation 9-2, and in T ₁ (b,k) and T ₂ (b,k+d), two transmitting antennas and two frequency indices k, k+d to send. In other words, the space diversity effect and frequency diversity effect can be obtained.

(Equation 9-2)

The transmitting device 100 sets the phase rotation amount θ to a value close to π radians (180 degrees) or -π radians (-180 degrees), thereby improving the frequency diversity effect and increasing data throughput.

Furthermore, the transmitting device 100 may set the phase rotation amount θ to a value different from π radians (180 degrees). Thereby, the signal separation between the transmission signal of the transmission antenna 111a and the transmission signal of the transmission antenna 111b is facilitated. In addition, the data circulation will increase.

The method of applying a phase rotation different from π radians to transmission symbols in OFDM has been disclosed in Non-Patent Document 2 as a PH (Phase Hopping) technology. However, the transmission device 100 of the present disclosure is different from the case of Non-Patent Document 2 in that single carrier transmission is used and the symbol order is reversed in the second transmission stream process. In this way, the signal separation between the two transmission signals becomes easier. In addition, a relatively high frequency diversity effect can be obtained.

The transmitting device 100 may also set the phase rotation amount θ to, for example, -7π/8 radians (d is -224), -15π/16 radians (d is 240), and the like.

The transmitting F/E circuits 110a and 110b include digital and analog filters, D/A converters, and RF (wireless) circuits. The transmitting F/E circuit 110a converts the transmitted data v1 (including the GI(p) and t1(b,n) signals shown in FIG. 8) output from the data symbol buffer 108a into a wireless signal, and outputs it to the transmitting antenna 111a. The transmission F/E circuit 110b converts the transmission data v2 (including ^{the signals of GI *} (-p) and t2 (b, -n) shown in FIG. 8) output from the phase rotation unit 109 into a wireless signal, and outputs it To the transmitting antenna 111b.

The transmitting antenna 111a transmits the wireless signal output from the transmitting F/E circuit 110a. The transmitting antenna 111b transmits the wireless signal output from the transmitting F/E circuit 110b. In other words, the transmitting antennas 111a and 111b respectively transmit different wireless signals.

In this way, the transmitting device 100 performs precoding on the two transmission stream data, and then performs sign order inversion and phase rotation on one of the transmission stream data. In this way, the space diversity effect and the frequency diversity effect can be improved. In addition, the error rate in data communication can be reduced, and the data circulation can be increased.

FIG. 7 is a diagram showing the structure of the receiving device 200. As shown in FIG.

The receiving antennas 201a and 201b respectively receive wireless signals. The processing of the received signal in the receiving antenna 201a is referred to as the first receiving RF chain processing. The first receiving RF chain processing is performed by the receiving F/E circuit 202a, the time zone synchronization unit 203a, and the DFT unit 205a. The processing of the received signal by the receiving antenna 201b is referred to as the second receiving RF chain processing. The second receiving RF chain processing is performed by the receiving F/E circuit 202b, the time zone synchronization unit 203b, and the DFT unit 205b.

The receiving F/E circuit 202a, 202b includes, for example, an RF circuit, an A/D converter, a digital filter, an analog filter, and a down-sampling processing unit, and converts wireless signals into digital baseband signals.

The time zone synchronization units 203a and 203b will synchronize the timing of receiving packets. Furthermore, the time zone synchronization unit 203a and the time zone synchronization unit 203b may exchange timing information with each other to obtain timing synchronization between the first receiving RF chain processing and the second receiving RF chain processing.

The channel estimation unit 204 uses the received signal processed by the first receiving RF chain and the received signal processed by the second receiving RF chain to calculate the frequency response of the wireless channel between the transmitting device and the receiving device. _{That is, H 11} (k), H ₁₂ (k), H ₂₁ (k), and H ₂₂ (k) in Fig. 1 are calculated for each frequency index k.

The DFT units 205a and 205b divide the received data into DFT blocks and perform DFT. The DFT block is, for example, 512 symbols. FIG. 8 is a diagram showing a method of dividing the received data into DFT blocks in the DFT sections 205a and 205b.

Set the received data processed by the first receiving RF chain (input data to the DFT section 205a) as y ₁ (n), and set the received data processed by the second receiving RF chain (input data to the DFT section 205b) as y ₂ (n). _{Next, the processing of y 1} (n) will be explained using FIG. 8. In addition, _{the processing for y 2} (n) is the same.

As mentioned above, the transmitting device 100 uses two transmitting antennas 111a and 111b to transmit two wireless signals (transmission data v1 and transmission data v2 shown in FIG. 8). In addition, the two wireless signals have the following situation: a direct wave and a plurality of delayed waves are respectively generated in the channel, and reach the receiving antennas 201a and 201b.

Furthermore, in addition to the direct wave and the delayed wave, the received signal may also include, for example, diffracted waves and scattered waves, respectively.

DFT is the DFT portion 205a of the first block contains the time determined as the data block transmission of information v1 t _{1 (1,} n), and the transmission of information data blocks v2 t ₂ (1, n) of the direct wave and delay Wave. Denote the DFT calculation result of the first DFT block as Y ₁ (1,k). As already mentioned, k represents a frequency index, and is, for example, an integer of 1 or more and 512 or less.

Similarly, the DFT calculation results of the b-th DFT block in the DFT sections 205a and 205b are respectively expressed as Y ₁ (b, k) and Y ₂ (b, k) (b is an integer greater than or equal to 1).

The receiving device 200 uses the MMSE weight calculation unit 206, the MMSE filter unit 207, the inverse phase rotation unit 208, the IDFT (inverse DFT) unit 209a, the IDFT and symbol order inversion unit 209b, and the inverse precoding unit 210 to calculate the Estimated values of the transmitted modulation symbols s ₁ (n) and s _{2 (n).} Next, the method of calculating _{the estimated values of the transmitted modulation symbols s 1} (n) and s ₂ (n) will be described.

(算式10) _{The output signals Y 1} (b,k) and Y ₂ (b,k) of the DFT sections 205a and 205b of the receiving device 200 are values using channels and are expressed as Equation 10.

(Equation 10)

Here, T ₁ (b, k) is a signal obtained by performing DFT on the symbol block of the transmitting device 100 (t _{1 (b, n) in Equation 8).} T ₂ (b, k) is a signal obtained by performing DFT on the symbol block of the transmitting device 100 (t _{2 (b, n) in Equation 8).} Z ₁ (b,k) is a signal obtained by DFT of the noise in the first RF chain. Z ₂ (b,k) is a signal obtained by DFT of the noise in the second RF chain.

(算式11)When Formula 10 is expressed as a matrix, it becomes Formula 11.

(Equation 11)

(算式12)In Equation 11, the channel matrix H _2x2 (k) is formulated as in Equation 12.

(Equation 12)

(算式12-1)The MMSE weight calculation unit 206 calculates the weight matrix W _2x2 (k) according to Equation 12-1.

(Equation 12-1)

In Equation 12-1, H ^H represents the complex conjugate transpose of the matrix H. Furthermore, σ ² is the dispersion of the noise Z ₁ (b,k) and Z ₂ (b,k). In addition, I _{2 ×2} is an identity matrix with 2 rows and 2 columns.

(算式12-2)The MMSE filter unit 207 calculates ^{the estimated values T ^} ₁ (b, k) and T ^{^} ₂ (b, k) of _{T 1} (b,k) and T ₂ (b,k) using Equation 12-2. In addition, the processing on the estimated value T ^{^} ₁ (b,k) is referred to as the first received stream processing, and the processing on T ^{^} ₂ (b,k) is referred to as the second received stream processing.

(Equation 12-2)

The calculation of Equation 12-2 is called the MMSE method. MMSE MMSE filter unit according to embodiment 207, comprising from t to the transmission of information v1 ₁ (b, n), it is included in the transmission data t ₂ (b, n) v2 of, and directly affect the respective delayed waves are mixed In the resulting received data y1 and y2 (refer to FIG. 8), _{the estimated values of the data symbols t 1} (b,n) and t ₂ (b,n) after the phase rotation are obtained. However, the MMSE filter unit 207 uses channel estimation values (estimated values of channel frequency response) H ₁₁ (k), H ₁₂ (k), H ₂₁ (k), and H ₂₂ (k), and It is easy to calculate, and as shown in the formula 12-2, the frequency region signal is calculated.

(算式12-3)The reverse phase rotation unit 208 performs a process opposite to that of the phase rotation unit 109 in FIG. 3. The processing of the phase rotation unit 109 is equivalent to the processing of shifting the frequency indices k and -k by the frequency division d in the frequency region, as shown in FIG. 6F. Here, d is calculated by Equation 9-1. Then, the inverse phase rotation unit 208 shifts the frequency region signal of the second received stream output from the MMSE filter unit 207 by -d. In other words, the reverse phase rotation unit 208 performs the processing of Equation 12-3 in the frequency region.

(Equation 12-3)

(算式12-4)Furthermore, the receiving device 200 may swap the IDFT unit 209a, the IDFT and the sign order reversal unit 209b, and the reverse phase rotation unit 208, and apply the reverse phase rotation after IDFT the output from the MMSE filter unit. In this case, the reverse phase rotation unit 208 performs the processing of Equation 12-4 in the time zone.

(Equation 12-4)

That is, although the inverse phase rotation unit 208 applies inverse phase rotation to the second received stream data, since the symbol sequence is reversed by the IDFT and the symbol sequence inversion unit 209b, the matrix determined by the sum formula 9 is performed. The same processing as the multiplication of P.

The IDFT unit 209a performs IDFT on the first received stream data output from the reverse phase rotation unit 208. In addition, the IDFT and symbol order reversal unit 209b performs IDFT on the second received stream data output from the reverse phase rotation unit 208, and reverses the sign order for each DFT block.

(算式12-5)The inverse precoding unit 210 multiplies the first received stream data and the second received stream data by the inverse matrix of the precoding matrix G used by the precoding unit 105 in FIG. 3 to calculate s1(b,n), s2(b ,n) Estimated value. The processing of the inverse precoding unit 210 is shown in Equation 12-5.

(Equation 12-5)

The data demodulation units 211a and 211b demodulate the estimated values of s1(b,n) and s2(b,n) output from the inverse precoding unit 210 to calculate the estimated value of the bit data.

The decoding units 212a and 212b perform error correction processing using LDPC codes on the estimated value of the bit data.

The stream integration unit 213 integrates the first received stream data and the second received stream data, and notifies the MAC unit 215 as the received data.

The header data extraction unit 214 extracts header data from the received data to determine, for example, MCS (Modulation and Coding Scheme) and the phase rotation amount θ used by the phase rotation unit 109 in FIG. 3. In addition, the header data extraction unit 214 may also control: the precoding matrix G applied to the inverse precoding unit 210, IDFT, and the sign inversion process in the sign order inversion unit 209b, and the use of the inverse phase rotation unit 208 The amount of phase rotation θ.

In the receiving device 200, since the MMSE filter unit 207 uses the transmission signals T ₁ (b,k) and T ₂ (b,k) obtained by frequency shifting the second transmission stream data for estimation, it is possible to Get a higher frequency diversity effect. In addition, the reception error rate can be reduced, and the data circulation can be increased.

<Effects of the first embodiment> In the first embodiment, the transmitting device 100 adds the complex conjugate of GI to the second precoding symbol, reverses the order of the symbols, and applies phase rotation (phase change). The complex conjugate is the complex conjugate of the GI added to the first precoding symbol.

In this way, in the MIMO channel, a higher frequency diversity effect can be obtained. In addition, the error rate of communication data can be reduced, and the data circulation can be increased.

(Embodiment 2) In Embodiment 1, a description is given for a case where the transmission device 100 performs π/2-BPSK modulation in the data modulation sections 104a and 104b to perform MIMO transmission. In the second embodiment, the description is directed to the following case: the transmitting device 300 (refer to FIG. 9) switches a plurality of data modulation methods (for example, π/2-BPSK modulation and π/2-BPSK modulation) in the data modulation sections 104a and 104b π/2-QPSK modulation), and perform MIMO transmission.

Fig. 9 is a diagram showing the structure of a transmitting device 300 according to the second embodiment. In addition, the same reference numerals are assigned to the same components as those in FIG. 3, and the description is omitted.

The data modulation units 104c and 104d perform data modulation corresponding to the control of the MAC unit 101 on the coded data output by the coding units 103a and 103b.

Next, an example in which the precoding unit 105a switches the precoding process according to π/2-BPSK modulation and π/2-QPSK modulation will be described.

Fig. 10A is a diagram showing an example of a constellation diagram of π/2-QPSK modulation. _{The modulation symbols s 1} (m) and s ₂ (m) output from the data modulation units 104c and 104d have any value of +1, -1, +j, and -j, respectively. Furthermore, the constellation diagram of π/2-BPSK modulation is shown in Figure 4A.

(算式13)The precoding unit 105a changes the precoding matrix according to the data modulation method used in the data modulation units 104c and 104d, and performs the precoding process shown in Equation 13.

(Equation 13)

When π/2-BPSK is used in the data modulation units 104c and 104d, the precoding unit 105a uses, for example, the precoding matrix G shown in Equation 2, Equation 2-3, or Equation 2-5.

(算式14)When π/2-QPSK is used in the data modulation units 104c and 104d, the precoding unit 105a uses the precoding matrix G shown in Equation 14, for example.

(Equation 14)

When the precoding unit 105a precodes the π/2-BSPK symbol using Equation 2, the constellation diagram is the same as π/2-QPSK (see FIG. 4C). In addition, when the precoding unit 105a precodes the π/2-QSPK symbol (see FIG. 10A) using Equation 14, the constellation diagram is the same as that of 16QAM (see FIG. 10B).

The number of symbol candidates for π/2-BPSK is 2, the number of symbol candidates for π/2-QPSK is 4, and the number of symbol candidates for π/2-16QAM is 16. That is, by performing precoding, the number of symbol candidate points in the astrological chart will increase.

The processing of the second transmit RF chain will vary depending on the modulation method and the type of the precoding matrix G. When π/2-BPSK is used in the data modulation sections 104c and 104d, and the precoding matrix G shown in Equation 2, Equation 2-3, or Equation 2-5 is used in the precoding section 105a, the transmitting device Similar to the transmitting device 100 in FIG. 3, 300 uses the complex conjugate GI adding unit 106b and the sign order inverting unit 107 to perform the second transmission RF chain processing.

The complex conjugate GI adding unit 106b is a complex conjugate of adding GI to the _{output x 2 (m) of the precoding unit 105a.} The sign order reversal unit 107 performs sign order reversal processing on _{the output x 2} (n) of the complex conjugate to which GI is added.

When π/2-QPSK is used in the data modulation sections 104c and 104d, and the precoding matrix G shown in Equation 14 is used in the precoding section 105a, the transmission device 300 is different from the transmission device 100 in FIG. 3, and is The second transmission RF chain processing is performed by the GI adding unit 106c.

The GI adding unit 106c _{adds a GI to the output x 2} (m) of the precoding unit 105a, which is the same as the GI added by the GI adding unit 106a in the first RF chain processing.

Furthermore, the GI adding unit 106c may add a GI (GI2) different from the GI (GI1) added by the GI adding unit 106a. It is also possible to use mutually orthogonal sequences (cross correlation is 0) for GI1 and GI2. For example, the Ga64 sequence specified in the 11ad specification (see Non-Patent Document 1) may be used in GI1, and the Gb64 sequence specified in the 11ad specification may also be used in GI2.

The combination of π/2-BPSK modulation and the precoding matrix G of Equation 2, Equation 2-3, or Equation 2-5 is referred to as the first precoding method type. The combination of π/2-QPSK modulation and the precoding matrix G of Equation 14 is referred to as the second precoding method type. Furthermore, the method for determining the type of the first precoding method and the type of the second precoding method will be described later.

In the case of the first precoding method type, the selection unit 112a selects the output of the data symbol buffer 108a, and the selection unit 112b selects the output of the symbol order inversion unit 107.

In the case of the second precoding method type, the selecting unit 112a selects the output of the GI adding unit 106a, and the selecting unit 112b selects the output of the GI adding unit 106c.

In addition, the selection unit 112a may be arranged at a later stage of the GI adding unit 106a. In addition, the selection unit 112b may be arranged in the latter stage of the precoding unit 105a.

Next, the reason why the transmission device 300 changes the second transmission RF chain processing in accordance with the precoding method will be described.

In the first type of precoding method, as shown in Equation 2-2, Equation 2-4, or Equation 2-6, x ₁ (b,n) and x ₂ (b,n) have a complex conjugate relationship, And further to form a constant factor relationship. Therefore, as shown in FIGS. 5B and 5C, in the frequency region, the signal processed by the second transmission RF chain is a signal obtained by inverting the frequency of the signal processed by the first transmission RF chain, and is the same as that of the first transmission RF chain. The signal processed is a complex conjugate relationship.

On the other hand, in the second precoding method type, x ₁ (b, n) and x ₂ (b, n) do not have a complex conjugate relationship. Therefore, as shown in FIGS. 11A and 11B, in the frequency region, the signal processed by the first transmission RF chain and the signal processed by the second transmission RF chain are transmitted at the same frequency. For example, X ₁ (b,k) and X ₂ (b,k) are transmitted at the same frequency, and X ₁ (b,-k) and X ₂ (b,-k) are transmitted at the same frequency .

(算式15)When the complex number b that satisfies Equation 15 exists, it corresponds to the first precoding method type.

(Equation 15)

From the above considerations, in the first precoding method type, the transmitting device 300 adds the complex conjugate GI to the second transmit RF chain process and reverses the sign order. That is, the selection unit 112b selects the output from the sign order inversion unit 107. On the other hand, in the second precoding method type, the same GI as the first RF chain processing is added to the second RF chain processing, and the sign order is not reversed. That is, the selection unit 112b selects the output from the GI encoding unit 106c.

Thereby, regardless of the type of data modulation method and precoding matrix, the transmitting device 300 can realize the phase rotation θ given by the phase rotation unit 109 as shown in FIG. 6E and FIG. 6F (and by using the formula θ The frequency diversity effect of d) obtained by 9-1 conversion.

In π/2-BPSK, by using the precoding matrix of Equation 2, the precoded horoscope will be equivalent to QPSK (see Fig. 4B). This situation corresponds to the first precoding method type. In addition, in π/2-QPSK, by using the precoding matrix of Equation 14, the precoded horoscope will be equivalent to 16QAM (see FIG. 10B). This situation corresponds to the second precoding method type.

Furthermore, the selection units 112a and 112b can also select the input data according to the type of precoding method in π/2-BPSK modulation.

In addition, the transmission device 300 may also transmit using the same transmission parameters as π/2-QPSK and π/2-16QAM in transmission without precoding. The transmission parameter includes, for example, the setting value of the backoff operation point (Backoff) of the RF amplifier of the transmission F/E circuit 110a, 110b. In other words, the transmitting device 300 can also use either Equation 2 or Equation 14 to perform precoding in accordance with the modulation method. This allows transmission without changing the configuration of the transmission F/E circuits 110a and 110b. The reason will be explained below.

In general millimeter wave communication, the setting value of the backward operating point of the RF amplifier in the transmitting F/E circuit can be appropriately set and changed according to the transmitting astrological diagram configuration (Figure 10A, Figure 10B, etc.). For example, in 16QAM as shown in FIG. 10B, since the peak power (PAPR) relative to the average power becomes larger, the backoff operation point (Backoff) of the RF amplifier is set to be larger, and the RF amplifier is not Saturate the signal. In addition, since the configuration of the astrogram of the transmission signal is changed by performing the precoding process, the setting of the transmission F/E circuit will be changed.

On the other hand, in the transmitting device 300 of the present embodiment, the precoding process is performed by using, for example, Equation 2 and Equation 14, so that it is different from the astrogram configuration before the precoding process, but is different from the conventional one. Modulate the same star chart configuration. In other words, regardless of the presence or absence of precoding processing, the transmission signal has a conventional astrogram configuration, so it becomes unnecessary to change the configuration and setting of the transmission F/E circuit, and the control becomes easier.

<Effects of Embodiment 2> In Embodiment 2, when the first precoding symbol and the second precoding symbol are in a complex conjugate relationship, the transmission device 300 adds the complex GI to the second precoding symbol. The complex conjugate of the GI is the complex conjugate of the GI appended to the first precoding symbol, and a phase rotation (phase change) is applied.

In this way, a plurality of data modulation methods can be switched in the MIMO channel, and a higher frequency diversity effect can be obtained. In addition, the error rate of communication data can be reduced, and the data circulation can be increased.

(Modification of Embodiment 2) As explained in Embodiment 2, in the case of π/2-BPSK modulation, the symbol order inverting unit 107 signs the data symbol and the GI symbol in the transmitting device 300. MIMO transmission with reverse order. In the modification of the second embodiment, it will be explained that the GI adding sections 106d and 106e of the transmitting device 400 (refer to FIG. 12) add different sequences (for example, orthogonal sequences) for each stream for MIMO transmission. .

FIG. 12 is a diagram showing the configuration of a transmitting device 400 according to a modification of the second embodiment. In addition, the same components as those in FIG. 9 will be assigned the same reference numerals, and the description will be omitted.

The GI adding units 106d and 106e are arranged at a later stage than the selection units 112a and 112b and the phase rotation unit 109. Unlike the transmission device 300 of FIG. 9, regardless of the modulation method, the transmission device 400 may add GI symbols defined for each stream.

13 and 14 are diagrams showing an example of the transmission symbol format of the output (v3, v4) of the GI adding units 106d and 106e of the transmission device 400. FIG. 13 shows the case where the modulation of the data symbol is changed to π/2-BPSK modulation, and FIG. 14 shows the case where the modulation of the data symbol is other than the π/2-BPSK modulation.

The GI adding unit 106d divides the precoding symbol x ₁ (m) into data blocks of 448 symbols each, and adds 64 symbols of GI (GI ₁ (p)) to the front of each data block. GI is a symbol sequence modulated by π/2-BPSK on a conventional sequence. Furthermore, the GI adding unit 106d adds 64 symbols of GI to the end of the last data block. In this way, the transmission symbol v3 as shown in FIG. 13 and FIG. 14 can be generated. In addition, the number of these symbols is only an example, and the number of symbols other than these may be used in this embodiment.

Similarly, the GI adding unit 106e divides the precoding symbol x ₂ (m) into data blocks of 448 symbols each, and adds 64 symbols of GI to the front of each data block (GI ₂ (p)) , And append a 64-symbol GI after the last data block. In this way, the transmission symbol v4 as shown in FIG. 13 and FIG. 14 can be generated. The GI added by the GI adding unit 106e may have a different sequence from the GI added by the GI adding unit 106d.

When the receiving device 200 receives the transmission signal of the transmitting device 400 having the format shown in FIG. 13 and FIG. 14, as shown in the first embodiment, the MMSE equalization is performed using Equation 12-2 and the receiving process is performed.

The receiving device 200 can also compare the GI symbols (the GI-related part of the output of the MMSE filter unit 207) that has been MMSE equalized with the conventional GI symbols to detect the error of the channel estimation matrix, and perform channel estimation. The correction of the measurement matrix. When GI ₁ (p) and GI ₂ (p) are orthogonal sequences, the correlation between _{GI 1} (p) estimated by MMSE equalization and the conventional GI ₁ (p) can be calculated. In this calculation, the residual error of MMSE equalization can be reduced, and the value of, for example, the phase offset can be calculated with high accuracy. Therefore, the channel estimation matrix can be corrected with high accuracy and the reception performance can be improved.

Next, another method for the MMSE filter unit 207 of the receiving device 200 to receive the transmission signal of the transmitting device 400 having the format of FIG. 13 and FIG. 14 will be described.

(算式16)The receiving device 200 generates GI ₁ (p) and GI ₂ (p) replica signals by using Equation 16. Here, the so-called duplicate signal refers to the estimated value of the signal received by the receiving antenna _{when conventional patterns (such as GI 1} (p) and GI _{2 (p)) are transmitted, and it is} It is calculated by multiplying the conventional pattern by the channel matrix (refer to Equation 12).

(Equation 16)

In Equation 16, X _G1 (k) and X _G2 (k) are signals (the frequency region signals of GI) obtained by performing DFT on the GI time zone signals (symbols) GI ₁ (p) and GI _{2 (p).} In addition, Y _G1 (k) and Y _G2 (k) are frequency region signals when the receiving device 200 receives GI ₁ (p) and GI _{2 (p).} By assigning the mark "^" to Y _G1 (k) and Y _G2 (k) to show that they are estimated values.

(算式17)The receiving device 200 uses Equation 17 to _{subtract Y^G1 (k) from the received signal Y 1} (b,k) to estimate the data signal component Y^ _D1 (k) included in the received signal, and from Y ₂ (b,k) Subtract Y^ _G2 (k) to estimate the data signal component Y^ _D2 (k).

(Equation 17)

(算式18)The receiving device 200 sets the estimated data signal components Y^ _D1 (k) and Y^ _D2 (k) as inputs and performs MMSE equalization, thereby calculating the estimated value T^ _D1 (k) of the transmitted data symbol And T^ _D2 (k).

(Equation 18)

Although the calculation process performed in equation 18 is the same as that in equation 12-2, the following points are different: the input Y ₁ (b,k) and Y ₂ (b,k) of equation 12-2 contains data In the case of the signal component of GI, the input Y^ _D1 (k) and Y^ _D2 (k) of Equation 18 are the signal components that include the data of the signal component minus GI.

Since the MMSE filter unit 207 receives the transmission signal from the transmitter 400, the GI of each stream is not related to the complex conjugate and time sequence inversion. Therefore, in the demodulation of the GI symbol, it is necessary to obtain and implement The same frequency diversity effect of form 1 is difficult. According to this, there may be a situation where the inter-symbol interference from the GI symbol to the data symbol remains after being equalized by the MMSE, and the reception performance is degraded.

Here, the MMSE filter unit 207 uses Equation 16, Equation 17, and Equation 18 to subtract the symbol replica of GI from the received signal when receiving the transmission signal of the transmitting device 400, and perform MMSE Equalize. In other words, reduce the influence of GI to perform MMSE equalization of data symbols.

_{The receiving device 200 uses the estimated values T ^ D1} (k) and T ^ _D2 (k) of the transmitted data symbol generated by the MMSE filter unit 207 using Equation 18, and performs the same embodiment including inverse phase rotation and inverse precoding 1 and Embodiment 2 are the same receiving processing.

<Effects of the modification of the second embodiment> In the modification of the second embodiment, the transmitting device 400 performs the second precoding when the first precoding symbol and the second precoding symbol are in a complex conjugate relationship. Sign to reverse the order of signs and apply phase rotation (phase change). In addition, different GIs are inserted between the first precoding symbol and the second precoding symbol.

In this way, a plurality of data modulation methods can be switched in the MIMO channel, and a higher frequency diversity effect can be obtained. In addition, the error rate of communication data can be reduced, and the data circulation can be increased.

<Summary of the embodiment> The transmitting device of the first aspect of the present disclosure includes a precoding unit that performs precoding processing on the first baseband signal and the second baseband signal to generate the first precoding signal and the second precoding signal. Encoding signal; sequence reversing section, which reverses the sequence of the symbol sequence constituting the second pre-encoding signal to generate a second reversal signal; phase changing section, generating a second phase change signal, and the second phase change signal It is a signal in which the phase of the second inversion signal is changed; and the transmitting unit transmits the first precoding signal and the second phase change signal from different antennas.

The transmitting device of the second aspect of the present disclosure, in the transmitting device of the first aspect, includes: a first addition section for adding a first conventional signal to the first precoding signal; and a second addition section for adding To the second precoding signal, a second conventional signal having a complex conjugate relationship with the aforementioned first conventional signal is added.

The third aspect of the transmission device of the present disclosure is that in the second aspect of the transmission device, the sequence reversing unit reverses the sequence of the symbol sequence constituting the second conventional signal, and the sequence is reversed in the reversed second The inverted second precoding signal is connected to the conventional signal to generate the aforementioned second inverted signal.

In the transmitting device of the fourth aspect of the present disclosure, in the transmitting device of the second aspect, the sequence reversing unit reverses the sequence of the symbol sequence that has connected the second conventional signal to the second precoding signal , And generate the aforementioned second inversion signal.

The transmitting device of the fifth aspect of the present disclosure is the transmitting device of any one of the second to fourth states, and includes: a third addition part for adding a third conventional signal to the second precoding signal; and The selection unit selects whether to make the second inversion signal and the second signal to which the third conventional signal has been added based on the relationship between whether the first fundamental frequency signal and the second fundamental frequency signal are complex conjugate. Any one of the precoding signals is input to the aforementioned phase changing unit.

The transmission method of the sixth aspect of the present disclosure is to perform precoding processing on the first baseband signal and the second baseband signal to generate the first precoding signal and the second precoding signal, and to form the aforementioned second precoding The sequence of the symbol sequence of the signal is reversed to generate a second inversion signal, and a second phase change signal that performs phase change on the second inversion signal is generated, and the first precoding signal is transmitted from different antennas. It is the same as the aforementioned second phase change signal.

The receiving device of the seventh aspect of the present disclosure includes: a receiving unit, which uses different antennas to receive the first received signal and the second received signal; and a demodulation unit, from the first received signal and the second received signal Generate the first baseband signal and the second baseband signal. In the first received signal and the second received signal, the first precoding signal and the second phase change signal are included. The first precoding signal is the transmission The device performs precoding processing on the first baseband signal and the second baseband signal to generate a signal, and the second phase change signal is the following signal: The baseband signal is pre-encoded to generate a second pre-encoded signal, and the sequence of the symbol sequence constituting the second pre-encoded signal is reversed to generate a second inverted signal, and the generated second is inverted The signal is generated by changing the phase of the signal.

The receiving method of the eighth aspect of the present disclosure uses different antennas to receive the first received signal and the second received signal, and generates the first baseband signal and the second received signal from the first received signal and the second received signal. 2 Base frequency signal. The first received signal and the second received signal include the first precoding signal and the second phase change signal. The first precoding signal is the transmission device's response to the first And the signal generated by precoding the second baseband signal, the second phase change signal is the following signal: the transmitting device performs precoding processing on the first baseband signal and the second baseband signal, and A second precoding signal is generated, and the sequence of the symbol sequence constituting the generated second precoding signal is reversed to generate a second inverted signal, and the generated second inverted signal is phase-changed to generate a signal .

Although various embodiments have been described above with reference to the drawings, the present disclosure is of course not limited to the examples. Obviously, anyone with ordinary knowledge in the technical field to which the present invention belongs should be able to think of various changes or amendments within the scope of the patent application, and it should be understood that these also belong to this disclosure. Technical scope. In addition, it is also possible to arbitrarily combine the constituent elements in the above-mentioned embodiments within a range that does not deviate from the gist of the present disclosure.

In each of the above embodiments, although the present disclosure is an example constructed by using hardware for explanation, the present disclosure can also be implemented by software in cooperation with hardware.

In addition, each functional block used in the description of each of the above-mentioned embodiments is typically realized as an LSI having an integrated circuit with input terminals and output terminals. These can be individually integrated into one chip, or a part or all of them can be integrated into one chip. Here, although it is made as LSI, depending on the degree of integration, there may be cases called IC, System LSI (System LSI), Super LSI (Extra Large LSI), and Ultra LSI (Ultra Large LSI).

In addition, the method of integrated circuitization is not limited to LSI, and it can also be realized by a dedicated circuit or a general-purpose processor. It is also possible to use a programmable FPGA (Field Programmable Gate Array) after the LSI is manufactured, and a reconfigurable processor (Reconfigurable Processor) that can reconfigure the connection or setting of the battery inside the LSI.

In addition, if there is an integrated circuit technology that can replace LSI due to advances in semiconductor technology or other technologies derived from it, of course, this technology can also be used to integrate functional blocks. The present invention may have the possibility of application of biotechnology and the like. Industrial availability

The present disclosure can be widely used in a communication system that transmits modulated signals from a plurality of antennas.

100, 300, 400‧‧‧Transmitting device 101, 215‧‧‧MAC unit 102‧‧‧Stream generation unit 103a, 103b‧‧‧Encoding unit 104a, 104b, 104c, 104d‧‧‧Data modulation unit 105, 105a ‧‧‧Precoding part 106a, 106c, 106d, 106e‧‧‧GI addition part 106b‧‧‧Complex conjugate GI addition part 107‧‧‧Symbol sequence reversal part 108a, 108b‧‧‧Data symbol buffer 109‧ ‧‧Phase rotation part 110a, 110b‧‧‧Transmitting F/E circuit 111a, 111b‧‧‧Transmitting antenna 112a, 112b‧‧‧Selecting part 200‧‧‧Receiving device 201a, 201b‧‧‧Receiving antenna 202a, 202b‧ ‧‧Receiving F/E circuit 203a, 203b‧‧‧Time zone synchronization unit 204‧‧‧Channel estimation unit 205a, 205b‧‧‧DFT unit 206‧‧‧MMSE weight calculation unit 207‧‧‧MMSE filter unit 208‧ ‧‧Inverse phase rotation unit 209a‧‧‧IDFT unit 209b‧‧‧IDFT and symbol order inversion unit 210‧‧‧Inverse precoding unit 211a, 211b‧‧‧Data demodulation unit 212a, 212b‧‧‧Decoding unit 213 ‧‧‧Stream Integration Unit 214‧‧‧Header Data Extraction Unit H11(k), H12(k), H21(k), H22(k)‧‧‧Channel

Fig. 1 is a diagram showing an example of the configuration of the MIMO communication system according to the first embodiment. Fig. 2 is a diagram showing an example of the amplitude component of the frequency response. Fig. 3 is a diagram showing an example of the configuration of the transmitting device of the first embodiment. 4A is a diagram showing an example of a constellation of π/2-BPSK (Binary Phase Shift Keying) with an odd symbol index. Fig. 4B is a diagram showing an example of a π/2-BPSK astrological chart with an even symbol index. Fig. 4C is a diagram showing an example of a star map of the output data of the precoding unit. Fig. 5A is a diagram showing an example of a GI adding method. FIG. 5B is a diagram showing an example of a DFT signal in which DFT is performed on a symbol block with GI added to a pre-encoded symbol. FIG. 5C is a diagram showing an example of a DFT signal when DFT is performed on a symbol block ^{with GI* added to a pre-encoded symbol.} FIG. 6A is a diagram showing an example of the sign order reversal processing in the sign order reversal unit. Fig. 6B is a diagram showing another example of the sign order reversal processing in the sign order reversal unit. FIG. 6C is a diagram showing an example of a DFT signal when DFT is performed on a symbol block with GI added to a pre-encoded symbol. FIG. 6D is a diagram showing an example of an inverted DFT signal when DFT is performed on an inverted symbol. FIG. 6E is a diagram showing a DFT signal in which the symbol is DFT performed after the phase is rotated for each symbol block. FIG. 6F is a diagram showing a DFT signal in which the symbol is DFT performed after the phase is rotated for each symbol block. Fig. 7 is a diagram showing an example of the configuration of the receiving device. FIG. 8 is a diagram showing a method of dividing the received data into DFT blocks in the DFT section. Fig. 9 is a diagram showing the structure of a transmitting device according to the second embodiment. Fig. 10A is a diagram showing an example of a constellation diagram of π/2-QPSK modulation. Fig. 10B is a diagram showing an example of a constellation diagram of 16QAM modulation. FIG. 11A is a diagram showing an example of the DFT signal processed by the first transmit RF chain (chain). FIG. 11B is a diagram showing an example of the DFT signal processed by the second transmit RF chain (chain). Fig. 12 is a diagram showing the configuration of a transmission device according to a modification of the second embodiment. Fig. 13 is a diagram showing an example of a GI adding method according to a modification of the second embodiment. Fig. 14 is a diagram showing another example of the GI adding method according to the modification of the second embodiment.

100‧‧‧Sending device

101‧‧‧MAC Department

102‧‧‧Stream Generation Department

103a, 103b‧‧‧Coding Department

104a, 104b‧‧‧Data Modulation Department

105‧‧‧Precoding section

106a‧‧‧GI Additional Department

106b‧‧‧Additional part of complex conjugate GI

107‧‧‧Symbol sequence reversal part

108a, 108b‧‧‧Data symbol buffer

109‧‧‧Phase Rotation Unit

110a, 110b‧‧‧Sending F/E circuit

111a, 111b‧‧‧transmitting antenna

Claims (7)

A transmitting device comprising: a signal processing circuit that performs precoding processing on a first baseband signal and a second baseband signal to generate a first precoding signal and a second precoding signal, and adding to the first precoding signal The first conventional signal performs sequence reversal processing on the symbol sequence constituting the second precoding signal, and performs the aforementioned sequence on the symbol sequence constituting the second conventional signal having a complex conjugate relationship with the first conventional signal Invert processing, and connect the inverted second precoding signal to the inverted second conventional signal to generate a second inverted signal, thereby from the first fundamental signal and the second The baseband signal generates the first transmission signal and the second transmission signal; and the transmission circuit transmits the aforementioned first transmission signal and the aforementioned second transmission signal from different antennas, respectively. For example, the transmitting device of claim 1, wherein the signal processing circuit performs phase change processing on the second inverted signal to generate a second phase change signal, thereby generating the aforementioned first baseband signal and the aforementioned second baseband signal The first transmission signal and the aforementioned second transmission signal. For example, the transmitting device of claim 1, wherein the signal processing circuit performs the sequence reversal processing on the symbol sequence in which the second conventional signal is connected to the second precoding signal to generate the second reversal signal , Thereby generating the first transmission signal and the second transmission signal from the first base frequency signal and the second base frequency signal. Such as the sending device of any one of claims 1 to 3, wherein The signal processing circuit adds a third conventional signal to the second precoding signal, and selects whether to reverse the second signal based on whether the first fundamental frequency signal and the second fundamental frequency signal are complex conjugate. Any one of the retransmission signal and the second precoding signal to which the third conventional signal has been added undergoes a phase change process, thereby generating the first transmission signal from the first baseband signal and the second baseband signal Send the signal with the 2nd above. A transmission method, including: performing precoding processing on a first baseband signal and a second baseband signal to generate a first precoding signal and a second precoding signal, and adding a first knowledge to the first precoding signal The signal is subjected to sequence reversal processing on the sequence of symbols constituting the second precoding signal, and sequence reversal processing is performed on the sequence of symbols constituting the second conventional signal in a complex conjugate relationship with the first conventional signal, And connect the inverted second precoding signal to the inverted second conventional signal to generate a second inverted signal, thereby generating from the first fundamental frequency signal and the second fundamental frequency signal The first transmission signal and the second transmission signal; and the aforementioned first transmission signal and the aforementioned second transmission signal are respectively transmitted from different antennas. A receiving device, comprising: a receiving unit that uses different antennas to receive a first received signal and a second received signal; and The demodulation unit generates a first baseband signal and a second baseband signal from the first received signal and the second received signal. The first received signal and the second received signal include the first precoding signal and the second received signal. 2 Inverted signal. The first precoding signal is a signal generated by the transmitting device performing precoding processing on the first baseband signal and the second baseband signal, and the first conventional signal is added. The inversion signal is the following signal: the second precoding signal is generated by the transmitting device performing the precoding process on the first baseband signal and the second baseband signal, and the second precoding signal is generated to form the generated second precoding signal. The symbol sequence of the coded signal is subjected to sequence reversal processing, and the sequence of the symbol sequence of the second conventional signal constituting the complex conjugate relationship with the first conventional signal is subjected to the sequence reversal processing, and the sequence is reversed in the reversed first 2 The conventional signal is connected to the previously inverted second precoding signal to generate a signal. A receiving method includes: receiving a first received signal and a second received signal with different antennas; and generating a first baseband signal and a second baseband signal from the first received signal and the second received signal, and The first received signal and the second received signal include a first precoding signal and a second inverted signal. The first precoding signal is used by the transmitting device to compare the first baseband signal and the second baseband signal. The first conventional signal is added to the signal generated by precoding the frequency signal, The second inversion signal is the following signal: the second precoding signal is generated by the transmitting device performing precoding processing on the first baseband signal and the second baseband signal, and forming the generated The sequence of symbols of the second precoding signal is subjected to sequence reversal processing, and the sequence of symbols of the second conventional signal constituting the complex conjugate relationship with the first conventional signal is subjected to the sequence reversal processing described above, and the sequence is reversed. The converted second conventional signal is a signal generated by connecting the aforementioned inverted second precoding signal.

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