TWI511441B - Apparatus for signal receving and method for signal receiving - Google Patents

Description

Signal receiving device and signal receiving method

The present invention relates to a signal receiving apparatus and a signal receiving method.

In general, when the transmitter and the local oscillator of the receiver are not synchronized, signal distortion such as carrier frequency offset (CFO) and carrier phase error may be caused. In the current multi-carrier transmission and reception system, the signals are mostly modulated by Orthogonal Frequency Division Multiplexing (OFDM), and the receiving end needs to be from the in-phase path (In-phase). Path) and the quadrature path receive the complex signal, and the time domain signal is converted into the frequency domain signal by the Fast Fourier Transform (FFT) and then the correlation demodulation process is performed. Therefore, the receiving end can further compensate the above-mentioned carrier frequency offset and phase difference by analyzing the complex signal to estimate the strength and phase of the signal.

Conversely, in a single-carrier transmit and receive system, receive Binary Phase Shift Keying (BPSK) When the signal is used, the receiving end only needs to carry a simple demodulation transformer and related processing circuit. Ideally, the data content carried by the signal can be obtained without using a Fast Fourier Transform (FFT) circuit, and there is no need to add a complex number. Dedicated circuit for demodulation. Such receivers have the advantage of being less complex than other receiving systems. However, the estimation of the carrier frequency offset and the compensation of the phase difference become a thorny problem because of the lack of phase information.

The present invention provides a signal receiving apparatus and a signal receiving method for receiving a single-carrier real number signal of Binary Phase Shift Keying (BPSK), and compensating for problems caused by carrier frequency offset and phase difference. .

A signal receiving apparatus of the present invention is adapted to receive a single carrier real number signal, wherein the single carrier real number signal is modulated by a Binary Phase Shift Keying (BPSK) modulation method. The method includes: a radio frequency circuit, a analog/digital converter, a frequency down unit, and a base frequency processing circuit. The radio frequency circuit receives a single carrier real number signal. The analog/digital converter is coupled to the RF circuit, receives the single carrier real number signal and converts it into a digital intermediate frequency real number signal, wherein the digital intermediate frequency real number signal is located at an intermediate frequency. The frequency down unit is coupled to the analog/digital converter, and converts the digital intermediate frequency real signal into a fundamental frequency signal according to a synchronization signal, wherein the synchronization signal is located at a local intermediate frequency. The baseband processing circuit is coupled to the down-converting unit, receives the baseband signal and processes the baseband signal to obtain a digital data. among them, The frequency reduction unit performs a carrier frequency offset estimation and compensation algorithm, estimates an offset frequency value of the intermediate frequency, and adjusts the frequency value of the local intermediate frequency according to the offset frequency value to compensate the digital intermediate frequency real signal. Conversion with the baseband signal. And, the frequency reduction unit performs a carrier phase difference correction algorithm to dynamically adjust the offset frequency value to correct a phase difference between the intermediate frequency real signal and the synchronization signal.

The signal receiving method of the present invention is applicable to a signal receiving apparatus, wherein the signal receiving apparatus receives a single carrier real number signal, wherein the single carrier real number signal is modulated by a two-bit phase offset modulation method. The signal receiving method includes the following steps. First, the single carrier real number signal is received. Then, the converted single carrier real number signal is a digital intermediate frequency real number signal, wherein the digital intermediate frequency real number signal is located at an intermediate frequency. Then, the digital intermediate frequency real signal is converted into a fundamental frequency signal according to a synchronization signal, wherein the synchronization signal is located at a local intermediate frequency. Furthermore, performing a carrier frequency frequency offset estimation and compensation algorithm, estimating an offset frequency value of the intermediate frequency, and adjusting the frequency value of the local intermediate frequency according to the offset frequency value to compensate the digital intermediate frequency real number The conversion between the signal and the fundamental signal. Next, a carrier phase difference correction algorithm is performed to dynamically adjust the offset frequency value to correct a phase difference between the intermediate frequency real signal and the synchronization signal. And, processing the baseband signal to obtain a digital data.

Based on the above, the present invention provides a signal receiving apparatus and a signal receiving method for compensating for a carrier frequency offset and a carrier phase difference on a baseband signal by a carrier frequency offset estimation and compensation algorithm and a carrier phase difference correction algorithm.

The above described features and advantages of the invention will be apparent from the following description.

10‧‧‧Signal receiving device

110‧‧‧RF circuit

120‧‧‧ Analog/Digital Converter

130‧‧‧down frequency unit

131‧‧‧Numerical Controlled Oscillator

132‧‧‧Digital Down Converter

133‧‧‧Processing unit

1331‧‧‧High frequency rejection unit

610‧‧‧Absolute Value Calculator

620‧‧‧ low pass filter

630‧‧Sampling unit

1332‧‧‧ Differentiator

1333‧‧‧Signal selection unit

710‧‧‧Relay unit

720‧‧‧No mutual exclusion or gate

730‧‧‧Delay unit

740‧‧‧Selector

140‧‧‧ fundamental frequency processing circuit

R[n]‧‧‧ single carrier real number signal

‧‧‧Digital IF real signal

r [ n ]‧‧‧ fundamental frequency signal

r _O [ n ]‧‧‧Overcompensated baseband signal

SS‧‧‧Synchronization signal

OSS‧‧‧Overcompensated sync signal

CS‧‧‧Control signal

RD‧‧‧ residual low frequency signal

Z‧‧‧ indication signal

Y‧‧‧Selection signal

Y _pre ‧‧‧delay selection signal

X‧‧‧ digital signal

A‧‧‧Maximum amplitude of the data signal

HS1‧‧‧ first harmonic

I ₁ ‧‧‧First estimated interval

f _OC ‧‧‧Overcompensation value

f _offset ‧‧‧offset frequency

f _MAX ‧‧‧Maximum offset frequency

f _U ‧‧‧unit frequency value

‧‧‧target frequency value

f _compens ‧‧‧dynamic estimation frequency

f _const ‧‧‧frequency constant

S801~S805‧‧‧Steps

FIG. 1 is a functional block diagram of a signal receiving apparatus according to an embodiment of the invention.

FIG. 2 is a functional block diagram of a frequency down unit according to an embodiment of the invention.

FIG. 3A is a frequency spectrum diagram showing absolute values of a fundamental frequency signal according to an embodiment of the invention.

FIG. 3B is a frequency spectrum diagram illustrating the absolute value of the compensated baseband signal according to an embodiment of the invention.

FIG. 4A is a diagram showing a relationship between a first harmonic and an estimated interval according to an embodiment of the invention.

FIG. 4B is a diagram showing a relationship between a first harmonic and a tone signal in an estimation interval according to an embodiment of the invention.

FIG. 5 is a partial functional block diagram of a processing unit according to an embodiment of the invention.

FIG. 6 is a functional block diagram showing a high frequency reject unit according to an embodiment of the invention.

FIG. 7 is a functional block diagram of a signal selection unit according to an embodiment of the invention.

FIG. 8 is a flow chart showing the steps of a signal receiving method according to an embodiment of the invention.

Hereinafter, a detailed description will be given of how the present invention analyzes a single-carrier real number signal modulated by BPSK, and further proposes a method of carrier frequency offset compensation and carrier phase difference cancellation.

FIG. 1 is a functional block diagram of a signal receiving apparatus according to an embodiment of the invention. In the present embodiment, the signal receiving apparatus 10 is configured to receive a BPSK modulated single carrier real number signal. Referring to FIG. 1, the signal receiving apparatus 10 includes a radio frequency circuit 110, an analog/digital converter 120, a down-converting unit 130, and a baseband processing circuit 140. The radio frequency circuit 110 receives a single carrier real number signal R(t). The analog/digital converter 120 is coupled to the radio frequency circuit 110, receives the single carrier real number signal R(t) and converts the single carrier real number signal R(t) into a digital intermediate frequency real number signal. Digital IF real number signal Located at a known intermediate frequency. The down-converting unit 130 is coupled to the analog/digital converter 120 and converts the digital intermediate frequency real signal according to the synchronization signal. Converted to a baseband signal r [ n ], where the sync signal is at a local intermediate frequency. The baseband processing circuit 140 is coupled to the frequency down unit 130, receives the baseband signal r [ n ], and processes the baseband signal r [ n ] to obtain digital data, that is, the data content carried in the single carrier real number signal R[n]. .

The frequency down unit 130 performs a carrier frequency offset estimation and compensation algorithm to estimate the digital intermediate frequency real number signal. An offset frequency value at the intermediate frequency, and adjusting the frequency value of the local intermediate frequency according to the offset frequency value to compensate the digital intermediate frequency real signal Conversion with the baseband signal r [ n ]. And the frequency down unit 130 performs a carrier phase difference correction algorithm to dynamically adjust the offset frequency value to correct the intermediate frequency real number signal A phase difference from the sync signal. Hereinafter, the implementation of the carrier frequency offset compensation algorithm and the carrier phase difference cancellation algorithm described above will be described in detail by way of embodiments and drawings.

2 is a functional block diagram of a frequency down unit corresponding to an embodiment of the present invention. Referring to FIGS. 1 and 2 , the down-converting unit 130 includes a numerically controlled oscillator 131 , a digital down converter (DDC) 132 , and a processing unit 133 . The numerically controlled oscillator 131 produces a synchronization signal SS, wherein the synchronization signal is located at a local intermediate frequency. The digital down converter 132 is coupled to the analog/digital converter 120 and the numerically controlled oscillator 131, and converts the digital intermediate frequency real signal according to the synchronization signal SS. Is the fundamental frequency signal r [ n ]. The processing unit 133 is coupled to the digital down converter 132 and the numerically controlled oscillator 131, and transmits the control signal CS to the numerically controlled oscillator 131. The control signal CS includes a predefined frequency value of a local intermediate frequency and a dynamic estimation. The sum of the measured frequency values (ie, the corrected intermediate frequency value). The frequency value of the predefined local intermediate frequency is the expected frequency value of the local frequency, and the offset frequency value is obtained by the carrier frequency offset estimation and compensation algorithm, and is corrected by the carrier phase difference correction algorithm. The method dynamically adjusts the offset frequency value.

Wherein, when the numerically controlled oscillator 131 receives the control signal CS, the numerically controlled oscillator adjusts the frequency value of the synchronization signal SS to the frequency value indicated by the control signal CS (ie, a predefined frequency value of a local intermediate frequency and a dynamic Estimate the sum of the frequency values).

Ideally, a predefined IF frequency value and a digital IF real signal The frequency of the intermediate frequency is located at the same value (ie, the offset frequency value is zero). In this way, the digital down converter 132 can successfully lock the digital intermediate frequency real signal by using the synchronization signal SS. And generate a fundamental signal r [ n ] with good signal quality. However, since there is often a slight gap between the signal transmitting end and the signal receiving device 10, the difference is the above-mentioned carrier frequency offset (that is, the offset frequency value is not equal to 0).

Here, the fundamental frequency signal represents the following equation (1): r [ n ]= m [ n ]×cos(2 πf _offset T _s n + θ _offset ), where|m[n]|=|Acos(2 πf _BPSK T _s n )| (1) where m[n] represents the data signal carried by the fundamental frequency signal, that is, if there is no carrier frequency offset, the fundamental frequency signal r [ n ] will be equal to the data signal m[n]. A is the maximum amplitude of the data signal m[n], and f _BPSK is the data frequency in the data signal m[n], which is the data frequency of BPSK. The latter term in equation (1) is the error term caused by the carrier frequency offset and the phase difference, f _offset is the offset frequency, and θ _offset is expressed as the phase difference. Since the frequency value of the data frequency f _BPSK of the data signal m[n] is much larger than the frequency value of the offset frequency f _offset , the fundamental frequency signal r[n] shown in the above equation (1) is like the data signal m[n] The envelope of cos(2πf _offset T _s n + θ _offset ) envelops, so that the amplitude intensity of the data signal m[n] is controlled by the offset frequency f _offset and the magnitude is variable, and even the phase inversion may occur. . In this case, it is difficult for the back-end baseband processing circuit 140 to correctly determine the data content of the data signal m[n] when processing the baseband signal r[n] .

In the present invention, in order to eliminate the influence of the carrier frequency offset and the phase difference, the carrier frequency offset estimation and compensation algorithm is used to estimate the real IF signal. The frequency value f _{offset of the} envelope (ie, the offset of the intermediate frequency of the previous stage), after the offset frequency value f _{offset is} minimized, the carrier phase difference correction algorithm dynamically compensates the phase difference to improve The signal strength of the signal r[n] . Among them, in order to avoid the undue influence of the data in the data signal m[n] on the estimated content, the following estimation is mostly calculated by the absolute value of the fundamental frequency signal | r[n] | On the premise that the frequency value of the data frequency f _BPSK is much larger than the frequency value of the offset frequency f _offset (ie, the offset frequency value), the absolute value of the fundamental frequency signal | r[n] | can be exploited by Taylor expansion (Taylor) Expansion) (1) is as follows: Where | r _H [n] | indicates that the frequency is not less than the sum of all high frequency terms of the data frequency f _BPSK .

The carrier frequency offset estimation and compensation algorithm is described below. First, the processing unit 133 sets an overcompensated frequency value to the sum of the predefined intermediate frequency value and the overcompensated value f _OC , and transmits a control signal CS including the overcompensated frequency value to the numerically controlled oscillator 131. Then, when the numerically controlled oscillator 131 receives the control signal CS, the overcompensated synchronizing signal OSS is generated based on the overcompensated frequency value, and the compensated synchronizing signal OSS is transmitted to the digital down converter 132. When the digital down converter 132 receives the compensated synchronization signal OSS, the digital down converter 132 generates an overcompensated base frequency signal r _O [n] according to the overcompensated frequency value in the overcompensated synchronization signal OSS. And, the processing unit 133 obtains the frequency value of the offset frequency based on the overcompensated baseband signal r _O [n] .

The advantage of the overcompensated signal r _O [n] is that, compared to the spectrum of the absolute value | r[n] | of the fundamental frequency signal, the absolute value of the overcompensated fundamental frequency signal | r _O [n] | The distance between the individual harmonics is large, so the processing unit 133 will more easily lock the absolute value of the compensated fundamental frequency signal | r _O [n] | each harmonic in the spectrum, without the distance between the harmonics being too small And misjudged.

3A is a frequency spectrum diagram of an absolute value | r[n] | of a baseband signal according to an embodiment of the present invention, and FIG. 3B is an absolute value of an overcompensated baseband signal according to an embodiment of the invention. | Spectrogram of r _O [n] | 3A and 3B, it can be found that the distance between all the harmonics in the absolute value of the overcompensated baseband signal shown in FIG. 3B is widened to the overcompensation value f _OC plus the offset frequency f _offset Double the distance.

The processing unit 133 sets an estimation interval for one of the harmonics, and finds the frequency value of the harmonic in the interval. When the frequency value of the harmonic is known, the frequency value of the harmonic can be obtained. Shift frequency value (ie, the estimated frequency value of the offset frequency). For example, in the embodiment shown in FIG. 3B, the processing unit 133 sets an estimation interval for the first harmonic HS1, and the frequency value of the first harmonic HS1 in the spectrum shown in FIG. 3B is the overcompensation value f _OC plus The offset frequency f _offset is twice. In the case where the over-compensation value f _{OC is} known, the offset frequency value can be calculated using the frequency value of the first harmonic HS1 after obtaining the frequency value of the first harmonic HS1. .

The overcompensation value f _OC must be large enough so that the estimated intervals of the first harmonic and the second harmonic do not overlap. If the maximum value of the offset frequency f _offset is set to f _MAX , the relationship that the two adjacent estimation intervals cannot overlap each other is:

From equation (3), it is calculated that the overcompensated value f _OC must be greater than or equal to three times the offset frequency maximum value f _MAX . The estimated interval for each harmonic is: I _m =[2 m ( f _OC - f _MAX ), 2 m ( f _OC + f _MAX )] (4)

For example, FIG. 4A is a diagram showing a relationship between a first harmonic and an estimated interval according to an embodiment of the invention. Referring to FIG. 4A, the estimated interval I ₁ corresponding to the first harmonic HS1 is between 2 ( f _OC - f _MAX ) and 2 ( f _OC + f _MAX ). In the present embodiment, i.e., the processing unit 133 to estimate the interval I ₁ corresponding to the first harmonic frequency estimation value HS1 and thereby obtain an offset frequency f _offset.

FIG. 4B is a diagram showing a relationship between a first harmonic and a tone signal in an estimation interval according to an embodiment of the invention. In the present embodiment, the processing unit 133 places a plurality of single tone signals f _k in the estimation interval I ₁ . Wherein, the tone signals f _{k are} evenly distributed in the estimation interval I ₁ , and are separated from each other by twice the unit frequency value 2 f _U . The selection of the unit frequency value f _U will affect the accuracy of the frequency value at which the offset frequency is estimated. The value of k is between 0 and K, which is the index value of the tone signal f _k , and the value of K corresponds to the number of tone signals f _k . Therefore, when the processing unit 133 receives the compensated baseband signal r _O [n] , the processing unit 133 takes the absolute value of the overcompensated baseband signal r _O [n] . Next, the processing unit 133 calculates a plurality of correlation values between the absolute value | r _O [n] | of the compensated baseband signal and the tone signals f _k .

Among them, the tone signal f _k has a cosine sub-signal i[n, k] and a sine sub-signal q[n, k], respectively . The index value of the tone signal f _k closest to the first harmonic HS1 Then can be expressed as:

among them The correlation value of the cosine sub-signal i [ n , k ] of the tone signal f _k and the first harmonic HS1 can be expressed as:

The correlation value of the sinusoidal signal q [ n , k ] of the tone signal f _k and the first harmonic HS1 can be expressed as:

Where N is the number of sampling points. It is worth noting that the above correlation values can be calculated by using the angle formula to save the memory for storing each temporary tone signal. For example, when the content of the equation (5) is not calculated using the angle formula, the processing unit 133 requires a temporary memory of 2*N*(K+1) size, and when the content of the equation (5) is calculated using the angle formula, , you only need 2*(K+1) size of scratch memory.

The processing unit 133 finds an index value of the maximum correlation value in the equation (5) (among them, ),and The corresponding tone signal frequency is . Processing unit 133 and according to the target frequency value The estimated frequency value is generated. Target frequency value Can be expressed as:

When the target frequency value is obtained After that, the offset frequency can be derived as:

Get the offset frequency The processing unit 133 can shift the frequency The frequency value is added to the predefined intermediate frequency to obtain the corrected local intermediate frequency value, and then the local intermediate frequency value is transmitted to the numerically controlled oscillator 131 via the control signal CS. It is worth noting that it can be observed from Fig. 4B that the target frequency value may be caused due to the selection of the unit frequency value f _{U and} the like. The frequency value of the first harmonic HS1 is still slightly slightly different, making the offset frequency Still a little inaccurate. Therefore, in the present invention, the carrier phase difference correction algorithm is additionally used to reduce the above-mentioned difference or the influence of the phase difference.

FIG. 5 is a partial functional block diagram of a processing unit according to an embodiment of the invention. Referring to FIG. 5, in the embodiment, the processing unit 133 includes a high frequency reject unit 1331, a differentiator 1332 coupled to the high frequency signal canceling unit 1331, and a signal selecting unit 1333 coupled to the differentiator 1332.

In the present invention, the operational concept of the carrier phase difference correction algorithm is similar to the operation principle of a phase lock loop (PLL), including the following actions. First, the high frequency rejecting unit 1331 continues to receive the fundamental frequency signal r [ n ] from the digital down converter 132, and filters out a high frequency component of the fundamental frequency signal r [ n ] to obtain a residual low frequency signal RD. Next, the differentiator 1332 receives the residual low frequency signal RD from the high frequency rejecting unit 1331, and differentiates the fundamental frequency signal RD to obtain the indication signal Z. The indication signal Z is used to indicate the slope of the current sampling point of the residual low frequency signal RD, that is, to obtain the state in which the current residual low frequency signal RD is increasing or decreasing. Then, the signal selection unit 1333 receives the indication signal Z, Z and frequency _f compens generating a dynamic estimation instruction signal, so that the processing unit 133 may be adjusted according to the estimated frequency value a frequency value of the dynamic estimation of the frequency _f compens.

FIG. 6 is a functional block diagram showing a high frequency reject unit according to an embodiment of the invention, and an embodiment of the high frequency reject unit 1331 in the embodiment shown in FIG. 5 is provided. Referring to FIG. 6, the high frequency reject unit 1331 includes an absolute value calculator 610, a low pass filter (LPF) 620, and a down sample unit 630. The absolute value calculator 610 receives the fundamental frequency signal r[n] and calculates the absolute value of the fundamental frequency signal | r[n] |, and the purpose of taking the absolute value of the fundamental frequency signal r[n] in the present embodiment is to remove The influence of the data signal m[n] on the estimation is the same as in the previous embodiment.

The low-pass filter 620 is coupled to the absolute value calculator 610, and receives the absolute value of the fundamental frequency signal | r[n] |, and filters out the high-frequency component of the absolute value of the fundamental frequency signal | r[n] | 2) High frequency term | r _H [n] |. Therefore, for the low pass filter 620, the requirements of the following formula (10) are included:

The down sampling unit 630 is coupled to the low pass filter 620, receives the absolute value of the fundamental frequency signal | r [ n ]|, and downsamples the absolute value of the fundamental frequency signal | r [ n ]| to obtain the residual low frequency signal RD. The significance of downsampling is that the sampling points in the residual low frequency signal RD can be reduced, so that the operating frequency of the signal selecting unit 1333 can be lowered, and the cost can be reduced. In addition, the downsampling action has the function of low-pass filtering.

FIG. 7 is a functional block diagram of a signal selection unit according to an embodiment of the invention. Referring to FIG. 7, the signal selection unit 1333 includes a relay unit 710, a NXOR gate 720, a delay unit 730, and a selector 740. The relay unit 710 receives the indication signal Z and converts the indication signal Z into a digital signal X. As previously mentioned, the value of the indication signal Z represents that the current residual low frequency signal RD is in an incrementing or decrementing state, and the relay unit 710 further converts the incremental or decreasing state transition in a two-bit manner, wherein The relationship between the input signal (indication signal Z) of the relay unit 710 and the output signal (digital signal X) is as shown in the following table.

That is to say, when the value of the indication signal Z is positive, indicating that the current residual low frequency signal RD is in an increasing state, the relay unit 710 outputs 1. When the value of the indication signal Z is negative, indicating that the current residual low frequency signal RD is in a decreasing state, the relay unit 710 outputs 0.

The multiplexer or gate 720 has a first input end, a second input end, and an output end, wherein the first input end is coupled to the relay unit 710 and receives the digital signal X, the second input end is coupled to the delay unit 730, and the output end is coupled to the delay unit 730. The selection signal Y is output. The delay unit 730 is coupled between the second input end and the output end of the mutex or the 720, the delay selection signal Y is the delay selection signal Y _pre , and the delay selection signal Y _{pre is} transmitted to the mutex or the gate 720 The second input.

Briefly, the mutex or gate 720 outputs the selection signal Y based on the digital signal X and the selection signal of the previous time point, that is, the delay selection signal Y _pre . The table of truth values for mutual exclusion or gate 720 is shown in Table 2 below:

The selector 740 is coupled to the output of the mutex or gate 720, and outputs a dynamic estimated frequency value f _compens according to the selection signal Y. In the present embodiment, when the selection signal Y is equal to 1, the selector 740 outputs the offset frequency. The sum of the frequency constant f _const as the dynamic estimated frequency value f _compens or offset frequency The difference output with the frequency constant f _const is the dynamic estimated frequency value f _compens . After obtaining the dynamic estimated frequency value f _compens , the processing unit 133 may add the dynamic estimated frequency value f _compens to the predefined intermediate frequency frequency value to obtain the corrected local intermediate frequency frequency value, and transmit the control signal CS. The corrected local intermediate frequency value is passed to the numerically controlled oscillator 131.

Therefore, referring to Table 2 above, when the current fundamental frequency DC signal RD is in progress, referring to Table 2 above, when the current residual low frequency signal RD is in a decreasing state (digital signal X is 0), and the previous time point output offset frequency When the sum of the frequency constant f _const (Y _pre is 0), the signal selecting unit 1333 continuously outputs the offset frequency The difference from the frequency constant f _const (Y is 1). When the current residual low frequency signal RD is in an increasing state (the digital signal X is 1), and the previous time point outputs the offset frequency When the difference between the negative value and the frequency constant value f _const (Y _pre is 1), the signal selecting unit 1333 also outputs the offset frequency. The difference between the negative value and the frequency constant f _const (Y is 1).

When the current residual low frequency signal RD is in a decreasing state (digital signal X is 0), and the previous time point output offset frequency The signal selection unit 1333 outputs an offset frequency when the difference from the frequency constant f _const (Y _pre is 1) The negative value is compensated by the sum of the frequency constants f _const (Y is 0). And when the current residual low frequency signal RD is in an increasing state (the digital signal X is 1), and the previous time point output offset frequency When the sum of the frequency constant f _const (Y _pre is 0), the signal selecting unit 1333 continuously outputs the offset frequency The sum of the negative value and the frequency constant f _const (Y is 0).

The above offset frequency Obtained by the aforementioned carrier frequency offset estimation and compensation algorithm. The frequency constant value f _const is a constant value and is set in advance in the signal selecting unit 1333. The magnitude of the frequency constant f _const depends on the estimation error of the previous stage carrier frequency offset estimation and compensation algorithm (ideally the estimated error is less than the unit frequency value f _U ). For example, setting a frequency difference Δf is an offset frequency obtained by transmitting the carrier frequency offset estimation and compensation algorithm described above. The difference between the frequency value and the actual offset frequency f _offset . In this embodiment, the frequency constant f _const must be greater than the frequency difference Δf described above. In this way, the dynamic estimation frequency value f _compens can be switched (ie, The frequency of the residual frequency offset ( f _offset + f _compens ) is adjusted such that the influence of the dynamic estimated frequency value f _compens is large enough to compensate for the phase difference of the fundamental frequency signal r [ n ].

In addition, in this embodiment, the frequency constant value f _const also needs to meet the following two criteria, and the criterion one is a switching symmetry criterion, as shown in the following formula (11):

Such a criterion setting is mainly to avoid the sum of the negative value of the offset frequency and the frequency constant ( And the difference between the negative value of the offset frequency and the frequency constant ( The ratio is disparity, resulting in an asymmetry in the ability to compensate for the fundamental frequency signal r[n] .

The other is the switching speed criterion, as shown in the following equation (12)

Where M is the switching period and T _s is the signal sampling period

Since the sine wave or the cosine wave changes the polarity of the slope (increment/decrement state) every quarter cycle, there should be a sufficient number of samples per quarter cycle so that the signal selection unit 1333 does not make the signal selection unit 1333 less dynamic. Switch the dynamic estimated frequency value f _compens or misjudge the polarity of the slope. However, in actual conditions, there should be more sampling points in the quarter cycle to provide sufficient sampling data to the high frequency rejection unit for filtering, reducing the probability of high frequency components causing carrier phase error correction.

The carrier frequency offset estimation and compensation algorithm is implemented by a processor included in the processing unit 133, and the processor is coupled to the signal selection unit 1333 in FIGS. 5 and 7 to transmit an offset frequency. To signal selection unit 1333. The processor may be a fixed hardware circuit or a firmware solution (ie, the general purpose processor cooperates with executing a specific code). The hardware circuit (ie, signal selection unit 1333) for implementing the carrier phase difference correction algorithm in FIG. 7 may also be replaced by a firmware solution, and the present invention is not limited to the above embodiments.

The invention also provides a signal receiving method, which is suitable for a signal receiving device, wherein the signal receiving device receives a single carrier real number signal, wherein the single carrier real number signal is modulated by BPSK modulation.

FIG. 8 is a flow chart showing the steps of a signal receiving method according to an embodiment of the invention. Referring to FIG. 8, first, at step S801, a single carrier real number signal is received. Then, in step S802, the single-carrier real number signal is converted into a digital real IF signal, wherein the digital real IF signal is located at an intermediate frequency. Next, in step S803, the digital real IF signal is converted into a fundamental frequency signal according to a synchronization signal, wherein the synchronization signal is located at a local intermediate frequency. In step S804, the carrier frequency offset estimation and compensation algorithm is performed, an offset frequency value of the intermediate frequency is estimated, and the frequency value of the local intermediate frequency is adjusted according to the offset frequency value to compensate the digital The conversion between the frequency real signal and the fundamental signal. Next, in step S805, a carrier phase difference correction algorithm is executed to dynamically adjust the offset frequency value to correct a phase difference between the intermediate frequency real signal and the synchronization signal. Finally, in step S806, the baseband processing circuit demodulates the signal to obtain a digital data. For the detailed implementation of the method, reference may be made to the description of the embodiment shown in FIG. 1 to FIG. 7 , and details are not described herein.

In summary, the present invention provides a signal receiving apparatus and a signal receiving method. The signal receiving device is a receiver that receives a BPSK modulated single carrier signal, and the device has a lower complexity than a general complex signal receiver. However, a problem that carrier frequency offset estimation and compensation and carrier phase difference correction are difficult is accompanied. To solve this problem, a signal receiving method is proposed in the present invention. In this method, the harmonic spacing of the fundamental frequency signal is widened by the aid of the overcompensation procedure, so that the carrier frequency offset estimation and compensation algorithm can more accurately estimate the offset frequency and compensate it. Then the carrier phase difference correction algorithm dynamically adjusts the carrier instantaneous phase to reduce the carrier phase difference, so that the proposed receiving device When receiving a BPSK modulated single carrier signal, it is affected by the carrier frequency offset and phase difference on the signal.

10‧‧‧Signal receiving device

110‧‧‧RF circuit

120‧‧‧ Analog/Digital Converter

130‧‧‧down frequency unit

140‧‧‧ fundamental frequency processing circuit

R[n]‧‧‧ single carrier real number signal

‧‧‧Digital IF real signal

r [ n ]‧‧‧ fundamental frequency signal

Claims (16)

A signal receiving apparatus is adapted to receive a single carrier real number signal, wherein the single carrier real number signal is modulated by a Binary Phase Shift Keying (BPSK) method, including: An RF circuit receives the single carrier real number signal; a analog/digital converter coupled to the RF circuit receives the single carrier real number signal and converts it into a digital intermediate frequency real number signal, wherein the digital intermediate frequency real number signal is located in one a frequency down frequency unit, coupled to the analog/digital converter, converts the digital intermediate frequency real number signal into a fundamental frequency signal according to a synchronization signal, wherein the synchronization signal is located at a local intermediate frequency; and a fundamental frequency processing a circuit coupled to the down-converting unit, receiving the baseband signal and processing the baseband signal to obtain a digital data, wherein the down-converting unit performs a carrier frequency offset estimation and compensation algorithm to estimate the intermediate frequency An offset frequency value of the frequency, and adjusting a frequency value of the local intermediate frequency according to the offset frequency value to compensate between the digital intermediate frequency real signal and the base frequency signal Converter; and a frequency reduction unit performs carrier phase correction algorithm, the frequency of dynamically adjusting the offset value to correct the real intermediate frequency signal and a phase of the synchronization signal. The signal receiving device of claim 1, wherein the frequency down unit comprises: a numerically controlled oscillator that generates the synchronization signal, wherein the synchronization signal is located a local intermediate frequency, a digital down converter (DDC) coupled to the analog/digital converter and the numerically controlled oscillator, and converting the digital intermediate real signal to the fundamental frequency according to the synchronization signal And a processing unit coupled to the digital down converter and the numerically controlled oscillator to transmit a control signal to the numerically controlled oscillator, wherein the control signal includes the corrected intermediate frequency frequency value, wherein the correction The subsequent intermediate frequency value is a sum of a frequency value of a predefined local intermediate frequency and a dynamically estimated frequency value, and the processing unit dynamically corrects the offset frequency value to obtain the dynamic estimated frequency value, wherein The digital down converter adjusts the local intermediate frequency of the synchronization signal according to the corrected intermediate frequency value in the control signal to convert the digital intermediate frequency real signal into the fundamental signal. The receiving device of claim 2, wherein the carrier frequency offset compensation and estimation algorithm comprises: the processing unit setting an overcompensated frequency value, wherein the overcompensated frequency value is in the predefined local a sum of a frequency value of the frequency frequency and an overcompensated frequency value, and transmitting the control signal including the overcompensated frequency value to the numerically controlled oscillator; when the numerically controlled oscillator receives the control signal, according to the overcompensated frequency The value generates an over-compensation synchronization signal, and transmits the over-compensated synchronization signal to the digital down-converter; when the digital-down converter receives the over-compensated synchronization signal, the digital-down converter converts according to the over-compensated synchronization signal Obtaining an over compensated baseband signal; The processing unit obtains the offset frequency value according to the overcompensated baseband signal. The signal receiving device of claim 3, wherein: when the processing unit receives the overcompensated baseband signal, the processing unit takes an absolute value of the overcompensated baseband signal and calculates an absolute value. a plurality of correlation values between the overcompensated baseband signal and the plurality of single tone signals; and the processing unit determines that the largest correlation value among the correlation values is a maximum correlation value, and obtains An index value of the tone signals corresponding to the maximum correlation value, and the offset frequency value is calculated according to the index value. The signal receiving device of claim 4, wherein: the overcompensated baseband signal comprises a residual low frequency signal and n harmonic signals; the monophonic signals are distributed in an estimated interval, spaced apart from each other a unit frequency value having a sinusoidal sub-signal and a cosine sub-signal, wherein a first one of the n harmonic signals is located in the estimated interval; and one of the monophonic signals And the correlation value between the compensated baseband signals is a product of the sinusoidal sub-signal of one of the monophonic signals and the first harmonic signal of the overcompensated baseband signal, and integrated to obtain an absolute value and the tones The sum of the absolute values of the integral of the product of the cosine sub-signal of one of the signals and the first harmonic signal of the overcompensated baseband signal. The signal receiving device of claim 2, wherein the processing unit comprises: a high frequency rejecting unit; a differentiator coupled to the high frequency rejecting unit; a signal selecting unit coupled to the differentiator; and the carrier phase difference correction algorithm comprising: the high frequency rejecting unit receiving the baseband signal and filtering the baseband signal a high frequency component obtains a residual low frequency signal; the differentiator receives the residual low frequency signal from the high frequency rejecting unit, and differentiates the residual low frequency signal to obtain an indication signal; and the signal selecting unit receives the indication signal, and The offset frequency value is adjusted according to the indication signal to obtain the dynamic estimated frequency value. The signal receiving device of claim 6, wherein the high frequency rejecting unit comprises: an absolute value calculator, receiving the fundamental frequency signal, and calculating an absolute value of the fundamental frequency signal; a low pass filter, The absolute value calculator is coupled to receive the absolute value of the baseband signal, and filter the high frequency component of the absolute value of the baseband signal; a downsampling unit coupled to the low pass filter to receive the baseband signal The absolute value of the fundamental frequency signal is downsampled to obtain the residual low frequency signal. The signal receiving device of claim 6, wherein the signal selecting unit comprises: a relay unit, receiving the indication signal, and converting the indication signal into a digital signal; and a mutual exclusion or gate (NXOR gate) ) having a first input and a second input And an output end, wherein the first input end of the mutex or the gate is coupled to the relay unit and receives the digital signal, and the output of the mutex or the gate outputs a selection signal; a delay unit coupled to the Between the second input end and the output end of the mutex or the gate, delaying the selection signal as a delay selection signal, and transmitting the delay selection signal to the second input terminal of the non-mutation or gate; a selector, according to Selecting a signal to output the dynamic estimated frequency value, wherein the selector selects a sum of a negative value of the offset frequency value and a frequency constant value or a negative value of the offset frequency value and the frequency constant value according to the selection signal The difference output is the dynamic estimated frequency value. A signal receiving method is applicable to a signal receiving apparatus, wherein the signal receiving apparatus receives a single carrier real number signal, wherein the single carrier real number signal is modulated by a two-bit phase offset modulation method, and the signal receiving is performed. The method includes: receiving the single carrier real number signal; converting the single carrier real number signal into a digital intermediate frequency real number signal, wherein the digital intermediate frequency real number signal is located at an intermediate frequency; and converting the digital intermediate frequency real number signal according to a synchronization signal a baseband signal, wherein the synchronization signal is located at a local intermediate frequency; performing a carrier frequency frequency offset estimation and compensation algorithm, estimating an offset frequency value of the intermediate frequency, and adjusting according to the offset frequency value a frequency value of the local intermediate frequency to compensate for conversion between the digital intermediate frequency real signal and the base frequency signal; performing a carrier phase difference correction algorithm to dynamically adjust the offset frequency value to correct the intermediate frequency real number a phase difference between the signal and the synchronization signal; The baseband signal is processed to obtain a digital data. The signal receiving method according to claim 9, wherein the step of converting the digital intermediate frequency real signal into the fundamental frequency signal according to the synchronization signal comprises: dynamically correcting the offset frequency value to obtain the dynamic estimated frequency value. Generating a control signal, wherein the control signal includes the corrected intermediate frequency frequency value, wherein the corrected intermediate frequency frequency value is a predetermined frequency value of a local intermediate frequency and the dynamic estimated frequency value And generating a synchronization signal according to the control signal; adjusting a frequency value of the local intermediate frequency to the corrected intermediate frequency frequency value; and converting the digital intermediate frequency real signal to the fundamental frequency signal according to the synchronization signal. The signal receiving method according to claim 10, wherein the carrier frequency offset compensation and estimation algorithm comprises: setting an overcompensation frequency value to a frequency value of the predefined local intermediate frequency and a And overcompensating the sum of the frequency values; generating an overcompensated synchronization signal according to the overcompensated frequency value; generating an overcompensated baseband signal according to the overcompensated frequency value in the overcompensated synchronizing signal; and obtaining the overcompensated baseband signal according to the overcompensated baseband signal The offset frequency value. The signal receiving method according to claim 11, wherein the step of obtaining the offset frequency value according to the overcompensated baseband signal comprises: taking an absolute value of the overcompensated baseband signal, and calculating an absolute value Overcompensation Compensating for a plurality of correlations between the baseband signal and the plurality of single tone signals; and determining that the largest correlation value among the correlation values is a maximum correlation value, and finding the maximum correlation An index value of the monophonic signals of the value, and the frequency value is estimated according to the index value. The signal receiving method of claim 12, wherein the overcompensated baseband signal comprises a residual low frequency signal and n harmonic signals; the monophonic signals are distributed in an estimated interval and spaced apart from each other a unit frequency value having a sinusoidal sub-signal and a cosine sub-signal, wherein a first one of the n harmonic signals is located in the estimated interval; and one of the monophonic signals And the correlation value between the compensated fundamental frequency signals is a product of the sinusoidal sub-signal of one of the monophonic signals and the absolute value of the overcompensated baseband signal, and the absolute value and the cosine of one of the monophonic signals are taken The sum of the absolute values of the product of the sub-signal and the absolute value of the over-compensated baseband signal. The signal receiving method according to claim 11, wherein the carrier phase difference correction algorithm comprises: filtering a high frequency component of the baseband signal to obtain a residual low frequency signal; and differentiating the residual low frequency signal to obtain a And indicating the signal; and adjusting the offset frequency value according to the indication signal to obtain the dynamic estimated frequency value. The signal receiving method of claim 14, wherein the step of filtering the high frequency component in the baseband signal to obtain the residual low frequency signal comprises: Calculating an absolute value of the baseband signal; filtering the high frequency component of the absolute value of the baseband signal; and downsampling the absolute value of the baseband signal to obtain the residual low frequency signal. The signal receiving method of claim 14, wherein the step of adjusting the estimated frequency value according to the indication signal to the dynamic estimated frequency value comprises: converting the indication signal to a digital signal; according to the digital signal, And a delay selection signal, outputting a selection signal; and outputting the dynamic estimation frequency value according to the selection signal, wherein the dynamic estimation frequency value is a sum of a negative value of the offset frequency value and a frequency constant value, or The difference between the negative value of the offset frequency value and the constant value of the frequency.