TWI433446B - Rotor position estimation method of interior permanent magnet synchronous motor - Google Patents

Description

Method for estimating shaft angle of built-in permanent magnet synchronous motor

The invention relates to an estimation technique for a shaftless angle detecting component, and particularly relates to a shaft angle/speed estimation method for a built-in permanent magnet synchronous motor under a three-phase four-switch inverter architecture.

Motor drive systems use a wide range of powers, ranging from a few watts to several kilowatts, ranging from high-performance position drivers for robots to adjustable speed drives that adjust pump flow rates. In order to apply the vector control to the motor drive system, the conversion relationship between the a-b-c axis and the d-q axis coordinate must be calculated by detecting the angle of the rotary shaft.

Generally, the method of detecting the angle of the rotating shaft is to utilize the salient pole effect of the built-in permanent magnet synchronous motor, and the inductance of the stator changes with the angle of the rotating shaft. Therefore, a high-frequency carrier current can be injected on the stator side of the motor to detect a change in the rotor position, and the high-frequency carrier current is not affected by the rotational speed. Conventional techniques are used for low speed operation, high speed operation, and even start at rest. However, in addition to causing torque chatter, this high frequency carrier current must also add an additional carrier current generator.

The Republic of China Patent Publication No. 483218 is an estimation method for the position (angle) of the three-phase AC motor shaft, which discloses measuring the sub-phase current change rate in the switching state of the inverter, and calculating the motor shaft angle. According to the third item of the request of the technology, the frequency converter is composed of six power switches and a flywheel diode, and two zero voltage vectors and six non-zero voltage vectors are obtained by switching the switches; As can be seen from the content of item 6 of the claim, the conventional estimation method must use the three-phase current rate of change under the zero voltage vector to estimate the angle of the shaft. It can be seen that the conventional technology requires a frequency converter structure composed of six power switches when detecting the position of the rotating shaft of the motor, and requires a current rate of change of the zero voltage vector.

See Figure 1. 1 is a conventional technique. Under the architecture of a six-switch inverter, the switching strategy of the inverter adopts current hysteresis control, and the switching pattern distribution simulation diagram of the motor at the rotational speed command 50 rpm. Where Switch Mode 1 and Switch Mode 8 are zero voltage vectors, and Switch Mode 2 to Switch Mode 7 are non-zero voltage vectors.

See Figure 2. 2 is a prior art diagram of a switching mode distribution simulation diagram of a zero-voltage vector switching strategy at a rotational speed command of 50 rpm under the architecture of a six-switch inverter. If the switching mode is not changed periodically and the frequency converter is switched out of the zero voltage vector, the conventional method cannot be directly applied to the inverter architecture that can only switch out the non-zero voltage vector. In addition, as can be seen from Figure 2, forcing the frequency converter to switch to the zero voltage vector too frequently will greatly reduce the utilization of other non-zero voltage vectors.

The conventional method requires the use of six power switches and a three-phase current rate of change under a zero voltage vector. However, under the original switching strategy of the inverter, the probability of occurrence of the zero voltage vector is very low compared to the non-zero voltage vector, so the calculation is quite difficult. It can be seen that the main disadvantage of the prior art is that the current rate of change of the zero voltage vector is required, and the number of power switches cannot be reduced.

How to solve the conventional technology without using zero voltage vector, how to reduce the number of voltage vectors, to achieve a simplified operation, and can be applied to the industrial application of the estimated angle of the shaft, this is a problem to be overcome.

The invention provides a method for estimating a rotating shaft angle, which is suitable for estimating a rotating shaft angle in a built-in permanent magnet synchronous motor, and the estimating method comprises: maintaining a original switching switching strategy in a three-phase four-switching inverter structure And reading a first phase current and a second phase current of the stator; calculating a three-phase current change rate in different switch switching modes according to the first phase current and the second phase current; and then calculating a three-phase current The difference in rate of change is used for matrix conversion to estimate the angle of the shaft.

In an embodiment of the invention, the wiring of the built-in permanent magnet synchronous motor is a three-phase Y wiring, and the three-phase four-switch inverter has four switching states, and generates four different and non-zero a voltage vector; and the first phase current and the second phase current are stator two-phase currents of the built-in permanent magnet synchronous motor.

In an embodiment of the invention, the difference between the three-phase current change rates is defined as calculating a difference between the three-phase current change rates in the two non-zero voltage vector switching modes, and each of the switch switching states has a corresponding Stator voltage equation.

In an embodiment of the invention, the difference in the three-phase current change rate satisfies the following formula relationship:

In formula (A), Di _as | _Vij represents the a-phase stator current change rate difference of the motor under two non-zero voltage vectors; Di _bs | _Vij represents the b-phase stator current change rate of the motor under two non-zero voltage vectors Di _cs | _Vij represents the c-phase stator current change rate difference of the motor under two non-zero voltage vectors; v _asi - v _asj represents the a-phase stator voltage difference of the motor under two non-zero voltage vectors; v _bsi - v _bsj represents the b-phase stator voltage difference of the motor under two non-zero voltage vectors; and L _AA , L _AB , L _AC , L _BA , L _BB , L _BC , L _q are the self-inductance and mutual inductance related parameters of the motor .

In an embodiment of the invention, the three-phase four-switch inverter outputs a non-zero voltage vector at each switching, and at least the step of reading the first phase current and the second phase current is required. Switch more than twice.

In an embodiment of the invention, the difference in the three-phase current change rate includes a stator current change rate difference of the motor a phase, the b phase, and the c phase.

In an embodiment of the invention, the stator current change rate difference of the a phase, the b phase, and the c phase of the motor satisfies the following formula relationship:

In formulas (B) to (D), Di _as | _Vij , Di _bs | _Vij and Di _cs | _Vij represent the stator current change rate difference of the a phase, b phase and c phase of the motor under different voltage vectors, respectively; i _as , i _bs , i _cs represent the three-phase stator current of the motor respectively; the subscript Vi represents the switching mode of the three-phase four-switch inverter under the voltage vector V _i ; and the subscript Vj represents the three-phase four-switch inverter The switching mode is under the voltage vector V _j .

In an embodiment of the invention, at least three non-zero voltage vectors are required to calculate the three-phase current rate of change.

In an embodiment of the invention, in the step of converting using the difference of the three-phase current change rate, a sine function and a cosine function are synthesized from the difference between the plurality of sets of three-phase stator current change rates, and then An arctangent function operation is performed with the sine function and the cosine function to estimate the axis angle.

In an embodiment of the invention, the difference between the plurality of sets of three-phase stator current change rates must be linear independent vectors with each other.

Based on the above, the present invention applies a three-phase stator voltage mathematical model of a built-in permanent magnet synchronous motor. In the four switch switching modes, it is not necessary to measure the current change rate under the zero voltage vector. The three-phase stator voltage equation is calculated, and the stator voltage equation is a nonlinear function that contains information on the electrical angle and the double electrical angle of the motor. Under reasonable assumptions, the phase reduction of the three-phase stator voltage equation under two different non-zero voltage vectors can eliminate one-time electrical angle information and resistance voltage drop associated with the motor, and retain the information of the double electrical angle. The information of the current change rate difference, and then through the designed matrix operation, can obtain the double electrical angle correlation function under the static coordinate axis. This double electrical angle correlation function is related to the position of the rotating shaft, so that the information of the rotating shaft angle of the built-in permanent magnet synchronous motor can be further estimated.

The above described features and advantages of the invention will be apparent from the following description.

Reference will now be made in detail be made to the embodiments of the invention

In this embodiment, a shaft angle/speed estimation method which is simple and easy to implement in a built-in permanent magnet synchronous motor is proposed, and the current switching change strategy of the inverter and the current variation without using a zero voltage vector are not changed. Under the conditions of the rate, the characteristics of the inverter switching and the rate of current change can still be used.

4 is a block diagram of a built-in permanent magnet synchronous motor speed control system according to an embodiment of the present invention. See Figure 4. The wiring of the built-in permanent magnet synchronous motor is a three-phase Y wiring. Since the general motor can be a balanced three-phase motor 1, the digital signal processor 4 can obtain the two phase currents i _a , i _{b of the} motor 1 detected by the current sensor 2 via the analog/digital converter 3. In this way, the current i _c of the other phase can be calculated. Then, the three-phase stator current change rate can be calculated via the digital signal processor 4. The axis estimation program is still based on the stator current change rate difference under different voltage vectors. Through the designed matrix operation, the corresponding three-axis to two-axis can be found in the stator current change rate difference under different voltage vectors. Coordinate transformation matrix; further through the inverse tangent function operation, the rotation angle can be further estimated . The estimated angle can then be calculated via the differentiator 6 to calculate the estimated shaft speed. . Then, the shaft speed Speed command with shaft Calculate AC and DC current commands via speed controlled component 7 And the AC and DC current commands are via the current controlled component 8. Convert to three-phase current command And transmitting to the component 9 of the pulse width modulation control to generate a trigger signal for controlling the power switches T _{a 0} , T _{a 1} , T _{b 0} , T _{b 1} , thereby determining the switching mode of the four power switches. Thereafter, a trigger signal is sent to the frequency converter (inverter) 10 to complete a closed loop drive system.

The method for estimating the rotor angle/speed of the embodiment of the present invention is applicable to a built-in permanent magnet synchronous motor digitalized driving system. In the existing digital drive system and three-phase four-switch inverter architecture, only the current signal of the motor needs to be measured, and the estimation method is written into the digital signal processor and executed in one program, which can replace the traditional shaft. Angle detector.

A detailed description of the estimation method of the angle of the shaft is as follows :

The three-phase stator equivalent voltage equation of the built-in permanent magnet synchronous motor can be expressed as follows:

among them

In the formulas (1) to (4), v _as , v _bs , and v _cs are the a-phase, b-phase, and c-phase stator voltages of the motor, respectively, i _as , i _bs , and i _cs are the phase a of the motor, respectively. , b phase, c phase stator current, r _s is stator resistance, L _q is q axis equivalent inductance, L _A is the inductance DC coefficient of the motor, L _B is the inductance AC coefficient of the motor, L _ls is the leakage inductance, λ _as λ _bs and λ _cs are the extended flux flux chains of phase a, phase b, and phase c, respectively, which are defined as follows:

λ _as = L _AA i _a + L _AB i _b + L _AC i _c + λ _m cos θ _re formula (5);

among them

Where λ _m is the rotor permanent magnet equivalent to the stator side flux linkage, and θ _re is the motor shaft angle (electrical angle). The torque and speed equations of the motor can be expressed as follows:

Where P _o is the pole pair of the motor, L _d is the d-axis inductance, L _q is the q-axis inductance, i _d is the d-axis current, i _q is the q-axis current, T _e is the electromagnetic torque of the motor, J _m It is the shaft inertia of the motor, ω _rm is the shaft mechanical speed of the motor, B _m is the shaft friction coefficient of the motor, and T _L is the load torque.

3 is a simulation diagram of four voltage vector distributions of a conventional three-phase four-switch inverter at a rotational speed command of 200 rpm. See Figure 3. In the three-phase four-switch inverter, under the switching strategy of current hysteresis control, the four voltage vectors appear randomly and irregularly. The frequency converter (inverter) has four switching modes by switching between four power switches.

FIG. 5 is a schematic diagram of four voltage vectors of a conventional three-phase four-switch inverter. See Figure 5. Four different voltage vectors are represented by V ₁ , V ₂ , V ₃ and V ₄ , and the four voltage vectors and corresponding switching states are as shown in Table 1 below.

The embodiment of the invention does not need to measure the current change rate under the zero voltage vector, and does not need to change the original switch switching strategy of the frequency converter, and only needs to measure the current change rate under four non-zero voltage vectors. In order to facilitate understanding of the four voltage vectors and the corresponding switching mode, when the voltage vector is V ₁ , the power switches T _a0 and T _{b0 are} turned on as shown in FIG. 6; when the voltage vector is V ₂ , the power switches T _a1 and T _{B0 is} turned on as shown in FIG. 7; when the voltage vector is V ₃ , the power switches T _a0 and T _{b1 are} turned on as shown in FIG. 8; when the voltage vector is V ₄ , the power switches T _a1 and T _{b1 are} turned on as shown in FIG. 9 . Show.

Note: Each phase has a pair of power switches, and each pair of power switches can control the operation of the upper arm and the lower arm. When "1" represents the upper arm of the power switch is turned on, and when "0" represents the lower of the power switch The arm is turned on.

In order to explain the basic principle of the axis estimation of the embodiment of the present invention, in the three-phase four-switch inverter architecture, the switching state of the voltage vector V ₁ is taken as an example. At this time, the power switches T _{a 0} and T _{b 0} are in a conducting state as shown in FIG. 6 . In the voltage vector V ₁ state, and using equations (1) to (16), the a-phase and b-phase stator voltage equations and the three-phase current-slope balance equations of the built-in permanent magnet synchronous motor can be obtained, respectively:

Equation (19) to (21), the subscript represents the change amount V1 is limited in voltage vector V ₁ state.

Similarly, when the switching state of the three-phase four-switch inverter under the voltage vector V ₂ is taken as an example, the power switches T _{a 1} and T _{b 0} are turned on as shown in FIG. 7 , in the state of the voltage vector V ₂ . And using equations (1) to (16), the a-phase and b-phase stator voltage equations and the three-phase current-slope balance equations of the built-in permanent magnet synchronous motor are obtained, respectively:

In the formulas (22) to (24), the subscript symbol V2 indicates that the amount of change is limited to the state of the voltage vector V ₂ .

Similarly, when the switching state of the three-phase four-switch inverter under the voltage vector V ₃ is taken as an example, the power switches T _{a 0} and T _{b 1} are turned on as shown in FIG. 8 , in the state of voltage vector V ₃ . And using equations (1) to (16), the a-phase and b-phase stator voltage equations and the three-phase current-slope balance equations of the built-in permanent magnet synchronous motor are obtained, respectively:

In the formulas (25) to (27), the subscript symbol V3 indicates that the amount of change is limited to the state of the voltage vector V ₃ .

Similarly, when the switching state of the three-phase four-switch inverter under the voltage vector V ₄ is taken as an example, the power switches T _{a 1} and T _{b 1} are turned on as shown in FIG. 9 , in the state of the voltage vector V ₄ . And using equations (1) to (16), the a-phase and b-phase stator voltage equations and the three-phase current-slope balance equations of the built-in permanent magnet synchronous motor are obtained, respectively:

In the formulas (28) to (30), the subscript symbol V4 indicates that the amount of change is limited to the state of the voltage vector V ₄ .

When the three-phase four-switch inverter is switched, it is assumed that the change in the angle of the inductance to the rotation axis does not change during the adjacent sampling interval, that is, the following conditions are true:

Since the three-phase current is balanced, that is, i _a + i _b + i _c =0, the sum of the current change rate differences under different voltage vectors is zero, that is, the following conditions are true:

Di _as | _Vij + Di _bs | _Vij + Di _cs | _Vij =0, i ≠ j and i , j =1, 2, 3, 4 Equation (32).

Among them, the three-phase stator current change rate difference under different voltage vectors is defined as:

It can be seen from equation (32) that Di _as | _Vij , Di _bs | _Vij and Di _cs | _Vij are linearly dependent, and the corresponding phase a and sum of two different voltage vectors can be selected via equations (19) to (31). The b-phase stator voltage equation and the formula (32) can eliminate the influence of the stator resistance voltage drop and the back electromotive force, and the relationship between the obtained three-phase stator current change rate differences is arranged in a matrix manner as follows:

The corresponding 3×3 matrices in equation (36) are all reversible, so the relationship between Di _as | _Vij , Di _bs | _Vij and Di _cs | _Vij and the angle of the shaft and the voltage V _d can be obtained. In order to obtain that the three stator current change rate differences selected under different voltage vectors are linearly independent and the found relationship matrix is reversible, there are many combinations that satisfy such a combination. For convenience of explanation, only the following combinations are listed. Change as a reference.

For example, if the combination of stator current change rate differences under three different voltage vectors V ₁ , V ₂ , V _{4 is selected} , the formula (37) can be satisfied:

Where Di _as | _{V 12} and Di _bs | _{V 12} can use equation (36) to sort out the relationship matrix:

Substituting the formulas (8) to (16) into the formula (39) to solve Di _as | _{V 12} and Di _bs | _{V 12} is:

For the same reason, it can be obtained by formula (36):

Substituting equations (8)-(16) into equation (42) to solve Di _bs | _{V 14} is:

Rearrange the derived Di _as | _{V 12} , Di _bs | _{V 12} and Di _bs | _{V 14} as follows:

among them

The relationship matrix of Di _α , Di _β and Di _dc can be obtained from equations (44) to (46):

It can be seen from the formula (50) that Di _α and Di _β can be calculated by the formula (50), and the inverse tangent function of the Di _α and Di _β equations can be used to obtain the rotation axis angle θ _re :

It is worth mentioning that the embodiment of the present invention is a three-phase stator voltage mathematical model of a built-in permanent magnet synchronous motor, a three-phase stator voltage equation in four switching modes, wherein the stator voltage equation is a nonlinear function, including The electric motor doubles the electrical angle and the double electrical angle information. Under reasonable assumptions, the three-phase stator voltage equation subtraction under two different non-zero voltage vectors can eliminate one-time electrical angle information and resistance related to the motor. The voltage drop, and retain the information of the second electrical angle, the obtained current change rate difference information, and then through the designed matrix operation, can obtain the double electrical angle correlation function under the static coordinate axis, this function is related to the position of the rotating shaft, so The shaft angle information of the built-in permanent magnet synchronous motor can be further estimated.

In addition to the formula (38), there are many possible combinations that satisfy the formula (37), and 23 embodiments are exemplified below. The relationship between the current change rate difference and the conversion matrix of Di _α and Di _β under different voltage vectors is the formula (52) ~ (74), please refer to the following:

It can be seen from equations (52)-(74) that Di _α and Di _β can be obtained by calculating the current change rate difference under different voltage vectors through the designed matrix operation. Finally, it can be easily obtained by formula (51). Estimating the angle θ _re of the shaft and then differentiating the angle of the obtained shaft, the shaft speed ω _re can be obtained as:

As can be seen from the above detailed description, the method for estimating the angle of the rotating shaft of the embodiment of the present invention can realize the position control and speed control of the shaftless detecting element in the built-in permanent magnet synchronous motor. Repeating the above analysis means, we can deduce the relationship between the current change rate difference of any three non-zero voltage vectors and the conversion matrix of Di _α and Di _β , and do not need to use any zero voltage vector current change. rate.

It is worth mentioning that under the three-phase four-switch frequency conversion architecture, zero voltage vector cannot be generated. Therefore, the conventional technique cannot be directly applied to the topology of the three-phase four-switch of this embodiment.

In the estimation method of the rotation angle of the embodiment, it is known from the prior art that the current change rate difference of the formulas (33) to (35) is the information that can be obtained, and then, by the corresponding conversion matrix operation, Di _α can be obtained. And the value of Di _β , and then the formula (51) can estimate the angle of the shaft. In other words, the current change rate of the a-phase and the b-phase is recorded under each switching, and the current change rates of the a-phase and the b-phase under other different voltage vectors are used to obtain the a-phase and the b-phase under three different voltage vectors. The current change rate difference is calculated by using a pre-designed transformation matrix to calculate Di _α and Di _β , and finally the angle of the shaft can be estimated by the formula (51).

Since the three-phase four-switch inverter (inverter) is switched at the switch, the four switching modes of the switch appear randomly. Here is another example for illustration.

Assuming that the kth switch is switched to the voltage vector V ₃ , the current change rate of the a phase and the b phase can be measured by a conventional technique. with Assuming that the switching state of the switch adjacent to the kth time before the kth time is the voltage vector V ₄ , then with Information; Di _as | _{V 34} and Di _bs | _{V 34} can be obtained using equations (33) to (34). In general, the switching frequency of the inverter switch can reach the number of switching times of 10,000 times in one second, so different voltage vectors must appear in the vicinity of the kth time, assuming a voltage vector V ₂ , so The information can be obtained by using equation (34) to obtain Di _bs | _{V 32} . Finally, the calculation of the formula (74) and formula (51) can estimate the information of the angle θ _re of the shaft.

Based on the description of the above embodiments, FIG. 10 is a flowchart of a method for estimating a shaft angle according to an embodiment of the present invention. The method for estimating the angle of the shaft of the embodiment may include the following steps: reading the phase currents i _as and i _bs (step S1005); calculating the rate of change of the current (Step S1010); calculating the current change rate difference using Equation (33) and Formula (34) (Step S1015), wherein Equation (33) and Formula (34) are respectively equivalent to Formula (B) in [Invention] And the formula (C); using the formula (50) to the formula (74) to obtain Di _α and Di _β (step S1020); and calculating by the formula (51) (Step S1025).

However, the above-described embodiments are merely illustrative of the present invention and are not intended to limit the actual application of the present invention.

About the simulation results:

The embodiment of the invention uses the computer set software Matlab as the simulation platform to simulate the built-in permanent magnet synchronous motor drive system powered by the three-phase four-switch inverter to verify the feasibility of the proposed method. Several simulation results demonstrate that the method of the embodiments of the present invention is achievable. FIG. 11 is a schematic diagram of a Di _α and Di _β waveform synthesized by a matrix operation of a built-in permanent magnet synchronous motor powered by a three-phase four-switch inverter in a rotation speed command of 200 rpm according to an embodiment of the present invention. It can be seen from Fig. 11 that the proposed method can successfully extract the component signals Di _α and Di _{β from the} current change rate difference signal and synthesize them into a circular trajectory.

FIG. 12 is a schematic diagram showing the a-phase stator voltage distribution of a built-in permanent magnet synchronous motor powered by a three-phase four-switch inverter in a rotation speed command of 200 rpm according to an embodiment of the present invention. FIG. 13 is a schematic diagram showing the c-phase stator voltage distribution of a built-in permanent magnet synchronous motor powered by a three-phase four-switch inverter in a rotation speed command of 200 rpm according to an embodiment of the present invention. 12 and 13, the output voltage of the three-phase four-switch inverter is the same as that in Table 1, that is, the stator voltages of the a-phase and the c-phase are consistent with the derivation results in Table 1.

FIG. 14 is a comparison diagram of the actual value and the estimated value of the shaft speed of the built-in permanent magnet synchronous motor supplied by the three-phase four-switch inverter in the rotation speed command 200 rpm according to the embodiment of the present invention. FIG. 15 is a comparison diagram of the actual value and the estimated value of the shaft angle of the built-in permanent magnet synchronous motor powered by the three-phase four-switch inverter in the rotation speed command 200 rpm according to the embodiment of the present invention. Referring to FIG. 14 and FIG. 15, it is shown that the simulation results of estimating the angle/speed of the rotating shaft in the position control and the medium speed control are idealized in the embodiment of the present invention.

FIG. 16 is a comparison diagram of the actual value and the estimated value of the shaft angle of the built-in permanent magnet synchronous motor powered by the three-phase four-switch inverter according to the embodiment of the present invention at a rotational speed command of 50 rpm. FIG. 17 is a diagram showing an estimated angle error of a rotating shaft at a rotational speed command of 50 rpm for a built-in permanent magnet synchronous motor powered by a three-phase four-switch inverter according to an embodiment of the present invention. Referring to FIG. 16 and FIG. 17, it is shown that the simulation results of estimating the angle of the rotating shaft in the position control and the low speed control are idealized in the embodiment of the present invention.

FIG. 18 is a comparison diagram of the actual value and the estimated value of the shaft angle of the built-in permanent magnet synchronous motor powered by the three-phase four-switch inverter in the embodiment of the present invention at a rotation axis command of 140 degrees. FIG. 19 is a diagram showing an estimated angle error of a rotating shaft of a built-in permanent magnet synchronous motor powered by a three-phase four-switch inverter according to an embodiment of the present invention. Referring to FIG. 18 and FIG. 19, it is shown that the simulation result of estimating the angle of the rotating shaft of the embodiment of the present invention tends to be idealized.

It can be seen from the correlation simulation results of FIG. 14 to FIG. 19 that the technology of the embodiment of the present invention is applicable to the built-in permanent magnet synchronous motor of the industry, and the angle of the shaft can be estimated in both the position control and the low speed control.

In summary, the embodiment of the present invention uses four non-zero voltage vectors generated by a three-phase four-switch inverter to measure the current change rate difference signal under two different non-zero voltage vectors, and estimates The shaft angle of the built-in permanent magnet synchronous motor. Embodiments of the present invention have at least the following advantages over the prior art:

(1) It is not necessary to use a frequency converter architecture consisting of six power switches;

(2) The current rate of change under zero voltage vector is not used for measurement;

(3) There is no need to change the switching strategy of the original inverter switch;

(4) Easy to calculate.

Although the present invention has been disclosed in the above embodiments, it is not intended to limit the present invention, and those skilled in the art can make some changes and refinements without departing from the spirit and scope of the present invention. The scope of the invention is defined by the scope of the appended claims.

1. . . electric motor

2. . . Current sensor

3. . . Analog/digital converter

4. . . Digital signal processor

5. . . Rotor angle estimation component

6. . . Differentiator

7. . . Speed control unit

8. . . Current controlled components

9. . . Pulse width modulation control component

10. . . Inverter (inverter)

i _a , i _b , i _c . . . Three-phase stator current

. . . Three-phase stator current command

T _a0 , T _a1 , T _b0 , T _b1 . . . Power switch

V ₁ , V ₂ , V ₃ , V ₄ . . . Voltage vector

. . . Straight axis current command

. . . Axial current command

‧‧‧Reducing angle of the shaft

‧‧‧Reducing speed of shaft

‧‧‧Rotary speed command

S1005~S1025‧‧‧Reducing method of shaft angle

FIG. 1 is a conventional technique. Under the architecture of a six-switch inverter, the switching strategy of the inverter is to use current hysteresis control, and the switching pattern of the motor is simulated at a speed command of 50 rpm.

2 is a prior art diagram of a switching mode distribution simulation diagram of a zero-voltage vector switching strategy at a rotational speed command of 50 rpm under the architecture of a six-switch inverter.

Figure 3 is a conventional three-phase four-switch inverter. The switching strategy of the inverter is current hysteresis control. When the motor is at the speed command 200 rpm, the four voltage vector distributions are simulated.

4 is a block diagram of a built-in permanent magnet synchronous motor speed control system according to an embodiment of the present invention.

FIG. 5 is a schematic diagram of four voltage vectors of a conventional three-phase four-switch inverter.

6 is a schematic diagram showing an equivalent circuit of a switching state of a voltage vector V ₁ and a built-in permanent magnet synchronous motor in a three-phase four-switch inverter structure according to an embodiment of the present invention.

FIG. 7 is a schematic diagram showing the switching state of the switching under the voltage vector V ₂ and the equivalent circuit of the built-in permanent magnet synchronous motor in the three-phase four-switch inverter structure according to the embodiment of the present invention.

FIG. 8 is a schematic diagram showing the switching state of the switch under the voltage vector V ₃ and the equivalent circuit of the built-in permanent magnet synchronous motor under the three-phase four-switch inverter structure according to the embodiment of the present invention.

FIG. 9 is a schematic diagram showing the switching state of the switching under the voltage vector V ₄ and the equivalent circuit of the built-in permanent magnet synchronous motor in the three-phase four-switch inverter structure according to the embodiment of the present invention.

FIG. 10 is a flowchart of a method for estimating a rotation angle of an embodiment of the present invention.

FIG. 11 is a schematic diagram of a Di _α and Di _β waveform synthesized by a matrix operation of a built-in permanent magnet synchronous motor powered by a three-phase four-switch inverter in a rotation speed command of 200 rpm according to an embodiment of the present invention.

FIG. 12 is a schematic diagram showing the a-phase stator voltage distribution of a built-in permanent magnet synchronous motor powered by a three-phase four-switch inverter in a rotation speed command of 200 rpm according to an embodiment of the present invention.

FIG. 13 is a schematic diagram showing the c-phase stator voltage distribution of a built-in permanent magnet synchronous motor powered by a three-phase four-switch inverter in a rotation speed command of 200 rpm according to an embodiment of the present invention.

FIG. 14 is a comparison diagram of the actual value and the estimated value of the shaft speed of the built-in permanent magnet synchronous motor supplied by the three-phase four-switch inverter in the rotation speed command 200 rpm according to the embodiment of the present invention.

FIG. 15 is a comparison diagram of the actual value and the estimated value of the shaft angle of the built-in permanent magnet synchronous motor powered by the three-phase four-switch inverter in the rotation speed command 200 rpm according to the embodiment of the present invention.

FIG. 16 is a comparison diagram of the actual value and the estimated value of the shaft angle of the built-in permanent magnet synchronous motor powered by the three-phase four-switch inverter according to the embodiment of the present invention at a rotational speed command of 50 rpm.

FIG. 17 is a diagram showing an estimated angle error of a rotating shaft at a rotational speed command of 50 rpm for a built-in permanent magnet synchronous motor powered by a three-phase four-switch inverter according to an embodiment of the present invention.

FIG. 18 is a comparison diagram of the actual value and the estimated value of the shaft angle of the built-in permanent magnet synchronous motor powered by the three-phase four-switch inverter in the embodiment of the present invention at a rotation axis command of 140 degrees.

FIG. 19 is a diagram showing an estimated angle error of a rotating shaft of a built-in permanent magnet synchronous motor powered by a three-phase four-switch inverter according to an embodiment of the present invention.

S1005~S1025. . . Rotating axis angle estimation method flow chart steps

Claims (4)

An estimation method for a shaft angle in a three-phase four-switch inverter architecture, which is suitable for estimating a shaft angle in a built-in permanent magnet synchronous motor, wherein the estimation method comprises: while retaining the original switch switching strategy, Reading a first phase current and a second phase current of the stator; calculating a three-phase current change rate according to the first phase current and the second phase current; and performing a matrix operation by using a difference between the three-phase current change rates According to the estimated angle of the rotating shaft, wherein the difference of the three-phase current change rate is defined as a difference between the three-phase current change rates in the two non-zero voltage vector switching modes. The method for estimating a rotation angle of a three-phase four-switch inverter structure according to the first aspect of the patent application, wherein the connection of the built-in permanent magnet synchronous motor is a three-phase Y connection, and the three-phase four-switch The frequency converter has four switch switching states and generates four different and non-zero voltage vectors; and the first phase current and the second phase current are built into the three-phase four-switch inverter architecture A two-phase current of a stator of a permanent magnet synchronous motor under a non-zero voltage vector. The method for estimating the angle of the rotating shaft in the three-phase four-switch inverter architecture as described in claim 1, wherein the difference in the three-phase current change rate satisfies the following formula relationship: In formula (A), Di _as | _Vij represents the a-phase stator current change rate difference of the motor under two non-zero voltage vectors; Di _bs | _Vij represents the b-phase stator current change rate of the motor under two non-zero voltage vectors Di _cs | _Vij represents the c-phase stator current change rate difference of the motor under two non-zero voltage vectors; v _asi - v _asj represents the a-phase stator voltage difference of the motor under two non-zero voltage vectors; v _bsi - v _bsj represents the b-phase stator voltage difference of the motor under two non-zero voltage vectors; and L _AA , L _AB , L _AC , L _BA , L _BB , L _BC , L _q are the self-inductance and mutual inductance related parameters of the motor . The method for estimating a shaft angle in a three-phase four-switch inverter architecture according to claim 1, wherein the difference in the three-phase current change rate includes a stator current of the a phase, the b phase, and the c phase of the motor The rate of change difference satisfies the following formula relationship: In formulas (B) to (D), Di _as | _Vij , Di _bs | _Vij and Di _cs | _Vij represent the stator current change rate difference of the a phase, b phase and c phase of the motor under different voltage vectors, respectively; i _as , i _bs , i _cs represent the three-phase stator current of the motor respectively; the subscript Vi represents the switching mode of the three-phase four-switch inverter under the voltage vector V _i ; and the subscript Vj represents the three-phase four-switch inverter The switching mode is under the voltage vector V _j .