UZ4449 UMCD-2006-0025 19504twf.doc/g 九、發明說明： 【發明所屬之技術領域】 • 本發明是關於一種混頻哭，日4主<2,丨β 、 吧用0。’且特別是關於一種使用摺 • 疊式電感電容疊接之次譜波混頻器。 【先前技術】 ，直接降頻接收器（direci_c〇nversi〇n敗以或 . hO—麗i·)在架構上採用一次的降頻動作，直接將 ㈣訊號降至基頻訊號，因此又稱為零中頻接收器(zer〇_IF 此種接收器所產生的多相位本地振盈訊號之振 盪頻率，與射頻訊號的振盪頻率非常接近，因此不僅避免 了鏡像雜訊(image noise)的干擾，且伴隨鏡像雜訊干擾的 消失，射頻訊號與多相位本地振盪訊號進行混波之前就不 品預置一鏡像抑制濾波器（image rejecti〇n 丨ter)。如此一 來’相較於其它接收器架構-例如差外差接收器 (superheterodyne receiver)，直接降頻接收器具有架構簡 單、單晶片化等優點，而逐漸地在現今的收發器（transceiver) • 中嶄露頭角。 直接降頻接收器雖具有單晶片化等優點，但其架構仍 存在一些缺點必須去克服。其中直流偏移(DC offset)為其 衍生的問題之一。直流偏移的產生主要是由於混頻器 (mixer)和低雜訊放大器（I· n〇ise ampiifier，簡稱LNA)的 輸入端(在此統稱為射頻訊號的輸入端），與混頻器接收多 相位本地振盪訊號的輪入端之隔絕度(isolation)並非無限 大’因此當多相位本地振盪訊號經由穿隧(feedthr〇ugh)效 5 1324449 UMCD-2006-0025 195〇4twf.doc/g 應，出現在射頻訊號的輸入端時，將和原本的多相位本地 振盪訊號進行自我混波(sdf_mixing)進而形成直流偏移。此 外’偶次階失真(even_〇rder distortion)也是一個值得注意的 問題，因為直接降頻接收器進行混波之前不具有鏡像抑制 濾波器，因此伴隨在射頻訊號附近的干擾訊號，經由非線 性電路所產生的偶次階失真，也會經由穿隧效應而直接傳 送到混頻器的輸出，進而影響真正想接收的射頻訊號。UZ4449 UMCD-2006-0025 19504twf.doc/g IX. Description of the invention: [Technical field to which the invention pertains] • The present invention relates to a type of mixing crying, day 4 main < 2, 丨β, bar 0. And especially with respect to a sub-spectral mixer that uses folded-stacked inductors and capacitors. [Prior Art], the direct down-conversion receiver (direci_c〇nversi〇n defeated or .hO-Li i·) uses a frequency-down action on the architecture to directly reduce the (four) signal to the fundamental frequency signal, so it is also called Zero-IF receiver (zer〇_IF) The oscillation frequency of the multi-phase local oscillation signal generated by this receiver is very close to the oscillation frequency of the RF signal, so it not only avoids the interference of image noise. And with the disappearance of the image noise interference, the image rejection signal (image reject) is not preset before the RF signal and the multi-phase local oscillation signal are mixed. Thus, compared with other receivers. Architecture - such as the superheterodyne receiver, the direct down-conversion receiver has the advantages of simple architecture, single-chip, etc., and gradually emerges in today's transceivers. Direct down-conversion receivers have The advantages of single-waferization, but there are still some shortcomings in the architecture that must be overcome. Among them, DC offset is one of the problems derived from it. Due to the mixer and the input of the low noise amplifier (LNA), which is collectively referred to as the input of the RF signal, the mixer receives the multi-phase local oscillation signal. The isolation of the input is not infinitely large. Therefore, when the multi-phase local oscillation signal passes through the tunneling effect, the input of the RF signal is shown in the signal of the singularity of the singularity of the singularity. At the end, it will self-mix (sdf_mixing) with the original multi-phase local oscillation signal to form a DC offset. In addition, the even-order distortion (even_〇rder distortion) is also a significant problem because of direct down-conversion reception. The image does not have an image rejection filter before mixing, so the even-order distortion generated by the nonlinear circuit accompanying the interference signal near the RF signal is directly transmitted to the output of the mixer via the tunneling effect. In turn, it affects the RF signal that you really want to receive.
為了解決上述問題，直接降頻接收器採用如圖丨所示 的傳統次谐波混頻器（sub-harmonic mixer，簡稱SHM)，來 提供良好的隔絕度給多相位本地振盪訊號與射頻訊號的輸 入端。傳統次諧波混頻器是由如圖2所示之吉伯特混頻器 (Gilbert mixer)衍生而來的。繼續參照圖丄與圖2，其中多 相位本地振盪訊號L01包括相移量分別為〇〇、9〇〇、、In order to solve the above problem, the direct down-converter uses a traditional sub-harmonic mixer (SHM) as shown in Figure , to provide good isolation to multi-phase local oscillator signals and RF signals. Input. The conventional subharmonic mixer is derived from the Gilbert mixer shown in Figure 2. Continuing to refer to FIG. 2 and FIG. 2, wherein the multi-phase local oscillation signal L01 includes phase shift amounts of 〇〇, 9〇〇, and
謂〇之本地《訊號LQ1—0。〜咖—加，μ相位本地振 盪訊號L02則包括相移量分別為〇g與18〇〇之本地振盪訊 號L02 一 0〇與L02—180〇。吉伯特混頻器中的每一個N型電 曰曰體MN9〜MN12，务·分別以兩個相互並聯的n型電晶體 取代，並將提供給吉伯特混頻器之多相位本地振盪訊號 L02之振盪頻率，降低至原本振盪頻率的〇 5倍，以形成 由相互並聯的NM0S電晶體所接收的多相位本地振盪訊 號L01，將可次變成如圖1所示的傳統次諧波混頻器。例 如MN9由MN1與MN2取代，且MN1與MN2接收相差 為180G的本地振盪訊號L01J)G與L〇i-180〇之振盪頻率， 是MN9所接收的本地振盪訊號l〇2_〇q的2倍。如此一來， 6 1324449 UMCD-2006-0025 19504twf.doc/g 採用傳統次諧波混頻器之直接降頻接收器，不僅可以將多 相位本地振盪訊號之振盪頻率操作在〇5倍的射頻訊號之 振盪頻率，且還可保有吉伯特混頻器所具有的良好隔絕度。 然而，上述之傳統次諧波混頻器在實際的晶片化過程 中，往往會因製程技術所造成的元件不匹配，進而造成電 路在不對_情況下’影響傳統切波混頻賴能提供的 隔絕度。因此，如何利用原有的電路架構來提高傳統次諧 波混頻器輸人端的隔絕度，以降低多相位本地振盛訊號的 洩漏(leakage)訊號，與干擾訊號在射頻訊號輸入端所^成 的偶次階失真，已是直接降頻接收器在應用上所面臨的最 大隱憂。 【發明内容】 有鑑於此’本發明的目的是在提供—種次譜波滿頻 器，利用諧振電路將洩漏訊號導向至第一電壓或第二電壓 的方式，讓讀波混頻H具錢好的隔絕度，進而^決使 用次譜波混頻器之直接降頻接收器在應用上，所面臨的直 流偏移與偶次階失真的問題。 本發明的另-目的是提供—種下轉換器，利用次谐波 混頻器所具㈣良好闕度，減少$難號所造成自我混 波與偶次階失真’進而提昇使用下轉換器之工作特性。 。。為達成上述及其他目的，本發明提出一種次譜波滿頻 器’用以將多相位本地振盛訊號與射頻訊號進行混頻，以 產生基頻訊號。該次舰混頻H包括絲放大單元、'電流 緩衝單元、以及城單元。絲放大單元將㈣訊號放大。 7 1324449 UMCD-2006-0025 19504twf.doc/g 電流緩衝單元耦接至差動放大單元，用以將差動放大單元 之輸出訊號之增益放大。切換單元耦接至電流緩衝單元， 用以依據多相位本地振盪訊號，而將電流缓衝單元之輸出 訊號切換至基頻訊號。 依照本發明的較佳實施例，上述之差動放大單元包括 * 第一諧振電路，電流緩衝單元則包括第二諧振電路。為了 * 讓次諧波混頻器達到將多相位本地振盪訊號與射頻訊號進 行混頻，以產生基頻訊號的目的。首先，差動放大單元將 射頻訊號放大，且放大後的射頻訊號利用第一諧振電路導 向至電流緩衝單元，而此時本地振盪訊號所產生的洩漏訊 號，也將利用第一諧振電路導向至第一電壓。之後，轉接 至差動放大單元的電流緩衝單元，將差動放大單元之輸出 訊號之增益放大，並利用第二諧振電路將電流緩衝單元之 輸出訊號導向至切換單元，且第二諧振電路此時也將本地 振盈sfl號所產生的茂漏訊號，導向至第二電壓。最後，切 換單元依據多相位本地振盪訊號，而將電流緩衝單元之輸 籲 出訊號切換至基頻訊號。 ] 上述之第一諧振電路與第二諧振電路之共振頻率，與 射頻訊號之振盪頻率相同，且第一電壓可為一操作電壓， 第二電壓可為一接地電壓。 在一較佳實施例中，上述之次諧波混頻器適用於直接 降頻接收器。 從另一觀點來看，本發明另提出一種下轉換器，用以 將射頻訊號轉換至基頻訊號。下轉換器包括訊號2生器與 8 1324449 UMCD-2006-0025 19504twf.doc/g 次譜波混頻器。其中次"t皆波混頻器包括差動放大單元 “ 流缓衝單元、以及切換單元。訊號產生器用以提供—夕= 位本地振盪訊號。耦接至訊號產生器的次諧浊 曰叹此頻态則用It is said that the local "signal LQ1-0. ~Caf-plus, μ phase local oscillation signal L02 includes local oscillation signals L02 - 0〇 and L02-180〇 with phase shift amounts of 〇g and 18〇〇, respectively. Each of the N-type electric bodies MN9 to MN12 in the Gilbert mixer is replaced by two n-type transistors connected in parallel with each other, and the multi-phase local oscillation provided to the Gilbert mixer is provided. The oscillation frequency of the signal L02 is reduced to 〇5 times of the original oscillation frequency to form a multi-phase local oscillation signal L01 received by the NM0S transistors connected in parallel with each other, which can be converted into the conventional subharmonic mixture as shown in FIG. Frequency. For example, MN9 is replaced by MN1 and MN2, and MN1 and MN2 receive the oscillation frequency of local oscillation signals L01J)G and L〇i-180〇 which are different from 180G, which is 2 of the local oscillation signal l〇2_〇q received by MN9. Times. In this way, 6 1324449 UMCD-2006-0025 19504twf.doc/g The direct down-conversion receiver using the traditional subharmonic mixer can not only operate the oscillation frequency of the multi-phase local oscillation signal at 〇5 times the RF signal. The oscillation frequency, and can also maintain the good isolation of the Gilbert mixer. However, in the actual wafer-forming process, the above-mentioned conventional subharmonic mixers often have component mismatch caused by the process technology, which in turn causes the circuit to affect the traditional CW mixing. Insulation. Therefore, how to use the original circuit architecture to improve the isolation of the traditional subharmonic mixer input, to reduce the leakage signal of the multi-phase local Zhensheng signal, and the interference signal at the input of the RF signal Even-order distortion is the biggest concern for direct down-conversion receivers in applications. SUMMARY OF THE INVENTION In view of the above, the object of the present invention is to provide a sub-spectral full-frequency device, and to use a resonant circuit to direct a leakage signal to a first voltage or a second voltage, so that the read wave mixing H has money. Good isolation, and in turn, the problem of DC offset and even order distortion faced by the direct down-conversion receiver using the sub-spectral mixer. Another object of the present invention is to provide a down-converter that utilizes a subharmonic mixer to have (4) good latitude and reduce the self-mixing and even-order distortion caused by the $-signal, thereby improving the use of the downconverter. Working characteristics. . . To achieve the above and other objects, the present invention provides a sub-spectral full frequency unit </ RTI> for mixing a multi-phase local oscillating signal with an RF signal to generate a fundamental frequency signal. The submarine mixing H includes a wire amplifying unit, a current buffer unit, and a city unit. The wire amplification unit amplifies the (four) signal. 7 1324449 UMCD-2006-0025 19504twf.doc/g The current buffer unit is coupled to the differential amplifying unit for amplifying the gain of the output signal of the differential amplifying unit. The switching unit is coupled to the current buffer unit for switching the output signal of the current buffer unit to the base frequency signal according to the multi-phase local oscillation signal. In accordance with a preferred embodiment of the present invention, the differential amplifying unit comprises a first resonant circuit and the current buffering unit comprises a second resonant circuit. In order to enable the subharmonic mixer to mix the multiphase local oscillator signal with the RF signal to generate the baseband signal. First, the differential amplifying unit amplifies the RF signal, and the amplified RF signal is guided to the current buffer unit by the first resonant circuit, and the leakage signal generated by the local oscillation signal is also guided to the first resonant circuit by using the first resonant circuit. A voltage. Afterwards, the current buffer unit is switched to the differential amplifying unit to amplify the gain of the output signal of the differential amplifying unit, and the output signal of the current buffer unit is guided to the switching unit by the second resonant circuit, and the second resonant circuit The leak signal generated by the local vibration sfl number is also directed to the second voltage. Finally, the switching unit switches the output buffer signal of the current buffer unit to the base frequency signal according to the multi-phase local oscillation signal. The resonant frequency of the first resonant circuit and the second resonant circuit is the same as the resonant frequency of the RF signal, and the first voltage may be an operating voltage, and the second voltage may be a ground voltage. In a preferred embodiment, the subharmonic mixer described above is suitable for direct downconversion receivers. From another point of view, the present invention further provides a down converter for converting the RF signal to the baseband signal. The downconverter includes a signal 2 generator and 8 1324449 UMCD-2006-0025 19504twf.doc/g sub-spectral mixer. The second "t-wave mixer includes a differential amplifying unit "flow buffer unit, and a switching unit. The signal generator is used to provide - eve = local oscillation signal. The second harmonic sigh coupled to the signal generator. This frequency is used
以將多相位本地振盪訊號與射頻訊號進行混頻，以產生義 頻訊號。其中次諧波混頻器產生基頻訊號的過裎包括，, 用差動放大單元將射頻訊號放大。接著，耦接至差動放大 單元的電流缓衝單元，在將差動放大單元之輪出訊號之增 益放大。最後’利用耦接至電流緩衝單元與訊號產生°哭^ 切換單元，依據多相位本地振盈訊號，而將電流緩衝單一' 之輸出訊號轉換成基頻訊號。藉此，下轉換器就可達到$ 一射頻訊號轉換至一基頻訊號的目的。 ： 上述之下轉換器，在一較佳實施例中，訊號產生哭包 括本地振盪器與相位偏移器。本地振盪器用以產生 振盪訊號，讓串接在本地振盪器與切換單元之間的相位偏 移器’可以將本地振盪訊號轉換成數個不同相移量的本地 振盪訊號，以輸出作為多相位本地振盪訊號。The multi-phase local oscillation signal is mixed with the RF signal to generate a frequency signal. The subharmonic mixer generates a fundamental frequency signal including, and the differential amplifying unit amplifies the RF signal. Then, the current buffer unit coupled to the differential amplifying unit amplifies the gain of the round-trip signal of the differential amplifying unit. Finally, the switching signal is coupled to the current buffer unit and the signal generating unit, and the output signal of the current buffer is converted into the fundamental frequency signal according to the multi-phase local vibration signal. In this way, the down converter can achieve the purpose of converting a RF signal to a fundamental frequency signal. : In the above preferred converter, in a preferred embodiment, the signal is generated by a local oscillator and a phase shifter. The local oscillator is used to generate an oscillating signal, and the phase shifter connected between the local oscillator and the switching unit can convert the local oscillating signal into a plurality of local oscillating signals of different phase shift amounts to output as multi-phase local Oscillation signal.
在一較佳實施例中，上述之下轉換器 接收器。 適用於直接降頻 本發明因採用差動放大單元與電流緩衝單元組合之^ 構，讓次諧波混頻器可利用第一諧振電路與第二諧振^ 路，達到將洩漏訊號導向至第一電壓或第二電壓之功效 如此一來’隨著次諧波混頻器隔絕度的提升，使用次1皆 混頻器之直接降頻接收器，所面臨的直流偏移與偶=二/类 真也將大幅度地降低。 白 9 1324449 UMCD-2006-0025 19504twf.doc/g 為讓本發明之上述和其他目的、特徵和優點能更明顯 易懂，下文特舉較佳實施例，並配合所附圖式，作詳細說 明如下。 ° 【實施方式】 圖3為根據本發明一較佳實施例之次譜波混頻器結構 示思圖，包括差動放大單元3〇1、電流緩衝單元302、以及 切換單元303。電流緩衝單元302耦接於差動放大單元3〇1 鲁與切換單元30:3之間。差動放大單元301將射頻訊號放大 後，次諧波混頻器300利用電流緩衝單元3〇2將差動放大 單元302之輸出訊號之增益放大，以讓切換單元3〇3依據 多相位本地振盪訊號，將電流緩衝單元3〇2之輸出訊號切 換至基頻訊號。如此一來，次諧波混頻器3〇〇就可以達到 將多相位本地振盪訊號與射頻訊號進行混頻，進而產生義 頻訊號的功效。 圖4為根據本發明較佳實施例之次諧波混頻器詳細電 路圖。對照圖3來看’其中輸入端31〇a與31〇b對應圖^ • 中，差動放大單元之差動輸入端310。輸出端32〇a與 對應圖3中，動放大單元301之差動輸出端320。輪 出端330a與330b對應圖3中，電流緩衝單元3〇2之差^ 輸出端330。輸出端340a與340b對應圖3中，切換單__ 3〇3之差動輸出端340。輸入端350aa與350ab、以及輪入 端350ba與350bb則對應圖3中’切換單元303之多相位 本地振盪訊號輸入端350a與350b。 1324449 UMCD-2006-0025 19504twf.doc/g 如圖4所示，差動放大單元301包括諧振電路4(n、N 型電晶體MN41與MN42、電阻R41與R42 '電容C41與 C42、以及電感L41。電流缓衝單元302包括p型電晶^ M41與MP42、以及諧振電路402。切換單元3〇3包=n 型電晶體MN43〜MN410、以及電阻R43與R44。電阻R41 • 與R42之弟一々而輕接至弟一電壓（比如操作電壓。n * 型電晶體M41與MP42之>及極搞接至譜振電路，n型 電晶體M41與MP42之源極耦接至第二電壓（比如接地電 壓）’ N型電晶體M41與MP42之閘極則分別耦接至電^ C41與C4之第一端。電感L41串接在電容C41之第二端 與電容C4之第二端之間。電容C4i與C42之第一端分^ 耦接至電感L41之第一端與第二端，且電容C41與 之弟一端分別拉線構成輸入端310a與310b。P型電晶體 M41與MP42之源極分別耦接至n型電晶體M41與m^42 之汲極’P型電晶體M41與MP42之閘極耦接至第二電壓。 譜振電路402串接在P型電晶體M41與MP42之閘極與第 • 二電壓之間。電阻R43與R44之第一端耦接至第一電壓’ . 電阻R43與R44之第二端則分別拉線構成輸出端34〇a與 340b。N型電晶體MN43與MN44、以及_47與議48 之汲極轉接至電阻R43之第二端。N型電晶體MN45與 MN46、以及MN49與MN410之汲極耦接至電阻R44之第 一知。且N型電晶體MN43-MN46之源極輕接至p型電晶 體M41之汲極。N型電晶體MN47〜MN410之源極耦接至 P型電晶體M42之汲極。 11 U24449 UMCD-2006-0025 19504twf.doc/g 上述之諧振電路401包括電感L42與L43、以及電容 C45與C46。諧振電路402則包括電感L44與L45、以及 電容C47與C48。電感L42與L43、以及電容C45與C46 之第一端耦接至第一電壓。電感L42與電容C45之第二端 搞接至N型電晶體MN41之汲極。電感L43與電容C46 之第二端耦接至N型電晶體MN42之汲極。電感L44與電 容C47之第一端耦接至P型電晶體MP41之汲極。電感L45 φ 與電容C48之第一端柄接至p型電晶體MP42之汲極。電 感L44與L45、以及電容C47與C48之第二端耦接至第二 電壓。 繼續參照圖4來看本實施例之工作原理。差動放大單 元301中的電感L41與L42 ’在此提供低阻抗路徑，以形 成N型電晶體MN41與MN42的直流偏壓電流。此時，經 由電容C43與C44之第二端所接收的射頻訊號，藉由操作 上相當於差動轉導的N型電晶體MN41與MN42放大。為 了將放大後的射頻訊號經由輸出端320a與320b傳送至電 鲁流緩衝單元302，本實施例將諧振電路4〇1之共振頻率操 作在射頻訊號之振盪頻率。由於譜振電路401操作在共振 頻率下相當於一高阻抗，操作在共振頻率外相當於一低阻 抗。因此，放大後的射頻訊號將可被導向至電流緩衝單元 302。不僅如此，由於次諧波混頻器3〇〇之多相位本地振盪 訊號之振盪頻率為射頻訊號之振盪頻率的〇5倍。因此， 由多相位本地振盪訊號經由穿隧效應出現在輸入端3i〇a 1324449 UMCD-2006-0025 19504twf.doc/g 與310b的洩漏訊號’或是由干擾訊號經由祚線性電路所產 生的偶次階失真’也將被諧振電路401導向至第一電壓。 之後的電流緩衝單元302利用P型電晶體M41與 MP42之源極接收放大後的射頻訊號。電感L44與L45提 供低阻抗路徑’形成P型電晶體MP41與MP42的直流偏 壓電流。此時，連接成共閘極組態的P型電晶體MP41與 - MP42 ’除了有助於電流緩衝單元302之隔絕度，並單增益 φ 放大由差動放大單元301所輸出的訊號。為了將由P型電 晶體MP41與MP42所放大的射頻訊號，導向至切換單元 303 ’在此諧振電路4〇2採取如同諧振電路4〇1 一樣的做 法’將共振頻率操作在射頻訊號之振盪頻率，讓電流緩衝 單το 302之輸出訊號導向至切換單元3〇3的同時，諧振電 路402也可將由多相位本地振盪訊號經由穿隧效應出現在 輸入端310a與310b的洩漏訊號，或是由干擾訊號經由非 線性電路所產生的偶次階失真，導向至第二電壓。 最後，多相位本地振盪訊號所包含的本地振盪訊號 LO4__00、L04—90〇、LO4—1800、以及 L04—270〇，分別經由 • 切換單元303内的n型電晶體MN43與MN49之閘極、 MN45與MN48之閘極、MN44與MN410之閘極、以及 MN46與MN47之閘極所接收。此時’操作特性相當於開 關的N型電晶體MN43〜MN41〇，則依據本地振盪訊號 L〇4_00、LO4—900、L04—1800、以及 LO4—2700，將電流緩 衝單元302之輸出訊號切換至基頻訊號。其中本地振盪訊 13 1^24449 UMCD-2006-0025 I9504twf.d〇c/g 號 L04J)〇、L〇4_9〇o、L〇4J8〇0、以及 L〇4—27〇◦之相移 量分別為0度、90度、180度、以及27〇度。 圖5〜圖8為本實施例實現在現今CM〇s製程技術下 的貫際：!：測結杲。本實施例在操作電壓VC。為1V、射頻 ^ 訊號之振盪頻率為5.2GHz、多相位本地訊號之振盪頻率為 2.6GHz之條件下。如圖5所示，次諧波混頻器3〇〇在頻率 ' 為1〇MHz下的雜訊指數(NoiseFigure)為17_3dB。且如圖6 _ 所不，次諧波混頻器300在接收功率為15.5(1Βπι之多相位 本地訊號下，於輸入端310a與310b測量到功率為 -65.154dBm的洩漏訊號。此洩漏訊號將造成次諧波混頻器 300輸出功率約為_i〇〇.7dBm的直流偏移，但此直流偏移 卻未超過無線區域網路(WLAN)接收器所規範的背景雜訊 (noise floor)。換而言之，此時次諧波混頻器3〇〇所形成的 直流偏移將掩蓋在背景雜訊中，而不會影響到電路本身的 工作性能。此外，如圖7所示的，於次諧波混頻器3〇〇之 輸入端310a與310b，只測量到非常地微小的2倍洩漏訊 _ 號(功率僅為-l〇9.934dBm)，由此可言正明本發明所提出的次 , 諧波混頻器3〇〇之輸入端具有良好的隔絕度。為了更進一 步了解本發明之電路性能，圖8列出了本實施例與電機電 子工程師協會(IEEE, Institute of Electrical and ElectronicIn a preferred embodiment, the lower converter receiver is described above. Applicable to direct down-conversion The invention adopts a combination of a differential amplifying unit and a current buffering unit, so that the subharmonic mixer can utilize the first resonant circuit and the second resonant circuit to guide the leakage signal to the first The effect of the voltage or the second voltage is such that with the increase in the isolation of the subharmonic mixer, the direct down-conversion receiver using the second-order mixer is faced with the DC offset and even = two/class It will also be greatly reduced. The above and other objects, features, and advantages of the present invention will become more apparent from the aspects of the appended claims appended claims as follows. [Embodiment] FIG. 3 is a structural diagram of a sub-spectral wave mixer according to a preferred embodiment of the present invention, including a differential amplifying unit 3'1, a current buffering unit 302, and a switching unit 303. The current buffer unit 302 is coupled between the differential amplifying unit 3〇1 and the switching unit 30:3. After the differential amplifying unit 301 amplifies the RF signal, the subharmonic mixer 300 amplifies the gain of the output signal of the differential amplifying unit 302 by using the current buffer unit 3〇2, so that the switching unit 3〇3 is based on multi-phase local oscillation. The signal switches the output signal of the current buffer unit 3〇2 to the baseband signal. In this way, the subharmonic mixer can achieve the function of mixing the multi-phase local oscillation signal and the RF signal to generate the frequency signal. 4 is a detailed circuit diagram of a subharmonic mixer in accordance with a preferred embodiment of the present invention. Referring to Fig. 3, the input terminals 31A and 31B correspond to the differential input terminal 310 of the differential amplifying unit. The output terminal 32A corresponds to the differential output terminal 320 of the dynamic amplification unit 301 in Fig. 3. The wheel ends 330a and 330b correspond to the difference between the current buffer unit 3〇2 and the output terminal 330 in Fig. 3. The output terminals 340a and 340b correspond to the differential output terminal 340 of the single __3〇3 in FIG. The inputs 350aa and 350ab, and the wheel terminals 350ba and 350bb correspond to the multi-phase local oscillator signal inputs 350a and 350b of the 'switching unit 303' in Fig. 3. 1324449 UMCD-2006-0025 19504twf.doc/g As shown in FIG. 4, the differential amplifying unit 301 includes a resonant circuit 4 (n, N type transistors MN41 and MN42, resistors R41 and R42 'capacitors C41 and C42, and an inductor L41) The current buffer unit 302 includes p-type electro-crystals M41 and MP42, and a resonance circuit 402. The switching unit 3〇3 includes n-type transistors MN43 to MN410, and resistors R43 and R44. Resistor R41 • Like the brother of R42 And lightly connected to a voltage (such as operating voltage. n * type transistor M41 and MP42 > and extremely connected to the spectrum circuit, the source of the n-type transistor M41 and MP42 is coupled to the second voltage (such as Grounding voltage) 'The gates of the N-type transistors M41 and MP42 are respectively coupled to the first ends of the capacitors C41 and C4. The inductor L41 is connected in series between the second end of the capacitor C41 and the second end of the capacitor C4. The first ends of the capacitors C4i and C42 are coupled to the first end and the second end of the inductor L41, and the capacitor C41 and the other end of the cable are respectively connected to form the input terminals 310a and 310b. The source of the P-type transistors M41 and MP42 The gates of the drain-P-type transistors M41 and MP42, which are respectively coupled to the n-type transistors M41 and m^42, are coupled to the second voltage. The circuit 402 is connected in series between the gates of the P-type transistors M41 and MP42 and the second voltage. The first ends of the resistors R43 and R44 are coupled to the first voltage '. The second ends of the resistors R43 and R44 are respectively pulled. The lines form the output terminals 34A and 340b. The N-type transistors MN43 and MN44, and the drains of _47 and 48 are switched to the second end of the resistor R43. The N-type transistors MN45 and MN46, and MN49 and MN410 The drain is coupled to the first resistor R44, and the source of the N-type transistor MN43-MN46 is lightly connected to the drain of the p-type transistor M41. The source of the N-type transistor MN47~MN410 is coupled to the P-type The drain of the transistor M42. 11 U24449 UMCD-2006-0025 19504twf.doc/g The above resonant circuit 401 includes inductors L42 and L43, and capacitors C45 and C46. The resonant circuit 402 includes inductors L44 and L45, and capacitor C47 and C48. The first ends of the inductors L42 and L43 and the capacitors C45 and C46 are coupled to the first voltage. The second end of the inductor L42 and the capacitor C45 are connected to the drain of the N-type transistor MN41. The inductor L43 and the capacitor C46 The second end is coupled to the drain of the N-type transistor MN42. The first end of the inductor L44 and the capacitor C47 are coupled to the drain of the P-type transistor MP41. The first end of the inductor L45 φ and the capacitor C48 is connected to the drain of the p-type transistor MP42. The inductors L44 and L45, and the second ends of the capacitors C47 and C48 are coupled to the second voltage. The working principle of this embodiment will be further described with reference to FIG. 4. Inductors L41 and L42' in differential amplifying unit 301 here provide a low impedance path to form the DC bias current of N-type transistors MN41 and MN42. At this time, the radio frequency signals received through the second ends of the capacitors C43 and C44 are amplified by the N-type transistors MN41 and MN42 which are equivalent to the differential transconductance. In order to transmit the amplified RF signal to the DC buffer unit 302 via the output terminals 320a and 320b, the present embodiment operates the resonant frequency of the resonant circuit 4〇1 at the oscillation frequency of the RF signal. Since the operation of the spectral circuit 401 corresponds to a high impedance at the resonant frequency, the operation is equivalent to a low impedance outside the resonant frequency. Therefore, the amplified RF signal will be directed to the current buffer unit 302. Moreover, the oscillation frequency of the multi-phase local oscillation signal of the subharmonic mixer is 〇5 times of the oscillation frequency of the RF signal. Therefore, the leakage signal generated by the multi-phase local oscillation signal via the tunneling effect at the input terminal 3i〇a 1324449 UMCD-2006-0025 19504twf.doc/g and 310b or the even-time generated by the interference signal via the 祚 linear circuit The order distortion 'will also be directed by the resonant circuit 401 to the first voltage. The subsequent current buffer unit 302 receives the amplified RF signal using the sources of the P-type transistors M41 and MP42. Inductors L44 and L45 provide a low impedance path to form a DC bias current for P-type transistors MP41 and MP42. At this time, the P-type transistors MP41 and -MP42' connected to the common gate configuration contribute to the isolation of the current buffer unit 302, and the single gain φ amplifies the signal output by the differential amplifying unit 301. In order to direct the RF signal amplified by the P-type transistors MP41 and MP42 to the switching unit 303 'where the resonant circuit 4〇2 takes the same approach as the resonant circuit 4〇1', the resonant frequency is operated at the oscillation frequency of the RF signal, While the output signal of the current buffer το 302 is directed to the switching unit 3〇3, the resonant circuit 402 can also display the leakage signal of the multi-phase local oscillation signal on the input terminals 310a and 310b via the tunneling effect, or the interference signal. The even order distortion generated by the nonlinear circuit is directed to the second voltage. Finally, the local oscillation signals LO4__00, L04-90〇, LO4-1800, and L04-270〇 included in the multi-phase local oscillation signal are respectively passed through the gates of the n-type transistors MN43 and MN49 in the switching unit 303, MN45. The gates of MN48, the gates of MN44 and MN410, and the gates of MN46 and MN47 are received. At this time, the operational characteristics correspond to the N-type transistors MN43 to MN41 of the switch, and the output signals of the current buffer unit 302 are switched to the local oscillation signals L〇4_00, LO4-900, L04-1800, and LO4-2700. Baseband signal. Among them, local oscillation signal 13 1^24449 UMCD-2006-0025 I9504twf.d〇c/g No. L04J) 〇, L〇4_9〇o, L〇4J8〇0, and L〇4-27〇◦ phase shift amount respectively It is 0 degrees, 90 degrees, 180 degrees, and 27 degrees. FIG. 5 to FIG. 8 are schematic diagrams of the implementation of the current CM〇s process technology in the present embodiment: !: measurement. This embodiment operates at a voltage VC. The oscillation frequency of the 1V, RF ^ signal is 5.2GHz, and the oscillation frequency of the multi-phase local signal is 2.6GHz. As shown in Fig. 5, the noise figure of the subharmonic mixer 3 at the frequency '1 〇 MHz is 17_3 dB. And as shown in FIG. 6 _, the subharmonic mixer 300 measures a leakage signal with a power of -65.154 dBm at the input terminals 310a and 310b under a multi-phase local signal having a received power of 15.5 (1 Β πι.) The output power of the subharmonic mixer 300 is about _i 〇〇 .7 dBm DC offset, but the DC offset does not exceed the noise floor specified by the wireless local area network (WLAN) receiver. In other words, the DC offset formed by the subharmonic mixer 3〇〇 will be masked in the background noise without affecting the performance of the circuit itself. In addition, as shown in Figure 7. At the input terminals 310a and 310b of the subharmonic mixer 3, only a very small 2 times leakage signal (the power is only -l 〇 9.934 dBm) is measured, thereby being able to explain the present invention. At the proposed time, the input of the harmonic mixer has good isolation. In order to further understand the circuit performance of the present invention, FIG. 8 lists the present embodiment and the Institute of Electrical and Electronics Engineers (IEEE, Institute of Electrical). And Electronic
Engineers)於1998年固態電路會刊第33卷第Η期 (Solid-state Circuits，V0L.33，N0.12)中所發表的期刊（圖 8 中以期刊[1]表示）、2000年射頻暨無線會刊第219頁至第 222頁(RAWCON，pp.219-222)中所發表的期刊（圖8中以 14 1324449 UMCD-2006-0025 19504twf.doc/g 期刊[2]表示）、2004年微波與無線元件會刊第14卷第7 期（Microwave and Wireless Components Letters，VOL. 14, N0.7)中所發表的期刊（圖8中以期刊[3]表示）、以及2004 年固態電路會刊弟39卷第6期(Solid-state Circuits，VOL.39, NO.6)中所發表的期刊（圖8中以期刊[4]表示）之比較結 果。由圖8可顯示出本發明之次諧波混頻器具有良好的隔 絕度’以至於與現今期刊所發表的論文相比較下，本發明 鲁 不論是在輸入端三階交錯點（Input 3rd order intercept point ’ IIP3)、輸入主而一階交錯點(Input 2rd order intercept point ’ IIP2)、LOR(local oscillator rejection)、或是針對所 產生的洩漏訊號與直流偏移，都具有良好特性。其中圖8 中所附註的符號*表示該期刊所發表的次譜波混頻器具有 線性化電路，符號+表示該期刊所發表的次譜波混頻器假 設射頻訊號輸入端之隔絕度為50dB。 從另一觀點來看，圖9為依據本發明較佳實施例之下 轉換器結構示意圖，包括次諧波混頻器3〇〇與訊號產生器 _ 501。次諧波混頻器300則包括差動放大單元3〇1、電流緩 衝單元302、以及切換單元303。其中次諧波混頻器3〇〇 耦接至訊號產生器501。電流緩衝單元3〇2耦接至差動放 大單元301。切換單元303耦接至電流緩衝單元3〇2與訊 號產生器501。下轉換器在達到將射頻訊號轉換至基頻訊 號的過程中’包括先利用差動放大單元3〇1將所接收的射 頻訊號放大。之後，再將經由電流緩衝單元3〇2單增益放 大差動放大單元301之輸出訊號。藉此，讓切換單们 15 1324449 UMCD-2006-0025 I9504twf.doc/g 依據^產生器501所提供的多相位本地振盤訊號，將電 流緩衝單元302之輸出訊號轉換成基頻訊號。 夕„上述之訊號產生器501包括本地振盪器510與相位偏 ^态520。相位偏移器52〇串接在本地振盪器與切換 •^元303之間。其中本地振逢器51〇用以產生本地振盈訊 ^以便讓相位偏移器52G將所接㈣的本地振盛訊號， ^成數個不同相移量的本地㈣訊號，以輸出作為多相 • 地振盛訊就。至於圖9實施例中，次譜波混頻器300 之^㈣、電路架構、以及相關電路特性，則包含在圖 〜圖二貫施例中，在此就不多加敘述。 亓細二上本發明因採用差動放大單元與電流緩衝單 -吨:ί Ϊ構’讓錢波㈣器在湘第―譜振電路與第 導向至第―電愿或第二電壓之情 得必,ιΓ提兩次諧波混頻器之隔絕度。如此一來，使 接降頻、;混頻，相關電路’例如下繼^ 而大幅地提升本身電路性能，：喝， 而言，1所而陟尤/、疋針對直接降頻接收器 雖缺本 =1_移與偶讀失真將大财地降低。 ΡΡ ^ 么月已以較佳實施例揭露如上，然豆並非用 ‘ίπΓ明’任何熟習此技藝者，在不脫離本發明之俨神 範圍當視後附之申rt動與潤飾’因此本發明之保護 【圖二二申_範圍所界定者為準。 圖1為傳統次财混_之結構示意圖。 1324449 UMCD-2006-0025 19504twf.doc/g 圖2為傳統吉伯特混頻裔之結構不意圖。 圖3為根據本發明一較佳實施例之次諧波混頻器結構 示意圖。 圖4為根據本發明較佳實施例之次諧波混頻器詳細電 路圖。 圖5〜圖7為用以說明圖4實施例電路特性之實際量測 結果。 圖8為圖4實施例與現今期刊之相關特性比較表。 圖9為根據本發明較佳實施例之下轉換器結構示意 圖。 【主要元件符號說明】 301 :差動放大單元 302 :電流緩衝單元 303 :切換單元 310、350a與350b :差動輸入端 320、330、340 :差動輸出端 310a 與 310b、350aa 與 350ab、350ba 與 350bb :輸入 端 320a 與 320b、330a 與 330b、340a 與 340b、：輸出端 401、402 :諧振電路 MN1 〜MN12、MN41 〜MN410 : N 型電晶體 MP41與MP42 : P型電晶體 R11 與 R12、R41 〜R44 :電阻 C41〜C48 :電容 L41〜L45 :電感 17Engineers) Journal published in Solid-state Circuits (V0L.33, N0.12) in the 1998 issue of Solid State Circuits (indicated by the journal [1] in Figure 8), 2000 RF Journal published in pp. 219-222 (RAWCON, pp. 219-222) of the Wireless Journal (in Figure 8, 14 1324449 UMCD-2006-0025 19504twf.doc/g Journal [2]), 2004 Journal published in Microwave and Wireless Components Letters (VOL. 14, N0.7) (in Journal 8 [3]), and in 2004 Solid State Circuits The results of the journals published in Solid-state Circuits (VOL.39, NO.6) (indicated by the journal [4] in Figure 8). It can be seen from Fig. 8 that the subharmonic mixer of the present invention has good isolation', so that compared with the paper published in the current journal, the present invention is in the third-order interlacing point on the input side (Input 3rd order). Intercept point ' IIP3), Input 2rd order intercept point 'IIP2', LOR (local oscillator rejection), or for the generated leakage signal and DC offset, have good characteristics. The symbol * in the note of Figure 8 indicates that the sub-spectral mixer published in the journal has a linearization circuit, and the symbol + indicates that the sub-spectral mixer published in the journal assumes that the isolation of the RF signal input is 50 dB. . From another point of view, Figure 9 is a schematic diagram of the structure of the converter in accordance with a preferred embodiment of the present invention, including a subharmonic mixer 3 讯 and a signal generator _ 501. The subharmonic mixer 300 includes a differential amplifying unit 301, a current buffer unit 302, and a switching unit 303. The subharmonic mixer 3〇〇 is coupled to the signal generator 501. The current buffer unit 〇2 is coupled to the differential amplification unit 301. The switching unit 303 is coupled to the current buffer unit 3〇2 and the signal generator 501. The down converter, in the process of converting the RF signal to the baseband signal, includes first amplifying the received RF signal by using the differential amplifying unit 3〇1. Thereafter, the output signal of the differential amplifying unit 301 is amplified by the single gain of the current buffer unit 3〇2. In this way, the switch unit 15 1324449 UMCD-2006-0025 I9504twf.doc/g converts the output signal of the current buffer unit 302 into a baseband signal according to the multi-phase local vibration disk signal provided by the generator 501. The above signal generator 501 includes a local oscillator 510 and a phase offset 520. The phase shifter 52 is connected in series between the local oscillator and the switching element 303. The local oscillator 51 is used. A local vibration signal is generated so that the phase shifter 52G converts the local (4) local vibration signal into a plurality of local (four) signals of different phase shifts, and outputs the signal as a multiphase/ground vibration signal. As shown in FIG. In the embodiment, the (4), circuit architecture, and related circuit characteristics of the sub-spectral wave mixer 300 are included in the embodiment of the figure to the second embodiment, and are not described here. Differential Amplifier Unit and Current Buffer Single-Ton: Ϊ Ϊ 让 'Let Qian Bo (4) in Xiangdi - spectrum circuit and the first to the first - electric or second voltage must be, Γ Γ two harmonics The isolation of the mixer. In this way, the connection frequency reduction, the mixing, the related circuit 'for example, the next step ^ greatly improve the performance of the circuit itself: drink, in terms of 1 and then / / 疋Although the direct down-conversion receiver lacks this = 1_shift and even-read distortion will reduce the wealth. ΡΡ ^ 么月 has The preferred embodiment discloses the above, but the bean is not used by any of the skilled artisans, and the invention is protected by the invention without departing from the scope of the invention. The scope defined by _ scope shall prevail. Figure 1 is a schematic diagram of the structure of the traditional sub-mixed _ 1324449 UMCD-2006-0025 19504twf.doc/g Figure 2 is a schematic diagram of the structure of the traditional Gilbert mixer. Figure 3 is based on FIG. 4 is a detailed circuit diagram of a subharmonic mixer according to a preferred embodiment of the present invention. FIG. 5 to FIG. 7 are diagrams for explaining the embodiment of FIG. Actual measurement results of circuit characteristics Figure 8 is a comparison table of related characteristics of the embodiment of Figure 4 and current journals. Figure 9 is a schematic diagram of the structure of the converter according to a preferred embodiment of the present invention. [Key element symbol description] 301: Poor Dynamic amplifying unit 302: current buffering unit 303: switching units 310, 350a and 350b: differential input terminals 320, 330, 340: differential output terminals 310a and 310b, 350aa and 350ab, 350ba and 350bb: input terminals 320a and 320b, 330a and 330b, 340a 340b, output terminals 401, 402: resonant circuits MN1 to MN12, MN41 to MN410: N-type transistors MP41 and MP42: P-type transistors R11 and R12, R41 to R44: resistors C41 to C48: capacitors L41 to L45: inductance 17