TWI321392B - A high-efficiency current-source inverter using resonant technique - Google Patents

Description

1321392 IX. INSTRUCTIONS: [Technical Fields of the Invention] The technical field to which the present invention relates includes power electronics, DC/AC conversion technology, and green energy technology, but mainly uses DC power sources (such as fuel cells, wind power, Solar energy, etc.) is used as an input power source to convert it to an AC power source to provide emergency power or general AC electrical load. The device controls the on-time of the clamp switch in each cycle by the error of the AC sinusoidal voltage command and the feedback voltage. The current source is generated by the coupled inductor, and the output capacitor is charged by the full-bridge switching positive and negative cycle guidance, and the voltage rise or fall is adjusted to accumulate the AC sine wave output voltage. [Prior Art] At present, the products that convert DC power into 60Hz AC voltage on the market are roughly divided into two categories; the first type is an inverter applied to an AC motor, which uses a coil inductance characteristic of the motor to generate a sinusoidal pulse width modulation voltage waveform. Approximate sinusoidal current, this architecture cannot be applied to resistive or capacitive loads. Strictly speaking, inverters cannot supply general household appliances and computer products. The second category is the product launched for the shortcomings of the former. The typical product is the uninterruptible power equipment (UPS) as its structural diagram is shown in Figure 1 (a). Compared with the former class, the output increases the LC filter circuit of the inductor series capacitor, and increases the feedback control to make the output voltage fixed to overcome the load and input voltage variation. The biggest advantage of this type of product is that the architecture is simple and the cost is low, and the battery and Charge and discharge circuits to provide emergency power supplies other than utility power. At present, Taiwan's UPS 1321392 - technology and product share, such as Delta Electronics and Feirui, has been ranked among the best in the world. Even so, there are still a number of features that still need to continue to improve. First, the LC filter circuit has many load limitations: the first point filter inductor runs through the entire output current, and considers the half-power frequency (_3dB) response in the offset second-order resonant circuit, the inductance value and the capacity are large, and the inductor used in the commercially available UPS is generally used. The value is approximately in the w/f range, and increased filter inductance increases product weight and energy conversion losses. The second 'point, applied to the voltage across the inductor I · along / 汾 is the difference between the DC voltage and the output capacitance, the value is the smallest near the sinusoidal peak, the limited filter inductance value, resulting in the positive Φ string voltage peak turning point distortion, resulting High harmonic components, even if the filter voltage is increased, cannot be avoided. This inductor provides filtering, but at the same time limits the ability to adjust for non-linear loads and transient load changes. Third, some loads will endanger the drive circuit, such as half-wave rectifying load or high-inductive load, mainly due to the positive and negative half-cycle waveform symmetry of the LC filter circuit, and the high-inductive load changing the frequency response of the second-order filter, output sine If the voltage is too low, the DC voltage level must be increased and the system may be over-pressed and burned. Fourth, the voltage waveform distortion rate of non-resistive loads is also commonly referred to as Total Harmonic Distortion (THD), which is much higher than resistive load. This is due to the fact that the previously designed second-order filter circuit characteristics cannot satisfy non-resistive loads such as capacitive, inductive and non-linear loads. All in all, the current products of the inverter cannot fully provide various types of loads. It is not only inconvenient for the consumers, but also the users can distinguish the types of loads. Only the luck can be used to buy the uninterrupted equipment. Is a feasible way. In addition, the switching loss increases as the switching frequency increases, 1321392 - system efficiency is thus reduced. Many manufacturers have begun to cite various flexible switching techniques. The technology is applied to high-power IGBT switching components, and several documents have been shown to reduce PWM switching losses, increase switching frequency, and improve output voltage waveform. Compared to the traditional sinusoidal pulse width modulation voltage waveform, the sinusoidal voltage of the current source inverter, most of the control current source charges the capacitor to accumulate the sinusoidal voltage, can withstand various load and frequency changes, but why the market is less There is such a product, the reason for the analysis is that the inductance of the current source is too large, and the control inductance loop and the flexible switching technology are not easy to realize, so the efficiency is not high. The resonant circuit with flexible switching is a frequently cited technology. The voltage value of the voltage resonance type and the voltage value of the current resonance are prone to abnormal conditions during the resonance process. Once the voltage exceeds the rated value, the semiconductor switch is damaged immediately, so the development The voltage gain is not affected by the load of the resonant circuit [1], [2]. However, since the energy utilization rate of the spectral vibration circuit is too low, nearly half of the energy will flow back to the power supply measurement, so follow-up experts have proposed a semi-resonant voltage source inverter with a voltage clamping circuit to reduce the switching stress and use the pulse width modulation. Variable technology to control the inverter output waveform P]-[5]. When the switch is turned on, it needs to be matched with the semi-resonant period. Therefore, some resonant circuits [6], [7] are developed, and the resonant waveform is limited to the switch on or off crossover interval. Therefore, the switching frequency can be fixed. The period does not have to match the resonance time. Recently, some scholars have studied to place the filter inductor between the DC power supply side and the full-bridge inverter. As shown in Figure 1(b), the inductor value is reduced and the dynamic response capability is improved to form a current source inverter structure. Improve the shortcomings of traditional voltage source architectures. However, the switching voltage value generated by the current cutoff is twice as large as the system voltage, which jeopardizes the switching components. Therefore, the withstand voltage value of the switching component must be relatively increased, and the cost is greatly increased, which is not conducive to the architecture of the technology_7_Application . Recently, a current source flexible switching architecture [8] has been proposed to solve this problem. This is applied to the solar power system to feed the commercial power. The reverse current has zero voltage switching (Zer.v. 1 mail dream: Zero Current Switching) , ZCS) effect, reducing the power factor of the city is close to one, but the volume and quantity of the inductor still have to be set to a low frequency (5 kHz), resulting in a high chopping output voltage. However, the f is set to direct feed. The mains supply is not directly supplied to the load. The power supply for this system is better than the current voltage source reverse flow. 2 Continued to propose a voltage clamp and flexible switching dual-effect inverter machine. The three-phase AC motor driver uses the secondary side of the transformer to make the current, the overall inductance value is reduced and the conversion efficiency is high, the base is large, and the current source architecture is characterized. However, the current source inductance loop is too concentrated: C:, not easy to 'theory On the transformer copper loss will be very high, but the loss of the transformer q. In addition to the voltage waveform ripple high, this architecture::: point is the switch has to withstand four times the DC power supply voltage, the switch cost over the drive Dongzi Xiang for the induction horse It is not conducive to the application of consumer products. Bu Xin developed a sinusoidal voltage inverter with low-frequency high-order resonance [10]. This kind of wood structure is not simple, and it uses the characteristic of geometric mean frequency ((3) then this river (10) Frequency) In the open loop state, the voltage regulation rate has a good performance, but (4) the vibration inductance capacity is too large and the three-cut wave component is too high is the lack of electric current clamped and flexible switching efficiency current source sinusoidal voltage inverter [11], Π 2 The low-inductive inductive transformer—the secondary side acts as the current source inductor, which reduces the voltage of the switch and the quadruple DC power supply by a factor of two, and the four bridges of the full bridge are switched to the voltage and zero current. However, the four The switch must be 1321392 to cause the wheel to be powered out; the short circuit = is the period of the reverse phase switching ' _ 冓 全 full bridge switch series diode blocking = same ::: line: _ car performance, this is the majority == muscle Secondly, the current source needs to reduce the amount of electricity, but to ensure that the clamp switch is turned on when electricity is turned on; ::: for 1, the chopping current is high, and because of the switching frequency 4 = plus; 'Order. After taking' is a sandwich that greatly increases the face factor pressure, and = difficult Winding method, and the secondary side winding electricity;:= Two-point output electric grinding one-third, this high circulating current is not introduced into the output end, resulting in light load conversion efficiency is particularly low. =:=r on the limit system _, and := Load direct-to-frequency switching charge control, cumulative sinusoidal power (4) tut all power semiconductors_components with zero- or zero-current flexible switching characteristics, thereby improving the output efficiency of the inverter. The present invention improves the principle and comparison of the prior art The functions are as follows: · The electric clamp technology, the inductive current energy transfer and the parallel resonance technology make the power switching switch and the diode have flexible switching characteristics, and the highest conversion efficiency is greater than the coffee; 2. Using the electric capping technology of the present invention, the withstand voltage specification of all power semiconductor switch components can be reduced, and the coffee is approximately equal to or less than the input DC power supply 1321392 voltage, which is much lower than the above document. 3. Regenerative brake function: Since the full bridge switch of the present invention does not require a series diode and provides an uncontrolled brake diode circuit, the AC energy can be reversed to the DC power supply terminal, further clamping all systems. The voltage of the component. 4. The coupling inductor capacity and volume required by the present invention are smaller than the general current source architecture, and the coupled inductor can operate in a continuous current mode and can quickly adjust the current to supply the load, and the switching frequency can help reduce the capacity of the output capacitor. . When the power semiconductor switch trigger signal is turned on, if the secondary side winding current has dropped to zero before the trigger signal is turned on, it is a discontinuous current mode (DCM), so naturally forming a conduction has a ZCS phenomenon; similarly, In the case of Continuous Current Mode (CCM), the limited coupling inductance energy transfer effect still forms conduction with ZCS characteristics. 5. The power semiconductor switch of the full-bridge inverter can omit the four serial two-body of the conventional architecture, and use the parallel resonance characteristic to handle the polarity crossover of the output voltage. It does not need to pass any switching elements, so it can be omitted. Conventional current source inverter commonly used series diode; 6. The output terminal can omit the conventional high-capacity series filter inductor, the current source directly charges the output load and the filter capacitor, so it can accept various inductive, capacitive, nonlinear And the transient load, and the output voltage waveform distortion rate and Fourier spectrum analysis are better than the traditional pulse width modulation architecture. Remarks: References [1] FJ Lin, RY Duan, and JC Yu, aAn ultrasonic motor drive 12 1321392 using a current-source parallel-resonant inverter with energy feedback, Trans. Power Electron., vol. 14, no. Pp. 31-42, 1999.

[2] FJ Lin, RY Duan, RJ Wai and CM Hong, 4iLLCC resonant inverter for piezo-electric ultrasonic motor drive, 5, IEE Proc. Electric Power Appl, vol. 146, no. 5, pp. 479-487, 1999 .

[3] L. Malesani, P. Tenti, P. Tomasin, and V· Toigo, “High efficiency quasi-resonant DC link three-phase power inverter for full-range FWM, ^ IEEE Trans. Ind. Αρρί, vol. 31 , pp. 141-147, 1995.

[4] V. V. Deshpande, and S. R. Doradla, “A new topology for parallel resonant DC link with reduced peak voltage, IEEE Trans. Ind.

Appl., vol. 32, pp. 310-307, 1996.

[5] S. Chen, and T. A. Lipo, 44A novel soft-switched PWM inverter for AC motor drivers, IEEE Trans. Power Electron. ^ vol. 11, pp. 653-659, 1996.

[6] P. C. Theron, and J. A. Ferreira, “The zero voltage switching partial series resonant converter, ^ IEEE Trans. Ind. Αρρί, vol 31, pp. 879-886, 1995.

[7] C. S. Moo, YC Chuang, and CR Lee, “A new power - factor -correction circuit for electronic ballasts with series-load resonant inverter, 5, IEEE Trans. Power Electron., vol. 13, pp. 273-278, 1998.

[8] R. Itoh, K. Ishizaka, H. Oishi and H. Okada, “Soft-switched current-source inverter for single-phase utility interfaces,” 13 1321392

Electron. Lett., vol. 37, pp. 1208-1209, 2001.

[9] H. Ishikawa, and Y· Murai, “A novel soft-switched PWM current source inverter with voltage clamped circuit/5 IEEE Trans. Power Electron., vol. 15, no. 6, pp. 1081-1087, 2000 .

[10] RJ Wai, RY Duan, JD Lee, and LW Liu, "High-efficiency fuel cell power inverter with soft-switching resonant technique, 5, IEEE Transactions on Energy Conversion, vol. 20, no. 2, pp. 485 -492, 2005.

[11] R. J. Wai and R. Y. Duan, "High-efficiency power conversion for low power fuel cell generation system, IEEE Trans. Power Electronics, vol. 20, no. 4, pp. 847-856, 2005.

[12] RJ Wai, RY Duan, and LW Liu, UA current-source sine wave voltage driving circuit via voltage-clamping and soft-switching techniques, ROC Conference on Electrical Power Engineering, PartC-3, pp. 749-753, 2003 .

[13] RJ Wai and RY Duan, “High-efficiency DC/DC converter with high voltage gain,” IEE Proc. Electric Power Applications, vol. 152, no. 4, pp. 793-802, July 2005. As shown in FIG. 2, the high-efficiency resonant current source inverter disclosed in the present invention includes a DC power supply 1〇1 which is slightly higher than the AC peak; the clamp circuit 102: is composed of two clamp switches η, Qiao , two clamping diodes Di, A 'one brake diode / 3⁄4 and a clamp capacitor & a coupling circuit i〇3, consisting of _He Lei Xian-rectifier diode sc and: secondary side Winding ~ and full bridge door closing + 反 inverting circuit 104: by four groups - drive circuit '; Γ; ί (10) ^ drive signal. When the negative feedback control and output of the six switches are half-cycled, the current is supplied by the DC power supply (8) and the inductance Z; the K / ", Γ 2, and the full-bridge switching of the light-synchronous circuit 103 are transmitted by the spectrum. The circuit 104 is sinusoidal negative half cycle output === electricity. Similarly, the voltage spectrum inverting circuit 104 is to be generated: the box switch of the input circuit 102 and the output capacitor Q are reversely charged and simultaneously triggered to conduct, and the high frequency switching current source circuit is the ancient crying. κ, coupling The primary winding 々 of the D circuit 103 is between the voltage core of 01, the voltage of the clamp capacitor c of the clamp circuit 102, and the output capacitance of the spectral inverter circuit 1〇4. It is necessary to use the current source as the medium to achieve the effect of the output-= around the leakage inductance, to achieve the full power semi-conductivity = electricity (four) system; spectrum oscillator circuit 1 (^ four = switch η Responsible for guiding the positive half cycle of the current source current path 1 guide Wangqiaoguan 7Γ and 7Γ is the negative half cycle path; the charm inverter circuit W vibration inductance 4 and the output capacitor Q form - group second-order parallel inductor current behind Electrical characteristics, by Wei inductor U output capacitor ^ 1321392 voltage t polarity commutation. Invented "high efficiency (10) vibrating wire reactor" waveform timing. Circuit operation mode, as shown in Figure 3 and Figure 4, respectively In the following, the following two principles will be used to describe the process of red, and in order to simplify the circuit analysis, the voltage drop of all the switches and diodes is ignored. In addition: =) Solution: The slightly proper noun is not too long. The circuit belongs to the figure number “口...=路101). If it is omitted, it can be clearly defined by the following description: Mode 1: Time (〇1) Clamp switch 7], r2 is turned on for a while, and switch 7 is turned on. After the period - (10) (four) electrically coupled inductors, the secondary winding 4 is known, from the straight-electric U's through the clamp Close the coincidence, the capacitor c. And the inductive state of the primary side of the inductor 7 ~ 'into the spectral inverting circuit of the off state, so that the pole:: two = two can be expressed as ^ is the timing, then the primary winding core Trans-voltage νω

Ld · did !dt = vLd = vm + vco - v . ° (1) it is the input ΐ DC power supply voltage, the output AC voltage, the equal-number current valley again:, - the secondary winding, so the current marriage is not too High, can: side, and effectively reduce the conduction cycle of the system switch = loss. In addition, the 'clamp switch' is called VTl - Vco - VD2 (2.a) 16; T2 = Vc〇~ VD, ; T2 = Vc〇~ VD, (2.b) To push the diode A and & the cross-pressure ~ and % as follows (3.a) (3.b) (4) VD\ = ν〇〇- VT2 VD2 = vc〇- Vr, output AC voltage V . It can be calculated as V〇 - it^d ~ Ll ~ i. ● dt is the AC load current. In this mode, it is the spectral inductance 々 current and clamp capacitor C. Voltage'. Discharge to zero: special mode ^: time "~~" clamp diode ΑA lead 35 '瞀 C ° discharge to near zero volts, according to equation (3.a) and formula (3.b, = diode The end-span voltage is reduced from zero to volts by reverse bias, and then the two voltages of the two clamps are switched to form a turn-on voltage. At this time, the trigger signals of the switch 7] and Γ2 are still j-channel and then parallel with each other, the average is 82 : 2: J f off series diodes. 10 Qiao knife to the output capacitor Q charging electric core type 2. Day 〇 〇 2 ~ '3) clamp switch; and ^ trigger signal cutoff = clamp open, and Mingfa After the signal is cut off, the full bridge switch (and & still temple, pass, coupled inductor primary winding, limited leakage inductance freewheeling factor, its diameter converter material, C. and ... charge the clamp capacitor & output capacitor Because 'so when clamping capacitor c. When the voltage injury rises: only a slight rise. According to equations (2.a) and (2 b), the clamp-on voltage is equal to the clamp diode turn-on voltage plus capacitor ~ When the voltage rises gradually, it means that the switch is turned off and the voltage is switched, and the energy of the charge is clamped. Cycle mode fM. Ll transport, .,. Output capacitor Q, clamp capacitor c. Electric ^ can be expressed as (5)

Jd-dt t2<t<t4 At this time, the cross-voltage νω of the primary winding core can be expressed as vLd=Vm-vco-v〇_ (6) According to the above formula, the light-inductive inductance-secondary winding & At the point, the positive and negative voltages are induced to the secondary winding. In the same situation, the rectifying diode is still reversed due to the rotation of the electric motor, and the secondary winding has no current path. Mode 4: Time (?3~ phantom-coupled inductor secondary current begins to conduct. The local mode starts from the clamp capacitor Ce voltage %. When calculating according to equation (6), the voltage polarity of the primary winding core of the coupled inductor begins to reverse. The positive voltage at the non-polar point and the inductance to the secondary winding are also the same, so the rectifying diode of the coupling circuit is in a forward-biased state. According to the law of flux inactivation, the secondary winding L/ begins to generate current. During this period, the secondary side winding ^ current flows along with the primary side winding core into the resonant inverter circuit, and its relationship can be expressed as L(^d(max) - Ldid + / Π\ where '~(stomach &gt The highest value of the current G 'both currents ^ and 匕 one long, the current mode stops at the current & fall to zero, and the current reaches the highest point. Due to the clamping capacitor C, the leakage energy can be fully absorbed, and the magnetic Passing the secondary winding and releasing 'leakage inductance' has little effect on the system. To simplify the analysis, the leakage inductance energy is not considered for the time being. [13] Let the light-combined inductance first and second winding turns are respectively 1321392 and then The turns ratio π can be expressed as %仏(8) During the mode, the secondary winding voltage -V£/ is equal to V. The winding Ld voltage vu is reversible on the primary side v (9). According to the Khrhew voltage law, the equation of this mode can be expressed as follows VIN ~Vco+ — ~ Vo=() η (10) Finishing the above formula gives Vco = 1^ΪΝ +( l)V η (11) Since V is the AC voltage, add the absolute value to simplify the analysis ^co=VIN+{Xln-l)\ Vo\ (12) The second term of the upper equal sign is a negative number. Since the clamp capacitor voltage Ve() is across the clamp switch, the highest voltage value occurs when the output AC voltage is zero. • The switch withstand voltage can be obtained. Specifications are

Vn(max) = V7'2(max) = ^1N (13) Therefore, the clamp switch of this architecture has the same withstand voltage as the input voltage, and the same bridge has four switches 7; +, 77, 7; + and 7; Because the output is a capacitor and has the voltage clamping performance of the flywheel diode, the switch withstand voltage and the output AC voltage V. the same. Mode 5: Time 〇4~〇current 9 starts to fall from the highest point. When the primary side winding k current G drops to zero, the coupled inductor energy is transmitted through the secondary side winding current & to the output capacitor 仏, so the current is the most 19 1321392 - The high point begins to fall. This mode ends when the trigger switch 7] and Γ2 trigger signal are turned on. Mode 6: Time 〇 5~?. The clamp switch 7;, Γ 2 trigger signal is turned on at this time, the clamp switch 7; Γ 2 trigger signal is turned on, if the secondary side winding current b has dropped to zero before the start of the mode, then the current discontinuous mode (Discontinue Current Mode, DCM), so naturally forming conduction has a ZCS phenomenon. Similarly, if the current continuous mode (CCM), the limited primary winding & leakage inductance and the energy transfer of the two sides of the winding, like the same electrical characteristics in mode four, so still form conduction φ with ZCS characteristics . Further, in the resonant inverter portion, the current Q of the resonant inductor A is delayed by an electrical angle of about 90 degrees from the output AC voltage, so when the AC voltage V is output. When the voltage L is near the highest value, the current L is the highest value, and the energy of the output capacitor q can be extracted to complete the zero-crossing commutation. This process does not need to be switched and controlled, so the series diode commonly used in general current sources can be omitted. At the same time, in order to avoid the high gain voltage of the second-order resonance occurring at no load, the resonant frequency should avoid 60Hz. In order to ensure the success of the commutation, to avoid omitting the short circuit current of the flywheel path caused by the series diode, the resonant inductor current energy must be higher than the output capacitor potential energy, which can be expressed as ^l}LL(mdix.) ^ ^o (max) (1 4) The path of the series diode of the full-bridge switch of the general current source architecture to block the load feedback to the DC power supply must be used as an AC resistance brake circuit. When the output voltage is higher than the DC power supply voltage, the brake diode _ body 1) 3 of the present invention naturally turns on and provides an energy feedback path, which occurs when the load * is the flywheel inertia of the AC motor, and the inductive load changes the resonant frequency. 20 1321392 'The smart voltage is on the side; (4) The car diode draws the energy of the 至 to the DC power supply voltage. • Absorbs all the switching voltages except the box. The regenerative brake function effectively utilizes the energy of the source and energy to improve energy efficiency. . As can be seen from the above description, most of the switching diodes and switches have the ZCs and zvs characteristics when turned on and off (4). Therefore, in theoretical analysis, the present invention can achieve high conversion efficiency. Although the present invention has been disclosed in the foregoing preferred embodiments, it is not intended to limit the invention to those skilled in the art, and various changes and modifications may be made without departing from the spirit and scope of the invention. The scope of the invention is defined by the scope of the appended claims. [Embodiment] All the power semiconductor switches of the main components of the present invention are selected from MOSFETs, and the number is the signal conduction resistance / ^ (four) ^ ^ ^ ' withstand voltage 250V and rated current 44A, and the package form is TO-247. According to the equation (13), the highest switching voltage specification can be obtained. The purpose of the embodiment of the present invention is to control the output voltage to a peak value of 156V, convert it into an effective value of ll〇v, and set a rated output specification of AC 110V60Hz 500W. The other parameter design and component selection of the present invention are as follows: DC input voltage: 170V AC output voltage: AC ll〇V 60Hz Switching frequency: 50kHz Coupling inductor 7; primary winding core and secondary winding & : 100 W and 200 you, primary and secondary winding turns % and %: 8.5 and 12 匝, core-EE55, 21 1321392 - shank combination coefficient k = 0.98. Clamp switch 7],: Γ 2 and full bridge switch 7; +, Γ; , 7; +, 77 : POWER MOSFET IRFP264, F ^ = 250F, /? 颂 tear) = 60mQ, / D = 44d, TO-247 output capacitor Q and resonant inductor 4: 6.8W and 1.034 / /

Clamping Capacitor Q: 0.04wF Clamping Diode A, /) 2 and QC: SFA1604G and SFA1608G Inverter Resonant Frequency: 60Hz Figure 5 shows the high-efficiency resonant current source inverter φ disclosed by the present invention. The measured waveform diagram, Figure 5 (a) is the output voltage and coupled inductor 7 when no load; the secondary side winding current &, the output voltage is accumulated by the frequency 50kHz high frequency switching according to the PI proportional integral method, The chopping voltage of the trace high frequency component has been improved, and the current G is small. Output voltage V. The area near the zero value; since the commutation energy is completely provided by the resonant inductor A, the current & is almost zero, which proves that the high efficiency resonant current source of the present invention has an extremely low energy loss when no load is present. Figure 5(b) shows the waveform of the output voltage and the resonant inductor current L. The resonant inductor 4 has an inductance of 1.034//, and the current Q peak value is 0.39A. The capacitance is 30.5VAR. Although the inductance is high, the actual value is actually The volume is quite small. In Figure 5(c), the output capacitor C with an output power of 400W and the terminal voltage V. And the primary side winding Zy current. In order to simplify the analysis of the average value of the primary winding current Q at steady state, the clamp capacitor voltage % is not considered at this time, according to the Khreshofer voltage and current law.

Vo=VLL=LL-diLL/dt = VCL (15) " and U ~ ^CL + hL + Zo (16) 22 1321392 The output voltage νβ is AC 110V 60Hz, so v0 can be expressed as 156sin377i' flowing through the load /? (assumed to be 3 Ω), the output capacitor Q and the resonant inductor 4 currents tb and L can be expressed as;

156sin377r R

5.2sin377/A (17)

156sin377i XCL

=Q · 156 cos377i = 0.39 sin (377/ + 90°) A z.cx =

156sin377i

X

= 0.39 sin (377 卜 90.) A (18) (19)

LL φ, so the output voltage is near the zero value, the current flowing through the load is zero, the current G and Q are the same but the phase difference is 180 degrees, so the voltage commutation energy of the output capacitor (^ can be completely provided by the resonant inductor A, Therefore, it is not necessary to provide the current path required for commutation by the input. The current G of the primary winding A/ of the coupled inductor is almost zero, and only the energy loss of the resonance process needs to be supplied. After the commutation is completed, the switch 7 is clamped; During the turn-on of the trigger signal, according to equation (18), the charging current of the output capacitor Q is proportional to the cos function, so the current core is the highest; in addition, the output voltage v. The peak value only needs to supply the current of the load ,, The current G is related to the light weight of the load. In Figure 5(d), the full bridge switch 7; the voltage να+ and the current ί of the +, due to the 60Hz low frequency switching of the full bridge switch, the current G of the full bridge switch is turned on and off, the switch The two terminals exhibit zero voltage state, no switching loss at all, and the full bridge switch output is a capacitor, and the circuit provided by the flywheel diode makes the switch withstand voltage clamped the same as the output AC voltage V. Figure 5 (e ) is the voltage of the clamp switch 7; the voltage νη and the current ζη, the figure shows that the clamp switch voltage % has a voltage clamping characteristic when it is cut off. Since the clamped capacitor is electrically connected and connected across the clamp switch, the highest voltage value occurs at the output. AC 23 1321392 - When the voltage is zero, the voltage required by the waveform display switch is consistent with the derivation, program (13). As shown in the actual waveform of Figure 5, the resonant inverter circuit has effectively controlled the output voltage V. The zero-crossing waveform, and almost no current flows through all the switches in the vicinity of the crossover region. The high-efficiency resonant current source inverter of the present invention is clamped to the same voltage as the input voltage of all power semiconductor switches. The voltage clamping performance is equal to or lower than the power supply voltage, and the voltage clamping performance is better than that of the conventional inverter [11]. Fig. 6 is a diagram showing the measured waveform of the high-frequency operation of each component of the high-efficiency resonant current source inverter φ disclosed by the present invention, wherein 6(a) shows the coupled inductor 7 operating in continuous mode; —, the secondary side winding currents 6 and 9 crossover waveforms, limited leakage inductance on both sides of the winding, when clamped When the voltage is calculated, according to equation (6), the coupled inductor 7; the voltage polarity of the secondary winding ~ begins to be inverted, and the positive voltage is at the non-polar point, and the secondary winding & is also the same, so the coupling circuit The rectifying diode D/ is in a forward bias state. According to the law of magnetic flux immortality, the secondary winding & begins to generate current, during which the secondary winding & current & follows the primary winding 4 current core • With the inflowing resonant inverter circuit, the relationship between the two winding currents changes according to equation (7). The crossover changes, as shown in Figure 6(b), the primary winding current rises and the secondary winding & current Z/descent, and Fig. 6(c) shows the primary side winding current drop and the secondary side winding & current ^ rise. Figure 6(d) shows the voltage vD/, current & waveform of the rectified diode. The waveform of the rectifier diode V# waveform has a flexible switching effect when it is turned on and off. There is zero voltage and zero current switching; in addition, the voltage of the rectified diode and the surge voltage generated by the high voltage oscillation are reduced to 560 volts by adding the cushioning circuit 24 1321392 (snubber) compared with the numerical simulation of ~660 volts. The voltage can be reduced by the vibration damping circuit to reduce the diode withstand voltage specifications. Figure 6 shows the voltage vn and current Z n waveform of the clamp switch 5, and the clamp switch 5 is connected to the ^ (E > flexible switching effect at the end of the cut-off plus a lower-rate semiconductor switch with a withstand voltage specification can improve the present Conversion efficiency of the invention. _ Figure 7 shows the high-efficiency spectral-type current source counter-cursor disclosed in the present invention. The voltage and current waveforms of the unloaded and various loads are measured and divided into two ^ Fourier frequency analysis to calculate the total Harmonic distortion rate (THD). Figure 7 (a) is no load when 'the fundamental frequency (6 〇 Hz) value is 40.6dB35 is the same as the AC voltage, the total wave distortion rate (and) is L 〇%. Figure 7(b) The load is a nonlinear rectifying load (h_, C = 47〇MF) 'where the fundamental frequency (6 〇 is 40.3 guilty 'total spectral distortion rate (D = 8%). Figure 7 (c) negative salt load (^50ω, ζ = 5〇μ〇, measured total harmonic distortion rate (THD) is ancient ^ 13.92%, higher than the international standard value of 5%. Figure 7 (d) is the figure % = The output voltage pattern after the carrier shape is improved. If the round-out capacitor & both ends are connected in parallel (10), the total spectral wave loss is fully supplied to the inductive load. Inductive Negative:: Raw Waveform Distortion == 7(e) is the inverter's instantaneous load 3 The voltage of this electro-plate load ν'Κ waveform diagram, according to the experiment shown in the instant: Output voltage ~ peak part only a little distortion According to the various wave of the liver of Fig. 7, the difficult current source current reflux device disclosed by the invention can be connected with negative inductance of inductive, capacitive, nonlinear and transient changes, = waveform_wave distortion rate and Fourier spectrum analysis is superior to the above-mentioned reference 0 25 1321392 - Figure 8 shows the high efficiency resonant current source inverter disclosed in the present invention. and the conversion efficiency diagram of reference [11], which shows the present invention. The high-efficiency resonant current source inverter has a conversion efficiency higher than 97%, which is superior to the reference [11] at light load and heavy load, especially at light load, using I-commutation commutation technology and secondary winding The 丄/current is all introduced into the output terminal, which greatly reduces the circulating current component of the switching current, so the conversion efficiency can be increased up to about 8%. The present invention implements a high-performance current sinusoidal voltage conversion circuit through a circuit, and the comprehensive features are as follows: φ 1 Using coupled inductors, secondary side energy The transfer mode controls the current source, so that all semiconductor switches and diodes have flexible switching characteristics, the maximum conversion efficiency is greater than 97%, and the total harmonic distortion (THD) of the inverter of the architecture is low after various load tests. 2% or less; 2. Using voltage clamping technology, the voltage specification of power semiconductor switching components can be reduced, and all switches are subjected to voltages equal to or less than the power supply voltage; 3. The coupled inductor capacity and volume are smaller than the general current source architecture, and its energy All transmitted to the output, no circulation problem; • 4. Coupled inductor operation in continuous current mode, can increase the switching frequency to reduce the sense of value, while the switch still has a flexible switching effect; the circuit is implemented as a component flexible switching characteristics summary table 1 is shown. 26 1321392

Table 1 Flexible switching characteristics of main components Zero voltage switching (zvs) Zero current switching (zcs) Component symbol turn-on turn-off turn-off Ά, τ2 〇〇〇Τ:, Ta-, Tb+, Tb- 〇〇〇〇D', D2 D3 〇〇〇〇Df 〇〇〇〇27

[Simple diagram of the diagram] shows the conventional sinusoidal voltage inverter architecture diagram. The block diagram of the high efficiency spectral oscillator current source inverter circuit disclosed in the present invention is shown. The waveform timing diagram of each point of the high efficiency resonant current source inverter disclosed in the present invention is shown. The working mode diagram of the high efficiency spectral oscillator current source inverter disclosed in the present invention is shown. The 60 Hz measured waveform of each component of the high efficiency resonant current source inverter disclosed in the present invention is shown. The measured waveforms of the high frequency operation of each component of the high efficiency resonant current source inverter disclosed in the present invention are shown. The measured waveform diagram of the efficiency spectrum oscillator current source inverter disclosed in the present invention is unloaded and various loads. The conversion efficiency diagram of the high efficiency resonant current source inverter disclosed in the present invention and reference [11] is shown. [Main component symbol description] 1〇ι: DC power supply 1〇2: Clamp circuit 103: Coupling circuit 104: Spectral inverter circuit 105: Control and drive circuit 7]: Power semiconductor switch for clamp circuit (referred to as clamp switch) 28 1321392 , r2 : Power semiconductor switch for clamp circuit (referred to as clamp switch) . 7; +, 77, 7; + and 7; -: power semiconductor switch of resonant inverter (referred to as full bridge switch) 7; Transformer for magnetizing current (referred to as coupled inductor) A:: Coupling inductor 7; Coupling coefficient (referred to as coupling coefficient) b: Primary winding of coupled inductor C (referred to as primary winding)

Zy: coupled inductor 7; secondary winding (abbreviated as secondary winding) k: resonant inductor φ of resonant inverter circuit Θ: first clamped diode of clamp circuit A: second clamped diode of clamp circuit Body A: Brake diode of clamp circuit A: Rectifier diode of coupling circuit Ci: Output capacitor of resonant inverter circuit G: Capacitor of clamp circuit

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Claims (1)

1321392 X. Scope of application: 1 A high-efficiency spectral oscillator current source inverter consisting of a clamp circuit consisting of two clamp switches, two clamp diodes, a brake diode and a clamp capacitor. Mainly to control the current and rebound energy of the coupled inductor; a coupling circuit consisting of one of the coupled inductor, the secondary winding and a rectifying diode, the primary winding limits the current of the DC voltage source, and turns it on during the conduction period. The stored energy is transferred to the secondary winding by the coupled inductor and continues to be discharged to the resonant. Inverter circuit; a resonant inverter circuit: consisting of four full bridge switches, an output capacitor and a resonant inductor, mainly guiding The primary winding current is output to the output capacitor to accumulate the AC output voltage; a control drive circuit: the closed-loop control of the sine wave command voltage and the output voltage, and finally the driving signals required for the two clamp switches and the four full-bridge switches The coupled inductor circuit is a buffer circuit between the input DC power supply and the resonant inverter circuit, and the voltage difference between the two voltage sources is coupled across the coupling The inductor is represented by a current source, and the current source current flows into the resonant inverter circuit through the two symmetric switch and the clamped diode combination path of the powder circuit; the spectral backflow is that the circuit will come from the DC power supply terminal. The current is directed to the output capacitor and controls the voltage polarity of the output capacitor to generate an AC voltage. The clamp circuit not only interrupts the current of the inverter circuit, but also clamps the two winding characteristics of the coupled inductor with voltage clamping, so that all power semiconductor switches Both the diode and the diode have flexible switching characteristics; the clamping circuit can reduce the power semiconductor 30 1321392, wherein in the voltage clamping circuit, the brake diode mainly provides an energy feedback path with an output voltage higher than the DC power supply voltage, which occurs when the load is The flywheel inertia of the AC motor and the high voltage caused by the inductive load changing the resonant frequency; the brake diode draws its energy to the DC power supply voltage terminal for absorption, in addition to clamping all switching voltages, and effectively utilizing the source-to-load return energy . 32