TW202130322A - Pulse detector and blood pressure detector thereof - Google Patents

Abstract

A pulse detector and a blood pressure detector thereof are provided. The pulse detector is used for detecting pulsations and includes a microwave sensor and an amplitude demodulator connected with the microwave sensor. The microwave sensor generates an electric field and senses the variation of the electric field which is interfered by the pulsations so as to obtain sensing signals corresponding to the pulsations. The amplitude demodulator demodulates the amplitude of the sensing signals to detect the pulsations according to the amplitude variation of the sensing signals.

Description

Pulse wave detector and blood pressure detector

The present invention relates to a pulse measurement technology, in particular to a pulse wave detector and a blood pressure detector.

There are many ways to measure the pulsation state, such as photoplethysmography (PPG). The blood pressure measurement technology is usually based on the cuff measurement method, but for long-term monitoring, there is also a cuff-less measurement technology under development. Generally, the electrocardiogram and PPG are used to measure the pulse arrival time and use Dual PPG measures the pulse transmission time, dual radars are used to measure the pulse transmission time, etc., and the blood pressure value is calculated based on the measured time.

However, these measurement methods are still not suitable for long-term measurement for users in an active state. For example, the electrode patch may make the user feel uncomfortable, and the sensor needs to be close to the body surface. Moreover, the high cost of measurement equipment used in the current technology is not conducive to popularization.

In view of this, an embodiment of the present invention provides a pulse wave detector including a microwave sensor and an amplitude demodulator coupled to the microwave sensor for detecting the pulse state. The microwave sensor emits an electric field and senses that the electric field is disturbed by the pulse wave state to obtain a sensing signal. The amplitude demodulator demodulates the amplitude of the sensing signal, so as to detect the pulsation state according to the amplitude change of the sensing signal.

An embodiment of the present invention also provides a blood pressure detector, which includes the above-mentioned two pulse wave detectors and a data analysis device. The data analysis device synchronously receives the signals of the two pulse wave detectors to obtain the pulse transmission time of the blood, and estimates the blood pressure value according to the pulse transmission time.

In summary, according to the pulse wave detector and blood pressure detector of the embodiment of the present invention, for the sensing signal of the microwave sensor, the amplitude demodulator can be used to detect the pulse state, which can reduce the hardware cost , While simplifying the complexity of signal processing.

1 and 2 together, FIG. 1 is a block diagram of the pulse wave detector 100 according to the first embodiment of the present invention, and FIG. 2 is a schematic diagram of the use state of the pulse wave detector 100 according to some embodiments of the present invention. The pulse wave detector 100 includes a microwave sensor 120 and an amplitude demodulator 130. The pulse wave detector 100 can be used to detect the pulse state of an object (such as a solid or fluid), such as heartbeat, vibration, and so on. The data analysis device 500 is coupled to the pulse wave detector 100, and can perform further data analysis and application of the detection result. The data analysis device 500 can be, for example, a computing device such as a single chip, a microprocessor, a personal computer, a smart phone, a tablet computer, and a network server. The microwave sensor 120 serves as the sensing unit 200, and the sensing unit 200 is set at or close to the target to be detected. The microwave sensor 120 may be, for example, a Split-Ring Resonator (SRR). As an example of implementation, please refer to FIG. 5 and FIG. 12 for its structure, which will be further described later.

Here, the pulsation when the blood 320 of the human body flows is taken as the target to be detected as an example. The sensing unit 200 is arranged on or close to the surface of the human body, for example, it can be attached to the skin or fixed on clothes. The blood vessel 310 will undergo contraction and diastole changes due to the pulsation of the blood 320 (compared to the blood vessel 310'). Correspondingly, the muscle 330 around the blood vessel 310 will also change in contraction and relaxation (compared to the muscle 330'). The microwave sensor 120 is configured to emit an electric field E and can sense that the electric field E is disturbed by a pulsation state to obtain a sensing signal. Specifically, the microwave sensor 120 has a resonant cavity, and the electric field E formed by the resonant cavity is directly or indirectly affected by the target to be detected and changes. In other words, because the blood 320, the blood vessel 310 and the surrounding muscles 330 are all conductive materials, they will affect the electric field E distribution and cause the resonance frequency to change. Therefore, the pulse state can be detected by detecting the change amplitude of the resonance frequency. . In this example, the blood 320 and the blood vessel 310 are under the muscle 330 and are far away from the microwave sensor 120. Therefore, the disturbance induced by the electric field E mainly comes from the contraction and relaxation of the muscle 330.

Referring to FIG. 3, it is a schematic diagram of a sensing signal according to some embodiments of the present invention. As shown on the left side of FIG. 3, the sensing signal S1 shows the measurement result of the blood vessel 310 and the muscle 330 in the contracted state; the sensing signal S1' shows the measurement result of the blood vessel 310 and the muscle 330 in the diastolic state. It can be seen that the sensing signals S1 and S1' are not only different in frequency, but also different in amplitude. Since the cost of the instrument for recognizing the amplitude is lower than that of the frequency recognizing method, and can have higher sensitivity, the embodiment of the present invention uses the amplitude to recognize the pulse wave change. The amplitude demodulator 130 is coupled to the microwave sensor 120 to demodulate the amplitudes of the sensing signals S1, S1', so as to detect the pulsation state according to the amplitude changes of the sensing signals S1, S1' (here, blood 320). pulsation). As shown on the right side of FIG. 3, it is the output result of the sensed signals S1 and S1' passing through the amplitude demodulator 130. The change of the waveform w1 shown can correspond to the change of the blood flow 320 caused by the heartbeat. Here, the amplitude demodulator 130 may be, for example, an envelope detector (Envelope Detector).

4, which is a block diagram of the pulse wave detector 100 according to the second embodiment of the present invention. The difference from the aforementioned first embodiment is that in addition to the microwave sensor 120, the sensing unit 200 also includes a zero point filter 140, which is coupled between the microwave sensor 120 and the amplitude demodulator 130 for Increase the frequency response zero point (Zero) in the sensing signal S1, S1'. The zero point filter 140 may be, for example, a Complementary Split-Ring Resonator (CSRR), a Bandstop Filter (Bandstop Filter), or a Notch Filter (Notch Filter). Here, the zero point filter 140 is an example of a complementary slit ring resonator as an example of implementation. Please refer to FIG. 5 and FIG. 12 for its structure, which will be further described later. Referring again to Figure 3, as shown on the left side of Figure 3, the sensing signal S2 is the result of adding the frequency response zero point Z to the sensing signal S1; the sensing signal S2' is the result of adding the frequency response zero point Z to the sensing signal S1' result. It can be seen that due to the increase of the frequency response zero point Z, the slope change between the frequency response poles (Pole) P, P'and the frequency response zero point Z is steeper than that of the first embodiment, achieving the effect of enhancing the sensitivity. From the right side of Figure 3, we can see that the change of the waveform w2 is more obvious.

5, which is a front view of the first implementation aspect of the sensing unit 200 according to the second embodiment of the present invention. Here, a printed circuit board manufacturing technology can be used to fabricate the sensing unit 200, which includes a substrate 400, a slit ring 410, two microstrip lines 421 and 422, and a complementary slit ring 430. The slit ring 410 and the two microstrip lines 421 and 422 are made of metal. The back surface of the substrate 400 is a metal ground surface, and a complementary crack ring 430 is formed by the missing metal portion. The slit ring 410, the microstrip line 421 and part of the microstrip line 422 constitute the microwave sensor 120, and the other part of the microstrip line 422 and the complementary slit ring 430 constitute the zero point filter 140. The slit ring 410 and the two microstrip lines 421 and 422 are located on the front side of the substrate 400, and the complementary slit ring 430 is located on the reverse side of the substrate 400. In FIG. 5, the complementary slit ring 430 is represented by a dashed line.

The slit ring 410 is located between the two microstrip lines 421 and 422, and the slit ring 410 is separated from the two microstrip lines 421 and 422 by a gap G1, respectively. Here, the two microstrip lines 421 and 422 are T-shaped, but the embodiment of the present invention is not limited to this. The T-shaped lateral ends 4211 and 4221 of the two microstrip lines 421 and 422 are close to the slit ring 410. The end of the T-shaped central portion 4212 of the microstrip line 421 is used to feed high frequency signals. The T-shaped central part 4212, 4222 gradually changes from a wide band at the end to a narrow band. The wide band width is denoted as W1, and the narrow band width is denoted as W2. Here, the slit ring 410 is generally a rectangular ring shape (the length of the long side and the width of the short side are marked as L1 and L2, and the line width is marked as W3), but the embodiment of the present invention is not limited to this shape, and the slit ring 410 is parallel There is a gap 411 (the gap between the gaps 411 is marked as G2). The slit ring 410 has two T-shaped portions 412 (the length of the lateral ends is denoted as L3) with opposite lateral ends on both sides of the notch 411.

The complementary slit ring 430 also has an outer shape similar to that of the slit ring 410, and is also roughly rectangular in shape (the length of the four sides is marked as L4, and the line width is marked as W4), but the embodiment of the present invention is not limited to this shape, and has a gap 431 (The gap of the gap 431 is marked as G3). The complementary slit ring 430 has two inwardly extending L-shaped portions 432 on both sides of the notch 431 (the length of the extension is marked as L5). Here, the rear section of the T-shaped central portion 4222 of the microstrip line 422 traverses the complementary slit ring 430 and passes through the position corresponding to the gap 431. As shown in FIG. 5, the slit ring 410 and the complementary slit ring 430 are sequentially arranged along the extension direction of the microstrip lines 421 and 422. As an example, the value of each parameter can be as listed in Table 1.

Table 1 L1 12 mm W2 0.7 mm L2 6 mm W3 1.2mm L3 4 mm W4 0.75 mm L4 11 mm G1 0.3 mm L5 2mm G2 1.2mm W1 1.7mm G3 0.5 mm

。電容C_CSRR 和電感L_CSRR 為互補裂隙環430的共振電容和電感值。電感L為微帶線422的等效電感。6, which is an equivalent circuit diagram of the first implementation aspect of the sensing unit 200 according to the second embodiment of the present invention. Among them, the resistances R _muscle and R _sub are the loss resistances of the muscle 330 and the substrate 400 respectively. The capacitance C _sub is the parasitic capacitance of the substrate 400. The capacitance C _g is the coupling capacitance between the slit ring 410 and the microstrip lines 421 and 422, respectively. The capacitance C _res and the inductance L _res are the resonance capacitance and inductance values of the slit ring 410. The capacitance C _{air -gap} is the equivalent capacitance between the sensing unit 200 and the muscle 330. The capacitance C _muscle is the equivalent capacitance of the muscle 330. The total capacitance C _total can be expressed as C _res +(C _{air -gap} ||C _muscle ). Therefore, the resonance frequency is

. The capacitance C _CSRR and the inductance L _CSRR are the resonance capacitance and inductance values of the complementary slit ring 430. The inductance L is the equivalent inductance of the microstrip line 422.

Referring to FIG. 7, it is a schematic diagram of the use state of the blood pressure detector according to some embodiments of the present invention. The blood pressure detector can use the aforementioned two pulse wave detectors 100 for blood pressure detection. Specifically, the sensing units 200 of the two pulse wave detectors 100 are respectively arranged at positions adjacent to the carotid artery and the radial artery to measure the pulsation of the blood 320 at these two parts respectively. However, the embodiment of the present invention is not limited to these two measurement positions, and the sensing units 200 of the two pulse wave detectors 100 can be placed in the same closed circulation (Closed Circulation) appropriate position. As shown on the left side of Figure 7, the upper waveform is the measurement result of the carotid artery, and the lower waveform is the measurement result of the radial artery. The data analysis device 500 can find out the pulse transit time (PuL2e Transit Time) PTT from the difference between the two measurement results. After finding the pulse transmission time PTT, the blood pressure value can be estimated according to some conversion formulas between the pulse transmission time PTT and blood pressure (such as formula 1 and formula 2). Formula 1 is the calculation formula of systolic blood pressure (SBP). Formula 2 is the calculation formula of diastolic blood pressure (DBP). Among them, a _s , b _s , a _d , and b _d are correction constants. However, the embodiments of the present invention are not limited to these calculation formulas. For example, the natural logarithm (Ln) in formula 1 and formula 2 can be changed to common logarithm (Log). Alternatively, the regression analysis method can be used to obtain other correlation formulas between pulse transit time PTT and blood pressure.

SBP=a _s Ln(PTT)+b _s (Equation 1)

DBP=a _d Ln(PTT)+b _d (Equation 2)

Referring to FIG. 8, it is the difference between the pulse transit time PTT measured by the blood pressure detector and the actual systolic blood pressure measured by blood pressure measurement according to the first implementation aspect of the sensing unit 200 according to the second embodiment of the present invention. Related graphs. It can be seen that there is a significant correlation between pulse transit time PTT and systolic blood pressure (correlation coefficient R=-0.79). Among them, the round symbol indicates the measurement result in the resting state, and the cross symbol indicates the measurement result in the recovery state after exercise, a total of 56 data points.

Referring to FIG. 9, it is the difference between the pulse transit time PTT measured by the blood pressure detector and the diastolic blood pressure measured by the actual blood pressure measurement according to the first implementation aspect of the sensing unit 200 according to the second embodiment of the present invention. Related graphs. It can also be seen that there is a significant correlation between pulse transit time PTT and diastolic blood pressure (correlation coefficient R=-0.82). Among them, the round symbol indicates the measurement result in the resting state, and the cross symbol indicates the measurement result in the recovery state after exercise, a total of 56 data points.

10 and FIG. 11, the first embodiment of the sensing unit 200 according to the second embodiment of the present invention is used as a blood pressure detector to obtain the estimated value of systolic blood pressure and diastolic blood pressure. Statistics chart of Bland-Altman Analyses. According to the aforementioned formula 1, the pulse transmission time PTT is converted into an estimated value of systolic blood pressure, and the reference value of systolic blood pressure measured by a commercially available blood pressure meter, and the statistics are shown in Figure 10. Similarly, the pulse transit time PTT is converted into an estimated value of diastolic blood pressure according to the aforementioned formula 2, and the reference value of diastolic blood pressure measured by a commercially available blood pressure meter is statistically shown in Figure 11. It can be seen that, as shown in Figure 10 and Figure 11, the values all fall within the confidence interval of the mean ±1.96 standard deviations.

12 and 13 are respectively a front view and a rear view of the substrate 400 of the second embodiment of the sensing unit 200 according to the second embodiment of the present invention. The main difference from the first embodiment is that in the second embodiment, the zero point filter 140 can not only increase the frequency response zero point Z, but also provide a microwave sensing function. Through the through hole 401, the sensing area of the microwave sensor 120 is led to the same side and the same area as the sensing range of the zero point filter 140, so that the sensing effect can be improved.

Refer to Figure 12 and Figure 13 together. The slit ring 450, the microstrip line 461, and part of the microstrip line 462 constitute the microwave sensor 120 (here, also a slit ring resonator), and the other part of the microstrip line 462 and the complementary slit ring 470 constitute the zero point filter 140 ( Here is also a complementary slit ring resonator). The slit ring 450 and the two microstrip lines 461 and 462 are located on the front side of the substrate 400, and the complementary slit ring 470 is located on the reverse side of the substrate 400. The slit ring 450 and the two microstrip lines 461 and 462 are made of metal. The back surface of the substrate 400 is a metal ground surface, and a complementary crack ring 470 is formed by the missing metal portion. In FIG. 12, the complementary slit ring 470 is shown in dashed lines, and the above-mentioned elements on the front side are not shown in FIG. 13.

The slit ring 450 is located between the two microstrip lines 461 and 462, and the slit ring 450 is separated from the two microstrip lines 461 and 462 by a gap G4. Here, the two microstrip lines 461 and 462 are T-shaped, and respectively have T-shaped lateral ends 4611, 4621 and T-shaped central portions 4612, 4622, but the embodiment of the present invention is not limited thereto. The T-shaped lateral ends 4611, 4621 of the microstrip lines 461, 462 are close to the slit ring 450 (the line width of the T-shaped lateral ends 4611, 4621 is denoted as W5). The T-shaped transverse ends 4611 and 4621 each have an extension section at both ends, which respectively extend along the periphery of the slit ring 450. The spacing between the T-shaped lateral ends 4611, 4621 is denoted as G5. The end of the T-shaped center 4612 of the microstrip line 461 is used to receive high-frequency signals. The T-shaped central part 4612, 4622 gradually changes from a wide band at the end to a narrow band. The width of the broadband is marked as W6. The length of the T-shaped central portion 4612 of the microstrip line 461 is marked as L6; the length of the T-shaped central portion 4622 of the microstrip line 462 is marked as L7. Here, the slit ring 450 is roughly in a rectangular ring shape (the length of the long side and the width of the short side are marked as L8 and L9, and the line width is marked as W7), but the embodiment of the present invention is not limited to this shape. The slit ring 450 is not limited to this shape. There is a gap 451 (the gap between the gaps 451 is marked as G6). Here, the slit ring 450 has two outwardly extending L-shaped portions 452 (the length of the extension is denoted as L10) on both sides of the notch 451.

As shown in FIG. 13, the complementary slit ring 470 also has a shape similar to that of the slit ring 450, and is also roughly rectangular in shape (long side length and short side width are marked as L11 and L12, respectively, and the line width is marked as W8). The embodiment of the invention is not limited to this shape, and has a gap 471 (the gap between the gaps 471 is marked as G7). The complementary slit ring 470 has a convex portion 472 protruding outward on the opposite side of the notch 471, and the width of the convex portion 472 is marked as W9. The convex part 472 is provided with two belt-shaped parts 473 (the length is marked as L13, the width is marked as W10, the interval is marked as G8, and the distance between the belt-like part 473 and the convex part 472 is marked as G9). Each belt-shaped portion 473 is respectively coupled to the L-shaped portion 452 of the slit ring 450 through a through hole 401 (the diameter of the through hole 401 is denoted as D1). Here, the band 473 is made of metal. As shown in FIG. 12, the T-shaped central portion 4622 of the microstrip line 462 is serpentine and traverses the complementary slit ring 470. The slit ring 450 and the complementary slit ring 470 are respectively located on two sides of the substrate 400 and staggered up and down, and are coupled through the through hole 401. Through the through hole 401, the sensing area of the slit ring 450 is guided to the opposite side of the substrate 400 (that is, one side of the complementary slit ring 470), and the sensing area is interlaced between the slit ring 450 and the complementary slit ring 470 The position is the center, and a range is expanded outward. As an example, the value of each parameter can be as listed in Table 2.

Table 2 L6 16.1mm L13 2.2mm G4 0.3 mm L7 44.1mm W5 0.7 mm G5 5.8mm L8 9.2mm W6 1.7mm G6 1.2mm L9 8.5mm W7 1.2mm G7 1.2mm L10 2.5mm W8 1.2mm G8 1.2mm L11 9.2mm W9 0.3 mm G9 0.3 mm L12 8 mm W10 1.2mm D1 0.7 mm

FIG. 14 is a schematic diagram of the microwave sensor 120 of the second embodiment of the sensing unit 200 according to the second embodiment of the present invention. 12 and 14 together, the slit ring 450 uses a coplanar waveguide (CPW) technology, and is coupled to a complementary slit ring 470 via a through hole 401. The slit ring 450 can be regarded as a slit ring resonator integrated with a transmission line and a step impedance (Step Impedance). The inductance L _via is the equivalent inductance of the through hole 401. Since the slit ring 450 is a symmetrical structure, it can be analyzed based on an even-odd mode equivalent circuit. 15 and FIG. 16, which are respectively the odd-even mode equivalent circuit diagrams of the microwave sensor 120 of the second embodiment of the sensing unit 200 according to the second embodiment of the present invention. According to the transmission line theory, if the input impedance viewed from port P is infinite, the odd-even mode resonance can be expressed as Equation 3 and Equation 4, respectively. Among them, Z ₁ and Z ₂ are the characteristic impedances of the transmission line coupled on the front and back sides of the through hole 401, respectively, and θ ₁ and θ ₂ are the electrical lengths, respectively. As an example, Z ₁ is 60.8Ω, Z ₂ is 54.5Ω; at 2.67 GHz, θ ₁ is 83.4°, θ ₂ is 5.3°; L _via is 0.146 nH.

Z ₁ tanθ ₁ =Z ₂ cotθ ₂ -ωL _via (Equation 3)

Z ₁ cotθ ₁ = Z ₂ cotθ ₂ +ωL _via (Equation 4)

Since the complementary slit ring 470 is a complementary structure of the slit ring 450, its odd-even mode resonance analysis principle is the same as that described above, and will not be repeated here. It should be particularly noted here that, in some embodiments, the pulse wave detector 100 further includes a variable capacitor C _var coupled to the zero point filter 140 to change the zero point filter according to the capacitance value of the variable capacitor C _var The frequency response of 140 is zero point Z. Referring to FIG. 17, it is a front view of the second implementation aspect of the sensing unit 200 according to the second embodiment of the present invention. The variable capacitance C _var can be realized by a variable capacitance diode. Depending on the magnitude _{of the bias voltage V b applied to the variable capacitance diode, the capacitance value of the variable capacitance C var} can be controlled, thereby changing the frequency response zero point Z. In order to isolate the bias voltage V _b , an inductor L _{b is} added between the input terminal of the bias voltage V _{b and the variable capacitor C var} . In order to prevent the variable capacitor C _{var from being} short-circuited, a bypass capacitor C _{b is added} . The variable capacitor C _var is coupled to the ground surface on the reverse side of the substrate 400 through the through hole 402; the bypass capacitor C _b is coupled to the ground surface on the reverse side of the substrate 400 through the through hole 403. The through holes 402 and 403 are respectively placed inside and outside of the complementary slit ring 470. Through the through holes 402 and 403, the variable capacitor C _var , the bypass capacitor C _b and the inductance L _b can be arranged in the available space on the front surface of the substrate 400. As an example, _{the capacitance value of the variable capacitor C var} varies from 1.3 pF to 22.62 pF, the capacitance value of the _{bypass capacitor C b is 0.68 pF, the inductance value of the inductor L b} is 25 nH, and the frequency response zero point Z can be controlled. It is 2.2GHz～2.7GHz.

Referring to FIG. 18, it is a schematic diagram of the zero point filter 140 of the second implementation aspect of the sensing unit 200 according to the second embodiment of the present invention. Since the complementary slit ring 470 is coupled to the variable capacitor C _var and the bypass capacitor C _b , if the same resonant frequency is used as the purpose, the size of the complementary slit ring 470 can be reduced compared to the pure complementary slit ring 470. In addition, since the band portion 473 and the surrounding ground plane will also generate a capacitance C _n , it will also cause the resonance frequency to drop. However, since _{the capacitance value of the variable capacitor C var} can be adjusted, it can compensate for the resonance frequency caused by the _{capacitance C n} The impact of decline.

19, which is a frequency response diagram of the second embodiment of the sensing unit 200 according to the second embodiment of the present invention _{under different bias voltages V b.} It can be seen that applying bias voltages V _{b of} different voltage values (here, 0V～8V is taken as an example), the frequency response zero point Z can be changed.

Referring to FIG. 20, it is a schematic diagram of the frequency response zero point Z adjustment of the second implementation of the sensing unit 200 according to the second embodiment of the present invention. Comparing the left side of Figure 3 and Figure 19, it can be seen that if the frequency response zero point Z is not fixed, the closer to the frequency response zero point Z, the less obvious the amplitude change (as shown on the left side of Figure 3); and in response to the frequency offset (such as Moving from frequency f to frequency f') Adjust the frequency response zero point Z, as shown in Figure 19, adjusting from the frequency response zero point Z to the frequency response zero point Z', the amplitude change can be maintained to a considerable extent.

In some embodiments, the variable capacitor C _var can also be coupled to the microwave sensor 120 to change the frequency response pole of the microwave sensor 120. It can also respond to the frequency offset, so that the amplitude change can be maintained at a comparable level. degree.

Referring to FIG. 21, it is a block diagram of a blood pressure detector according to a second embodiment of the present invention. As mentioned above, two pulse wave detectors 100 are also used for blood pressure detection, and the detected signals are further analyzed by the data analysis device 500. Here, each constituent element is further explained. In addition to the aforementioned sensing unit 200 and amplitude demodulator 130, the pulse wave detector 100 may also include an amplifier 150. The amplifier 150 is coupled after the sensing unit 200 to amplify the signal sensed by the sensing unit 200, so that the subsequent amplitude demodulator 130 can better process the signal. In some embodiments, the amplifier 150 may be, for example, a Low Noise Amplifier (LNA). Here, if the sensing unit 200 has the aforementioned variable capacitance C _var , the blood pressure detector can provide a bias voltage V _b to adjust the capacitance value of the variable capacitance C _var. The blood pressure detector also includes a signal generator 600 and a power divider 700. The signal generator 600 is used to generate high-frequency signals, and the power divider 700 is used to output high-frequency signals from two output ports and input them to the microwave sensors 120 of the two sensing units 200 to generate resonance effects. Here, the power divider 700 may be, for example, a Wilkinson power divider. In some embodiments, the frequency of the high-frequency signal is 2.4 GHz˜2.5 GHz.

22, the second embodiment of the sensing unit 200 according to the second embodiment of the present invention is used as a blood pressure detector between the measured pulse transit time PTT and the actual systolic blood pressure measured by blood pressure measurement. Related graphs. It can be seen that there is a significant correlation between pulse transit time PTT and systolic blood pressure (correlation coefficient R=-0.79). Among them, the round symbol indicates the measurement result in the resting state, and the cross symbol indicates the measurement result in the recovery state after exercise, a total of 155 data points.

23, the second embodiment of the sensing unit 200 according to the second embodiment of the present invention is used as a blood pressure detector to measure the pulse transit time PTT and the diastolic blood pressure measured by a commercially available blood pressure meter Correlation graph between. It can also be seen that there is a significant correlation between pulse transit time PTT and diastolic blood pressure (correlation coefficient R=-0.72). Among them, the round symbol indicates the measurement result in the resting state, and the cross symbol indicates the measurement result in the recovery state after exercise, a total of 155 data points.

24 and 25, the second embodiment of the sensing unit 200 according to the second embodiment of the present invention is used as a blood pressure detector to obtain the estimated value of systolic blood pressure and diastolic blood pressure. The statistical chart of the German-Ultraman analysis. According to the aforementioned formula 1, the pulse transmission time PTT is converted into an estimated value of systolic blood pressure, and the reference value of systolic blood pressure measured by a commercially available blood pressure meter, and the statistics are shown in Figure 24. Similarly, the pulse transit time PTT is converted into an estimated value of diastolic blood pressure according to the aforementioned formula 2, and the reference value of diastolic blood pressure measured by a commercially available blood pressure meter is statistically shown in Figure 25. It can be seen that, as shown in Figure 24 and Figure 25, the values all fall within the confidence interval of the mean ±1.96 standard deviations.

In summary, according to the pulse wave detector and blood pressure detector of the embodiment of the present invention, for the sensing signal of the microwave sensor, the amplitude demodulator can be used to detect the pulse state, which can reduce the hardware cost , While simplifying the complexity of signal processing. According to the pulse wave detector of some embodiments, the microwave sensor is coupled to the zero point filter to increase the frequency response zero point in the sensing signal, which can improve the sensing sensitivity. According to some embodiments, the zero point filter is also coupled with a variable capacitor, so that the frequency response zero point can be adjusted, so that the state of different frequency offsets still maintains a considerable amplitude change, and the sensing sensitivity is maintained. In some embodiments, the variable capacitor can be changed to be coupled to the microwave sensor to adjust the frequency response pole and also maintain the sensing sensitivity. According to some embodiments, the microwave sensor and the zero point filter are located on both sides of the substrate, and are coupled via the through holes, so that the sensing areas of the microwave sensor and the zero point filter overlap, and the sensing effect is increased.

100: Pulse wave detector 120: Microwave sensor 130: Amplitude demodulator 140: Zero point filter 150: Amplifier 200: Sensing unit 310, 310': Blood vessel 320: Blood 330, 330': Muscle 400: Base plate 401, 402, 403: Through hole 410: slit ring 411: gap 412: T-shaped part 421, 422: microstrip line 4211, 4221: T-shaped transverse end 4212, 4222: T-shaped central part 430: complementary slit ring 431: gap 432: L-shaped part 450: slit Ring 451: Notch 452: L-shaped part 461: Microstrip line 462: Microstrip line 4611, 4621: T-shaped transverse end 4612, 4622: T-shaped center part 470: Complementary slit ring 471: Notch 472: Convex 473: Strip 500: data analysis device 600: signal generator 700: power divider E: electric field PTT: pulse transmission time S1, S1': sensing signal S2, S2': sensing signal Z, Z': frequency response zero point f, f': frequency w1, w2: waveform G1~G9: spacing L1~L13: length W1~W10: width D1: diameter R _muscle , R _sub : resistance C _g , C _res , C _sub C _air-gap , C _muscle , C _n , C _CSRR : capacitance C _var : variable capacitance C _b : bypass capacitor L _res , L _sub , L _via , L _b , L _CSRR : inductance P: port V _b : bias

[Fig. 1] is a block diagram of the pulse wave detector according to the first embodiment of the present invention. [Fig. 2] is a schematic diagram of the use state of the pulse wave detector in some embodiments of the present invention. [Figure 3] is a schematic diagram of sensing signals in some embodiments of the present invention. [Fig. 4] is a block diagram of the pulse wave detector according to the second embodiment of the present invention. [Fig. 5] is a front view of the first implementation aspect of the sensing unit according to the second embodiment of the present invention. [Fig. 6] is an equivalent circuit diagram of the first implementation aspect of the sensing unit of the second embodiment of the present invention. [Fig. 7] is a schematic diagram of the use state of the blood pressure detector according to some embodiments of the present invention. [Figure 8] is a correlation diagram between the pulse transmission time measured as a blood pressure detector and the actual systolic blood pressure measured by blood pressure measurement according to the first implementation aspect of the sensing unit according to the second embodiment of the present invention . [Fig. 9] is a correlation diagram between the pulse transmission time measured by the blood pressure detector and the diastolic blood pressure measured by blood pressure measurement according to the first implementation aspect of the sensing unit according to the second embodiment of the present invention . [Fig. 10] is a statistical diagram of Brand-Altman analysis of the estimated systolic blood pressure obtained by the blood pressure detector according to the first implementation aspect of the sensing unit according to the second embodiment of the present invention. [Fig. 11] is a statistical diagram of Brand-Altman analysis of the predicted value of diastolic blood pressure obtained by the blood pressure detector as the first implementation aspect of the sensing unit according to the second embodiment of the present invention. [Fig. 12] is a front view of the substrate of the second embodiment of the sensing unit of the second embodiment of the present invention. [Fig. 13] is a back view of the substrate of the second embodiment of the sensing unit of the second embodiment of the present invention. [Fig. 14] is a schematic diagram of the microwave sensor of the second implementation aspect of the sensing unit of the second embodiment of the present invention. [FIG. 15] and [FIG. 16] are respectively the odd and even mode equivalent circuit diagrams of the microwave sensor in the second implementation of the sensing unit of the second embodiment of the present invention. [Fig. 17] is a front view of the second implementation aspect of the sensing unit according to the second embodiment of the present invention. [Fig. 18] is a schematic diagram of the zero point filter of the second implementation aspect of the sensing unit of the second embodiment of the present invention. [Fig. 19] is a frequency response diagram of the second embodiment of the sensing unit according to the second embodiment of the present invention under different bias voltages. [Fig. 20] is a schematic diagram of frequency response zero point adjustment of the second embodiment of the sensing unit according to the second embodiment of the present invention. [Figure 21] is a block diagram of a blood pressure detector according to a second embodiment of the present invention. [Fig. 22] is a correlation diagram between the pulse transmission time measured by the blood pressure detector and the actual systolic blood pressure measured by blood pressure measurement according to the second implementation aspect of the sensing unit according to the second embodiment of the present invention . [FIG. 23] The second embodiment of the sensing unit according to the second embodiment of the present invention is the correlation between the pulse transit time measured as a blood pressure detector and the diastolic blood pressure measured by a commercially available blood pressure meter picture. [Fig. 24] is a statistical diagram of Brand-Altman analysis of the estimated systolic blood pressure obtained by the blood pressure detector as the second implementation aspect of the sensing unit according to the second embodiment of the present invention. [Fig. 25] is a statistical diagram of Brand-Altman analysis of the estimated value of diastolic blood pressure obtained by the blood pressure detector as the second implementation aspect of the sensing unit according to the second embodiment of the present invention.

100: Pulse detector

120: Microwave sensor

130: Amplitude demodulator

200: sensing unit

500: data analysis device

Claims (10)

A pulse wave detector for detecting a pulsation state. The pulse wave detector includes: A microwave sensor that emits an electric field and senses that the electric field is disturbed by the pulsation state to obtain a sensing signal; and An amplitude demodulator is coupled to the microwave sensor to demodulate the amplitude of the sensing signal, so as to detect the pulsation state according to the amplitude change of the sensing signal. The pulse wave detector according to claim 1, further comprising a zero point filter coupled between the microwave sensor and the amplitude demodulator to add a frequency response zero point to the sensing signal. The pulse wave detector according to claim 2, wherein the zero point filter is a complementary slit ring resonator. The pulse wave detector according to claim 2 further includes a variable capacitor coupled to the zero point filter to change the frequency response zero point of the zero point filter according to the capacitance value of the variable capacitor. The pulse wave detector according to claim 1, wherein the microwave sensor is a slit ring resonator. The pulse wave detector according to claim 1, wherein the amplitude demodulator is an envelope detector. The pulse wave detector according to claim 1, further comprising a variable capacitor coupled to the microwave sensor to change a frequency response pole of the microwave sensor according to the capacitance value of the variable capacitor. The pulse wave detector according to claim 1, wherein the microwave sensor is a slit ring resonator, and the pulse wave detector further includes a complementary slit ring resonator, the complementary slit ring resonator being coupled to Between the slotted ring resonator and the amplitude demodulator, the complementary slotted ring resonator and the slotted ring resonator are separately arranged on two sides of a substrate, and are coupled through at least one through hole. The pulse wave detector according to claim 8, wherein a microstrip line passes through the other side of the complementary slit ring resonator, and the microstrip line is coupled to the slit ring resonator. A blood pressure detector, comprising two pulse wave detectors according to any one of claim items 1 to 9 and a data analysis device. The data analysis device receives signals from the two pulse wave detectors to obtain A pulse transmission time of blood, and a blood pressure value is estimated based on the pulse transmission time.

Images