TW201336213A - Controller, power converter for controlling transformer, and load driving circuit thereof - Google Patents

Abstract

A circuit includes a transformer and a controller. The transformer includes a primary winding and a secondary winding, and operates in multiple switching cycles. A switching cycle includes a charging period and a discharging period. During the charging period, the transformer is powered by an input voltage and a current flowing through the primary winding increases. During the discharging period, the transformer discharges to power the load and a current flowing through the secondary winding decreases. The controller includes a pin that receives a first feedback signal indicating the input voltage during the charging period and receives a second feedback signal indicating an electrical condition of the secondary winding during the discharging period. The controller generates a first control signal and a second control signal to regulate the input voltage and a current flowing through the load, respectively.

Description

Controller for controlling transformer, power converter and load driving circuit thereof

The present invention relates to a controller, and more particularly to a controller for controlling a transformer and its power converter and load drive circuit.

The flyback converter is a switch mode regulated power supply that can be used in applications such as AC/DC adapters or battery chargers. A conventional flyback converter 100 is shown in FIG. The flyback converter 100 controls a transformer using the controller 120. The transformer includes a primary winding 104 coupled to a DC power source V _BB , a secondary winding 106 coupled to the load 112 , and an auxiliary winding 108 . Controller 120 controls switch 118 coupled in series with primary winding 104. When the switch 118 is turned on, current flows through the primary winding 104, and the magnetic core 124 of the transformer stores energy. When the switch 118 is turned off, the diode 110 coupled to the secondary winding is forward biased, and the energy stored in the core 124 is discharged through the secondary winding 106 to the capacitor 122 and the load 112. The error amplifier 114 compares the current flowing through the current monitoring resistor 111 with a reference current and generates a feedback signal FB. The feedback signal FB is transmitted to the controller 120 through the optical coupler 116. The controller 120 controls the switch 118 to adjust the output power of the transformer based on the feedback signal FB. A disadvantage of the conventional flyback converter 100 is that it is relatively large in size.

It is an object of the present invention to provide a load drive circuit comprising: a transformer comprising a primary winding receiving an input voltage and being coupled to a load a primary winding, the transformer operates in a plurality of cycles, wherein one of the plurality of cycles includes a charging phase and a discharging phase, wherein the transformer is powered by the input voltage and flows through the primary An electric current of the winding is increased, in which the transformer discharges to supply power to the load, and a current flowing through the secondary winding is reduced; and a controller coupled to the transformer, including a turn, wherein Receiving, in the charging phase, a first feedback signal indicating the input voltage, and wherein, in the discharging phase, the chirp receives a second feedback signal indicating a power state of one of the secondary windings, wherein The controller generates a first control signal according to the first feedback signal to adjust the input voltage, and generates a second control signal according to the second feedback signal to adjust a current flowing through the load.

The invention also provides a power converter comprising: a transformer comprising a primary winding receiving an input voltage, a primary winding connected to a load, and an auxiliary winding, the transformer operating in a plurality of cycles, wherein the plurality One cycle of the cycle includes a charging phase and a discharging phase, in which the transformer is powered by the input voltage, and the current flowing through the primary winding increases, and in the discharging phase, the transformer discharges Powering the load, and the current flowing through the secondary winding is reduced; a pair of resistors connected in series with each other, and the pair of resistors are coupled to the auxiliary winding; and a controller including a turn connected to the pair A common node between the resistors, the controller clamps a voltage on the common node to a predetermined voltage value during the charging phase.

The invention also provides a controller for controlling a transformer to supply a load, comprising: a first turn generating a first control signal to adjust an input voltage of the transformer; and a second turn generating a second control signal, Adjusting a current flowing through the load and operating the transformer in a plurality of cycles, wherein one of the plurality of cycles includes a charging phase and a discharging phase, wherein the transformer is subjected to the input voltage Powering, and a current flowing through the primary winding increases; during the discharging phase, the transformer discharges to power the load, and a current flowing through the secondary winding decreases; and a third turn is coupled to An auxiliary winding of the transformer, in the charging phase, the third receiving a first feedback signal indicating the input voltage; and in the discharging phase, the third receiving receives a state indicating a power state of the secondary winding a second feedback signal; wherein the controller generates the first control signal according to the first feedback signal, and generates the second control signal according to the second feedback signal.

A detailed description of the embodiments of the present invention will be given below. While the invention will be described in conjunction with the embodiments, it is understood that the invention is not limited to the embodiments. On the contrary, the invention is intended to cover various modifications, modifications and equivalents

In addition, in the following detailed description of the embodiments of the invention However, it will be understood by those of ordinary skill in the art that the present invention may be practiced without these specific details. In other instances, well-known methods, procedures, components, and circuits have not been described in detail in order to facilitate the invention.

2 shows a power converter 200 in accordance with an embodiment of the present invention. An illustrative block diagram. 4 is an illustrative waveform diagram of signals received or generated by power converter 200. Figure 2 will be described in conjunction with Figure 4.

In the example of FIG. 2, power converter 200 includes a controller 220 for controlling transformer 202. In an embodiment, transformer 202 includes a primary winding 204, a secondary winding 206, and an auxiliary winding 208. One end of primary winding 204 is coupled to DC input voltage V _BB and the other end is coupled to ground through switch 218 and resistor 230. Secondary winding 206 is coupled to load 212 through diode 210. In an embodiment, the auxiliary winding 208 is located on the side of the primary winding 204 of the transformer 202. One end of the auxiliary winding 208 is coupled to ground through a resistor 214 and a resistor 216, and the other end is coupled to ground.

Controller 220 controls transformer 202 by controlling a switch 218 coupled in series with primary winding 204. In an embodiment, controller 220 is powered by voltage _VDD generated by auxiliary winding 208. Resistor 230 provides a feedback signal FB1 indicating the current I _PR flowing through primary winding 204. The auxiliary winding 208 provides a feedback signal FB2 indicating the output voltage of the auxiliary winding 208 and further indicating the output voltage of the secondary winding 206. Therefore, the feedback signal FB2 can reflect whether the current I _SE flowing through the secondary winding 206 drops to a preset current value, such as zero. In an embodiment, the feedback signal FB2 is generated at a node between the resistor 214 and the resistor 216.

Power converter 200 also includes a signal generator 226 (such as oscillator 226) and a clamp circuit 228. When the switch 218 is turned on, the clamp circuit 228 clamps the voltage of the feedback signal FB2. In an embodiment, the controller 220 receives the reference signal PEAK and the reference signal SET. The reference signal PEAK determines the maximum current value I _PEAK of the current I _PR flowing through the primary winding 204. The reference signal SET has a reference voltage value V _SET . In another embodiment, the reference signal PEAK and the reference signal SET are generated by the controller 220.

The controller 220 receives the feedback signal FB1 and the feedback signal FB2, and generates a pulse signal (such as the pulse width modulation signal PWM1) according to the feedback signal FB1 and the feedback signal FB2 to control the switch 218. Controller 220 operates transformer 202 for a plurality of switching cycles by controlling switch 218 coupled in series with primary winding 204. In one embodiment, one switching cycle includes a charging phase T _ON , a discharging phase T _{DIS ,} and an adjustment phase T _ADJ , as shown in FIG. 4 . During the charging phase T _ON , the transformer 202 is powered by the input voltage V _BB and the current I _PR flowing through the primary winding 204 is increased. During the discharge phase T _DIS , the transformer 202 discharges power to the load 212 and the current I _SE flowing through the secondary winding 206 decreases.

More specifically, during the charging phase T _ON , the controller 220 turns on the switch 218, thereby causing the transformer 202 to receive the input voltage V _BB . When switch 218 is turned on, diode 210 coupled to secondary winding 206 is reverse biased and no current flows through secondary winding 206. Current I _PR flows through primary winding 204, switch 218, and resistor 230 to ground. The current I _PR increases linearly. Thus, during the charging phase T _ON , the core 224 of the transformer 202 stores energy. The clamp circuit 228 clamps the voltage of the feedback signal FB2 during the charging phase T _ON such that the voltage of the feedback signal FB2 is substantially zero.

During the discharge phase T _DIS , the controller 220 turns off the shutdown 218 and discharges the load 212 through the transformer 202. When the switch 218 is turned off, the diode 210 coupled to the secondary winding 206 is forward biased, and the energy stored in the core 224 is released through the secondary winding 206 to the capacitor 222 and the load 212. During the discharge phase T _DIS , the current I _SE flowing through the secondary winding 206 linearly decreases from a maximum current value I _SE-MAX to a predetermined current value (eg, 0). The maximum current value I _{SE-MAX of the} secondary winding 206 is determined by the maximum current value I _{PEAK of the} primary winding 204 and the turns ratio of the transformer 202.

In the adjustment phase T _ADJ , the switch 218 remains off. In an embodiment, no current flows through the primary winding 204 or the secondary winding 206 during the adjustment phase T _ADJ .

As shown by the waveform of the current I _SE flowing through the secondary winding 206 in FIG. 4, the average current I _OAVG output by the secondary winding 206 in one switching period T _S can be obtained from equation (1).

Where T _S =T _ON +T _DIS +T _ADJ .

In an embodiment, the length of time of the charging phase T _{ON and} the length of time of the discharging phase T _DIS may be determined by the inductance of the primary winding 204 and the secondary winding 206, the input voltage V _{BB ,} and the output voltage V _OUT across the load 212. The controller 220 determines that the adjustment phase T _ADJ has an appropriate length of time such that the ratio of the length of the discharge phase T _{DIS to} the length of the switching period T _S is constant, wherein the length of the switching period T _S is the charging phase T _ON , The total length of time during the discharge phase T _DIS and the adjustment phase T _ADJ . In equation (1), the maximum current value I _{SE-MAX of the} secondary winding 206 is determined by the maximum current value I _{PEAK of the} primary winding 204 and the turns ratio of the transformer 202. In one embodiment, since the maximum current value I _PEAK of the primary winding 204 is a predetermined value and the turns ratio of the transformer 202 is constant, the maximum current value I _SE-MAX of the secondary winding 206 is constant. Therefore, according to equation (1), if the ratio of the length of the discharge phase T _{DIS to} the length of the switching period T _S is constant (ie, T _S =k* T _DIS , k is a constant), the average of the secondary winding 206 The output current I _OAVG is also constant.

Therefore, even if the input voltage V _BB and the output voltage V _OUT may vary, as long as the ratio of the length of the discharge phase T _{DIS to} the length of the switching period T _S is constant, the average output current I _OAVG is also constant. In other words, by utilizing a filter (such as capacitor 222 coupled to load 212), power converter 200 can provide substantially constant output current to load 212. Here, "substantially constant" means that the output current varies only within a range, so current ripple caused by non-ideal conditions of the circuit components is negligible.

FIG. 3 is a schematic diagram showing the architecture of the controller 220 of FIG. 2. Figure 3 will be described in conjunction with Figures 2 and 4. The controller 220 causes the adjustment phase T _{ADJ to} have an appropriate length of time such that the ratio of the length of the discharge phase T _{DIS to} the length of the switching period T _S is constant. Thus power converter 200 can provide a substantially constant output current to load 212.

In one embodiment, controller 220 includes a signal generator (e.g., an oscillator 226), a comparator 314, a comparator 316, and a pulse signal generator 318 (e.g., pulse width modulation signal generator 318). The oscillator 226 generates a sawtooth wave signal SAW based on the feedback signal FB2. The feedback signal FB2 indicates the output voltage of the secondary winding 206. The comparator 314 compares the sawtooth wave signal SAW with the reference signal SET. The reference signal SET has a reference voltage level V _SET . Comparator 316 compares feedback signal FB1 with reference signal PEAK. The feedback signal FB1 indicates the current I _SE flowing through the primary winding 204. The reference signal PEAK determines the maximum current value I _PEAK of the current I _PR flowing through the primary winding 204. The pulse width modulation signal generator 318 is coupled to the comparator 314 and the comparator 316 and generates a pulse width modulation signal PWM1. The sawtooth signal SAW from the oscillator 226 can be used to control the duty cycle of the pulse width modulation signal PWM1. The pulse width modulation signal PWM1 controls the conduction state of the switch 318 to control the power of the transformer 202.

Controller 220 also includes a control signal generator 320. The control signal generator 320 generates a control signal CTRL based on the feedback signal FB2. Control signal CTRL is applied to oscillator 226. In an embodiment, if the voltage of the feedback signal FB2 is greater than the preset threshold TH (eg, TH>0V), the control signal CTRL is logic 1, otherwise the control signal CTRL is logic 0. In the example of FIG. 3, oscillator 226 includes current sources 302 and 304, switches 306 and 308, and capacitor 310. The voltage signal generated on the capacitor 310 is the sawtooth signal SAW. Depending on the conduction state of switches 306 and 308, capacitor 310 can be charged by current source 302 or discharged by current source 304.

If the voltage of the capacitor 310 rises to the reference voltage value V _SET , the controller 220 generates a pulse width modulation signal PWM1 having a first level to turn on the switch 218 (eg, PWM1 is a logic 1). In turn, the transformer 202 is operated in the charging phase T _ON . In one embodiment, the clamping circuit 228 forces the voltage of the feedback signal FB2 to be substantially zero, and the control signal CTRL has a first level (eg, a logic zero). Control signal CTRL controls switch 308 in oscillator 226. Control signal CTRL is coupled to switch 306 through NOT gate 312. In the example of FIG. 3, when control signal CTRL is logic 0, switch 306 is turned "on" and switch 308 is turned "off". Capacitor 310 is charged by the current of current source 302. Therefore, the voltage of the capacitor 310 (for example, the voltage of the sawtooth wave signal SAW) rises from the reference voltage value V _SET . At the same time, the current I _PR flowing through the primary winding 204 increases. Comparator 316 compares feedback signal FB1 with reference signal PEAK. When the voltage of the feedback signal FB1 reaches the voltage of the reference signal PEAK, the current I _PR flowing through the primary winding 204 is increased to the maximum current value I _PEAK , at which time the controller 220 _turns off the switch 318, thereby ending the charging phase T _{ON .} And start the discharge phase T _DIS . In particular, pulse width modulation signal generator 318 generates a pulse width modulation signal PWM1 having a second level to turn off switch 218 (eg, PWM1 is a logic 0). When the charging phase T _ON ends, the voltage of the capacitor 310 (for example, the voltage of the sawtooth wave signal SAW) rises to the first voltage value V ₁ as shown in FIG. In other words, the voltage of the capacitor 310 (for example, the voltage of the sawtooth wave signal SAW) is turned on from the reference voltage value V _SET to the first voltage value V ₁ .

During the discharge phase T _DIS , the switch 218 is turned off and the current I _SE flowing through the secondary winding 206 is reduced from the maximum current value I _SE-MAX . During the discharge phase T _DIS , the auxiliary winding 208 produces a substantially constant output voltage. This output voltage is divided by resistors 214 and 216. Therefore, in the discharge phase T _DIS , the voltage of the feedback signal FB2 (such as the voltage across the resistor 216) is proportional to the output voltage of the auxiliary winding 208, and is thus substantially constant. In one embodiment, the resistance of the resistor 214 and the resistor 216 are properly selected such that during the discharge phase T _DIS , the voltage of the feedback signal FB2 is greater than the preset threshold TH. When the voltage of the feedback signal FB2 is greater than the preset threshold TH, the control signal CTRL is logic 1, turning the switch 306 off and the switch 308 conducting. Capacitor 310 is discharged at the current of current source 304. Therefore, the voltage of the capacitor 310 drops from the first voltage value V ₁ .

When the voltage of the feedback signal FB2 representing the output voltage of the secondary winding 206 drops to the threshold value TH, indicating that the current I _SE flowing through the secondary winding 206 is reduced to a preset current value, the controller 220 ends the discharge phase. T _DIS starts the adjustment phase T _ADJ . In an embodiment, when the current I _SE flowing through the secondary winding 206 is reduced to substantially zero, the controller 220 ends the discharge phase T _DIS and initiates the adjustment phase T _ADJ . When the discharge phase T _DIS ends, the voltage of the capacitor 310 (for example, the voltage of the sawtooth wave signal SAW) drops to the second voltage value V ₂ as shown in FIG.

In the adjustment phase T _ADJ , since the voltage of the feedback signal FB2 drops to the threshold TH, the control signal CTRL becomes a logic 0. Switch 306 is turned "on" and switch 308 is turned "off". Capacitor 310 is again charged by the current of current source 302. The voltage of the capacitor 310 rises from the second voltage value V ₂ . During the adjustment phase T _ADJ , the switch 218 remains off and no current flows through the primary winding 204 or the secondary winding 206 . When the voltage of the sawtooth signal SAW rises to the reference voltage value V _SET , the controller 220 ends the adjustment phase T _ADJ and turns on the switch 218 to initiate the charging phase T _ON in the next switching cycle T _S . Specifically, the pulse width modulation signal generator 318 generates a pulse width modulation signal PWM1 having a first level to turn on the switch 218 (eg, PWM1 is a logic 1).

Assume that the capacitance value of the capacitor 310 is C ₁ , the current of the current source 302 is I ₁ , and the current of the current source 304 is I ₂ . At the end of the charging phase T _ON , the voltage of the sawtooth signal SAW (the voltage of the capacitor 310) can be expressed as:

At the end of the discharge phase T _DIS , the voltage of the sawtooth signal SAW can be expressed as:

At the end of the adjustment phase T _ADJ , the voltage of the sawtooth signal SAW can be expressed as:

Therefore, the length of the adjustment phase T _ADJ can be derived from equations (2)-(4), namely:

From equation (5), the relationship between the length of the adjustment phase T _{ADJ and} the length of the switching period T _S can be expressed as:

According to equation (6) can be obtained, the ratio of the length of time the total length of the discharge time period T _DIS the charging phase T _ON, and the discharging period T _DIS adjust the period T _{ADJ 1,} I ₂ determined by the current I. If the magnitudes of the currents I ₁ , I ₂ from the current sources 302, 304, respectively, are constant, the length of time of the discharge phase T _DIS is proportional to the length of time of the switching period T _S . Therefore, referring to equation (1), the average output current I _{OAVG of the} secondary winding 206 is substantially constant.

FIG. 5 shows a flowchart 500 of an exemplary method of controlling a transformer in accordance with an embodiment of the present invention. FIG. 5 will be described in conjunction with FIGS. 2, 3, and 4.

In step 502, transformer 202 operates for a plurality of switching cycles. One switching cycle includes a charging phase T _ON , a discharging phase T _{DIS ,} and an adjustment phase T _ADJ .

In step 504, the transformer 202 is powered with an input power source during the charging phase T _ON . In the charging phase T _ON , the switch 218 coupled in series with the primary winding 204 of the transformer 202 is turned on. In one embodiment, the length of time of the charging phase T _ON is controlled by monitoring the current flowing through the primary winding 204. Specifically, when the current flowing through the primary winding 204 increases to a preset maximum current value, the charging phase T _{ON is} ended (turning off the switch 218) and the discharging phase T _{DIS is} started.

At step 506, the load is supplied to the load using transformer 202 during the discharge phase T _DIS . In one embodiment, the length of time of the discharge phase T _DIS is controlled by monitoring the output voltage of the transformer 202 auxiliary winding 208. The output voltage of the auxiliary winding 208 can indicate whether the current flowing through the secondary winding 206 of the transformer 202 drops to a predetermined current value. Specifically, when the current flowing through the secondary winding 206 is reduced to a preset current value (eg, 0), the discharge phase T _{DIS is} ended and the adjustment phase T _{ADJ is} initiated. In an embodiment. When the output voltage of the auxiliary winding 208 is reduced to a predetermined voltage value, the current flowing through the secondary winding 206 is reduced to the preset current value.

In step 508, it determines the length of the time period T _ADJ adjusted such that the length of the discharge time period T _DIS with the charging phase T _ON, the ratio between the total duration of the discharge period T _DIS and adjusting period T _ADJ is constant. In one embodiment, the length of the adjustment phase T _ADJ is determined by an oscillator 226. The oscillator 226 generates a sawtooth wave signal SAW. During the charging phase T _ON , the voltage of the sawtooth signal SAW rises from the preset reference voltage value V _SET to the first voltage value V ₁ . In the discharge phase T _DIS , the voltage of the sawtooth wave signal SAW drops from the first voltage value V ₁ to the second voltage value V ₂ . In the adjustment phase T _ADJ , the voltage of the sawtooth signal SAW rises from the second voltage value V ₂ to a preset reference voltage value V _SET . When the voltage of the sawtooth signal SAW rises to the preset reference voltage value V _SET , the adjustment phase T _{ADJ is} ended and a new switching period T _{S is} initiated.

Accordingly, the present invention provides a circuit and method for controlling a power converter that can be used to power various loads. The power converter includes a transformer that operates over multiple switching cycles. At least one switching cycle includes a charging phase T _ON , a discharging phase T _{DIS ,} and an adjustment phase T _ADJ . The power converter can have an adjustment phase T _ADJ having a suitable length of time, such that the ratio of the length of the discharge phase T _{DIS to} the length of the switching period T _S is constant. The length of the switching period T _{S is} the total length of time in the charging phase T _ON , the discharging phase T _DIS and the tuning phase T _ADJ . Therefore, the average current output per switching cycle is substantially constant.

The power converter provided by the present invention can be applied to various occasions. In an embodiment, the power converter can provide a substantially constant current output to drive a light source such as a light emitting diode string. In another embodiment, the power converter can provide a substantially constant current to charge the battery. The power converter provided by the present invention is relatively small in size compared to conventional flyback converters including optocouplers and error amplifiers.

In addition, even if the change of the input voltage and the output voltage of the power converter may cause a change in the length of the charging phase T _ON and the discharging phase T _DIS , the power converter can automatically determine the length of the adjustment phase T _ADJ to maintain the discharging phase T The ratio of the length of the _{DIS to} the length of the switching period T _S is constant. Therefore, the power converter can automatically adjust to output a current having a constant average value. Moreover, it can be seen from equation (1) that the average value of the output current of the power converter is not affected by the inductance value of the transformer winding, thereby enabling more precise control of the output current.

FIG. 6 shows a schematic diagram of a load drive circuit 600 for driving a load 212 in accordance with one embodiment of the present invention. Elements in Figure 6 that are numbered the same as in Figure 2 have similar functions. In the example of FIG. 6, load drive circuit 600 is coupled to a power source 602 that produces an AC input voltage V _AC (eg, an AC voltage having a sinusoidal waveform). The load drive circuit 600 acts as a power converter that receives the AC input voltage V _AC and provides an output voltage V _OUT to power the load 212. The load 212 can be a light source (for example, a light emitting diode light source), but is not limited thereto.

In one embodiment, load drive circuit 600 includes a rectifier 603, a converter 604, a transformer 202, and a controller 620. In one embodiment, controller 620 includes 埠 VDD, 埠 DRV1, 埠 CS1, 埠 DRV2, 埠 CS2, and 埠 FB. The rectifier 603 rectifies the AC input voltage V _AC to provide a rectified voltage V _REC (eg, having a rectified sinusoidal waveform). Capacitor 605 acts as a filter to smooth the rectified voltage V _REC . Converter 604 is coupled between rectifier 603 and transformer 202 to convert rectified voltage V _REC to input voltage V _IN . In the embodiment of FIG. 6, converter 604 is a boost converter comprising an inductor L1, a diode D1, a capacitor C1, a resistor R1, and a switch 613. However, the present invention is not limited thereto, and the converter 604 may be of other types such as a buck converter or a buck-boost converter. Resistor R1 provides a monitor signal 656 indicative of the current flowing through inductor L1, and controller 620 receives monitor signal 656 via 埠CS1. Transformer 202 is powered by input voltage V _IN and produces an output voltage V _{OUT that} powers load 212. A capacitor 222 is coupled to the load 212 to filter out the ripple of the current I _LOAD flowing through the load 212. Controller 620 generates a switch control signal 654 on 埠DRV1 to regulate input voltage V _IN and a switch control signal 650 on 埠DRV2 to regulate current I _LOAD flowing through load 212.

In one embodiment, transformer 202 includes a primary winding 204, a secondary winding 206, an auxiliary winding 208, and a magnetic core 224. One end of primary winding 204 is coupled to converter 604 and the other end is coupled to ground via switch 218 and resistor 230. Secondary winding 206 is coupled to load 212 through diode 210 and capacitor 222. In one embodiment, one end of the auxiliary winding 208 is connected to ground through a resistor 614 and a resistor 616, and the other end is connected to ground. Controller The 埠FB of 620 is connected to a common node of resistor 614 and resistor 616.

Figure 7 is a waveform diagram of signals received or generated by the load drive circuit 600 of Figure 6 in accordance with one embodiment of the present invention. Figure 7 will be described in conjunction with Figure 6. The waveform shown in FIG. 7 sequentially shows the current I _PR flowing through the primary winding 204, the current I _SE flowing through the secondary winding 206, the voltage V _AUX of the non-identical end of the auxiliary winding 208, and the current flowing through the 埠FB of the controller 620. I _FB , voltage V _{FB at 埠 FB,} and switch control signal 650.

In one embodiment, controller 620 generates a switch control signal 650 to turn switch 218 on or off to operate transformer 202 for a plurality of cycles. In one embodiment, one cycle includes a charge phase T _ON and a discharge phase T _DIS . Alternatively, as in the embodiment shown in FIG. 7, one cycle includes a charging phase T _ON , a discharging phase T _{DIS ,} and an adjustment phase T _ADJ . In both cases, the switch control signal 650 turns the switch 218 on during the charging phase T _{ON and} turns off the switch 218 during the discharge phase T _DIS . Therefore, during the charging phase T _ON , the transformer 202 is powered by the input voltage V _IN and the current I _PR flowing through the primary winding 204 is increased. In one embodiment, during the charging phase T _ON , the resistor 230 generates a monitoring signal 652 indicative of the current I _PR . The 埠CS2 of the controller 620 receives the monitoring signal 652. During the discharge phase T _DIS , the transformer 202 is discharged to supply the load 212 and the current I _SE flowing through the secondary winding 206 is reduced.

During the charging phase T _ON and the discharging phase T _DIS , the transformer 202 provides a different feedback signal for the 埠 FB of the controller 620. Specifically, in one embodiment, during the charging phase T _ON , the voltage value V _{3 of the} voltage V _AUX is proportional to the voltage V _{IN of the} primary winding 204, which can be obtained from equation (7): V _AUX =V ₃ =−V _IN *(N _A /N _P ) (7)

Where N _A represents the number of turns of the auxiliary winding 208 and N _P represents the number of turns of the primary winding 204. As shown in equation (7), the voltage V _AUX is a negative voltage value during the charging phase T _ON . In one embodiment, controller 620 clamps voltage _{FB of}埠_FB to a predetermined voltage value (eg, 0 volts) to prevent voltage V _{FB from} dropping below zero volts. Thus, in one embodiment, during the charging phase T _ON , the voltage V _FB is equal to 0 volts. Therefore, current I _FB flows from 埠FB through resistor 614 to auxiliary winding 208. The current value I _{3 of the} current I _FB can be obtained by the equation (8): I _FB = I ₃ = V _IN * (N _A / N _P ) / R ₆₁₄ (8)

Where R ₆₁₄ represents the resistance of the resistor 614. Since (N _A /N _P )/R ₆₁₄ is a substantially constant constant, the current value I _{3 of the} current I _FB is proportional to the voltage V _IN .

During the discharge phase T _DIS , the auxiliary winding 208 senses the electrical state of the secondary winding 206. In particular, in one embodiment, when the current I _SE flowing through the secondary winding 206 decreases, the voltage V _AUX on the auxiliary winding 208 has a positive voltage value V ₄ (eg, V ₄ =V _OUT *(N) _A / N _S )), where N _S represents the number of turns of the secondary winding 206. When the current I _{SE is} reduced to a preset current value (eg, 0 amps), the voltage V _AUX produces a negative edge. Resistor 614 and resistor 616 divide voltage V _AUX to provide a voltage V _FB that is proportional to voltage V _AUX . Therefore, at the discharge phase T _DIS , the voltage V _{FB of 埠 FB} indicates whether the current I _SE flowing through the secondary winding 206 is reduced to a preset current value.

Therefore, during the charging phase T _ON , the current I _FB flowing through the 埠FB is proportional to the input voltage V _IN . In the discharge phase T _DIS , the voltage V _{FB of 埠 FB} indicates whether the current I _SE is reduced to a preset current value. Advantageously, the controller 620 receives the first feedback signal I _FB indicating the input voltage V _IN and the second feedback signal V _FB indicating the power state of the secondary winding 206 through the same _FB . Therefore, the number of turns of the controller 620 is reduced, thereby reducing the size and cost of the load drive circuit 600.

In one embodiment, the controller 620 according to the switch control signal 654 to control first port DRV1 feedback signal to regulate the voltage V _IN (e.g., the voltage V _IN is adjusted to the target voltage). In addition, controller 620 controls switch control signal 650 of 埠DRV2 based on the second feedback signal to adjust current I _LOAD (eg, to maintain current I _LOAD at a constant current value). The operation of controller 620 will be further described in FIG.

In one embodiment, capacitor 605 has a smaller capacitance value (eg, less than 0.5 microfarads) to help eliminate or reduce waveform distortion of rectified voltage V _REC (to correct the power factor of load drive circuit 600). The input voltage V _IN has a substantially constant voltage value through a converter 604 connected between the rectifier 603 and the transformer 202. Since the voltage V _{IN is} relatively stable, the ripple of the current I _LOAD is reduced.

FIG. 8 is a block diagram showing the structure of a controller 620 according to an embodiment of the present invention. Elements of the same reference numerals in Fig. 8 and Figs. 2, 3 and 6 have similar functions. FIG. 8 will be described in conjunction with FIGS. 3, 4, 6, and 7. As shown in FIG. 8, the controller 620 includes a voltage control unit 802 and a current control unit 804. The voltage control unit 802 monitors the current I _FB flowing through the 埠FB and generates a switch control signal 654 at the 埠DRV1 to regulate the input voltage V _IN . The current control unit 804 monitors the voltage V _FB on the 埠_FB and generates a switch control signal 650 at 埠 DRV 2 to regulate the output current I _OUT .

In one embodiment, current control unit 804 has a similar structure to controller 220 of FIG. The current control unit 804 includes a control signal generator 320, an oscillator 226, a comparator 314, a comparator 316, and a pulse width modulation signal generator 318. The control signal generator 320 generates a control signal CTRL based on the second feedback signal V _FB . The oscillator 226 receives the control signal CTRL and thereby generates a sawtooth wave signal SAW. The comparator 314 compares the sawtooth wave signal SAW with the reference signal SET. The comparator 316 compares the monitor signal 652 indicating the current I _PR of the charging phase T _ON with the reference signal PEAK1. The reference signal PEAK1 determines the peak current I _PEAK1 flowing through the primary winding 204. Pulse width modulated signal generator 318 is coupled to comparator 314 and comparator 316 and produces a switch control signal 650 (e.g., a pulse width modulated signal) to control switch 218.

Similar to the operation of controller 220, sawtooth signal SAW controls the duty cycle of pulse width modulation signal 650. Specifically, in conjunction with FIG. 3 and FIG. 4, in the charging phase T _ON , the sawtooth wave signal SAW increases from the voltage value V _{SET of the} reference signal SET, at which time the current I _PR flowing through the primary winding 204 increases. . When the voltage of the monitor signal 652 indicates that the current I _PR reaches the peak current I _PEAK1 (eg, when the sawtooth signal SAW reaches the voltage value V1), the switch control signal 650 turns off the switch 218 to end the charging phase T _ON and initiate the discharge phase. T _DIS . During the discharge phase T _DIS , the current I _SE flowing through the secondary winding 206 decreases, and the sawtooth wave signal SAW starts to decrease from the voltage value V1. When the current I _{SE is} reduced to a preset current value (eg, 0 amps) (ie, when the voltage V _FB produces a negative edge), the sawtooth signal SAW drops to a voltage value of V2. Thereby, the current control unit 804 ends the discharge phase T _DIS and starts the adjustment phase T _ADJ . In the adjustment phase T _ADJ , the sawtooth signal SAW rises from the voltage value V2. When the sawtooth signal SAW rises to the voltage value V _SET , the current control unit 804 turns on the switch 218 to start a new cycle.

Advantageous in that, according to equation (6), the current control unit 804 so that the length of the discharge time period T _DIS the charging phase T _ON, the ratio between the total length of the discharge time period T _DIS and adjusting period T _ADJ remains substantially constant Therefore, the current I _LOAD flowing through the load 212 remains substantially constant. The current control unit 804 can have other configurations and is not limited to the embodiment shown in FIG.

In one embodiment, voltage control unit 802 includes clamp circuit 810, current detector 808, and voltage regulator 818. As described in connection with FIG. 6, when switch 218 is closed, voltage V _AUX of auxiliary winding 208 is negative. The clamp circuit 810 is connected to the 埠FB, detects the voltage V _{FB of the 埠FB} , and clamps the voltage V _FB to a preset voltage value (for example, 0 volt) during the charging phase T _ON to prevent the voltage V _{FB from} dropping to 0 volts. the following. Thereby, the current I _FB flows from the current detector 808 through the clamp circuit 810 to the 埠FB.

In one embodiment, current detector 808 includes current mirror 812, resistor 814, and sample and hold circuit 816. Current mirror 812 mirrors current I _FB to produce a current I _{M that is} equal or proportional to current I _FB . Current I _M flows through resistor 814, so voltage V _M across resistor 814 is also proportional to current I _FB . According to equation (8), during the charging phase T _ON , the current I _FB is proportional to the input voltage V _IN . Therefore, the voltage V _M is proportional to the input voltage V _IN . Sample and hold circuit 816 samples the voltage V _M at the charging period T _ON, and the holding voltage V _H to produce the holding sampled values of the charging period T _ON of the voltage V _M. Therefore, during the discharge phase T _DIS and the adjustment phase T _ADJ , although the current I _FB drops to 0 amps, the hold voltage V _H still indicates the value of the input voltage V _IN .

For example, voltage regulator 818 includes error amplifier 820, comparator 822, comparator 823, or gate 828, oscillator 824, and pulse width modulation signal generator 826. The oscillator 824 generates a sawtooth signal V _SAW and a clock signal 850 (eg, a pulse signal). One input of the error amplifier 820 receives the reference signal V _REF indicating the target value of the input voltage V _IN , and the other input receives the hold voltage V _H . The error amplifier 820 amplifies the difference between the hold voltage V _H and the reference signal V _REF to generate an amplified voltage V _AMP . The comparator 822 compares the sawtooth wave voltage V _SAW with the amplification voltage V _AMP to generate a comparison voltage V _C1 . Comparator 823 will indicate the peak 656 and the monitoring current I _IND signal indicating a reference signal V _PEAK current flowing through the inductor L1 I _IND compared to generate comparison voltage V _C2. The OR gate 828 receives the comparison voltage V _C1 and the comparison voltage V _C2 and thereby generates a control signal 852.

The pulse width modulation signal generator 826 generates a switch control signal 654 based on the clock signal 850 and the control signal 852 to control the switch 613 to adjust the input voltage V _IN . In one embodiment, pulse width modulation signal generator 826 turns on switch 613 based on clock signal 850 and opens switch 613 in accordance with control signal 852. In particular, in one embodiment, the clock signal 850 is a pulse signal having a substantially constant frequency. Therefore, the on-time and off-time of the switch 613 are also substantially constant. Further, the hold voltage V _H indicating the input voltage V _IN determines the on-time of the switch 613. Therefore, the duty cycle of the switch control signal 654 is determined by the hold voltage V _H . For example, if the hold voltage V _{H is} greater than the reference voltage V _REF , it indicates that the input voltage V _{IN is} greater than the target voltage value (the target voltage value is indicated by the reference voltage V _REF ), and the duty cycle of the switch control signal 654 is reduced to reduce the input. Voltage V _IN . Conversely, when the hold voltage V _{H is} less than the reference voltage V _REF , it indicates that the input voltage V _{IN is} less than the target voltage value, and the duty cycle of the switch control signal 654 is increased to increase the input voltage V _IN . Therefore, the input voltage V _{IN is} adjusted to the target voltage value.

In one embodiment, the current I _IND flowing through the inductor L1 of the converter 604 has the function of overcurrent protection. For example, if the monitor signal 656 is greater than the reference voltage V _PEAK , then the current I _{IND is} greater than the peak current and the switch control signal 654 is turned off the switch 613. The voltage control unit 802 can have other configurations and is not limited to the embodiment shown in FIG.

FIG. 9 shows an operational flow diagram 900 of a load drive circuit (e.g., load drive circuit 600) in accordance with one embodiment of the present invention. Figure 9 will be described in conjunction with Figures 6-8. Although FIG. 9 discloses specific steps, these steps are merely illustrative, and the present invention is equally applicable to other steps or to some of the steps of the steps illustrated in FIG.

In step 902, the transformer (e.g., transformer 202) operates for a plurality of cycles. In one embodiment, one cycle includes a charge phase and a discharge phase. In another embodiment, one cycle includes a charge phase, a discharge phase, and an adjustment phase.

In step 904, during the charging phase, the transformer is powered by the input voltage and the current flowing through the primary winding of the transformer increases.

In step 906, during the discharge phase, the transformer discharges to supply power to the load, and the current flowing through the secondary winding of the transformer decreases.

In step 908, during the charging phase, the voltage of the controller 埠 (eg, 埠FB) connected to the auxiliary winding of the transformer is clamped at a preset voltage value (eg, 0 volts).

In step 910, during the charging phase, the 埠FB receives a first feedback signal indicative of the input voltage. In one embodiment, the first feedback signal includes a current flowing through 埠FB (eg, current I _FB flowing through 埠_FB ).

In step 912, the 埠FB receives a second feedback signal indicative of the secondary winding power state. In one embodiment, the second feedback signal includes the voltage of 埠FB (eg, voltage V _{FB of 埠FB} ).

The above detailed description and the accompanying drawings are only typical embodiments of the invention. It is apparent that various additions, modifications and substitutions are possible without departing from the spirit and scope of the invention as defined by the appended claims. It should be understood by those skilled in the art that the present invention may be changed in form, structure, arrangement, ratio, material, element, element, and other aspects without departing from the scope of the invention. Therefore, the embodiments disclosed herein are intended to be illustrative and not restrictive, and the scope of the invention is defined by the appended claims

100‧‧‧Reverse converter

104‧‧‧Primary winding

106‧‧‧Secondary winding

108‧‧‧Auxiliary winding

110‧‧‧ diode

112‧‧‧load

114‧‧‧Error amplifier

116‧‧‧Optocoupler

111‧‧‧current monitoring resistor

118‧‧‧ switch

120‧‧‧ Controller

122‧‧‧ Capacitance

124‧‧‧ magnetic core

200‧‧‧Power Converter

202‧‧‧Transformer

204‧‧‧Primary winding

206‧‧‧Secondary winding

208‧‧‧Auxiliary winding

210‧‧‧ diode

212‧‧‧load

214, 216‧‧‧ resistance

218‧‧‧ switch

220‧‧‧ Controller

222‧‧‧ Capacitance

224‧‧‧ magnetic core

226‧‧‧Signal Generator/Oscillator

228‧‧‧Clamp circuit

230‧‧‧resistance

302, 304‧‧‧ Current source

306, 308‧‧ ‧ switch

310‧‧‧ Capacitance

312‧‧‧NOT gate

314, 316‧‧‧ comparator

318‧‧‧Pulse signal generator / pulse width modulation signal generator

320‧‧‧Control signal generator

500‧‧‧flow chart

502, 504, 506, 508‧ ‧ steps

600‧‧‧Load Drive Circuit

602‧‧‧Power supply

603‧‧‧Rectifier

604‧‧‧ converter

605‧‧‧ Capacitance

613‧‧‧ switch

614‧‧‧resistance

616‧‧‧resistance

620‧‧‧ Controller

650‧‧‧Switch control signal

652‧‧‧Monitoring signal

654‧‧‧Switch control signal

656‧‧‧Monitoring signal

802‧‧‧Voltage Control Unit

804‧‧‧ Current Control Unit

808‧‧‧current detector

810‧‧‧Clamp circuit

812‧‧‧current mirror

814‧‧‧resistance

816‧‧‧Sampling and holding circuit

818‧‧‧Voltage regulator

820‧‧‧Error amplifier

822‧‧‧ comparator

823‧‧‧ comparator

824‧‧‧Oscillator

826‧‧‧ Pulse width modulation signal generator

828‧‧‧ or gate

850‧‧‧ clock signal

852‧‧‧Control signal

900‧‧‧Flowchart

902, 904, 906, 908, 910, 912 ‧ ‧ steps

The technical method of the present invention will be described in detail below in conjunction with the accompanying drawings and specific embodiments to make the features and advantages of the present invention more obvious. Wherein: Figure 1 shows a block diagram of a conventional flyback converter.

2 is an illustrative block diagram of a power converter in accordance with an embodiment of the present invention.

FIG. 3 is a schematic diagram showing the architecture of the controller in FIG. 2.

4 is an illustrative waveform diagram of signals received or generated by a power converter in accordance with an embodiment of the present invention.

FIG. 5 is a flow chart showing an exemplary method of controlling a transformer in accordance with an embodiment of the present invention.

Figure 6 is a schematic illustration of a load drive circuit in accordance with one embodiment of the present invention.

Figure 7 shows the load shown in Figure 6 in accordance with one embodiment of the present invention. A waveform diagram of a signal received or generated by a driver circuit.

Figure 8 is a block diagram showing the structure of a controller in accordance with one embodiment of the present invention.

Figure 9 is a flow chart showing the operation of a load driving circuit in accordance with one embodiment of the present invention.

600‧‧‧Load Drive Circuit

602‧‧‧Power supply

603‧‧‧Rectifier

604‧‧‧ converter

605‧‧‧ Capacitance

613‧‧‧ switch

614‧‧‧resistance

616‧‧‧resistance

620‧‧‧ Controller

650‧‧‧Switch control signal

652‧‧‧Monitoring signal

654‧‧‧Switch control signal

656‧‧‧Monitoring signal

Claims (27)

A load driving circuit comprising: a transformer comprising a primary winding receiving an input voltage and a primary winding connected to a load, the transformer operating in a plurality of cycles, wherein one of the plurality of cycles comprises a charging a phase and a discharge phase, in which the transformer is powered by the input voltage, and a current flowing through the primary winding is increased, during which the transformer discharges to supply power to the load, and flows through the current a current reduction of the stage winding; and a controller coupled to the transformer, including a turn, wherein the pick-up receives a first feedback signal indicative of the input voltage during the charging phase, and wherein a second feedback signal indicating a power state of the one of the secondary windings, wherein the controller generates a first control signal according to the first feedback signal to adjust the input voltage, and The second feedback signal generates a second control signal to regulate a current flowing through the load. The load driving circuit of claim 1, further comprising: a converter coupled between a power source and the primary winding, the converter converting an input AC voltage generated by the power source into the input voltage, And adjusting the input voltage according to the first control signal. The load drive circuit of claim 1, wherein the transformer further comprises: an auxiliary winding coupled to the 埠, during the charging phase, a voltage on the auxiliary winding and the input voltage on the primary winding Proportionate. The load driving circuit of claim 3, further comprising: a resistor coupled between the auxiliary winding and the crucible, wherein the controller clamps a voltage of the crucible to a preset during the charging phase A voltage value, and during the charging phase, a current value flowing through the resistor is proportional to the input voltage. The load drive circuit of claim 1, wherein the transformer further comprises: an auxiliary winding coupled to the 埠, wherein a voltage of the auxiliary winding indicates a current flowing through the secondary winding during the discharging phase Whether it falls to a preset value. The load driving circuit of claim 1, wherein the first feedback signal comprises a current flowing through the crucible, and the second feedback signal comprises a voltage on the crucible. The load driving circuit of claim 1, wherein the controller further comprises: a current mirror coupled to the cymbal, wherein a current flowing through the cymbal is mirrored during the charging phase to provide a current flowing through the resistor And a sample-and-hold circuit coupled to the current mirror to sample and maintain a voltage across the resistor to generate a hold signal indicative of the input voltage. The load driving circuit of claim 1, wherein the cycle further includes an adjustment phase, the second control signal causing a length of the discharge phase and the charging phase, the discharging phase, and the adjusting phase A ratio between the total length of time remains constant to adjust the current flowing through the load. The load drive circuit of claim 8, wherein the controller further comprises: a current control unit, when the second feedback signal indicates that the current flowing through the secondary winding drops to a predetermined current value The current control unit ends the discharge phase and initiates the adjustment phase. The load driving circuit of claim 8, wherein the controller further comprises: a signal generator for generating a sawtooth wave signal, wherein the sawtooth wave signal rises from a predetermined reference voltage value during the charging phase Go to a first voltage value, and in the discharging phase, the sawtooth wave signal drops from the first voltage value to a second voltage value, and in the adjusting phase, the sawtooth wave signal rises from the second voltage value to the The preset reference voltage value. The load driving circuit of claim 10, further comprising: a switch coupled in series to the primary winding, wherein the sawtooth wave signal rises from the preset reference voltage value to the first voltage value The controller turns on the switch, and wherein the controller turns off the switch when the current flowing through the primary winding increases to a predetermined peak current. The load driving circuit of claim 8 , further comprising: a switch coupled in series to the primary winding, wherein the controller turns on the switch during the charging phase, and disconnects the discharging phase and the adjusting phase switch. A power converter comprising: A transformer comprising a primary winding receiving an input voltage, a primary winding connected to a load, and an auxiliary winding, the transformer operating in a plurality of cycles, wherein one of the plurality of cycles includes a charging phase and a discharge phase in which the transformer is powered by the input voltage and a current flowing through the primary winding increases, and during the discharge phase, the transformer discharges to power the load and flow through the secondary a current of the winding is reduced; a pair of resistors connected in series with each other and the pair of resistors connected to the auxiliary winding; and a controller including a turn connected to a common node between the pair of resistors during the charging phase The controller clamps a voltage on the common node to a predetermined voltage value. The power converter of claim 13, wherein in the charging phase, a current flowing through the crucible is proportional to the input voltage; in the discharging phase, the voltage of the crucible is indicative of flowing through the secondary winding Whether the current drops to a preset value. The power converter of claim 14, wherein in the discharging phase, the pair of resistors divides a voltage on the auxiliary winding to supply the divided voltage to the 埠. The power converter of claim 14, wherein the pair of resistors comprises a first resistor connected to the auxiliary winding and a second resistor connected to a node having a reference voltage, and flowing through the cathode The current also flows through the secondary winding and the first resistor. The power converter of claim 14, wherein the controller further comprises: a current detector coupled to the NMOS, wherein the current detector mirrors the current flowing through the , to provide a current flowing through a third resistor, and the current detector maintains the current through sampling A voltage across the three resistors provides a hold signal, and the controller generates a control signal based on the hold signal to adjust the input voltage. The power converter of claim 13, wherein the cycle further comprises an adjustment phase, the controller causing a length of the discharge phase and the total duration of the charging phase, the discharging phase, and the adjusting phase A ratio between them remains constant to adjust a current flowing through the load. The power converter of claim 18, wherein the controller further comprises: a signal generator for generating a sawtooth signal, wherein the sawtooth signal rises from a predetermined reference voltage value to the charging phase a first voltage value, and in the discharging phase, the sawtooth wave signal drops from the first voltage value to a second voltage value, and in the adjusting phase, the sawtooth wave signal rises from the second voltage value to the pre- Set the reference voltage value. The power converter of claim 19, further comprising: a switch coupled in series to the primary winding, wherein the sawtooth signal rises from the predetermined reference voltage value to the first voltage value The controller turns on the switch, and when the current flowing through the primary winding increases to a predetermined peak current, the controller turns off the switch. For example, the power converter of claim 18, further comprising: A switch coupled in series to the primary winding, wherein the controller turns the switch on during the charging phase, and wherein the controller turns off the switch during the discharging phase and the adjusting phase. A controller for controlling a load to supply a load, comprising: a first turn generating a first control signal to adjust an input voltage of the transformer; and a second turn generating a second control signal to adjust a current flowing through the load and operating the transformer in a plurality of cycles, wherein one of the plurality of cycles includes a charging phase and a discharging phase, wherein the transformer is powered by the input voltage And a current flowing through the primary winding increases; during the discharging phase, the transformer discharges to supply power to the load, and a current flowing through the secondary winding decreases; and a third turn is coupled to the transformer An auxiliary winding, in the charging phase, the third receiving a first feedback signal indicating the input voltage; in the discharging phase, the third receiving a second indicating a state of the electrical energy of the secondary winding The feedback signal is generated, wherein the controller generates the first control signal according to the first feedback signal, and generates the second control signal according to the second feedback signal. The controller of claim 22, further comprising: a current mirror coupled to the third turn, wherein the current mirror mirrors a current flowing through the third turn to provide a flow through the third pass a current of the resistor; and a sample and hold circuit coupled to the current mirror, sampling and maintaining a voltage on the resistor to generate a hold signal indicating the input voltage number. The controller of claim 22, wherein the first feedback signal comprises the current flowing through the third turn, and the second feedback signal comprises a voltage on the third turn. The controller of claim 22, wherein the cycle further comprises an adjustment phase, the controller causing a length of the discharge phase and the total duration of the charging phase, the discharging phase, and the adjusting phase A ratio between the two remains constant to adjust a current flowing through the load. The controller of claim 25, further comprising: a signal generator for generating a sawtooth wave signal, wherein the sawtooth wave signal rises from a predetermined reference voltage value to a first voltage value during the charging phase And in the discharging phase, the sawtooth wave signal drops from the first voltage value to a second voltage value, and in the adjusting phase, the sawtooth wave signal rises from the second voltage value to the preset reference voltage value. The controller of claim 22, further comprising: a clamp circuit, wherein the clamp circuit clamps a voltage on the third turn to a predetermined voltage value.