201225594 六、發明說明： 【發明所屬之技術領域】 本發明的實施形態係有關於OFDM收訊裝置中的傳輸 路徑響應之推定。 【先前技術】 在地表波數位播送中是採用了，使用彼此正交之複數 載波的正交分頻多工方式（以下稱作OFDM方式）來作爲調 變方式。 一般來說，OFDM方式的訊號格式，係在傳輸訊號中 除了資料訊號以外還多工了分散導頻訊號（以下稱作SP訊 號），將其在時間方向及頻率方向上進行內插，使用內插 過的SP訊號來推定傳輸路徑響應，進行多重路徑失真等 之等化。 SP訊號，在被從播送台發送的狀態下，是在傳輸訊 號中朝時間方向及頻率方向分別隔開所定符號數而被間歇 性插入。在收訊裝置上，係從被送來之傳輸訊號中抽出 SP訊號，針對該SP訊號朝時間方向及頻率方向使用內插 濾波器而對資料訊號的所有符號進行內插，使用內插過的 所有SP訊號來推定傳輸路徑響應》 SP訊號的內插濾波器，係可讓含有多重路徑所致之 延遲時間較長的延遲波（稱作長延遲波）的較多延遲波成分 通過’同時可去除雜訊成分，較爲理想。 -5- 201225594 【發明內容】 [發明所欲解決之課題] 本發明所欲解決之課題在於提供一種，在多重路徑時 的SP訊號之內插濾波器中可讓含有長延遲波之延遲波成 分通過，同時提高雜訊去除性能而謀求收訊性能之提升的 OFDM收訊裝置。 [用以解決課題之手段] 實施形態的OFDM收訊裝置，係具備：濾波器部，係 使用複數濾波器特性而將傳輸路徑推定訊號在頻率方向上 進行頻帶限制：和等化部，係使用前記濾波器部之輸出而 將收訊訊號進行等化：和品質偵測部，係偵測出前記等化 部之輸出的收訊品質；和判定部，係使用所偵測出之收訊 品質而從前記複數濾波器特性中判定出最佳者；前記複數 濾波器特性係具有：所定帶寬之濾波器特性、和讓前記所 定帶寬內之一部分透通之複數濾波器特性；前記部分透通 之複數濾波器特性係含有2個以上之通過頻帶。 [發明效果] 若依據上記構成之OFDM收訊裝置，則可在多重路徑 時的SP訊號之內插濾波器中可讓含有長延遲波之延遲波 成分通過，同時提高雜訊去除性能而謀求收訊性能之提升 201225594 【實施方式】 以下’參照圖面而詳細說明實施形態。 [第1實施形態] 圖1係第1實施形態之OFDM收訊裝置的區塊圖。 於圖1中，OFDM收訊裝置100係具備有··天線1、 選台器2、A/D轉換器3、IQ解調電路4、FFT電路5、 FFT窗控制電路6、第1等化電路7、第1之SP時間內插 濾波器8、第1之SP頻率內插瀘波器9、錯誤訂正電路 1 〇、第1係數切換電路1 1、係數判定電路1 2。 OFDM收訊裝置100係例如，將依照OFDM方式之傳 輸訊號（以下稱作OFDM訊號），透過無線傳輸路徑而加以 收訊。此外，亦可透過有線傳輸路徑而收訊。 選台器2，係將已被天線1所接收之RF訊號，頻率 轉換成IF訊號，將IF訊號輸出至A/D轉換電路3。A/D 轉換電路3係對從選台器2所供給之IF訊號實施A/D轉 換，將數位的IF訊號輸出至IQ解調電路4。 IQ解調電路4，係藉由進行正交解調，而從A/D轉換 電路3所供給的IF訊號中，取得時間領域OFDM訊號。 IQ解調電路4係將時間領域OFDM訊號’輸出至FFT電 路5及FFT窗控制電路6。 FFT電路5，係基於從FFT窗控制電路6所供給之 FFT窗控制訊號，而從1個OFDM符號之訊號中’抽出有 效符號長之範圍的訊號。又，FFT電路5係藉由對已抽出 201225594 的時間領域之OFDM訊號進行FFT演算，而生成頻率領域 之OFDM訊號，輸出至第1等化電路7及第1之SP時間 內插濾波器8。 FFT窗控制電路6，係從收訊訊號之中偵測出主波的 時序，以其爲基準而使FFT輸出成爲最佳的方式，偵測出 FFT窗位置。FFT電路5係依照FFT窗位置而將時間軸的 OFDM訊號轉換成頻率軸上的訊號。FFT電路5的輸出係 爲圖2之訊號格式所示的訊號波配置。 在圖2所示的OFDM訊號的訊號格式例子中，包含有 :資訊符號S1、表示OFDM訊號之傳輸方式等的TMCC/AC 符號S2、表示OFDM訊號之尾端的CP符號S3、屬於SP 訊號之符號的SP符號S4之各符號。SP符號S4爲係例如 ，在頻率方向上爲1/3、及在時間方向上爲1/4的比率， 而被插入。201225594 VI. Description of the Invention: TECHNICAL FIELD OF THE INVENTION Embodiments of the present invention relate to estimation of a transmission path response in an OFDM receiving apparatus. [Prior Art] In the terrestrial wave digital broadcasting, an orthogonal frequency division multiplexing method (hereinafter referred to as OFDM method) of complex carriers orthogonal to each other is used as the modulation method. In general, the signal format of the OFDM system is multiplexed with a scattered pilot signal (hereinafter referred to as an SP signal) in addition to the data signal, and is interpolated in the time direction and the frequency direction. The inserted SP signal estimates the transmission path response, and equalizes the multipath distortion and the like. In the state of being transmitted from the broadcast station, the SP signal is intermittently inserted in the transmission signal by the predetermined number of symbols in the time direction and the frequency direction. On the receiving device, the SP signal is extracted from the transmitted transmission signal, and all symbols of the data signal are interpolated using the interpolation filter for the SP signal in the time direction and the frequency direction, and the interpolated signal is used. All SP signals are used to estimate the interpolation filter of the transmission path response SP signal, which allows more delayed wave components of the delayed wave (called long delay wave) with longer delay time caused by multiple paths to pass through. It is ideal to remove noise components. [Embodiment] [Problem to be Solved by the Invention] An object of the present invention is to provide a delay wave component containing a long delay wave in an interpolation filter of an SP signal in a multipath path. An OFDM receiving apparatus that improves the reception performance by improving the noise removal performance. [Means for Solving the Problem] The OFDM receiving apparatus according to the embodiment includes a filter unit that uses a complex filter characteristic to perform band limitation on a frequency direction in a channel estimation signal and an equalization unit. The output of the filter unit is equalized by the output of the filter unit: the quality detection unit detects the reception quality of the output of the preamplifier, and the determination unit uses the detected reception quality. And the best one is determined from the characteristics of the complex filter; the complex filter characteristic has a filter characteristic of a predetermined bandwidth and a complex filter characteristic that allows a portion of the bandwidth defined by the preamble to pass through; The complex filter characteristic contains two or more passbands. [Effect of the Invention] According to the OFDM receiving apparatus configured as above, the interpolation signal of the SP signal in the multipath can be passed through the delay wave component including the long delay wave, and the noise removal performance can be improved. Improvement of Communication Performance 201225594 [Embodiment] Hereinafter, embodiments will be described in detail with reference to the drawings. [First Embodiment] Fig. 1 is a block diagram of an OFDM reception device according to a first embodiment. In FIG. 1, an OFDM receiving apparatus 100 includes an antenna 1, a channel selector 2, an A/D converter 3, an IQ demodulation circuit 4, an FFT circuit 5, an FFT window control circuit 6, and a first equalization. The circuit 7, the first SP time interpolation filter 8, the first SP frequency interpolation chopper 9, the error correction circuit 1 〇, the first coefficient switching circuit 1 1 and the coefficient decision circuit 12 are provided. The OFDM receiving apparatus 100 transmits, for example, a transmission signal (hereinafter referred to as an OFDM signal) according to the OFDM scheme through a wireless transmission path. In addition, it can also be received via a wired transmission path. The selector 2 converts the RF signal received by the antenna 1 into an IF signal, and outputs the IF signal to the A/D conversion circuit 3. The A/D conversion circuit 3 performs A/D conversion on the IF signal supplied from the selector 2, and outputs the digital IF signal to the IQ demodulation circuit 4. The IQ demodulation circuit 4 obtains the time domain OFDM signal from the IF signal supplied from the A/D conversion circuit 3 by performing quadrature demodulation. The IQ demodulation circuit 4 outputs the time domain OFDM signal ' to the FFT circuit 5 and the FFT window control circuit 6. The FFT circuit 5 extracts a signal of a range of effective symbol lengths from the signal of one OFDM symbol based on the FFT window control signal supplied from the FFT window control circuit 6. Further, the FFT circuit 5 generates an OFDM signal in the frequency domain by performing FFT calculation on the OFDM signal of the time domain of 201225594, and outputs it to the first equalizing circuit 7 and the first SP time interpolation filter 8. The FFT window control circuit 6 detects the timing of the main wave from the received signal, and uses the FFT output as the reference to determine the FFT window position. The FFT circuit 5 converts the OFDM signal of the time axis into a signal on the frequency axis in accordance with the FFT window position. The output of the FFT circuit 5 is the signal wave configuration shown in the signal format of Figure 2. In the signal format example of the OFDM signal shown in FIG. 2, the information symbol S1, the TMCC/AC symbol S2 indicating the transmission mode of the OFDM signal, the CP symbol S3 indicating the end of the OFDM signal, and the symbol belonging to the SP signal are included. Each symbol of the SP symbol S4. The SP symbol S4 is inserted, for example, at a ratio of 1/3 in the frequency direction and 1/4 in the time direction.
第1之SP時間內插濾波器8，係從頻率領域之OFDM 訊號中抽出SP訊號，對SP訊號進行時間方向的內插，將 其輸出至第1之SP頻率內插濾波器9。第1之SP頻率內 插濾波器9，係對在時間方向上做過內插的SP訊號，再 進行頻率方向的內插，藉由該時間方向及頻率方向上被內 插過的SP訊號，而取得對應於全部資料的傳輸路徑響應 〇 第1之SP時間內插濾波器8，係如圖2所示，SP訊 號是以每4符號中出現1個的比率而存在，因此其他3符 號則是例如進行線性內插（將SP間的時間差予以等分的內 -8 - 201225594 插）而將値予以放入。 第1之SP頻率內插濾波器9，係如後述，使用根據 OFDM訊號的訊號品質偵測結果所決定的最佳濾波器係數 ，將SP訊號的頻帶對每一符號做限制，藉此以將SP訊號 在頻率方向上進行內插。 此處，SP訊號係爲相位或功率是被預先決定的已知 訊號，是爲了求出用來推定傳輸訊號失真所需的傳輸路徑 響應，而被當成傳輸路徑推定符號來使用。 第1等化電路7,係使用SP訊號所致之傳輸路徑推 定訊號，而將頻率領域之OFDM訊號進行等化。第1等化 電路7的輸出，係被供給至錯誤訂正電路10,進行錯誤訂 正之解碼處理而解碼出收訊資料。 將第1之SP頻率內插濾波器9的濾波器係數加以切 換的第1係數切換電路11，係具備：將圖3(a)的複數平移 模態予以依序切換而設定的通過頻帶平移部11a、和將圖 3(b)的複數通過頻帶模態予以依序切換而設定的通過頻帶 模態選擇部1 1 b。 係數判定電路12係具備：第2之SP頻率內插濾波器 125、第2係數切換電路124、第2等化電路121、訊號品 質偵測電路122、控制部123 » 將第2之SP頻率內插濾波器125的濾波器係數加以 切換的第2係數切換電路124,係具備：將和圖3 (a)同樣 的複數平移模態予以依序切換而設定的通過頻帶平移部 124a、與將和圖3(b)同樣的複數通過頻帶模態予以依序切 201225594 換而設定的通過頻帶模態選擇部124b。 圖3係第1實施形態中的濾波器係數控制之一例的說 明圖。 第1係數切換電路1 1，係如圖3所示，係使得以主波 位置爲基準而將中心頻率予以平移而成的複數濾波器特性 、和從已被平移之濾波器特性之中可讓含有主波位置之一 部分頻帶透通之複數濾波器特性，變成可被切換：會依照 從係數判定電路12所供給之平移量與通過頻帶模態而作 動。 係數判定電路12，係會依序產生圖3所示的濾波器係 數（濾波器模態），從使用各個頻率內插之等化輸出的調變 誤差比（以下稱作MER)，偵測出收訊品質，判定收訊品質 最佳的濾波器係數，將平移量與通過頻帶模態的判定結果 ，供給至第1係數切換電路11。 此外，於圖3(a)及（b)中，將時間軸上的主波與延遲波 (及先行波）加以表示的圖（橫軸表示延遲時間、縱軸表示功 率的圖），係並非表示實際偵測延遲側寫之結果，而是針 對訊號位準最大的主波，例如設置閾値，若將超過閾値者 實際偵測成爲主波位準，則可以想定在以該主波位置爲中 心的前後時間位置上，會有先行波或延遲波存在。因此， 在含有主波所存在之窄頻帶（以下稱作主波頻帶）的所定帶 寬內，例如想定有延遲波存在的情況，也是假想性地圖示 在時間軸上。由於相對於主波而言延遲波的存在較容易分 辨，因此變成類似於延遲側寫的圖，但在實施形態完全沒 -10- 201225594 有進行延遲側寫偵測，是將所能想到的眾多濾波器模態藉 由逐次改變濾波器係數而加以設定，作動成會決定出收訊 品質最佳的濾波器模態（濾波器特性）。這點在後述的圖6 及圖8中也是同樣如此。 關於主波，是針對在相同播送頻道中，包含主波、延 遲波及先行波的所有訊號波中，將功率位準最高的訊號波 ，定義爲主波。 又，在如圖3(a)般地使所定帶寬之平移模態往頻率方 向上移動後，如圖3(b)般地讓所定帶寬內之一部分透通的 帶有2個以上之通過頻帶之最佳濾波器特性的生成方法係 爲，從與主波頻帶相同之窄帶寬的通過頻帶模態是複數（ 圖3(b)中係爲2個）重疊的狀態起，以該主波頻帶之通過 頻帶模態爲中心而使重疊的另1個通過頻帶模態逐次遠離 的方式，換言之，控制濾波器係數而將重疊的另1個通過 頻帶模態從主波頻帶之通過頻帶模態的中心起分離開來的 方式，而生成帶有2個通過頻帶的濾波器特性（例如通過 頻帶模態6)。亦即，針對含有藉由係數控制而使複數窄帶 寬之濾波器頻帶在頻率方向上作爲濾波器模態而依序逐一 展開之過程之模態的所有濾波器模態做嘗試，決定出可讓 主波及延遲波順利通過濾波器通過頻帶而獲得良好訊號品 質結果的據波器模態。适點在後述的圖6(b)及圖8(b)中也 是同樣如此。 接著，參照圖3，再來說明係數判定電路丨2的動作。 圖3 (a)及（b)係爲第1實施形態中的濾波器係數控制之一例 -11 - 201225594 的說明圖。圖3(a)係表示將平移量逐次改變而嘗試複數平 移模態的動作，圖3(b)係表示在圖3(a)中選擇了平移模態 1後，在由該平移模態1所決定之所定帶寬內，逐次改變 通過頻帶模態而從後續的通過頻帶做嘗試，一直嘗試到含 有獨立的2個通過頻帶之通過頻帶模態爲止的動作。 於係數判定電路1 2中，控制部1 23係藉由濾波器係 數控制，首先會使圖3(a)所示的中心頻率做了平移而成的 濾波器模態1〜7之濾波器特性，被依序產生。當圖3所 示的主波及延遲波之延遲差的2波多重路徑被輸入時，2 波被收斂在濾波器通過頻帶內的平移模態1時，訊號品質 偵測電路1 22上的MER爲最小，而被判定爲收訊品質最 佳。接著，相對於平移模態1，如圖3(b)所示，依序產生 讓主波及其他一部分頻帶透通的複數通過頻帶模態1〜6 之濾波器特性。當如圖3(b)所示的2波多重路徑時，相較 於讓全體頻帶透通的平移模態1，僅讓主波及延遲波附近 透通的通過頻帶模態6的雜訊去除能力較高，因此訊號品 質偵測電路1 22上的MER較小，而最終被判定爲收訊品 質最佳。 圖4係爲關於可讓相對於主波而延遲差較大之延遲波 成分（稱作長延遲之延遲波成分）透通之廣頻帶之濾波器的 說明圖，圖5係爲關於可讓相對於主波而延遲差較小之延 遲波成分（稱作短延遲之延遲波成分）透通之窄頻帶之濾波 器的說明圖。 圖4(a)係圖示相對於主波而延遲差較大之延遲波的關 -12- 201225594 係，圖4(b)係圖示具有如圖4(a)之長延遲之延遲波成分的 傳輸訊號的頻率特性。此時的頻率特性係爲節拍的間隔較 短，以較快週期而變動之特性。因此爲了要能支援到此種 長延遲之延遲波，必須要有包含高頻率頻帶的廣頻帶之濾 波器。可是，若單純使用廣頻帶之濾波器，則會產生雜訊 成分增加之問題點。 圖5 (a)係圖示相對於主波而延遲差較小之延遲波成分 (稱作短延遲之延遲波成分），圖5(b)係圖示具有如圖5(a) 之短延遲之延遲波成分的傳輸訊號之頻率特性。此時的頻 率特性係爲節拍的間隔較長，以較慢的週期而變動之特性 。因此，此時，可用對應於低頻率頻帶的窄頻帶之濾波器 來支援。 圖6係第1實施形態中的濾波器係數控制之另一例的 說明圖。 於係數判定電路12中，控制部123係首先會使圖 6(a)所示的中心頻率做了平移而成的濾波器模態1〜7之濾 波器特性，被依序產生。當圖6所示的先行波、主波及延 遲波之延遲差的3波多重路徑被輸入時，3波被收斂在濾 波器通過頻帶內的平移模態4時，訊號品質偵測電路1 22 上的MER爲最小，而被判定爲收訊品質最佳。接著，相 對於平移模態4，如圖6(b)所示，依序產生讓主波及其他 一部分頻帶透通的複數通過頻帶模態1〜6之濾波器特性 。當如圖6(b)所示的3波多重路徑時，相較於讓全體頻帶 透通的平移模態4，僅讓先行波、主波及延遲波附近透通 -13- 201225594 的帶有3個通過頻帶之通過頻帶模態6的雜訊去除能力較 高，因此訊號品質偵測電路1 22上的MER較小，而最終 被判定爲收訊品質最佳。 如以上所述，藉由使用在所定頻帶內帶有1或複數個 通過頻帶之濾波器特性來搜尋收訊品質最佳的濾波器特性 ，即使多重路徑波的延遲差較大時，在延遲波並沒有寬廣 到濾波器通過頻帶全體的情況下，由於通過頻帶以外之雜 訊會被截去，因此可去除SP訊號亦即傳輸路徑推定訊號 的雜訊而提升收訊性能。 若依據第1實施形態，則當可等化之多重路徑延遲時 間範圍擴大時，即使多重路徑的延遲差較大的情況下，在 延遲波並沒有寬廣到濾波器通過頻帶全體時，就可去除 SP訊號之雜訊而提升收訊性能。 [第2實施形態] 圖7係第2實施形態之OFDM收訊裝置的區塊圖。和 _1的第1實施形態同一部分係標示同一符號並省略說明 〇 於圖7的第2實施形態中，與圖1的第1實施形態的 不同點在於，第1係數切換電路11A除了具備通過頻帶平 移部11a與通過頻帶模態選擇部lib以外，還具備用來將 _ 8(a)之複數帶寬予以依序切換而設定的帶寬切換部lie ’以及第2係數切換電路124A除了具備通過頻帶平移部 124a與通過頻帶模態選擇部124b以外，還具備用來將與 -14- 201225594 圖8 (a)同樣之複數帶寬予以依序切換而設定的帶寬切換部 124c。因此係爲，控制部123A也會控制帶寬切換部11c 與帶寬切換部124c之構成。其他構成則和圖1相同。 圖8係第2實施形態中的濾波器係數控制之一例的說 明圖。 圖7的第1係數切換電路11A，係如圖8(a)所示，從 帶寬較窄起依序具有帶寬1、帶寬2、帶寬3之濾波器特 性，是具有針對最廣的帶寬3以主波位置爲基準而使中心 頻率平移而成的複數濾波器特性。又，相對於帶寬3之濾 波器特性，從已被平移之濾波器特性之中可切換讓包含主 波位置之一部分頻帶透通的複數濾波器特性，是依照來自 係數判定電路1 2 A所供給之帶寬、平移量、通過頻帶模態 而作動。 圖7的係數判定電路12A，係具備第2之SP頻率內 插濾波器125、第2係數切換電路124A、第2等化電路 121、訊號品質偵測電路122 '控制部123A，會產生出圖 8所示的濾婢器係數（瀘波器模態），從使用各個頻率內插 之等化輸出的MER，偵測出收訊品質，判定收訊品質最佳 的濾波器係數，將判定結果供給至第1係數切換電路11A 〇 接著，參照圖8，再來說明係數判定電路12A的動作 〇 於係數判定電路12A中，控制部123A係首先會使圖 8 (a)所示，依序產生出針對帶寬1、帶寬2、帶寬3以主波 -15- 201225594 位置爲基準而使中心頻率做了平移而成的複數濾波器特性 。當圖8所示之延遲差的2波多重路徑被輸入時，2波被 收斂在濾波器通過頻帶內的帶寬3 +平移模態1時，MER 爲最小，而被判定爲收訊品質最佳。 接著，相對於帶寬3 +平移模態1之所定帶寬，如圖 8(b)所示，依序產生讓主波及其他一部分頻帶透通的複數 通過頻帶模態之濾波器特性。於圖8(b)中，由於帶寬2以 下的通過頻帶係在圖8 (a)中已經判定過了，因此只會搜尋 含有超過帶寬2之通過頻帶的情形。當如圖8所示的2波 多重路徑時，相較於讓全體頻帶透通的帶寬3 +平移模態1 ，僅讓主波及延遲波附近透通的通過頻帶模態4的雜訊去 除能力較高，因此MER較小，而最終被判定爲收訊品質 最佳。 於圖8(a)之判定中，多重路徑延遲波是收斂在帶寬1 時則帶寬1爲最小MER，多重路徑延遲波超過帶寬1但爲 帶寬2的情況下則帶寬2爲最小MER。當帶寬1或帶寬2 的濾波器特性被選擇時，濾波器的雜訊去除良好，因此可 以不必進行搜尋部分性通過頻帶的處理，故可省略之》 如以上所述，先以複數帶寬之濾波器特性來進行判定 ，當廣頻帶濾波器被選擇時則進行在頻帶內帶有1或複數 通過頻帶的濾波器特性之判定，藉此，當多重路徑延遲差 較小時就不需要搜尋全部濾波器模態，因而可削減消費電 力。又，即使延遲差較大時，在延遲波並沒有寬廣到濾波 器通過頻帶全體的情況下，仍可去除SP訊號之雜訊而提 -16- 201225594 升收訊性能》 若依據第2實施形態，則在多重路徑之延遲差較小時 可削減演算量而抑制消費電力，同時，即使多重路徑的延 遲差較大時，在延遲波並沒有寬廣到濾波器通過頻帶全體 的情況下，仍可去除SP訊號亦即傳輸路徑推定訊號之雜 訊而提升收訊性能。 [第3實施形態] 圖9係第3實施形態之OFDM收訊裝置中的係數判定 電路的另一實施例的區塊圖。和圖1的係數判定電路同一 部分係標示同一符號來說明。 在圖1的係數判定電路12中，雖然是將濾波器係數 依序切換而偵測收訊品質，但當移動收訊等之收訊訊號狀 態有所變動的情況下，收訊品質之差異究竟是收訊狀態之 變動所致還是濾波器係數之差異所致，有時候會造成誤判 〇 圖9所示的係數判定電路12B，係具備第2之SP頻 率內插濾波器125、第2係數切換電路124、第2等化電 路1 2 1、第1訊號品質偵測電路1 2 2所成之和圖1相同的 一組電路部以外’還新設置有第3之SP頻率內插濾波器 14、第3係數切換電路16、第3等化電路13、第2訊號 品質偵測電路1 5所構成的另一組電路部。 第3係數切換電路16係具備：與通過頻帶平移部 124a相同之通過頻帶平移部16a、與通過頻帶模態選擇部 -17- 201225594 124b相同之通過頻帶模態選擇部16b。因此係爲，控制部 123 B除了控制圖1所示之第1係數切換電路11及第2係 數切換電路124以外，還會控制第3係數切換電路16之 構成。其他構成則和圖1的係數判定電路相同》 在圖9中，將SP頻率內插濾波器、係數切換電路、 等化電路、及訊號品質偵測電路之電路部予以設置2組， 對於同一收訊訊號是以2個濾波器係數而根據被第2及第 3等化電路1 2 1及1 3等化過之訊號而以第1及第2訊號品 質偵測電路122及15分別偵測收訊品質，控制部123 B係 選擇收訊品質較佳者。藉由依序比較已被選擇之濾波器特 性時的品質和下個濾波器特性時的品質，而最終決定出最 佳的濾波器係數。 藉由以上構成，即使在移動收訊等收訊訊號狀態有所 變動的情況下，仍可決定最佳的濾波器係數。同樣的構成 係亦可對第2實施形態的OFDM收訊裝置適用。 [第4實施形態] 圖1 〇係第3實施形態之OFDM收訊裝置中的係數判 定電路的又再另1實施例的區塊圖。 圖10所示的係數判定電路12C，係在圖1中的係數 判定電路12中，在第2等化電路121及第2之SP頻率內 插濾波器125的各前段，分別配設有記憶體126及127。 其他構成則和圖1的係數判定電路相同。 在圖10中，是將FFT輸出訊號及時間內插後的SP訊 -18- 201225594 號分別保持在記憶體126及127中，對同一訊號依序切換 濾波器係數而偵測收訊品質，決定最佳濾波器。 藉由以上構成，即使在移動收訊等收訊訊號狀態有所 變動的情況下，仍可決定最佳的濾波器係數。同樣的構成 係亦可對第2實施形態的OFDM收訊裝置適用。 若依據本發明之實施形態的OFDM收訊裝置，則將朝 時間方向及頻率方向分散過的SP訊號，在時間方向及頻 率方向以SP訊號內插濾波器進行內插，使用內插過的SP 訊號來推定等化時所必須之傳輸路徑響應之際，作爲SP 訊號內插濾波器是使用在所定頻帶內帶有2個以上之通過 頻帶的濾波器特性，將複數通過頻帶模態依序切換而嘗試 ，決定會使收訊品質呈最佳的濾波器特性之通過頻帶，藉 此而可良好地進行SP訊號之內插。在傳輸訊號中有長延 遲之延遲波成分而對廣頻帶適用濾波器的情況下，仍可去 除雜訊之影響而提升多重路徑時的收訊性能。 此外，雖然說明了本發明的數個實施形態，但這些實 施形態係只是作爲提示的例子，並非意在限定發明的範圍 。這些新穎的實施形態，係可用其他各種形態來實施’在 不脫離發明要旨的範圍內，可進行各種省略、置換、變更 。這些實施形態或其變形，係被包含在發明之範圍或要旨 ，並且被包含在申請專利範圍中所記載之發明和其均等範 圍中。 【圖式簡單說明】 -19* 201225594 [圖1]第1實施形態之OFDM收訊裝置的區塊圖》 [圖2]OFDM訊號的傳輸格式方式的說明圖》 [圖3]第1實施形態中的濾波器係數控制之一例的說 明圖。 [圖4]可對應長延遲之延遲波成分的廣頻帶之濾波器 的說明圖。 [圖5]可對應短延遲之延遲波成分的窄頻帶之濾波器 的說明圖。 [圖6]第1實施形態中的濾波器係數控制之另一例的 說明圖。 [圖7]第2實施形態之OFDM收訊裝置的區塊圖。 [圖8]第2實施形態中的濾波器係數控制之一例的說 明圖。 [圖9]係數判定電路的另一實施例的區塊圖。 [圖10]係數判定電路的又再另一實施例的區塊圖。 【主要元件符號說明】 1 :天線 2 :選台器 3 : A/D轉換電路 4 : IQ解調電路 5 : F F T電路 6: FFT窗控制電路 7 :第1等化電路 20- 201225594 8 :第1之SP時間內插濾波器 9 :第1之SP頻率內插濾波器 1 〇 :錯誤訂正電路 1 1 :第1係數切換電路 1 la :通過頻帶平移部 1 1 b :通過頻帶模態選擇部 1 lc :帶寬切換部 1 2 :係數判定電路 1 3 :第3等化電路 14 :第3之SP頻率內插濾波器 1 5 :第2訊號品質偵測電路 1 6 :第3係數切換電路 1 00 : OFDM收訊裝置 12 1 :第2等化電路 122 :訊號品質偵測電路 1 2 3 :控制部 124 :第2係數切換電路 125 :第2之SP頻率內插濾波器 126及127 :記憶體 1 2 3 A :控制部 1 2 3 B :控制部 124a :通過頻帶平移部 124b :通過頻帶模態選擇部 124c :帶寬切換部 -21 201225594 12A :係數判定電路 12B :係數判定電路 1 2 C :係數判定電路 16a:通過頻帶平移部 16b :通過頻帶模態選擇部In the first SP time interpolation filter 8, the SP signal is extracted from the OFDM signal in the frequency domain, and the SP signal is interpolated in the time direction and output to the first SP frequency interpolation filter 9. The first SP frequency interpolation filter 9 is an SP signal that has been interpolated in the time direction, and then interpolated in the frequency direction, by the SP signal that is interpolated in the time direction and the frequency direction, And the transmission path response corresponding to all the data is obtained. The first SP time interpolation filter 8 is as shown in FIG. 2, and the SP signal exists in a ratio of one out of every four symbols, so the other three symbols are For example, linear interpolation (internal -8 - 201225594 for dividing the time difference between SPs) is performed and 値 is placed. The SP frequency interpolation filter 9 of the first embodiment limits the frequency band of the SP signal to each symbol by using an optimum filter coefficient determined according to the signal quality detection result of the OFDM signal, as will be described later. The SP signal is interpolated in the frequency direction. Here, the SP signal is a known signal whose phase or power is predetermined, and is used as a transmission path estimation symbol in order to obtain a transmission path response required for estimating transmission signal distortion. The first equalizing circuit 7 estimates the signal by using the SP signal, and equalizes the OFDM signal in the frequency domain. The output of the first-class circuit 7 is supplied to the error correction circuit 10, and the error correction decoding process is performed to decode the received data. The first coefficient switching circuit 11 that switches the filter coefficients of the first SP frequency interpolation filter 9 includes a pass band shifting portion that is set by sequentially switching the complex translation modes of FIG. 3(a) 11a and passband mode selection unit 1 1 b set by sequentially switching the complex number of FIG. 3(b) through the band mode. The coefficient determination circuit 12 includes a second SP frequency interpolation filter 125, a second coefficient switching circuit 124, a second equalization circuit 121, a signal quality detecting circuit 122, and a control unit 123. The second coefficient switching circuit 124 that switches the filter coefficients of the interpolation filter 125 includes a pass band shifting unit 124a that is set in order to switch the same complex translation mode as in Fig. 3(a). The same complex number in Fig. 3(b) is sequentially cut by the band mode selection unit 124b by the band mode in 201225594. Fig. 3 is an explanatory diagram showing an example of filter coefficient control in the first embodiment. The first coefficient switching circuit 1 1, as shown in FIG. 3, is a complex filter characteristic in which the center frequency is translated based on the position of the main wave, and the filter characteristics from the shifted filter characteristics are allowed. The complex filter characteristic including the partial band pass-through of one of the main wave positions becomes switchable: it is actuated in accordance with the amount of shift supplied from the coefficient decision circuit 12 and the pass band mode. The coefficient determination circuit 12 sequentially generates the filter coefficients (filter modes) shown in FIG. 3, and detects the modulation error ratio (hereinafter referred to as MER) of the equalized output using the interpolation of each frequency. The reception quality is determined, and the filter coefficient having the best reception quality is determined, and the translation result and the determination result of the pass band mode are supplied to the first coefficient switching circuit 11. In addition, in FIGS. 3(a) and 3(b), the main wave and the delayed wave (and the preceding wave) on the time axis are shown (the horizontal axis represents the delay time and the vertical axis represents the power). It indicates the result of the actual detection delay profile, but the main wave with the largest signal level, for example, setting the threshold 値. If the actual detection of the threshold is exceeded, the main wave position is considered to be centered on the main wave position. There are pre- or delayed waves in the position before and after. Therefore, in the case of a predetermined bandwidth including a narrow band (hereinafter referred to as a main wave band) in which a main wave exists, for example, a case where a delayed wave exists is also assumed to be imaginarily illustrated on the time axis. Since the existence of the delayed wave is easier to distinguish with respect to the main wave, it becomes a graph similar to the delayed side write, but in the embodiment, there is no delay. -10- 201225594 has delayed side-write detection, which is a lot that can be thought of. The filter mode is set by changing the filter coefficients one by one, and the action determines the filter mode (filter characteristic) that is optimal for the reception quality. This is also the same in FIGS. 6 and 8 which will be described later. Regarding the main wave, the signal wave having the highest power level is defined as the main wave among all the signal waves including the main wave, the delay wave, and the preceding wave in the same broadcast channel. Further, after shifting the translation mode of the predetermined bandwidth in the frequency direction as shown in FIG. 3(a), one of the predetermined bandwidths is allowed to pass through two or more passbands as shown in FIG. 3(b). The optimum filter characteristic is generated by a method in which a passband mode having the same narrow bandwidth as the main wave band is a complex number (two in FIG. 3(b)), and the main wave band is used. The mode in which the band mode is centered and the other overlapped band modes are successively moved away, in other words, the filter coefficients are controlled to overlap the other pass band mode from the main band of the pass band mode. The center is separated from each other to generate filter characteristics with two passbands (eg, by band mode 6). That is, an attempt is made to determine all filter modes including a mode in which a filter band of a complex narrow bandwidth is sequentially expanded one by one as a filter mode in the frequency direction by coefficient control, and it is determined that The main wave and the delayed wave pass through the filter smoothly through the frequency band to obtain a good signal quality result. The same applies to Figs. 6(b) and 8(b) which will be described later. Next, the operation of the coefficient determination circuit 丨2 will be described with reference to Fig. 3 . Fig. 3 (a) and (b) are explanatory diagrams of an example of the filter coefficient control in the first embodiment -11 - 201225594. Fig. 3(a) shows the action of changing the translation amount successively to try the complex translational mode, and Fig. 3(b) shows the translation mode 1 after the translation mode 1 is selected in Fig. 3(a). Within the determined bandwidth, the pass band mode is successively changed from the subsequent pass band, and the operation until the pass band mode of the two independent pass bands is attempted. In the coefficient determination circuit 12, the control unit 1 23 controls the filter characteristics of the filter modes 1 to 7 which are first shifted by the center frequency shown in Fig. 3(a) by the filter coefficient control. , is produced sequentially. When the two-wave multiple path of the delay difference between the main wave and the delayed wave shown in FIG. 3 is input, when the two waves are converged in the translation mode 1 in the filter pass band, the MER on the signal quality detecting circuit 12 22 is The smallest, and is judged to be the best reception quality. Next, with respect to the translation mode 1, as shown in Fig. 3(b), the filter characteristics of the complex passband modes 1 to 6 for allowing the main wave and other partial bands to pass through are sequentially generated. When the two-wave multipath is as shown in Fig. 3(b), the noise removal capability of the passband mode 6 that allows only the vicinity of the main wave and the delayed wave to pass through is compared to the translation mode 1 in which the entire band is transparent. Therefore, the MER on the signal quality detecting circuit 1 22 is small, and it is finally determined that the receiving quality is the best. 4 is an explanatory diagram of a wide-band filter that allows a delay wave component (referred to as a long-delayed delayed wave component) having a large delay difference with respect to a main wave to be transmitted, and FIG. 5 is a view on the allowable relative An explanatory diagram of a filter of a narrow band in which a delayed wave component (referred to as a short-delay delayed wave component) having a small delay difference is transmitted through a main wave. Fig. 4(a) is a diagram showing a delay wave having a large delay difference with respect to a main wave, and Fig. 4(b) is a diagram showing a delayed wave component having a long delay as shown in Fig. 4(a). The frequency characteristics of the transmitted signal. The frequency characteristic at this time is a characteristic in which the interval between beats is short and varies with a fast cycle. Therefore, in order to support such a long delay delayed wave, it is necessary to have a wide-band filter including a high frequency band. However, if a wide-band filter is used alone, there is a problem that the noise component increases. Fig. 5(a) is a diagram showing a delayed wave component (referred to as a short-delayed delayed wave component) having a small delay difference with respect to the main wave, and Fig. 5(b) is a diagram showing a short delay as shown in Fig. 5(a). The frequency characteristic of the transmitted signal of the delayed wave component. The frequency characteristic at this time is a characteristic in which the interval of the beat is long and varies with a slower cycle. Therefore, at this time, it can be supported by a filter of a narrow band corresponding to the low frequency band. Fig. 6 is an explanatory diagram showing another example of filter coefficient control in the first embodiment. In the coefficient determination circuit 12, the control unit 123 firstly generates the filter characteristics of the filter modes 1 to 7 in which the center frequency shown in Fig. 6(a) is translated. When the three-wave multipath of the difference between the preceding wave, the main wave, and the delayed wave shown in FIG. 6 is input, the three waves are converged in the translation mode 4 in the filter pass band, and the signal quality detecting circuit 1 22 The MER is the smallest and is judged to be the best reception quality. Next, with respect to the translation mode 4, as shown in Fig. 6(b), the filter characteristics of the complex passband modes 1 to 6 which allow the main wave and other partial bands to pass through are sequentially generated. When the 3-wave multipath is as shown in Fig. 6(b), only the vicinity of the preceding wave, the main wave, and the delayed wave is transmitted through the translation mode 4 that allows the entire band to pass through -13 - 201225594 with 3 The passband mode 6 of the pass band has a higher noise removal capability, so the MER on the signal quality detecting circuit 1 22 is smaller, and finally it is determined that the receiving quality is the best. As described above, by using filter characteristics with one or a plurality of passbands in a predetermined frequency band to search for filter characteristics that have the best reception quality, even if the delay difference of the multipath wave is large, the delayed wave In the case where the filter passes through the entire band, the noise outside the band is cut off, so that the SP signal, that is, the noise of the transmission path estimation signal can be removed to improve the reception performance. According to the first embodiment, when the multipath delay time range that can be equalized is expanded, even if the delay difference of the multiple paths is large, the delayed wave is not widened to the entire filter transmission band, and can be removed. The noise of the SP signal improves the reception performance. [Second Embodiment] Fig. 7 is a block diagram of an OFDM reception device according to a second embodiment. In the first embodiment of the first embodiment, the same reference numerals will be given to the first embodiment, and the description will be omitted. In the second embodiment of FIG. 7, the difference from the first embodiment of FIG. 1 is that the first coefficient switching circuit 11A has a pass. The band shifting unit 11a and the pass band mode selecting unit lib include a bandwidth switching unit lie' and a second coefficient switching circuit 124A which are provided to sequentially switch the complex bandwidth of _8(a), in addition to the pass band. In addition to the passband mode selection unit 124b, the translation unit 124a further includes a bandwidth switching unit 124c for sequentially switching the same complex bandwidth as that of FIG. 8(a) of FIG. Therefore, the control unit 123A also controls the configuration of the bandwidth switching unit 11c and the bandwidth switching unit 124c. The other components are the same as those in Fig. 1. Fig. 8 is an explanatory diagram showing an example of filter coefficient control in the second embodiment. The first coefficient switching circuit 11A of FIG. 7 has the filter characteristics of bandwidth 1, bandwidth 2, and bandwidth 3 from the narrow bandwidth as shown in FIG. 8(a), and has the widest bandwidth 3 A complex filter characteristic in which the main wave position is a reference and the center frequency is translated. Further, with respect to the filter characteristic of the bandwidth 3, the complex filter characteristic that allows the partial band of the main wave position to be transparent is switched from among the filter characteristics that have been translated, in accordance with the supply from the coefficient decision circuit 1 2 A The bandwidth, the amount of translation, and the mode of the band are activated. The coefficient determination circuit 12A of FIG. 7 includes a second SP frequency interpolation filter 125, a second coefficient switching circuit 124A, a second equalization circuit 121, and a signal quality detecting circuit 122' control unit 123A, and generates a map. The filter coefficient (chopper mode) shown in Fig. 8 detects the reception quality from the MER using the equalized output of each frequency interpolation, determines the filter coefficient with the best reception quality, and determines the result. The first coefficient switching circuit 11A is supplied to the first coefficient switching circuit 11A. Next, referring to Fig. 8, the operation of the coefficient determining circuit 12A will be described in the coefficient determining circuit 12A. The control unit 123A first generates the sequence as shown in Fig. 8(a). A complex filter characteristic is obtained in which the center frequency is translated based on the position of the main wave -15 - 201225594 for the bandwidth 1, the bandwidth 2, and the bandwidth 3. When the two-wave multipath with the delay difference shown in Fig. 8 is input, the two waves are converged in the bandwidth 3 + translation mode 1 in the filter pass band, the MER is the smallest, and it is judged that the reception quality is the best. . Next, with respect to the bandwidth of the bandwidth 3 + translation mode 1, as shown in Fig. 8 (b), the filter characteristics of the complex pass mode of the main wave and other partial bands are sequentially generated. In Fig. 8(b), since the pass band below the bandwidth 2 has been determined in Fig. 8(a), only the case where the pass band exceeding the bandwidth 2 is searched is searched. When the two-wave multipath is as shown in FIG. 8, the noise removal capability of the passband mode 4 that allows only the main wave and the delayed wave to pass through is compared to the bandwidth 3 + translation mode 1 that allows the entire band to pass through. It is higher, so the MER is smaller, and it is finally judged that the reception quality is the best. In the determination of Fig. 8(a), when the multipath delay wave converges at the bandwidth 1, the bandwidth 1 is the minimum MER, and when the multipath delay wave exceeds the bandwidth 1 but the bandwidth 2, the bandwidth 2 is the minimum MER. When the filter characteristics of bandwidth 1 or bandwidth 2 are selected, the noise of the filter is well removed, so it is not necessary to perform the process of searching for partial passbands, so it can be omitted. As described above, the filtering is performed with complex bandwidth. The characteristics of the device are determined. When the wideband filter is selected, the determination of the filter characteristics with 1 or complex passbands in the band is performed, thereby eliminating the need to search for all filters when the multipath delay difference is small. Modal mode, thus reducing power consumption. Moreover, even if the delay difference is large, if the delay wave is not broad enough to the entire filter transmission band, the noise of the SP signal can be removed and the reception performance can be improved by 16-201225594 liters according to the second embodiment. Therefore, when the delay difference of the multiple paths is small, the amount of calculation can be reduced to suppress the power consumption, and even if the delay difference of the multiple paths is large, the delay wave is not wide enough to the entire filter pass band. The removal of the SP signal, that is, the noise of the transmission path estimation signal, improves the reception performance. [Third Embodiment] Fig. 9 is a block diagram showing another embodiment of a coefficient determination circuit in an OFDM reception device according to a third embodiment. The same portions as those of the coefficient decision circuit of Fig. 1 are denoted by the same reference numerals. In the coefficient determination circuit 12 of FIG. 1, although the filter coefficients are sequentially switched to detect the reception quality, when the state of the reception signal such as the mobile reception or the like is changed, the difference in the reception quality is exactly It is caused by a change in the reception state or a difference in the filter coefficients, and sometimes causes a misjudgment. The coefficient determination circuit 12B shown in FIG. 9 includes the second SP frequency interpolation filter 125 and the second coefficient switching. The circuit 124, the second equalizing circuit 1 2 1 , and the first signal quality detecting circuit 1 2 2 are identical to the same circuit group as in FIG. 1 and a third SP frequency interpolation filter 14 is newly provided. The other set of circuit sections formed by the third coefficient switching circuit 16, the third equalizing circuit 13, and the second signal quality detecting circuit 15. The third coefficient switching circuit 16 includes a pass band shifting unit 16a similar to the pass band shifting unit 124a and a pass band mode selecting unit 16b similar to the passing band mode selecting unit -17-201225594 124b. Therefore, the control unit 123B controls the configuration of the third coefficient switching circuit 16 in addition to the first coefficient switching circuit 11 and the second coefficient switching circuit 124 shown in Fig. 1 . The other configuration is the same as the coefficient determination circuit of FIG. 1. In FIG. 9, the SP frequency interpolation filter, the coefficient switching circuit, the equalization circuit, and the circuit portion of the signal quality detecting circuit are set in two groups. The signal signals are detected by the first and second signal quality detecting circuits 122 and 15 respectively according to the signals equalized by the second and third equalizing circuits 1 2 1 and 13 by two filter coefficients. The quality of the control unit 123B selects the one with better reception quality. The optimum filter coefficients are finally determined by sequentially comparing the quality of the selected filter characteristics with the quality of the next filter characteristic. According to the above configuration, even when the state of the received signal such as the mobile reception or the like is changed, the optimum filter coefficient can be determined. The same configuration can be applied to the OFDM receiving apparatus of the second embodiment. [Fourth Embodiment] Fig. 1 is a block diagram showing still another embodiment of the coefficient determining circuit in the OFDM receiving apparatus according to the third embodiment. The coefficient determination circuit 12C shown in FIG. 10 is provided with a memory in each of the front stages of the second equalization circuit 121 and the second SP frequency interpolation filter 125 in the coefficient determination circuit 12 of FIG. 126 and 127. The other configuration is the same as the coefficient decision circuit of Fig. 1. In FIG. 10, the FFT output signal and the SP signal -18-201225594 inserted in the time slot are respectively held in the memories 126 and 127, and the filter coefficients are sequentially switched for the same signal to detect the reception quality. The best filter. According to the above configuration, even when the state of the received signal such as the mobile reception or the like is changed, the optimum filter coefficient can be determined. The same configuration can be applied to the OFDM receiving apparatus of the second embodiment. According to the OFDM receiving apparatus according to the embodiment of the present invention, the SP signal dispersed in the time direction and the frequency direction is interpolated by the SP signal interpolation filter in the time direction and the frequency direction, and the interpolated SP is used. When the signal is used to estimate the transmission path response necessary for equalization, the SP signal interpolation filter uses the filter characteristics of two or more passbands in a predetermined frequency band to sequentially switch the complex number through the mode. Attempts have been made to determine the passband of the filter characteristics that will optimize the quality of the reception, thereby enabling good interpolation of the SP signals. In the case where there is a long delay delay wave component in the transmission signal and a filter is applied to the wide band, the effect of noise can be removed to improve the reception performance in the multipath. In addition, although the embodiments of the present invention have been described, these embodiments are merely illustrative and are not intended to limit the scope of the invention. The present invention may be embodied in a variety of other forms, and various omissions, substitutions and changes can be made without departing from the scope of the invention. The invention or its modifications are intended to be included within the scope of the invention and the scope of the invention. [Brief Description of the Drawings] -19* 201225594 [Fig. 1] Block diagram of the OFDM receiving apparatus according to the first embodiment [Fig. 2] Explanation of the transmission format of the OFDM signal [Fig. 3] First embodiment An illustration of an example of filter coefficient control in the middle. Fig. 4 is an explanatory diagram of a wide-band filter that can correspond to a delayed-wave component of a long delay. Fig. 5 is an explanatory diagram of a filter capable of responding to a narrow band of a short-delay delayed wave component. Fig. 6 is an explanatory diagram showing another example of filter coefficient control in the first embodiment. Fig. 7 is a block diagram of an OFDM receiving apparatus according to a second embodiment. Fig. 8 is an explanatory diagram showing an example of filter coefficient control in the second embodiment. [Fig. 9] A block diagram of another embodiment of the coefficient decision circuit. [Fig. 10] A block diagram of still another embodiment of the coefficient decision circuit. [Main component symbol description] 1 : Antenna 2 : Selector 3 : A/D conversion circuit 4 : IQ demodulation circuit 5 : FFT circuit 6 : FFT window control circuit 7 : First equalization circuit 20 - 201225594 8 : 1 SP interpolation filter 9: 1st SP frequency interpolation filter 1 〇: Error correction circuit 1 1 : 1st coefficient switching circuit 1 la : Pass band shifting unit 1 1 b : Pass band mode selection unit 1 lc : bandwidth switching unit 1 2 : coefficient determination circuit 1 3 : third equalization circuit 14 : third SP frequency interpolation filter 15 : second signal quality detection circuit 1 6 : third coefficient switching circuit 1 00 : OFDM receiving device 12 1 : second equalizing circuit 122 : signal quality detecting circuit 1 2 3 : control unit 124 : second coefficient switching circuit 125 : second SP frequency interpolation filters 126 and 127 : memory Body 1 2 3 A : Control unit 1 2 3 B : Control unit 124a : Pass band shift unit 124 b : Pass band mode selection unit 124 c : Band switching unit - 21 201225594 12A : Coefficient determination circuit 12B : Coefficient determination circuit 1 2 C : coefficient determination circuit 16a: pass band shifting section 16b: passband mode selection section