TW201214999A - Methods and systems for noise and interference cancellation - Google Patents

Abstract

Signals propagating from an aggressor communication channel can cause detrimental interference in a victim communication channel. One or more noise cancellers can generate an interference compensation signal to suppress or cancel the interference based on one or more settings. A controller can execute algorithms to find preferred settings for the noise canceller(s). The controller can use a feedback signal (e.g., receive signal quality indicator) received from a victim receiver during the execution of the algorithm(s) to find the preferred settings. One exemplary algorithm includes sequentially evaluating the feedback resulting from a predetermined list of settings. Another algorithm includes determining whether to move from one setting to the next based on the feedback values for both settings. Yet another algorithm includes evaluating a number of sample settings to determine which of the sample settings result in a better feedback value and searching around that sample setting for a preferred setting.

Description

201214999 VI. Description of the Invention: [Technical Field] The present invention relates to a method for compensating between two or more communication elements between two or more communication channels or in a communication system System and method for signal interference. RELATED APPLICATIONS RELATED APPLICATIONS STATEMENT OF RELATED APPLICATIONS RELATED APPLICATIONS RELATED APPLICATIONS right. The present application also claims the benefit of U.S. Provisional Patent Application Serial No. 61/375,491, filed on Aug. 20, 2010. The present application also relates to U.S. Patent Application Serial No. [A. No. 07982.1 05 112] entitled "Cascaded Filter Based Noise and Interference Canceller" filed on the same date as the present application. The entire content of each of the aforementioned priority and related application is hereby incorporated by reference in its entirety. [Prior Art] [Summary of the Invention] Compensating for interference can improve signal quality, improve communication bandwidth or information carrying capacity, or improve receiver sensitivity. Communication channels can include transmission lines, printed circuit board (PCB) traces, flexible circuit traces 'electrical conductors, waveguides, sinks 201214999 Λ : flow lines, communication antennas, media providing signal paths, or active or passive circuits or circuit components ( Such as chopper, vibratory, diode, vc〇, pLL, amplifier, digital or mixed signal integrated circuit). Thus, the channel may include Global System for Mobile Communications (GSM) devices, processor "detectors, signal sources, H & 'connector' circuit traces or digital signal processing (DSP) chips (to name a few possibilities) The exemplary embodiments described herein may include high input power series filter (HIPCF) noise and interference cancellation devices. The exemplary HIPCF cancellers described herein may support selective cancellation, correction, addressing, or compensation. Interference, electromagnetic interference (face), noise (eg, phase noise, intermodulation products, and other interfering noise), clutter, or communication systems (such as high-speed digital communication systems enabled by portable electronic devices) - or other bad spectral components associated with multiple communication paths. For the purposes of this specification, the term "high power" refers to signals having up to about +33 _ (relative to - milliwatts). Decibel number) or greater power ratio. For example, the exemplary HIPCF canceller described herein can be connected to the output of a cellular telephone power amplifier having an output power of this magnitude. The HIPCF canceller can obtain a sample of the communication signal that imposes interference from the communication path of the infringing transmission device, and processes the sampled signal to produce an interference compensation signal. The HIPCF may deliver the interference compensation signal to or within the communication path of the victim receiver of the interferer to cancel, mitigate, suppress or otherwise compensate for the received interference. [Embodiment] 5 201214999 The same numerals are used throughout the drawings to indicate the same or corresponding (but not necessarily identical) elements, and the exemplary embodiments of the present invention are described in detail. 1 is a functional block diagram of a __communication system (10), in accordance with some demonstrative embodiments. Referring to Figure 1, a communication system 100 includes a transmitter 105 that transmits electromagnetic signals via a transmission antenna 115. A transmission path including one or more electrical conductors also couples the transmitter 105 to the transmission antenna 11 5 . In certain exemplary embodiments, transmitter 105 uses one or more communication standards or methods (such as = ball mobile communication system (GSM), code division multiple access (cdma), long term decision (LTE), broadband Code Division Multiple Access (W-CDMA) 'Digital Honeycomb System (DCS), Personal Communication Service (PCS) and Wireless Local Area Network (WLAN) transmit data to remote devices. It will be appreciated by those skilled in the art having the benefit of this disclosure that the communication system (10) described herein is not limited to the aforementioned communication standards and methods ' is readily available for use with many other types of signal transmission techniques. The transmission path between the transmitter 105 and the transmission antenna 丄〇5 is set by the power amplifier 11Ge power amplifier UG to adjust the power level of the signals before the output signals of the transmitters are propagated by the antenna 115. When the power amplifier 110 adjusts the power level of the signal, a poor spectral component can be introduced to the signal. For example, transmitting $1G5 can transmit a signal with a certain carrier frequency or a certain tone. The power amplifier m can introduce an intermodulation product having a frequency different from this carrier frequency or tone. Other components associated with transmitting $1〇5 can also introduce noise or other undesirable spectral components onto the signal. For example, transmitter 1〇5 can include a local blue blue and/or one or more upconverting mixers that can include a bad spectral component number, sometimes referred to as an out-of-band blocking signal 201214999. The out-of-band noise (introduced to the signal outside the band transmitted by the nickname of Gandenham. The communication system! 00 also includes a receiver 135 that passes through 120 and receives the receiving antenna 12 for 2〇 The receiving path 133 electrically coupled to the receiver 135 receives the signal. In some cases, the receiver 135 receives the same or different frequency bands in the feeder (9). Signals within the frequency band. For example, a mobile electronic device such as a mobile phone, a personal digital assistant (pDA) or a mobile computer may include a transmission via the one of the communication protocols discussed above, @1G5 and - Receiver (1) for communication in different frequency bands (such as a mobile TV spectrometer, a Bluetooth receiver, a microwave access global interworking (pour AX) receiver, or a global positioning system (Gps) receiver, in the illustrated embodiment, The receive path 133 includes an optional receive (rx) chopper. The receive filter (10) may include a bandpass chopper or other filter configuration that allows the passer 5 仏5 received by the antenna 12 在 in the frequency band of the receiver 135 to pass to the receiver 135 while blocking A signal outside the band of the receiver 135.

The frequency band of the receiver 135 can be near the frequency band of the transmitter 1〇5 such that the phase noise or other bad spectral components generated by the force amplifier 110 or the components placed along the transmission path 1G7 degrade the sensitivity of the receiver 135. . For example, the communication system (10) can be embodied in a mobile device having a CDMA, GSM or LTE transmitter 105 and as a mobile τ dictator of the receiver 135. Some types of CDMA and GSM transmitters 1〇5 transmit signals in the frequency band of approximately 8〇〇 MHz to 900 MHz, and certain types of LTE transmitters 105 operate in the frequency band of approximately 698 MHz to 798 MHz. These transmitted signals of 201214999 often include phase noise with a frequency between 45 0 MHz and 776 MHz, which falls within the receive band of some mobile TV tuners and many other communication devices. The phase noise is imposed on the path of the receiver 13 5 (e.g., from the transmitting antenna 11 $ to the receiving antenna 120), the phase noise can degrade the sensitivity of the receiver 135. Typically, a receive filter such as receive chopper 140 does not filter out-band noise because the noise is in the band of receiver 135 (and thus the pass band of receive filter 140). Therefore, phase noise can be passed through the receive filter 140 and degrade the sensitivity of the receiver 135. In order to prevent degradation of the sensitivity of the receiver 135 caused by in-band noise (having noise in the frequency band of the receiver 135) caused by the transmission from the transmission antenna 115 or near-band out-of-band noise, the communication system A HIPCF canceller 130 is coupled to the transmission path 107 by the sampling device 125 at the output of the power amplifier 110 at the input of the HIPCF canceller 13A. Sampling device 125 can include a capacitor (e.g., a sampling or sub-sampling capacitor), a resistor, a coupler, a coil, a transformer, a signal trace, or an antenna/detector. A sampling device having two tan or two terminals, such as a resistor, capacitor, coil, transformer or signal trace, can have a first turn electrically coupled to the transmission path 107 and be electrically coupled to the HIpCF canceller 13 The second end of the input. In an antenna/detector having substantially one terminal, the second terminal is formed by an electromagnetic field protruding from the device, thereby allowing the device to be positioned close to the transmission path 1 〇7. In the illustrated embodiment, the sampling cassette 125 is coupled to the transmission path 1〇7 at the output of the power amplifier 110. The home device 125 obtains a sample of the signal at the output of the power amplifier 8 201214999 110 ("sampling transmission number") and provides the sampled transmission signal to the HIPCF canceller 130. In a certain implementation For example t, the sampling device f 125 can produce a decay for the transmitted signal of the sample. For example, the amplitude of the sampled transmission signal can be lower than the signal at the output of the power amplifier 110 by 2 GdBe (relative to the decibel number of the carrier in some In the exemplary embodiment, the 'sampling device 125 includes a voltage controlled capacitor (variable reactor) for trimming the frequency dependent attenuation to a desired value, and thus compensating for gain fluctuations. In an example, the sampling device 125 includes - Dust-controlled varactor. The capacitance of the variable reactance can be adjusted via the control voltage. This control can be generated by the controller 235 (Fig. 2) of the mPCF eliminator # 13() and via one or more electrical conductors. 127 is transmitted to the sampling device η. The F-eliminating 3130 selectively suppresses or eliminates the power amplification n UG (or along the frequency having a frequency in or near the receiving band of the receiver 135 that would otherwise interfere with the sensitivity of the receiver (1). An interference signal generated by another component of the transmission path iG7 (for example, phase noise, intermodulation products, bad spectral components, etc.). The CF canceller 13Q obtains a sample of the signal output by the power amplifier ιι〇' and processes the sample The signal is transmitted to generate an interference compensation signal. The interference compensation signal suppresses or cancels the interference signal when applied to the input of the port 135. In certain exemplary embodiments, the HEP canceller 130 is used to include one or more electrical conductors. The feedback path is just the feedback obtained from the receiver (1) (such as the received signal quality indicator). The interference compensation signal. The exemplary HIPCF canceller 130 is described in more detail below in conjunction with Figures 2 through 31. The interference compensation signal is applied to the receiver 135 201214999 = the receiving path 133β. In some exemplary embodiments, the electrical conductor of 133 and the turn-off path along the turn-off path (four) into ten (4) benefits 130. Conductance 13 :°°” The electrical conductors of the electrical conductors are electrically connected to implement the elimination of the point 1 and the flexible circuit trace of the 'receiver ##134 can be connected to the flexible output of the HIPCF output. Road Traces. In some exemplary embodiments, an interference compensation signal may be applied to the receive path US of the receiver 135 using components such as a tracker 'sum sum, point, adder, or another suitable technique. The communication system 100 illustrated in FIG. 1 can transmit an electromagnetic signal 1 having a frequency within a first frequency range, and receive an electromagnetic signal having a frequency within a second frequency range. The frequency range can be close to the second frequency range or It even includes frequencies that overlap or are included in the second frequency range. In operation, the transmitter 105 transmits signals to the power amplifier 110 along the transmission path #107. The power amplifier 110 adjusts the strength of the signal received from the transmitter and outputs the intensity-adjusted signal to the transmission antenna 115. The transmission antenna 丨丨5 transmits the signal received from the power amplifier 。. One of the signals transmitted by the transmission antenna 115 is partially coupled to the receiving antenna 120 over the air. If received by the receiver 135, the signal from the transmit antenna ι 5 coupled to the receive antenna 120 can interfere with the sensitivity of the receiver 135 or degrade the sensitivity of the receiver 135. For example, a signal transmitted by the transmission antenna 115 having a frequency in or near the frequency band of the receiver 135 (e.g., appears as a phase noise tail generated by a power amplifier 〇 [〇] The modulated spectrum) can degrade the sensitivity of the receiver 135. In order to compensate for this interference or sensitivity degradation, the Hipcf canceller 〇3〇 obtains a sample of the signal output by the power amplifier 110 (via the sampling device 10 201214999 125 ) 'and processes the sampled transmission signal to generate an interference compensation signal, interference compensation 4s The interference imposed by the signal transmitted by the transmitting antenna 115 on the receiver 135 is compensated when applied to the input of the receiver 135. 2 is a block diagram of the hipcf cancellation piano 130 of FIG. 1 in accordance with some exemplary embodiments. The exemplary HIPCF canceller 13A includes a bandpass filter (input BPF) 205 that receives signal samples from the sampling device 125. In this exemplary embodiment, input BPF 205 includes an inductor li and two switchable capacitors C1 and C2. By adjusting the capacitance of one or both of the switchable capacitors ci and C2, the resonant frequency of the input bpf 205 can be white D. The switchable capacitors 〇1 and C2 are described in further detail below in conjunction with FIG. In certain exemplary embodiments, inductor L1 is a high Q inductor. The use of a high Q inductor can provide a performance advantage, such as providing additional attenuation to the signal outside the passband of the input BPF 205 and thus protecting subsequent components in the hipcf canceller 130, allowing for a linear exchange for lower miscellaneous The bottom limit. In certain exemplary embodiments, inductor Li is a low q inductor. In some exemplary embodiments, input BPF 205 includes a q enhancement circuit 29 to improve the quality factor (Q factor) of inductor L1. However, some q enhancement circuits can introduce noise or interference into the signal that passes through the input bpf 2〇5. The resonant frequency of the input BPF 205 can be tuned to the receive frequency (or near) of the receiver 135 to pass the interfering signal at this frequency that can be present on the sampled transmit signal and to block or filter out the infringing signal, such as by The tone or carrier signal transmitted by transmitter 1〇5 and other out-of-band blocking signals (with 201214999 signals having frequencies outside the band of the receiver). If the receiver i35 includes a -action τν tuner or other frequency adjustable device, the resonant frequency of the input BPF 205 can be adjusted to match the frequency of the current channel to which the tuner τν tuner is set. For example, the channel 〇 of the mobile τν tuner may have a receive frequency in the band of 686 should be 2 to 692 then 2. When the action τν is tuned to this frequency, the input BpF 2〇5 can also be automatically tuned to this frequency. If the mobile TV is subsequently tuned to another channel having a different receive frequency, the resonant frequency of the input BPF 205 can be adjusted to match the receive frequency of the new channel. For example, t, controller 235 can communicate with receiver 135 to obtain the current receive frequency for receiving H 135. In response, controller 235 can adjust switchable capacitors C1 and C2 such that the resonant frequency of input BPF 2〇5 is near or equal to the receive frequency. The input BPF 205 reduces the amplitude of the signal having a different frequency than the resonant frequency of the input BPF 2〇5. For example, if receiver 135 and transmitter 105 operate at different frequencies, input BpF 2〇5 can reduce the amplitude of the tone of the transmitted transmission k number. In some exemplary embodiments, the input b抒2〇5 reduces the amplitude of the tone of the transmitted signal at 824 减小 by approximately 13 dBc to 18 dBc′ and its center frequency is tuned to 749 ΜΗζ (corresponding to action) TV tuner channel 60). The output of the input BPF 205 is electrically coupled to a low noise amplifier (LNA) 210. The LNA 210 amplifies the signal output by the input BPF 2〇5 and passes the amplified signal to a second band pass filter (referred to herein as LNA-BPF 2 15). In certain exemplary embodiments, [ΝΑ 210 is a serial LNA. In this exemplary embodiment, LNA-BPF 215 includes an inductor L2 12 201214999 - and a switchable capacitor C3. In some exemplary embodiments, 'inductor L2 is a Q-inductor. In certain exemplary embodiments, inductor L2 is a low q inductor. In some exemplary embodiments, LNA-BPF 2 15 includes a q enhancement circuit 29 1 to improve the Q factor of the inductor [2]. In some exemplary embodiments, the Q factor of 'L2 is smaller than the Q factor of l1. In some exemplary embodiments, the Q factor of 'L2 is greater than the Q factor of L2. The resonant frequency similar to the input BPF 205 ' LNA-BPF 215 can be set to the receive frequency of the receiver 135 to pass the signal at this frequency and further filtered, the tone of the transmitted signal from the sample, and the out-of-band blocking signal. In some exemplary embodiments, the LNA-BPF 215 can reduce the amplitude of the tone at 824 MHz by about 13 dBc to 18 dBc, while its t-heart frequency is tuned to 749 MHz. The output of the LN A BPF 215 is electrically connected to a variable gain amplifier (VGA) 22 that adjusts the amplitude of the signal output by the lnA-BPF 2 1 5 . In some exemplary embodiments, VGA 220 includes a plurality of variable gain amplifiers for adjusting the amplitude of the signals received from LNA BpF 215. The vibrating field adjustment 彳5 output from the vga 22〇 is then passed to the third band pass filter (卩 enhanced BPF) 225 〇Q enhanced BPF 225 can include an inductor 匕3 and a switchable electric device 615 (FIG. 6), the inductor L3 and the switchable capacitor 615 are used to base the q-square strong bpf 225 to the receiving frequency of the receiver 135 to transmit any signal at a frequency and further sample the carrier signal of the wave sampling. (6) Out-of-band blocking m can be implemented in certain materials, and the inductor can be a Q inductor b (eg, 'out-of-chip) or a low-Q on-wafer inductor ^ 13 201214999 In some exemplary embodiments, Q-enhanced The BPF 225 also includes a Q enhancement circuit 292. In certain exemplary embodiments, the Q-enhanced BPF 225 includes a current switch (Fig. 6) to adjust its Q factor. In certain exemplary embodiments, the Q-enhanced BPF 225 may further reduce the amplitude of the remaining tone in the signal received from the VGA 220 at 824 MHz by up to 26 dBc or more, with the center frequency tuned to 749 MHz. . The output of the Q-enhanced BPF 225 is electrically coupled to the I/Q modulator 230. Although in the illustrated embodiment, a series of bandpass filters 205, 215, and 225 are used to filter noise and other signals having frequencies outside the frequency band of receiver 135, in addition to bandpass filters 205, 215, and 225. Other types of choppers may be utilized in addition to or in lieu of one or more of the bandpass filters 205, 215, and 225. For example, one or more high pass and/or low pass filters may be used in certain exemplary embodiments. The string pass filter 205, 2 1 5, 225 blocks or reduces the amplitude of the signal outside the receive band of the receiver 丨 35, which typically does not interfere with the sensitivity of the receiver. The signals in the receive band of the receiver 135 are passed through the bandpass filters 205, 215 and 225 to the I/Q modulator 230. These in-band signals are also amplified by the LNA 210 and VGA 220. The I/Q modulator 230 adjusts at least one of the phase, amplitude, and delay of the signal received from the Q-enhanced BPF 225 to generate an interference compensation signal that is reduced when applied to the receive path 133 of the receiver 135, The noise and/or interference present on the receive path 丨33 of the receiver 135 imposed by the signal transmitted by the transmit antenna 115 is suppressed, cancelled or otherwise compensated. In some exemplary embodiments, the interference compensation signal has a 180 degree phase shift relative to the phase of the in-band impurity 14 201214999 signal signal and is close to the amplitude of the in-band noise signal or the same amplitude as the in-band noise signal. The amplitude. Therefore, the interference compensation signal reduces or eliminates in-band noise signals. In some exemplary embodiments, feedback from the victim receiver's Receive 1 quality indicator is used (such as bit error rate (Ber), packet error rate (PER), received signal strength indicator (RSSI). , noise floor, signal-to-noise ratio (SNR), error vector magnitude (EVM), and positional accuracy (for GPS), etc. based on one stored in memory device 76 (FIG. 7) and executed by controller 235 Group instructions (eg, algorithms) tune the aforementioned parameters of amplitude, phase, and delay. An exemplary algorithm for determining the settings for s-round amplitude, phase, and delay is described below with reference to Figures 17 through 31. As shown in FIG. 7, in some exemplary embodiments, the HIPCf canceller 丨3 includes a power detector 745 coupled to the input of the I/Q modulator 230, such as a peak detector. Power detector 745 senses or measures the power level of the signal at the input of modulation variable 230 and provides an indication of the power level to controller 235. Controller 235 uses this power level value to modify the current of i Q-enhanced BPF 225 and thus Qmax for maintaining acceptable suppression of noise and/or interference imposed on receiver n5 by the signal transmitted by transmission antenna 115. . In certain exemplary embodiments, the lung CF eliminator 130 includes an analog/digital (A/D) converter 75A, which is a self-powered detector. Receiving a power level value and providing a digital representation of the power level value to the control

Hi35. The controller 235 performs a calibration routine to ensure that it is transmitted by the antenna 115a. The transmitted signal is imposed on the receiver 135 and the interference and the degree of interference are suppressed. An exemplary calibration routine is described below with reference to Figures 9-16. 15 201214999 Controller 235 may be implemented in the form of a microcontroller, microprocessor, computer, state machine, programmable device, control logic, analog and digital circuitry, or other suitable technique. Controller 235 can perform settings for adjusting each of band pass filters 205 215 and 225 and one or more processes for operating switchable capacitors C 1 through C3 and SCA 6 15 (FIG. 6) Or program. In an example, controller 235 automatically adjusts the resonant frequency of one or more of band pass filters 205, 215, 225 in response to changes in the frequency of receiver 135. For example, if receiver 135 includes a motion τν tuner, controller 235 adjusts the resonant frequencies of bandpass filters 205, 215, 225 to match or correspond to the receiver frequency. The controller 235 adjusts the resonant frequencies of the bandpass filters 2〇5, 215, 225 by adjusting the capacitances of the switchable capacitors C1 to C3 and SCA 615, respectively, as discussed below with reference to FIG. Controller 235 can also adjust or modify the settings of I/Q modulator 23, bandpass filters 205, 2丨5 and 225, and VGA 22〇 to account for environmental changes such as temperature, supply voltage, and antenna coupling changes. In some exemplary embodiments, controller 235 performs a - calibration routine (g 16 ) to identify acceptable settings based on such environmental changes and store the identified optimal settings for subsequent use. The algorithm may be embodied as software stored on controller 235 or on memory storage device 760. Alternatively, a hardware device (such as a discrete logic gate) algorithm can be implemented in one or more. The HIPCF canceller 130 also includes an auxiliary circuit 240. As shown in FIG. 7, the auxiliary circuit 240 & includes a temperature sensing $ 755, a "four" detector 745, one or more analog/digital converters 75A, a digital/analog converter, and for use by HIPCF 4 Other types of circuits used in addition to n 13Q. The auxiliary circuit 16 201214999-240 may also include - or a plurality of memory storage devices 760, such as ram, R〇M, and/or flash memory. The settings for each of the band pass filters 205, 215, and 225 can be stored on the memory storage device 760. Additionally, the settings for the UQ modulator 230 can be stored on the memory storage device 760. For example, the settings for each channel of the mobile TV tuner can be stored on the memory storage device 760. Some of the components or functions of the HIPCF canceller 13 can be embodied in an integrated circuit', e.g., as depicted by the wafer boundary 25〇 illustrated in FIG. For example, one or more of the 'switchable capacitors C1 to C3, LNA 210, VGA 220, Q enhanced BPF 225, I/Q modulator 23, controller 235, and auxiliary circuit 24 can be embodied in a single Integral circuit or multiple integrated circuits. Although inductors Li and L2 are illustrated as out-of-chip inductors in the illustrative embodiment of the description, other exemplary embodiments may use on-wafer inductors in band pass filters 205 and 215. The integrated circuits and/or inductors L1 and L2 can be mounted on mobile devices such as mobile phones as well as other communication devices. Single or multiple integrated circuits can be embodied in or on a complementary metal oxide semiconductor (CMOS). Referring to Figures 1 and 2, the HIPCF canceller 130 suppresses, cancels or otherwise compensates for in-band or near-band out-of-band (carrying in proximity to the receiver 135) imposed by the transmitter 105 via the transmit antenna 115 on the receiver 135. Frequency) Interference signal. That is, the HIPCF cancellation @ UG compensates for an interference signal transmitted by the transmission antenna ι 5 having a frequency in or near the frequency band of the receiver 135. The HIPCF canceller 130 obtains samples of the signals transmitted by the transmitters 1〇5 from the sampling device 125 and processes the samples to generate interference compensation. 201214999 Compensation signal 'Interference compensation signal when applied to the input of the receiver i 35 Compensate for the imposed interference signal. The exemplary HIPCF 130 includes three bandpass filters 2〇5, 215, and 225 that each filter, block, or reduce the transmitted signal of the sample received by the out-of-band self-sampling device 125 with respect to the receive frequency of the receiver ι35. The strength of the signal component. The component of the transmission nickname relative to the receiver 135 is used to generate an interference compensation signal. The special signal of the sampled transmission signal: at least one of the phase, amplitude and delay of E is adjusted by the I/Q modulator 230 to produce an interference compensation signal. Controller 235 can perform one or more calibration algorithms and/or one or more tuning algorithms to improve the level of interference compensation. Controller 235 may obtain feedback from power detector 745 or from receiver 135 and use this feedback during execution of the algorithm. These algorithms are discussed in detail below with reference to Figures 9 through 31.

3 is a block diagram 3D of certain components of the hipcf canceller 130 of FIG. 2, in accordance with some demonstrative embodiments. Specifically, Fig. 3 is a diagram showing an example of a transistor level diagram of an indirect input BPF 205, an exemplary LNA_BpF 215, and an exemplary LNA 21?. Referring to Figure 3, the input BpF 2〇5 includes a first on-off capacitor array (SCA) 3〇5 and a second SCA 31〇. Each of sca 3 〇 5, 310 t includes an array having a number of "n+1" capacitors, which typically include 1 or 2 standard capacitor sizes (single power reserve). Each of the SC A 305, 310 has a corresponding transistor switch (e.g., MOS transistor) for starting the capacitor. The resonant frequency of the input F 205 can be adjusted by selecting one or more of the capacitors from SC A 305 & 310. The capacitor can be selected by activating the turn-on 18 201214999 associated with each selected capacitor. For example, Zhuangshan recognizes that the capacitor ci〇 is selected by starting (or turning on) the switch Μ10. Each of the SCA 3fts Λ — 305, 310 may have a different capacitance value corresponding to a different resonant frequency for inputting the BPF τ τ τ ° 205, or with a * weighting value to cover the reception _ 135 The frequency band. In some exemplary embodiments, control 3 235 can initiate and deactivate the switches in sca: and 310 to select the resonant frequency for the input BpF 2G 5. SCA 305 and 310 can also provide a voltage divider function. This is especially true for a wideband τ tuner as a receiver 135

example. In certain exemplary embodiments, the heart. 5 and 31. An additional 15 dBc reduction in the amplitude of the transmitted signal with == (for example, 1:5), thus reducing the linearity requirement for subsequent stages or achieving when selecting a smaller ratio (eg, 1:1) More gain. This ratio can also be varied in video channels to flatten or adjust the overall gain over the entire mobile TV band. In order to configure the input BPF 205 for a high UHF (Ultra High Frequency) channel with a frequency close to the frequency of the GSM CDMA or LTE transmitter 1 〇 5 (such as channel 5 for the mobile TV tuner), it can be activated Switch M61 and cancel start switches M60 and M62. This provides a voltage divider between the capacitor in the first SCA 3〇5 and the capacitor in the second SC A 310. In order to have the configuration input BPF 205 for a low UHF channel (such as channel 26 of the action τ tuner) that can have a frequency between 542 MHz and 548 MHz, the switches M6 〇 and M 62 can be activated and the start-up can be initiated. Switch M61 disconnects second SCA 310 from input BPF 205. In some exemplary embodiments, this configuration reduces the attenuation of the sampled signal by 15 dB. This compensates for the frequency variation of the three bandpass filters 205, 2 1 5 and 225. In some exemplary embodiments, the integrated circuit 'coupled to the inductor L1' can bias the inductor of the input BPF 205 by a half of the vdd to maximize the input voltage swing without violating the integrated body. The specification of the circuit, while capacitor C4 provides a return path to the ground. In some exemplary embodiments, sufficient precautions are taken regarding the maximum breakdown voltage of the circuit (eg, by using a Zener diode for ESD, serial input stage, larger channel) The device, LDD MOSFET, etc.) can have a higher bias voltage for the inductor L1. The LNA-BPF 215 also includes an SCa 315 having a number of "n+1" capacitors. In this exemplary embodiment, each capacitor in SCA 315 includes a corresponding transistor switch (e.g., M〇s transistor) for activating the grid. Similar to the input BPF 205, the resonant frequency of the LNA-BPF 215 can be adjusted by adjusting one or more of the capacitors of the SCA315. In this exemplary embodiment, LN A 210 is a tandem LNA having two transistors M4 and M5. The series-connected LNA 210 can use frequency dependent degradation, which can be initiated at a high frequency by deactivating the start switch M7. This can be used for the following purposes: to increase the input linearity at high frequencies and to provide sufficient gain at low frequencies to maintain the low noise index of LNA 2 10 by activating the switch]V17. The capacitors and switches in each of SC A 3 05, 310, and 315 can be configured to avoid charge pumping due to their single-ended nature. As shown in FIG. 3, the MOS switches M10 to Min can be inserted between the capacitors C10 to Cln and the chip input terminal, 20 201214999 - the M 〇 S switches M20 to M2n are inserted in the capacitors C20 to C2n and coupled to the AC. This is achieved between capacitors C4 and between MOS switches M3 〇 to M3n interposed between the outputs of capacitors C3 0 to C3n and LNA 210. High q external inductors L1 and L2 can be used in HIPCF canceller 130 instead of on-wafer inductors to provide higher frequency selectivity. SCA 3〇5, 31〇, and 315 are used with sufficient tuning range to compensate for the two off-chip inductors and the spread spectrum of L2· and the parasitic capacitance associated with the printed circuit board. The integrated circuit having the components of the HipcF canceller 130 can have an input pin, an ac ground pin, and an LNA pull-up pin, each having a plurality of E S D diodes arranged in series to achieve a large signal swing. In some exemplary embodiments, one or more of band pass filters 2〇5, 215, and 225 are implemented as parallel resonant circuits. In some exemplary embodiments, one or more of band pass filters 205, 215, and 225 are implemented as a series resonant circuit. In some exemplary embodiments, one or more of band pass filters 2〇5, 2 1 5, and 225 are implemented as low pass filters rather than band pass filters. For example, if the primary carrier tone of the transmitter 1 〇 5 has a frequency greater than the frequency range used for interference suppression, the bandpass filters 205, 2 15 and 225 can be replaced with a low pass filter. By. In some exemplary embodiments, one or more of band pass filters 205, 2 15 and 225 are implemented as high pass filters instead of band pass filters. For example, if the primary carrier frequency of the transmitter 1〇5 has a lower frequency than the frequency range used for interference suppression, each of the bandpass filters 205, 215, and 225 can be replaced with a high-pass data filter. In some exemplary embodiments, a combination of low pass, high pass and band pass filters may be used in place of band pass filters 205, 215 and 225. 21 201214999 FIG. 4 depicts a spectrogram of a signal received at a victim receiver antenna (such as antenna 120 of FIG. i) in accordance with certain exemplary embodiments. Referring to Figures 1 and 4, spectrogram 400 shows the amplitude 4〇3 of the apostrophe received at antenna 120 plotted against signal frequency 4〇2. The spectrogram 4A includes a first peak value 4〇4 corresponding to the carrier frequency Ft of the infringing transmitter 105 and a second peak value 4〇5 corresponding to the channel frequency Fr of the victim receiver 135. The spectrogram 4A also includes a noise sideband 406 corresponding to the phase noise generated by the infringing transmitter 1〇5 or its poor spectral component. In some exemplary embodiments, the difference in amplitude between the second peak 405 and the noise sideband 406 does not correspond to the preferred signal to noise ratio (Snr) of the victim receiver for proper reception. 5 depicts an input at a victim receiver (such as receiver 135 of FIG. 1) after an in-band bad spectral component is removed by a hjpcf canceller (such as HIPCF canceller 130 of FIG. 1), in accordance with certain exemplary embodiments. The k that is received at the end is called the frequency 510. Referring to Figures 1 and 5, spectrogram 500 shows amplitude 403 of the signal received at receiver 135 plotted against signal frequency 4〇2. Spectrogram 500 includes a noise sideband 506 corresponding to phase noise or other undesirable spectral components produced by infringing transmitters 〇5. The noise sideband 506 differs from the noise sideband 406 of the spectrogram 400 in that the noise sideband 506 includes a notch 507 centered on the channel frequency FR of the victim receiver 135. This notch 507 is generated by the compensation provided by the interference compensation signal generated by the HIPCF canceller 130 and applied to the input of the receiver 135. In some exemplary embodiments, the SNR of the signal is improved by the amount corresponding to the depth of the notch 507. Thus, the notch 507 improves the signal SNR,

Therefore, the sensitivity of the victim receiver 135 is increased. For example, the elimination of improved phase noise or other undesirable spectral components by the HIPCF 22 201214999 canceller 130 results in better tracking of the deeper notch 5〇7 and thus the victim receiver 135. FIG. 6 is a block diagram of the 〇 enhancement type ρρρ 225 of FIG. 2, in accordance with some exemplary embodiments. Specifically, FIG. 6 is a transistor level diagram of the Q-enhanced BPF 225. The exemplary Q-enhanced type 225 includes - an inductor L3, a bypass switch (four), and a slot pair like 615. In certain exemplary embodiments, the electric salt is 1 k ^. Electrical J (device L3 is a low Q on-wafer spiral inductor. In some exemplary embodiments, Rayham Crush τ 1 λ an electric device L3 is a 鬲Q off-chip inductor. Similar to a bandpass filter, 205 and 215, the resonant frequency of the Q enhanced 225 can be set (eg, 'automatically by the controller 235') to the receiving frequency of the receiver 135 to transmit the in-band component and the chopping, blocking the transmission from the sampling. The base of the signal and the out-of-band blocking signal or the intensity reduction. The SCA 615 includes a number of "n+1" capacitors (10). In the illustrated embodiment, each capacitor C4A through C4n includes two corresponding transistor switches. (for example, M〇s transistor) for starting the capacitor. For example, capacitor C40 includes transistor switches M4 and M5 (in addition, Q-enhanced BPF 225 also includes two series connected in parallel with SCA 615 Voltage-controlled capacitors VC 1 and VC 2. In some exemplary embodiments, the voltage-controlled electric grids VC1 A VC2 are varactors. The two voltage-controlled capacitors VC1 and VC2 are disposed between the two.帛(6), which electrically couples voltage-controlled capacitors VC1 and VC2 to digital/analog (d/A) Converter 650. D/A converter 650 varies voltage levels of voltage-controlled capacitors Is VC1 and VC2 in response to signals received from controller 235. Controller 235 can be activated by one or more of capacitors C40 through C4n ( The resonant frequency of the q-enhanced BpF225 is adjusted by switching the switches M40 to M4n and M50 23 201214999 to M5n) and by controlling the voltage level at the center tap 655 and thus the capacitance of the voltage-controlled capacitor VC1 & VC2. Voltage-controlled capacitor VC1 And ¥(:2 enables the controller 235 to fine tune the resonant frequency of the Q-enhanced BPF 225. The exemplary Q-enhanced BPF 225 also includes a cross-coupled pair 620 of transistor switches M8 and M9 in parallel with the SCA 615. Cross-coupling A pair of negative resistors is provided to reduce the resistance of the LC slots formed by inductors L3, SCA 615 and voltage controlled capacitors VC1 and VC2. Q-enhanced BPF 225 includes a number of "n+1" current sources M6〇 to M6n (eg, binary weighted), each having a gate terminal eQ enhancement b that is electrically coupled together and electrically connected to a reference current (Ref_C), pf 225 also includes a number of "n+i" current switches M70 to M7n. Start by starting and undoing (eg The current switches M70 to M7n corresponding to the current switches M70 to M7n of the controller 235) select one or more of the current sources M60 to M6n, and the currents in the switches M8 and M9 can be adjusted to adjust the resistance of the LC tank 610. Therefore, the Q enhancement type can be adjusted. The Q factor of the BPF 225. For example, the Q factor of the q-enhanced BPF 225 can be adjusted to the desired level to improve or maximize the filtering of the out-of-band signal without the q-enhanced BPF 225 oscillating. The Q-enhanced BPF 225 also includes a bypass switch 670 having a resistor R8 that is electrically coupled to the two transistor switches M80 and M81 and disposed between the two transistor switches M80 and M81. As discussed in more detail with respect to Figures n through 15, switches M80 and M8 1 can be turned "on" or "on" during calibration of input BPF 205 and LNA-BPF 2 1 while the starting current sources M60 through M6n are deactivated. When the switches M80 and M81 are activated, the resistor Rg can be turned off for the lc 24 201214999 slot. The switch M80 and ]Vi8 1 are typically deactivated during the operation of the gauge.

FIG. 7 is another block diagram of a Cong CF canceller 13A depicting additional components of HIpCF丨3〇, in accordance with certain exemplary embodiments. The HIPCF 130, as exemplified in Figure 7, also includes bypass switches 72A and 725 for use during calibration of the HIPCF 130. In particular, the input includes a bypass switch 720 and the LNA 215 includes a bypass switch 725. The bypass switch 720 includes a transistor switch M82 and a resistor. Similarly, the bypass switch 725 includes a transistor switch M83 and a resistor R3. Each of the bypass switches 72〇, 725 and the Q-enhanced BpF bypass switch 670 during configuration of the HIPcf 130 (eg, using an automatic test equipment (ATE), on-stage measurement, or in-situ calibration) The band pass filters 205, 215, and 225 can be selectively tuned by activation and deactivation. The HIPCF canceller 13A also includes a buffer 77〇 disposed between the VGA 22A, the Q-enhanced BPF 225 and the I/Q modulator 23A. The auxiliary circuit 240 includes a power detector 745 electrically connected to the output of the buffer 77. The power detector 745 measures the power level of the sampled transmission signal at the output of the buffer 77, and will An indication of the measurement result is provided to the a/d converter 750. The A/D converter 75 turns the indication into a digital signal and provides the digital signal to the controller 23 5 . The auxiliary circuit 240 also includes a temperature sensor 755 having an output electrically coupled to the controller 235. The temperature sensor 755 is positioned on a wafer (integrated circuit) (the HIPCF canceller 13 is mounted or fabricated on the wafer) to measure the temperature of the wafer. Controller 235 can receive temperature measurement results from temperature sensor 755 and use these measurements for monitoring, calibration, and for 25 201214999 temperature compensation. In some exemplary embodiments, the output of temperature sensor 755 is coupled to an A/D converter (such as A/D converter 750 or a second A/d converter). In an exemplary embodiment having a shared A/D converter 750 for the power controller 745 and the temperature sensor, the controller 235 can request which of the two measurements (power or The temperature - the signal is supplied to the A/D converter. FIG. 8 depicts a functional block diagram of a communication system 8 (8), in accordance with certain exemplary embodiments. The exemplary communication system 8A includes two communication devices 8〇5 and 850' each having transmitters 81A and 855 and having receivers 820 and 865, respectively. The communication system 8A includes a first HIpcF canceller 88A for compensating for free transmission $810, the signal transmitted via the first antenna 825 being imposed on the input and/or interference on the input of the receiver 865. The communication system (10) 〇 also includes a second HIPCF canceller 885 for compensating for the noise and/or interference imposed by the free transmitter 855 via the second antenna 87〇 on the input of the receiver. Therefore, the communication system 8() includes an interference compensation circuit for protecting the two communication devices 8 〇 5 and 850. For example, the pass device 805 can be a cellular radio and the communication device 85 can be a wiFi radio. In this example, the cellular radio is protected from interference imposed on the cellular radio receiver by the ss number transmitted by the wiFi radio, and conversely, the WiFi radio is protected from free cellular radio transmission. The interference imposed by the signal on the WiFi receiver. The HIPCF requester 880 receives samples of the signals transmitted by the transmitter 81A via a sampling device 89 that is electrically coupled to the output of the power amplifier 815 of the transmitter, and processes the samples to produce an interference compensation signal. HIp(:F消26 201214999 divider 880 applies an interference compensation signal generated at cancellation point 833 to the input of receiver 865, and then, the interference compensation signal is cancelled, suppressed, or otherwise compensated for imposed on receiver 865. The HipcF Elimination 880 may include a controller similar to that of Figure 2 that controls $235, which is directed to one or more calibrations and/or multiple spectral algorithms to improve noise and/or: Compensation. The controller may receive feedback (such as "received signal quality does not match") and use this feedback to improve noise and/or interference compensation during execution of the algorithm. Similar to cancellation point 134, cancellation point 833 may be implemented as A polymeric electrical conductor, a consumer, a summing node, an adder, or other suitable technique. "Similarly, the HIPCF canceller 885 receives the transmitted by the transmitter 855 via a sampling device 895 that is coupled to the output of the work of the transmitter." A sample of the signals, and processing the samples to generate an interference compensation signal. The HIPCF canceller 885 applies the interference compensation W generated at the cancellation point 834 to the input of the receiver (4), and then, the interference compensation The signal cancels, suppresses, or otherwise compensates for noise and/or interference imposed on the receiver 82. The chat CF canceller 885 can include a control similar to the controller (3) of FIG. 2, which performs one or more calibrations and One or more tuning algorithms compensate for good Dfl and/or interference. The controller can receive feedback (such as "receive nickname quality indicator") and use the feedback to improve the noise and/or during execution of the algorithm. Or interference compensation. Similar to the elimination point 134, the elimination point can be implemented as a polymeric electrical conductor, a lighter, a summing node, an adder, or other suitable technique. Figure 9 (iv) a lookup table according to certain operational embodiments Referring to Figures 2, 7, and 9, the lookup table 9 can be stored in the memory device 760 of the HipcF canceller 13〇27 201214999. The exemplary lookup table 900 includes a center frequency setting 910 for inputting the Bpf 205. The center frequency setting 920 for the LNA-BPF 215 and the center frequency setting 930 for the Q-enhanced BPF 225. In the exemplary embodiment, the 'input BPF center frequency setting 91 〇 includes three frequency values (Freql, Freq2, and Freq3), has been targeted Equivalently characterized bandpass filters 205, 2 15 and 225 » For example, in the embodiment of the mobile τ Δ receiver 135 'the bandpass filters 205, 215 can be characterized at 450 MHz, 600 MHz and 770 MHz and Each of the input BPF center frequency settings 910 also includes a switched capacitor array setting (SCA_Input_BPFi to SCA-Input_BPF3) for each of the three frequency values (Freqi to Freq3). The switched capacitor array settings (SCA Input_BpF1 to SC A-Input_BPF3) control the resonant frequency control SC A 3 05 and SC A 3 10 for each of the frequencies (Freqi to Freq3) and thus the input BPF 205 for these frequencies the way. The input BPF center frequency setting 910 also includes temperature coefficient values (Tempcol to Tempco3) for each frequency value (Freql to Freq3). » Temperature coefficient values (Tempcol to Tempco3) are used by the controller 235 to adjust the SCA 305 based on temperature changes. And 3 1 0 settings. Similarly, the 'LNA-BPF center frequency setting 920 includes switching capacitor array settings (SCA - LNA - BPF1 to SCA - LNA - BPF3) for each of the three frequency values (Freqi to Freq3). The switched capacitor array setting (SCA_LNA_BPF1 - SCA-LNA_BPF3) controls each of the frequencies (Freqi to Freq3) and therefore for these frequencies.

LNA-BPF 215 Resonant Frequency Control SCA 315 Mode" LNA-BPF 28 201214999 The center frequency setting 920 also includes temperature coefficient values (Tempcol to Tempco3) for each frequency value (Freql to Freq3). These temperature coefficient values (Tempcol to Tempco3) are used by the controller 235 to adjust the setting of the SCA 3 15 based on the change in temperature. The Q-enhanced BPF center frequency setting 930 includes switching capacitor array settings (SCA_QE_BPF1 to SCA-QE_BPF3) for each of the three frequency values (Freq1 to Freq3), respectively. The switched capacitor array setting (SCA-QE-BPF1 to SCA-QE-BPF3) controls the resonant frequency control SC A 615 for each of the frequencies (Freql to Freq3) and thus the input bpf 2〇5 for these frequencies the way. The Q-enhanced BPF center frequency setting 920 also includes temperature coefficient values (Tempcol to Tempco3) for each frequency value (Freql to Freq3). These temperature coefficient values (Tempcol to Tempco3) are used by the controller 235 to set the S C A 6 15 based on the temperature change s. The Q-enhanced B p f center frequency setting 9 3 0 also includes DAC settings (DAC1 to DAC3) for voltage-controlled capacitors VC 1 and VC2 for each frequency (Freql to Freq3). The Q-enhanced BPF center frequency setting 920 also includes temperature coefficient values for each frequency value (Freql to Freq3) (CurrentTempcol to

CurrentTempco3). These temperature coefficient values (currentTempcol to

CurrentTempco3) is used by the controller 235 to adjust the setting of the current switches M70 to M7n and thus the bias current in the q-enhanced bpf 225 based on the change in temperature. The exemplary lookup table 900 also includes a seed value 940 for the j/Q modulator 23A. The seed value 940 includes the in-phase and quadrature (LQ) settings for the Ι/Q modulator 23〇 at each frequency (Freqi to Freq3) 29 201214999 ((I1, qi) to (13, Q3)) . Lookup Table 9〇〇 also includes miscellaneous settings of 95〇. The miscellaneous setting 950 includes a temperature for performing calibration of the HIPCF canceller 13〇, a process parameter of a manufacturing lot of the canceller 130, a temperature coefficient set by the DAC 65〇, a transistor switch M8 required to hold the Q-enhanced BPF 225, and The minimum current that M9 turns on and the threshold for the detected oscillation of the power detector 745 on the chip. The lookup table 900 is stored on the memory device 76 and is accessed by the controller 235 to adjust the settings of certain components within the HIPCF canceller 130 during normal operation and during the calibration and tuning procedures discussed below. Many of the settings in lookup table 9〇〇 are also filled during such calibration and tuning procedures' as discussed in further detail below. Circle 10 is a flow diagram depicting a method for calibrating certain components of HIPCF canceller 130 in accordance with certain exemplary embodiments. After the HlpcF canceller 130 is fabricated, for example, in an integrated circuit, in block 1〇〇5, during the ate or on-stage characterization procedure, the initial lookup table in FIG. 9 is filled in. set up. In block 1010, the value set for the lookup table 900 is loaded into the internal register of controller 235 during the application phase when the HIPCF canceller is powered. The controller can access the lookup table 900 and use the current temperature measurement results from the temperature sensor 755 and the channel frequency control pjjpcF canceller 13 调 to which the receiver modulates. An optional calibration routine can also be performed in block 1〇1〇 to calibrate band pass filters 205, 215 and 225 and/or I/Q modulator 23A. In block 1015, if the channel of the receiver 135 changes, the I/Q tune 30 201214999. The transformer 230 is recalibrated by the controller 23 5 . This calibration can improve noise and/or interference cancellation based on the receiver's received signal quality indicator and the cancellation algorithm described below. In block 1020, the controller 235 triggers the bandpass ferrites 2〇5, 215, 225 and I/ in response to a command from the user or in response to a temperature change exceeding a predetermined threshold (eg, 10 degrees Celsius). Calibration of the Q modulator 230. During the calibration procedure of method 1000, the value in lookup table 9〇〇 is updated. 11 is a diagram depicting a method for configuring a filter of a HIPCF canceller 130 for a desired center frequency (eg, for a mobile TV embodiment, 450 MHz, 600 MHz, or 770 MHz), in accordance with certain exemplary embodiments. Flow chart. In block 1 105, the input BPF 205 is calibrated. The lnA-BPF 215 and the Q-enhanced BPF 225 are bypassed by activating the bypass switches 725 and 670 and deactivating the startup bypass switch 72. A pilot tone or tuner signal is applied to the input of the HIPCF canceller 130 and the power level of the pilot tone or tuner signal is measured at the output of the HIPCF canceller 130. The settings of SCA 305 and SCA 310 are adjusted based on the measured power level until the power level reaches an acceptable level. The settings of 8匸8 3〇5 and SCA 3 10 corresponding to acceptable power levels are entered in lookup table 9〇〇 for later use by controller 235. Block 11〇5 is discussed in further detail below with reference to FIG. In block 1110, the LNA - BPF 215 is calibrated. Input bpF 205 and Q enhanced BPF 225 are bypassed by activating bypass switches 720 and 670 and deactivating startup bypass switch 725. In the case where the pilot tone or tuner signal is still applied to the rounding end of the HIPCF canceller 130, the setting of the SCA 3 15 is adjusted based on the measured power level until the measured power level is acceptable 31 The 201214999 level is up to now. The settings of SC A 3 1 5 corresponding to acceptable power levels are populated in lookup table 900 for later use by controller 235. Block Π 10 is discussed in further detail below with reference to FIG. In block 1 1 1 5, the Q-enhanced BPF 225 is calibrated. Block 1115 is discussed in further detail below with reference to FIG. In block 112, the temperature coefficients for bandpass filters 205, 215, and 225 are calculated. In some exemplary embodiments, the ATE (or on-board measurement device) calibrates the settings for each of the band pass filters 205, 2 15 and 225 for more than one temperature. For example, the bandpass filter can be calibrated at room temperature (eg, 27 °C) at 70 C and at 0 °C. Controller 235 can calibrate the temperature coefficient by taking the settings for each bandpass filter 2〇5, 21 5, 225 at each temperature. The temperature coefficients can be stored in the corresponding blocks in the lookup table 900 in blocks 910, 920, 930, and 950. In block 1125, the I and q seed values for the i/q modulator 23 are calibrated. In some exemplary embodiments, the ATE (or on-stage measurement device) may use settings similar to the circuit 1 depicted in FIG. Transmitter 105 can be activated and one or more of the cancellation algorithms discussed below can be performed for the desired center frequency to identify better or acceptable cancellation points. The (I, Q) setting corresponding to the identified cancellation point can be stored in the block 94 of the lookup table 9〇〇. After block 1125, method 1100 ends. Of course, the method 1100 can be executed more than once. For example, the method can be performed during ATE and then performed again after the wafer or system is placed in operation. FIG. 12 is a depiction of the use as illustrated in FIG. η, in accordance with certain exemplary embodiments. A method for calibrating the input BpF 2 〇 5 of the HIPCF canceller 13 2012 32 201214999 1105. In block 12〇5, the bypass switch $ is activated and the bypass switch 720 is deactivated. This bypasses the LNA_BpF 215 and the q-enhanced BPF 225 for the calibration of the input 2()5. In some exemplary embodiments, controller 235 operates bypass switches 670, 720, and 725 in response to commands configured to bandpass filters 2〇5, 2丨5, and 225. In block 1210, a pilot tone or spectrum modulator signal (e.g., a traveling TV signal) having a desired center frequency (e.g., 45 〇 MHz, 600 MHU 770 MHz) is applied to the input of the HIPCF cancellation H 130. In some exemplary embodiments, the HIPCF; the multiplexer 13{) 锁 the phase locked loop on the wafer generates a pilot tone or tuner signal. In some exemplary embodiments, the pilot tone or the scheduler signal is generated by reusing the phase locked loop of the receiver 135 (e.g., via one of the output pins of the receiver). In block 1215, the power level of the pilot or quad signal is measured at the output of the mpCF canceller (10). In some exemplary embodiments, the 'output power' of the pilot is measured using an ATE or on-stage characterization device. For example, an ATE or on-stage characterization device = 4: 4 analysis benefits. In some exemplary embodiments, the received signal t-indicator transmitted from the receiver 135 is used to measure the pilot tone or (quad). In some exemplary embodiments, a power detector is used = the output power level of the pilot tone or tuner signal is set. Set = 2: Medium: Controller 235 performs on SCA 3 ° 5 and ... 31. Or, s s, and measure the output power level from the track generated by each adjustment. Control, 235 can continue to enter = to the pilot or the output power level of the tuner signal reaches or exceeds 33 201214999 until the next, better or maximum level. ±(ra 卜 or other, controller 235 has entered a certain number of tidy and recorded guides, such as the gate and the worm's output of the k-hand is early (for example, in the memory device -^ # ^ . 'and identify the record with the best, better or most power level. X. ; output rate level. In some exemplary embodiments, controller 235 is flat In the daily plus or minus program, the glance for the SCA 305 and SCA 31 is used to determine the value of the beta τ (for example, for digital SCA, one least significant bit S ("LSB") or multiple lsb) In some such exemplary embodiments, binary algorithms (such as those illustrated in Figure 2 and discussed below) can be used to find better settings for SCA 305 and SCA 31. In block 1225, the controller 235 stores the desired center frequency and the SC~SCAJ: setting for the acceptable, preferred or maximum level in the memory 装9(10) in the memory device. For example, , you can store the desired ~ frequency in the block "Freql", and store the settings for sca 305 and SCA310 in the block. In "SCAj[nput_BpFi". After block 1225, method 1 1 〇 5 proceeds to block 1 1 1 0 as mentioned in Figure 11. Figure 13 is a depiction in accordance with some exemplary embodiments. A flowchart of a method 1110 for calibrating the LNA-BPF 2 15 of the HIPCF canceller 13 as mentioned in block 1110 of the figure. In block 丨3〇5, the bypass switch 720 is activated. 670 and deactivates the bypass switch 725. This bypasses the input BpF 2〇5 and Q-enhanced BPF 225 for calibration of the LNA_BPF 215. In block 1310, the controller 235 performs the setting of the SCA 315. Multiple adjustments' and measurement of the pilot tone or tuner 34 201214999 round-out power level generated by each adjustment. The controller 235 can continue to adjust until the 'output or adjustment of the output signal rate of the 5 signal I meets or exceeds the acceptable, better or maximum level. In addition or other, the control H 235 can be used to control the output power level of the pilot tone or tuner signal / for example, 'in the memory device f 760) and identify the output power level of the record with the best, better or highest power level. In some exemplary embodiments, controller 235 scans the set values for SCA 315 in a monotonically increasing or decreasing routine (e.g., for a digital SCA, one LSb or more LSBs at a time). In some exemplary embodiments, a binary algorithm (such as the algorithm illustrated in Figure 20 and discussed below) may be used to find a preferred setting for SCA 315. In block 1315, controller 235 stores the settings for SCA 315 corresponding to the acceptable, preferred or maximum levels in lookup table 900 in memory device 76A. For example, the settings for sc A 315 can be stored in the block "SCA-LNA_BPF1". After block 1315, method 111 continues to block 1115 as mentioned in FIG. 14A and 14B (collectively referred to as FIG. 14) depict a method 1115 for calibrating the Q-enhanced BPF 225 of the hipcf canceller 130, as mentioned in block 111 5 of FIG. 11, in accordance with certain exemplary embodiments. flow chart. In block 1405, bypass switches 720 and 725 are activated and the bypass switch 670 is deactivated. This bypasses the input BPF 205 and the LNA-BPF 215 for calibration of the q-enhanced BPF 225. In block 1410, a bias current is applied (for example, M70 to M7n) in order to maintain the transistor switches M8 and M9 and the current sources M60 to M6n in the pair 620 in operation and in operation to avoid 35 201214999. The amount of current applied to the current-free Q-enhanced BPF 225, such as 'from controller 235' to the current-switching switches M70 to M7n, may correspond to the "minimum current for QE" field of the look-up table_ value. In block 1415, controller 235 performs a setting for sca or an evening adjustment! And measure the output power level of the pilot tone or tuner signal generated by each adjustment. Control $ 235 to continue adjusting until the output power level of the pilot or modulator signal reaches or exceeds the acceptable, preferred or maximum level. Additionally or alternatively, control _235 may perform a certain number of adjustments and record the output power level of the pilot or modulator signal (as in memory device 760) and identify the best, better or highest power level. The output power level of the quasi-recorded. In some exemplary embodiments, controller 235 scans for the set value for SCA 615 in a monotonically increasing or decreasing procedure (e.g., for a digital SCA, one or more LSBs). In some exemplary embodiments, a binary calculus (such as the algorithm described in "20 and discussed below) may be used to find a preferred setting for SCA 615. In block H20, the controller 235 increases the amount of current applied to the AC-on-cell transistor %(10) by increasing the current switch to M7n setting. In some exemplary embodiments, the amount of current is increased by a few, for example, four LSBs. At block 142", the pilot tone or number is cut. In block 1430, controller 235 makes an inquiry as to whether there is any jitter generated by Q-enhanced BPF 225. In certain illustrative embodiments The inquiry includes the output of the HIPCF canceller 13A, the 201214999 rate level and the predetermined threshold value at the ATE, or (4) the threshold value in the miscellaneous value 950 of the lookup table. The power detector output threshold is compared. If the measured output power level is below the threshold, the controller 235 determines that there is no oscillation. If the (4)^235 decision does not exist or there is a sufficiently low oscillation, then the method ιι 15 continues to block 1435 where the control H 235 is again turned on the pilot tone or tuner signal and the pilot is tuned or The tuner signal is applied to the input of HIPCF. After the block, the method returns to block 1415. If the controller adds a decision to vibrate, then method ill5 proceeds to block 144. In 1440, controller 235 reduces the amount of current applied to current switches M7 〇 to M7n to a level prior to oscillating. In block 1445, control benefit 235 will correspond to the current level before the horn. The setting of sca is stored in the memory device i 76〇. For example, the setting for SCA615 can be stored in the block "sca_qe-Βρρι"^. In block 1450 the 'pilot or tuner signal is restarted and applied to the input of HIPCF canceller 130. The controller 235 performs - or a plurality of adjustments to the settings of the DAC 650 for biasing the voltage controlled capacitors VC1 and VC2, and the target is measured from the mother-adjusted The output power level of the pilot or tuner signal. The adjustment of the DAC 65〇 adjusts the voltage level at the voltage controlled capacitors VCM and VC2. Controller 235 may continue to adjust i until the output power level of the pilot tone or spectrum modulator signal exceeds an acceptable, preferred or maximum level. Additionally or alternatively, control; 235 may perform a certain number of adjustments and record the pilot power level of the pilot tone or the spectrometer signal (eg, in memory pack i 760), and the identification: 37 201214999 Best, The best or highest power level record _ < Han out power level. In some exemplary embodiments, 'controller 235' scans the set value for DAC 650 in the early or weekly increase or decrease program (eg, - one LSB or multiple LSBs long). In an embodiment, "using a binary algorithm (such as illustrated in Figure 20 and discussed below)

The method is to find a better setting for the DAC 650. In block (4) 5, the pilot tone adjustment or modulation I is cut off in __, and the output power level of the HIPCF cancellation ϋ 130 is measured. In block (4), the 'similar to block 1430' controller 235 makes an inquiry as to whether there is any vibration generated by the q enhanced type 225. If the controller determines that there is an oscillation, then method 1115 proceeds to block 147. If controller 235 determines that there is no oscillation, then method Π5 proceeds to block 1475. In block 1470, controller 235 reduces the current level used to bias the cross-coupled transistors M8, M9 by reducing, for example, a few LSBs by setting the current switches Μ7〇 to η7η. Method 1115 returns to block 1450 after the current level is lowered. In block 1475, controller 235 stores the settings for dac 650 and current switches M70 through M7n in lookup table 900 in memory device 76A. For example, the settings for current switches M7〇 to M7n can be stored in the track "Currentl", and the settings for the DAC 650 can be stored in the booth "DAC1". After block 1475, method 丨j 15 ends. Of course, the method can be repeated any number of times for any number of frequencies. For example, ° bandpass filter, wavers 205' 215 and 225 can be calibrated for two frequencies (Freq 1, Freq2 and Freq3). 38 201214999 FIG. 15 is a flow diagram of a method 1500 for calibrating the wheel of the HIPCF canceller 130 of FIG. 7 into the BPF 205, in accordance with certain exemplary embodiments. This method 1500 is an alternative to the method 11〇5 of FIG. In the block test T ’ revoke the start bypass _ 72 〇, 725 and 67 〇. For example, controller 235 can deactivate start bypass switches 72A, 725 and . In block 1510, Jiang: and Death & Face + applies a pilot tone or tuner signal having the desired center frequency to the input end of the HIPCF cancellation _(10). At block 1515, the pilot of the reflected or tuner signal (e.g., reflection coefficient or return loss) is reflected at the input of the HIPCF canceller 13A. For example, this measurement can be performed by a power detector or a spectrum analyzer. In block 1520, controller 235 performs one or more adjustments to the settings of SCA 3〇5 and SCA 31〇, and f measures the pilot or (4) signal of the reflection. The controller 235 can continue to adjust until the pilot of the reflection is adjusted or adjusted (4) (4) 'to reach or exceed the acceptable, preferred or minimum level. Additionally or alternatively, the controller 235 can perform a certain number of adjustments and record the reflected pilot tone or tuner signal (eg, 'in the memory device 76') and identify the best, preferred, or lowest level. Record the reflected pilot or tuner signal. In some exemplary embodiments, the controller 235 scans for settings such as 3〇5 and 31〇 in a monotonic addition or subtraction procedure (eg, for digital SCA, - times-LSB or more) LSB). In some exemplary embodiments, binary algorithms, such as those illustrated in circle 2 且 and discussed below, may be used to find preferred settings for SCA 3〇5 and Sca. In block 1525, controller 235 stores the desired center frequency and the settings for SCA 305 and SC A 310 that correspond to the preferred or minimum level of 39 201214999 in memory device 760. in. For example, the desired center frequency can be stored in M # r c t . Only in the "Freql" position, the settings for 305 and SCA 310 can be stored in the rotten position "SCA_Input_BPF1". 16 is a diagram depicting a method for judging a given frequency for a given frequency, in accordance with some demonstrative embodiments! Flow chart of 6(10). For example, the method 16 can be performed in response to the user applying the channel change to the action τν. In some exemplary implementations, the settings of the bandpass filters 205, 215, and 225 are used and used. The I and Q seed values for each channel that the receiver 135 can tune to. In some exemplary embodiments, lookup table 900 includes settings for each band pass filter 2〇5, 2丨5, and 225 and Q seeds for a predetermined number (eg, 3) of channel frequencies. value. For such embodiments that do not include settings for calibration of each channel frequency, method 1600 provides for calculating SCA switch settings for each of band pass filters 205, 215, and 225 at any of the applied selectable channel frequencies. An exemplary procedure. The exemplary method 16 considers the calibration values in the lookup table 900 for a predetermined number of channel frequency identifications and the actual temperatures measured by on-wafer temperature sensors such as the 'temperature sensor 75 5 . In block 1605, the controller 235 makes an inquiry as to whether or not to start determining the switch setting for each band pass filter 205, 2 1 5, 225. In some exemplary embodiments, controller 253 communicates with receiver 135 to determine if the receive frequency for receiver 135 has changed, for example, due to a change in channel for receiver 135. . If the reception frequency for the receiver has changed, the controller 235 determines to start determining the switch settings for each of the band pass filters 40 201214999, 205, 215, 225, and proceeds to block 161 (^ otherwise, method 16 0 0 remains in block j 6 〇 5. In some exemplary embodiments, controller 235 determines if the wafer ash temperature resident by HIpCF canceller 130 has changed. Controller 235 monitors the received temperature measurement to determine Whether the temperature has changed to a certain threshold. If the controller 235 determines that the temperature has changed by an amount equal to or exceeding the threshold value, the controller 235 determines to start determining for each band pass filter 2〇5, 2丨5. The switch of 225 is set and continues to block i 6 . Otherwise, method 丨 6 〇〇 remains in block 1 605. In some exemplary embodiments, controller 235 determines lookup table 90 ό Whether the setting or value in the lookup table 9 is changed or not has been updated. If the controller 235 determines that the lookup table has changed, the controller 235 determines to start determining the switch setting for each of the band pass filters 205, 215, 225. And continue to enter The process proceeds to block 1610. Otherwise, the method 1600 remains in block 16〇5. At block 1610, controller 235 receives the receive frequency ("target frequency") for receiver 135, from lookup table 9〇〇 Current calibration values for bandpass filters 205, 215, and 225, immediate or near instantaneous temperature measurements from temperature sensor 755, and temperature values during calibration from lookup table 900. In block 1615, The controller 235 makes an inquiry as to whether the target frequency is less than the frequency threshold. For example, in some mobile TV embodiments, 'this frequency threshold is set to be in the middle of the receive band corresponding to some of the action τν tuners. 600 MHz. If the target frequency is less than the frequency threshold, then method 161 5 proceeds to block 1 6 2 0. Otherwise, the method proceeds to block 41 201214999 block 1625. At block 1620, controller 235 calculates the variable " _", which indicates the difference between the target frequency and the frequency threshold. The controller 235 performs an interpolation procedure (eg, linear interpolation) using two or more of the calibration values in the lookup table 900. To judge For the setting of SCA 3〇5, 31〇, 315 and 615, the dac setting for voltage controlled capacitors VC1 and VC2 and the bias current switch setting for current switches M70 to M7n. For example, for each of the foregoing The component, controller 235, in linear interpolation calculation by 1) centroid, uses settings stored for a first frequency (such as Freqi) and stored for a second frequency (such as Freq2). Set to determine the settings for this component. In block 1625, the controller calculates a variable DeltaF indicating the difference between the target frequency and the second frequency value. In some exemplary embodiments, if the frequency threshold is 600 MHz, the second frequency value is 77 〇 MHz. These frequency values are exemplary and not limiting, and without departing from the scope and spirit of the present invention. Other frequency values can be used. Similar to block 1 62 〇, the controller performs an interpolation procedure using two or more of the calibration values in the lookup table 900 to determine for SCA3〇5. , 31〇, 315 and 615 settings, DAC settings for voltage controlled capacitors VC1 and VC2 and bias current switch settings for current switches M70 to M7n. For example, for each pre-it, the component controller 235 uses the settings stored for the first frequency (such as Freq2) and stored for use in the linear interpolation calculation by DeltaF. The setting of the two frequencies (such as Freq3) to determine the settings for this component. 42 201214999 As shown in blocks 1620 and 1625, depending on the target frequency, method 1 600 determines the settings for the components of hij>cf canceller 130 using two different sets of calibration settings, which enables controller 235 to The appropriate settings for these components are determined using the calibration settings most near the target frequency. In block 1630, controller 235 determines the temperature compensation by calculating a variable "DeltaTemp" that produces the difference between the actual temperature and the temperature at which the last calibration was performed (stored in position 95 of lookup table 9). The controller (9) also calculates the deviation value caused by the temperature difference for the setting for each component. Controller 235 uses the offset values to determine the final settings for the components at the target frequency. Control H 235 stores the final settings in the internal register for use in the operation of the # component. Note that the Μ modulator 230 and the 〇 setting may not be the temperature compensated in the method _ , = because the I&Q setting can be calibrated using one of the cancellation algorithms discussed below with reference to Figures 17 through 。. Figure 17 depicts an implementation layer (10) of a noise and/or interference cancellation method, in accordance with certain exemplary embodiments. These algorithms may use the feedback signal from the victim receiving benefit to determine the appropriate worker feedback signal for the HIPCF canceller 13 including the quality indicator (such as production, etc.) for the communication system - coffee, coffee, noise floor, Earning, EVM, and location-accurate algorithm execution layer 1740. In this example, the heterogeneous control layer 1730 and the performance in a certain example of the inconsistency, layer 171〇 to 1740

Receiving one of the following three components: 1) victim's fundamental frequency integrated circuit, 2) stand-alone microcontroller, or 3) HIPCF 43 201214999 1 3 0 on-chip control §§ 91 Γ ϊ» ^ 5 (or another control device). For ease of discussion, the respective functions will be performed by controller 235 to discuss layers 171 〇 to 1740 ° at the link control layer 1 7 1 G t, for quality analysis and testing the feedback signal to determine whether cancellation should be initiated to improve victim reception. The sensitivity of the device 135. Due to the nature of the action noise and/or interference cancellation provided by the HIPCF canceller 130, the HIPCF cancellation mode 13G can also eliminate the noise generated by the power amplifier 11 (or another component) at the input of the victim receiver 135. Simultaneously and/or interfere with the output of its own noise floor. Thus, the total noise floor seen by victim receiver 135 is Hlp (: F eliminator 13 〇, output noise floor of receive antenna 120, power amplifier noise and/or interference received by receive antenna 12 〇 And the sum of the phase of the power amplification mo and the gain adjustment of the noise floor (left, by the HIPCF canceller 13A), which in turn can affect the sensitivity of the victim reception °: 35. Therefore, it can be received based on the victim receiver 135 The inter-sequence noise and/or interference of the amplifier 110 determines the determination as to whether to initiate the HIPCF cancellation n 13G to improve the sensitivity of the victim receiver 135. Figure 1 8 Tat Green is directed to 8 according to certain exemplary embodiments. The channel bandwidth of MHz and the CDMA8〇〇 power amplifier's Chuan Hall down-regulates the sensitivity of the TV tuners to the received power amplifier noise. _ See Figure 18, Figure 18G ( ) includes: - the first song: "805 which depicts the action η when the HIPCF canceller 13 is not active. The white sensor sensitivity 'a second curve i 8 i 〇, which depicts the bribe eliminator less than 1%', Or suppress the action τν tuner sensitivity when power amplifier noise. Exemplary embodiment illustrated embodiment, the power amplifier is not present for less than _i74dBm / Hz 201214999. HIPCF advantage of noise canceller 130 starts, the

Maximum cancellation/sensitivity improvement (for example, a power amplification 130 of about -1 60 dBm/Hz is active, in the case of 'in this exemplary embodiment, about 10 dB), because it is received Power amplifier noise is typically much higher than the output noise floor of the HipcF canceller 130. In addition, the link control layer i7i detects whether a particular channel has been optimized in the past in the multi-channel system and passes the settings corresponding to the previous optimization from the memory to the HIpCF i 3〇. An indication of whether the channel has been previously optimized may be stored by the controller 235 in the memory at the end of the optimization or during the optimization. The quality of the received signal to be victimized may be evaluated with respect to feedback received from the receiver (e.g., BER, PER, RSSI, noise floor, SNR, EVM, and location accuracy, etc.) to determine whether to activate the HIPCF canceller 130. For example, if the feedback indicates that the received signal is higher than the combined noise floor (ie, the HipcF canceller 130, the output noise floor of the receiving antenna 120, the power amplifier noise and/or interference received by the receiving antenna 120, and The HIPCF 130 is activated by the phase of the power amplifier 11 and the noise floor of the gain adjustment (via the HIPCF canceller 130). This feature is illustrated in FIG. 19, which depicts an output SNR of a mobile TV tuner tuned at 746 MHz with a channel bandwidth of 8 MHz relative to having an input at victim receiver 135, in accordance with certain exemplary embodiments. Figure 1900 of the received TV signal strength of the received CDMA800 power amplifier phase noise at -161 dBm/Hz. Referring to Figure 19, a diagram 1900 includes a first curve 1905, 45 201214999 which depicts the action TV tuner output SNR when the HIPCF canceller 130 is not active; a second curve 1910 depicting the HIpcF canceller 13 〇 elimination or suppression Action TV tuner output for power amplifier phase noise

SNR. A third 1915 depicts the desired minimum SNR output for the mobile TV tuner. As illustrated in the illustrative embodiments, there may be no advantage of initiating the HIPCF noise canceller 130 when the received mobile TV signal is below _90 dBm. The illustrative link control layer 1710 includes several modes of operation. The controller 235 determines which mode of operation is active based on the signal quality of the signal received by the receiver 135. In some exemplary embodiments, the link control layer 171 includes four modes of operation - maximum cancellation Mode, limited cancellation mode, wait for signal mode, and no signal mode. In this exemplary embodiment, the signal strength received by the right is acceptable (e.g., above an acceptable threshold (-81 dBm in Figure 19)), then the controller 235 begins the maximum cancellation mode. Thereby, the noise and/or interference cancellation of the HIPCF canceller 130 is at the 鬲 level. If the received signal strength is low (eg, between an acceptable level threshold and a very low level threshold, between (1) and -9 dBm in Figure 19), controller 235 begins The limited cancellation mode of the HIPCF canceller 135, whereby the noise and/or interference cancellation level is subjected to a noise floor, and the ''spoon bundle'' receives a ## ratio indicating (for example) a very low signal threshold (eg, at In Figure 19, about _9 〇 dBm) is low, then controller 235 will begin to wait for an acceptable signal pattern, whereby controller 235 delays entry into subsequent = 720 to 174 〇 until the received signal meets or exceeds the threshold. 4 The controller 235 may enter the subsequent layers 172 至 to 1740 when the control _ 235 4 waits for the same signal mode in the standby signal mode. The 201214999 signal meets or exceeds the threshold. If there is no signal received at the receiver 135, the controller 235 begins a no signal mode whereby the HIPCF canceller 13 is not active. The link control layer 1 7 1 0 can also infer information about the time dependent transmission of the threshold, for example, when two threshold levels are passed in less than one second. The carrier can be transported into or under the bridge and all settings can be kept constant until the acceptable level of delivery indicates that the mobile device has returned from the tunnel or bridge. The operation of the link control layer 171 can then be resumed using the settings of the hold. The slave signal processing layer 172G includes a right-handed program that ensures the robustness and robustness of the feedback signal. The first procedure includes averaging a predetermined number of feedback values of the feedback signal prior to performing the noise cancellation algorithm. The second procedure includes correcting errors in the feedback during the execution of the noise cancellation algorithm. An exemplary error correction procedure includes the self-sequencer 135 paying two feedback values and calculating the difference between the two feedback values. If the difference is less than

Feed:: average!! A feedback value. Otherwise, the difference between the third feedback value and the third feedback value is obtained from the receiver (1). 2 This tolerance is the tolerance, the material 帛 H == the value of the value, and the similar number is repeated for a predetermined number of =.爷, and can be: 两个 于 于 令 令 许 许 许 许 ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' The elimination algorithm 235 can be set to ten;; a certain number of feedback values are selected. For example, the control sorts the ten feedback values and selects the five feedbacks in the middle. The average value of the 2012 999 疋 疋 疋 得以 得以 得以 得以 得以 得以 得以 得以 得以 且 且 且 且 且 且 且 且 且 且 且 且 且 且 且 且 且 且 且 且 且 且The third program of 1720 includes SNR averaging. This SNr averaging procedure involves calculating the average of the #reads used for different satellites (sv) (eg, GPS, DASS (Digital Audio Broadcasting Service) or Silver Satellite). SNR averaging may be performed only for satellites having a certain altitude level above the -height threshold to avoid erroneous decisions in the algorithm execution layer 174A. In algorithm control layer 1730, several user controls can be implemented to control the algorithms described in the algorithmic execution layer. - This user is controlled to compare the polarity of the two feedback values (e.g., before and after the change of the 1 and/or Q settings of the canceller 1). The polarity can be positive. (eg, a higher feedback value is better) or negative (eg, a lower feedback value is better). Positive polarity - some exemplary (4) material, carrier noise ratio (7) N) and repeater amplifier gain can be used. Some of the exemplary feedback signals that can be used for negative polarity are PER, dirty, error vector magnitude, noise floor level, adjacent channel power ratio, and adjacent channel leakage ratio. The shoulder algorithm enforcement layer 1740 includes the execution of one of several noise cancellation algorithms. These algorithms include adjusting the Q value of the HIpcF canceller 13 and "resolving the feedback signal generated from the 5 weeks to find the acceptable work and q values for the operation to eliminate n 13". The method includes a solid type binary algorithm (Fast Binary Algorithm (FBA) and Two-Bar Correction Shoulder Algorithm (BCA)), a minimum step size algorithm (8), a blind U algorithm (BSA), and a double-slope algorithm. (Dsa) and the search and play 48 201214999 - Algorithm (TSA). Figure 2G is a flow diagram depicting a fast one-ary algorithm for eliminating noise and/or interference according to some exemplary embodiments. In this exemplary job 2_, each of the values for the mpcF canceller 13A is reversed in order and is better feedback for the polarity decision as defined in the algorithm control layer j 73〇. The value is measured. FBA 2〇〇〇T starts at a start bit and sequentially advances each of the Ϊ and Q values until it reaches a predefined stop bit. In some examples In the embodiment 10, the start bit and the stop bit are selectable by the user. In 5, controller 235 selects a first value and a first value for operating HIPCF canceller 13. The first value may be the starting value, seed value, or range intermediate value from lookup table 900. In block 2〇1, the HIPCF canceller 130 applies the first value and the first Q value to the I/Q modulator 230. In the block 2015, the receiver 135 provides a feedback signal with a feedback value. To controller 235. The feedback value can be SNR, RSSI, carrier-to-noise ratio (C/N), RSSI, repeater amplifier gain, pER, BER, error vector magnitude, noise floor level, adjacent channel power ratio Or adjacent channel leakage ratio. After obtaining the feedback value from the receiver 135, the controller 235 stores the feedback value in the memory. In the block 2020, the controller 235 inverts the bit of the I value and will update it. The I value is transmitted to the HIPCF canceller 130. In response, the HIPCF canceller 130 applies the updated I value to the I/Q modulator 230. For example, the bit 2 of the I value 2071 can be self-valued "1" Reverse to the value "〇". In the first iteration of block 49 201214999 block 2020, controller 235 may invert the start bit of the j value. In each subsequent iteration, the next bit can be inverted until the stop bit is completed. In block 2025, controller 235 obtains an updated feedback value from receiver 135. In block 2030, controller 235 compares the updated feedback value to the stored feedback value to determine which of the two feedback values is better based on the polarity defined in algorithm control layer i 73〇. For example, if the polarity is positive and the updated feedback value is greater than the stored feedback value, the controller 23 5 will determine that the updated feedback value is better. Similarly, if the polarity is negative and the updated feedback value is greater than the stored feedback value, the controller 235 will determine that the stored feedback value is better. Controller 23 5 stores the better feedback value and sets the value to the I value that results in a better feedback value. Controller 235 also applies a threshold value that results in a better feedback value to HIPCF canceller 130. In block 2035, controller 235 inverts the threshold bit and transmits the updated Q value to HIPCF canceller 13A. In response, HIpcF eliminator 130 applies the updated Q value to I/Q modulator 23A. For example, the bit 2085 with a Q value of 2〇81 can be inverted from the value “丨” to the value “〇”. In the first iteration of block 2035, controller 235 can reverse the start bit of the threshold. In each subsequent iteration, the next bit can be inverted until the stop bit is completed. In block 2040, controller 235 obtains an updated feedback value from receiver 135. In block 2045, controller 235 compares the updated feedback value to the stored feedback value to determine which of the two feedback values is better based on the polarity defined in algorithm control layer 丨73〇. The controller 23 5 stores a better back 50 201214999 • The value is fed and the Q value is set to a value that results in a better feedback value. The controller 235 also applies a Q value that results in a better feedback value to the HIPCF canceller i 3 〇 In block 2050, controller 235 makes an inquiry as to whether there are more test bits in the ί and q values. For example, controller 253 may determine whether the previous multiplex of blocks 2020 to 2050 is evaluated. Stop bit. If there are more test bits, follow the "yes" branch and return to block 2〇2〇. At block 2020, another bit is inverted and evaluated for better feedback. Otherwise, follow " No" branches to block 2055. In block 2, 55, controller 235 operates the HIpcF canceller 13 using the final stored j and q values. In some exemplary embodiments, illustrated in Figure 2 FBA 2〇〇〇 may not cover each condition, and therefore, by assigning one or both of the Z value and the Q value to the start value (eg, the most significant bit (msb)) Improvements. Another improvement to FBA 2000 involves performing the BSA described below before performing FBA 2〇〇〇 Obtain a starting value for the work value and for the devaluation value. BCA is the change to the fast binary algorithm described in Figure 2〇 and described above. In the BCA, each of the I and q values One element is sorted backwards (as in the fast binary algorithm), and if the original value of the bit is "1", then the value is added to 1" (and therefore, the more significant bit is rounded to its immediate vicinity) , or if the original value is "〇Β, ΙI, ·»·", / "', the value is reduced by "1" (and therefore, the more effective position is caused by the pelvic kidney) 1 70 ). In both cases, the controller 235 will evaluate the feedback value to determine which - 佶r, a wide a value (depending on the block, I, I value or Q value) results in better feedback similar to FBA, resulting in better feedback values The I value and Q are worth storing and used to control the HIPCF canceller 130 when the shoulder algorithm is implemented. BCA can be opened 51 201214999 Starts with a starting bit, 3仏, phase, and traverses the parent bit until the stop bit is completed. In an exemplary embodiment, the start bit and stop bit may be selected by the . In some exemplary embodiments, if #BSA is not implemented before execution of the binary correction algorithm, the binary correction algorithm may begin at the beginning of the traversal. The bits that are only for the faces of the 1 and Q values are reversed, This is because there is no more significant bit borrowed into or from the MSB. After the MSB of the I value and the Q value has been completed, the feature of increasing or decreasing the value of the evaluated bit "1" may start at the second MSB. An engine for implementing bCA instead of FBA 2000 can be discussed with reference to Figure 21, which depicts a graph 2100 of j and Q values adjusted using a binary algorithm. Referring to Fig. 21, a point X1 indicates a curve of an initial I value and a Q value for the hipCF canceller ι3 。. As part of the binary algorithm, the MSB of the inverted I value travels from point X1 to point χ2. In this graph, it is determined that the feedback value at point Χ 2 is better than the feedback value at point χ丨. Therefore, the binary algorithm will hold the I value for point 2 and reverse the MS of the q value to proceed to point X3. At point X3, it is assumed that the decision feedback value is better at point χ3 than at point ’2, then the second MSB of the I value will be inverted in FB A 2000. This bit reversal will cause the algorithm to travel from point X3 to point A, which is farther away from the optimal point C, and will therefore have a feedback value that is worse than the feedback value of point C. In the binary correction algorithm, the feedback value will be tested at both points A and B by increasing or decreasing the value of the second MSB of the I value by "1" and thus affecting the MSB. Since point B is closer to the optimal point C, point B will result in a better feedback value than point A, and BCA will continue from point B instead of point X3. Therefore, BCA can be more accurate than fast binary 52 201214999. However, in this case

In the implementation of the Shiyi II, the BCA is more repetitive and more hardware. It is necessary to provide fine tuning according to some exemplary embodiments, according to some exemplary embodiments, and the minimum step size algorithm 22 (10) for eliminating noise and interference is provided. The spectrum, for example, after the binary algorithm has been executed. The MSA 2200 can follow the interference channel between the jammer/noise source (eg, power amplifier 110) and the victim receiving crying mm ". Receiver 135. For a given step size (eg, 1 LSB to 7LSB resolution), noise and/or interference cancellation can be achieved by sequentially incrementing the value and Q value (plus step) or decrement (reducing step size). In some exemplary embodiments, Or the decrement ceases at the maximum or minimum value for an appropriate J value or ^ (eg, range criterion). In some exemplary embodiments, the msa22 operation operates over a given number of repetitions or a period of time and may be used The I value and the 卩, value each oscillate around an ideal value or follow the change of the coupled channel. Referring to Figures 1, 2 and 22, in block 22〇5, the controller 235 selects the first I value and The first Q value is used to operate the HlpCF canceller 13. In block 2210, the HIPCF canceller 130 applies the first I value and the _q value to the I/Q modulator 230. In block 2215 The receiver 135 provides a feedback signal having a feedback value to the controller 235. The feedback value may be SNR, RSSI, carrier miscellaneous Signal ratio (C/N), repeater amplifier gain, PER, ber, error vector magnitude, noise floor level, adjacent channel power ratio or adjacent channel leakage ratio, etc. The feedback value is obtained from the receiver 135. Thereafter, the controller 235 stores the feedback value in the memory. 53 201214999 In block 2220, the controller 235 increments the j value by a given step size (eg, 1 LSB) and transmits the updated I value to the HIpCF. The canceller 13 is in response. The HIPCF canceller 130 applies the updated value of j to the "Q modulator 230. The controller 235 also obtains the updated feedback value from the receiver 135. In block 2225, the controller 235 The updated feedback value is compared to the stored feedback value to determine which of the two feedback values is better based on the polarity defined in the algorithm control layer 173. The controller 235 stores the better feedback value and the I value. The controller 235 also applies an I value that results in a better feedback value to the HIPCF canceller 13A. In block 2230, the controller 235 increments the Q value by a given step. Long (eg, an LSB) and the updated Q value is transmitted to the mpCF canceller 13A. In response, HIPCF canceller 130 applies the updated Q value to i/q modulation benefit 230. Controller 235 also obtains the updated feedback value from receiver 135. In block 2235, controller 235 will update the feedback value. Comparing with the stored feedback value to determine which of the two feedback values is better based on the polarity defined in the algorithm control layer 173A. The controller 235 stores the better feedback value and sets the Q value to result in better feedback. The value of the value. The controller 235 also applies a Q value that results in a better feedback value to the HIPCF canceller 13A. In block 2240, controller 235 will! The value is decremented by a given step size (e.g., an LSB) and the updated threshold is transmitted to the HIpCF canceller 13A. In response, HIPCF canceller 130 applies the updated value of 1 to "q modulator 230. Controller 235 also obtains an updated feedback value from receiver 135. In block 2245, controller 235 compares the updated feedback value to the stored feedback value to determine which of the two feedback values is better based on the polarity determination 54 201214999 defined in algorithm control layer 173A. Controller 235 stores the better feedback value and sets the value of I to the value of the value that results in a better feedback value. The controller plus also applies an I value that results in a better feedback value to the HIpCF canceller i 3 〇. In block 2250, controller 235 decrements the Q value by a given step size (e.g., an LSB) and transmits the updated Q value to mpCF canceller 13A. In response, the HIPCF canceller 13 will update (5 values are applied to the i/q modulator 230. The controller 235 also obtains the updated feedback value from the receiver 135. In block 2255, the controller 235 will update The feedback value is compared with the stored feedback value to determine which of the two feedback values is better based on the polarity defined in the algorithm control layer 丨 < 730. The controller 235 stores the better feedback value and sets the Q value to The Q value that results in a better feedback value. The controller 235 also applies a Q value that results in a better feedback value to the HIPCF canceller 丨 300. In block 2260, the controller 235 determines whether to continue repeating blocks 2220 through 2255. In some exemplary embodiments, the decision is based on a time period. If the time period has expired, the controller 235 determines not to continue. In some exemplary embodiments, the determination is based on the sensitivity of the receiver 135. Sex or based on the feedback value obtained at block 2255. In some exemplary embodiments, the 'determination is based on the number of repetitions performed. If the controller 253 decides to continue repeating the blocks 2220 to 2255' then follow The branch returns to block 2 of 20. No Then, a "no" branch is followed to block 226s. In block 2265, controller 235 operates the hipcF canceller 130° using the final selected I value and Q value. In some exemplary embodiments, 'self I value or Q The decision to change the value to the decrement is based on whether the previous feedback has rejected the new feedback value, ie, the new feedback value of 2012 201299 is not better than the previous feedback value. Although FBA 2000, BCA has been discussed above in the sequence of IQIQIQ change. 2100 and MSA 2200, but other sequences (eg, including IIQQIIQQ and IIIQQQIIIQqq) can also be used to implement FBA 2〇〇〇, bca 21〇〇, and MSA 2200.

When the signal condition is poor (for example, the acceptable initial Z and Q values are not available) or the victim receiver base frequency ic has a limited accuracy of BER or SNR as the feedback value, #BSA can be implemented. In this embodiment, the BSA can be executed to determine the start of the algorithm (FBA2〇〇〇' or MSA 2200) used in the discussion above! Value and starting threshold. Figure 23 depicts an I Q plane 2 class with μ sub-regions (with pseudo-random feedback values). The brain can be evaluated for multiple differences! And q pre-greening (e.g., from the lookup table) and selecting the pre-sample with the best feedback value. After determining the best pre-sample, the BSA can be transitioned to ship 2_, heart 2 (10) or touch 2200 and used for the pre-sample start point. Values and Q values are used as algorithms to implement several methods of BSA. In a method ten, the IA q value associated with the optimal feedback value is selected from a sample with a preset $ (for example, 4 or 16). In the case of a bit UQ value, four samples of the feedback value can be selected from the following positions in the UQ plane: I = (OxFF, 0x2FF, OxFF, 0x2FF) Q (0x2FF, 0x2FF, OxFF, OxFF) 56 201214999 . In the case of 10-bit I and Q values, sixteen samples of the feedback value can be selected from the following positions in the I and Q planes: I=(0x80, 0x80, 0x80, 0x80, 0x180, 0x180, 0x180, 0x180, 0x280 , 0x280, 0x280, 0x280, 0x380, 0x380, 0x380, 0x380) Q=0x80, 0x180, 0x280, 0x380, 0x80, 0x180, 0x280, 0x380, 0x80, 0x180, 0x280, 0x380, 0x80, 0x180, 0x280, 0x380) Many other locations are feasible where the location is exemplary rather than the scope and spirit. A second method for performing BSA includes obtaining a feedback value at each of four (or other number of) pre-set I and Q points. The maximum and minimum feedback values of the obtained feedback values can be identified. By a) averaging the minimum and maximum feedback values, or b) adding the user-selected deviation value to the minimum feedback value, determining the feedback threshold. After determining the feedback threshold, bsa can evaluate the closest The feedback values of the 丨 and Q points of the best fields outside the four I and Q points. For example, the BSA can use a user-specified step size to detect the nearest four! And the best of the Q points... point. When the same feedback matches or exceeds the feedback threshold, the BSA can be terminated and then transitioned to MSA 2200. The DSA will use an isosceles triangle approximation with two phases with opposite slopes to estimate the noise funnel curve. Figure 24 is a graph depicting a received signal 57 201214999 oral mismatch 2405 plotted against a 4 or 4 V value generated from the implementation of the DSA according to some (four) illustrative embodiments. The DSA may select four points (XI X4 )' along the noise funnel curve formed by the received signal centroid indicator 2405 and calculate the apex near the cancellation point c. The vertices can be calculated using the point-oblique gradient of the linear equation. Once the vertex is found, Μ A can be used to convert the vertices to the initial I and Q values to MSA 22〇〇. FIG. 25 is a flow diagram depicting a DSA 2500 for eliminating noise and/or interference, in accordance with certain exemplary embodiments. The figure "represents the received signal quality of curve 2' (d) relative to the axis (as opposed to the UQ axis in the case of Figure 3 in three dimensions) resulting from the implementation of the DSA 25A from 25 in accordance with certain exemplary embodiments. The graph of the indicator is shown in Fig. 26. The exemplary DSA 25GG uses an isosceles triangle approximation with two equal silk anti-slopes for the noise curve formed by the quality indicator (4). See Figure 25 and Figure 26. 'In block 25〇5, controller 235 selects a number of samples along the I or Q axis for the I and/or q values. For example, controller 235 may select four samples. In some exemplary implementations In the example, the controller 2 uses the BSA to select the location of the sample for the value and/or Q value. In block 2510, the controller 235 communicates the sample to the i/q modulator 230, and the I/ The Q modulator applies each of the samples one at a time." In block 2520, the controller 235 obtains a feedback value (such as a "received signal quality indicator") for each of the application samples, and Each feedback value and corresponding sample I and Q values are stored in the memory device 760. In some exemplary embodiments, controller 235 receives a "received signal quality indicator" for each sample from receiver 135. In block 2520, controller 235 compares the stored feedback values and identifies 58 201214999 • Do not better reward values. For example, in Figure 26, controller 235 identifies the point χι as resulting in a better feedback value. Let point X1 have 〗 and Q value (Hi) and feedback value 1. ' 1 In block 2525, by pre-setting the step size "STEP" (for example, the STEP -I value or the most significant bit (MSB) of the Q value or MSB/2 or MSB/4), the controller 235 Change the value of j to choose the other two points around the point χι. For example, controller 235 can select point ( 2 (eg, Ii + STEp, Qi) and X3 (eg, l _ STEp, Qi). The controller 235 communicates the samples χ2 and Χ3 to the Ι/Q modulator 23〇, and the I/Q modulator 23 应 is applied one at a time for the settings of the samples X2 and X3. For each sample, the controller receives a feedback value, for example, received from receiver 135. Let 回2 have a feedback value of Υ+ and Χ3 has a feedback value of γ. In block 2530, controller 235 calculates another sample point based on the double slope. In some exemplary embodiments, controller 235 calculates another sample using SL〇PE = (γ+ -/STEP. This equation represents the slope of connection point χ2 and the imaginary line 2610. It extends from point X3 and has a line Another straight line 2615 of 261 〇 opposite slope is illustrated in Figure 26. Lines 261 〇 intersect 2615 at point 2620. In block 2530, the controller calculates an I value below point 262 使用 using the following equation: I2 = I , - STEP * (Y+ - Y ) / (γ+ _ Υι). In block 2535, the controller communicates the I and Q values of (I2, Qi) to the ϊ/q modulator 230, and I/Q The modulator 230 applies I and (^ value. In block 2540, the controller 235 receives the feedback value of (?2, Q!) and stores the feedback value in the memory device 59 201214999 760. The feedback value of Q,) is γ 2. In block 2545, by the preset step size "STEp", the controller 235 selects the other two around the point illusion by varying the q value from the point (I Ql ). For example, the controller 235 can select points (I仏+ste?) and (I2, Qr STEP). The controller 235 communicates the samples to the I/Q modulator 230, and the I/Q modulator 230 - one place For each sample, the controller 235 receives a feedback value of the feedback value Y+ and (I2, Q丨_STEP) the feedback value is γ. In block 255〇, the controller 235 Calculation: Q2 = Ql - STEP * (Υ+ _ Υ〇 / (γ+ · γ2). In block 255 5, controller 235 communicates the I and q values of (Η, Q2) to I/Q modulation The I/Q modulator 230 applies I and the threshold. In block 256, the controller 235 receives the feedback value of (I2, Q2) and stores the feedback value in the memory device 760. The feedback value of (12, q2) is γ3. In block 2565, controller 235 reduces the size of sTEP. In this exemplary embodiment, the size of STEP is halved. However, other (eg, It is also possible to reduce the size. In block 257, the controller 235 makes an inquiry as to whether the size of the STEP is smaller than the threshold "STEPend". If the size of the STEP is smaller than the STEPEND, the DSA 25 00 continues to the area. Block 2580, at block 258, the controller 235 uses (l2, q2) as a starting point to start the MSA (eg, MS A 2200) » if the size of the STEP is not less than st Epend' then method 250 continues to block 2575. In block 2575 the controller assigns I2, q2, and γ2 to ^ and Y1, respectively. After block 2575, DSA 2500 returns to block 2525. 60 201214999 t There is an area where there is a better elimination point for the whole domain, and exemplary can be particularly useful. Preferably, the region eliminates the preferred UQ value of the feedback value "regional". For example, the view (such as MSA·: does not jump out of the closest area is better to eliminate the area of the point of removal. In order to implement the DSA 2500 for the upper part, the controller 235 can avoid sticking to the area better. The points are eliminated, while the MSA 2200 can be fine tuned to find global better cancellation points. Figure 27 is a flow diagram depicting noise cancellation and/or interference t TSA 2700, in accordance with certain exemplary embodiments. The graph along the ¥ Μ plane 28 (the elimination point of H is evaluated in Figure 2 and FIG. 28, in block 2705, controller 235), which is evaluated in the implementation of the TSA of FIG. 27 of some exemplary embodiments. A plurality of (e.g., four) samples of the ϊ-q plane 28〇1 are selected. In some exemplary embodiments, the controller 235 uses the location in the IQ plane 2801 to select samples for use in the block. The controller 235 communicates the settings for the selected samples to the f/Q modulator 230, and the I/Q modulator 23 applies the settings for each sample one at a time. In block 2715 The controller 235 receives a feedback value (for example, from the receiver 135) for each sample, and the feedback value and The corresponding settings are stored in the memory device 76. In block 272, the controller 235 compares the feedback value and identifies a better or better feedback value. XI in Figure 28 is a sample that results in a better feedback value. In block 27U, controller 235 selects the other four samples closest to χι by a predetermined step size "STEp" (eg, STEp = MSB/2 or MSB/4). For example, controller 235 may Select (h + STEp, Qi ), ( L -STEP'Q,), (Il, Qi + STEp) and (Ii Qi_sTEp). Control 61 201214999 235 will set the four settings to j/q modulator 23 〇, and the j/q modulator 230 applies the settings for each sample one at a time. The controller 253 receives the feedback for each sample and uses the feedback value for each sample and The settings of the mother are stored in the memory device 76. The controller 2 3 5 compares the feedback values for the four samples and identifies the preferred feedback value. Let X2 in Figure 28 be the preferred feedback value. In block 2730, controller 235 reduces the size of STEP. In this exemplary embodiment, the size of STEP is halved. Other sizes are also reduced. In block 2735, controller 235 makes an inquiry as to whether the size of STEP is less than the threshold "STEPend". If the size of STEP is less than STEPend' then TSA 2700 proceeds to block 2755, at block 2755. At this point, the controller 235 controls the I/Q modulator 230 using the settings (in + 1, Qn + i ). If only one repetition of the TSA is performed, the controller 235 uses the settings for the samples corresponding to the preferred feedback values in the block 2725 to control the I/Q modulator 230. If the size of STEP is not less than STEp_, then TSA 2?00 proceeds to block 2740. In block 2740, controller 235 selects the other four samples that are closest to the sample with the best stored feedback value. If it is the first iteration, the sample is X2 with (h, Q2). This sample is represented as χη because the block 2740 can be executed multiple times. For example, the controller 235 selects samples (^ + STEP, Qn), (In_STEP, Qn), (InQn + STEp), (n STEP ). The controller 235 communicates the four settings to the pQ modulator 23A, and the I/Q modulator 230 applies the settings for each sample one by one. The controller 235 receives the feedback value for each sample and stores the feedback value for each sample 62 201214999 and the settings for each sample in the memory device 760. In block 2745, controller 235 compares the feedback values for the four samples and identifies the preferred feedback value. In block 2750, controller 235 reduces the size of STEP' and the TSA returns to block 2735. Figure μ illustrates that the TSA 2700 uses four iterations (represented by points XI through X4) which identify one of the i_q planes 2801 that preferably eliminates the point 2 8 1 5 . The exemplary TSA 27A may be particularly useful when based on a previously preferred elimination point search for improved cancellation points and corresponding ι/Q settings (e.g., in response to changes in temperature). In such cases, block 27〇5 can be adapted to use the previous preferred Ι/Q setting instead of selecting four samples. The TSA 27〇〇 can cause the range of the search to be changed to the area in the I-Q plane 280 1 near the previously preferred cancellation point. The individual algorithms discussed above (eg, BSA, FBA, BCA, MSA, DSA, and TS A ) can be implemented as separate algorithms to determine acceptable j and threshold values. Alternatively, more of these algorithms can be used together to increase the speed of the assessment and achieve the desired accuracy. For example, the BSA can be executed to determine the first MSB or the first MSB and the second msb of both the ^ and Q values. After bsa, FBA or BCA can be performed to determine the very few bits between the I and Q values. Finally, the MSA can be implemented to fine tune both the I and Q values to achieve better feedback values and therefore better noise or interference cancellation. Multiple iterations of the algorithms may be enforced and/or the algorithms may be executed for a longer period of time to achieve better results. In some exemplary embodiments, algorithms for fine tuning (eg, 5 and TSA) are used in the "on" mode, in which case the controller 15 continues when the noise canceller is in normal operation. Perform these algorithms. This allows the controller 15 2012 63 201214999 to adjust the settings of the noise canceller to account for environmental changes, such as changes in temperature or operating conditions. In addition, the noise cancellers operating in parallel may each perform one or more of the algorithms simultaneously or sequentially. 29 is a diagram depicting a method 29 for discovering a preferred noise cancellation point for one of two noise cancellers disposed in a communication system (such as 'communication system 1'), in accordance with certain exemplary embodiments. The flow chart of 〇〇. For example, in an alternate embodiment communication system 1A may include two HipCF cancellers 130 in parallel. In the risk block 2905, a control device (such as the control stomach 235 of the two devices 130) is configured in a sequence for the (I, Q) setting of the two beta devices. For example, 'this sequence can be configured as (IninQnqn...I0i0Q0q0), the towel no and Qn...Q〇 represent the (I,Q) setting of the first-party canceller, and 丨...1ϋ and qn. ..q〇 indicates the usage:: 1Q setting of the canceller. The control device can then process the two cancellers with a single-eliminator of the configured sequence. Elimination: =: 91° The medium control device uses the sequence to perform one or more of the two discussed above (eg, BSA, FBA, BCa, MSa, 2915^^ to determine the preferred cancellation settings for the canceller). The control device stores the preferred cancellation settings in the memory. The mapping is depicted in the flowchart for the discovery of the method for the generation of the fourth generation method in the block side. — (such as the two noise cancellers... control the controller 235 that sets the two noise cancellers out) to find the better elimination point for the two, and for both 64 201214999 § The setting of the second of the hole eliminators remains unchanged. The above discussed

One of the elimination algorithms (e.g., 'BSA, FBA, BCA, MSA, DSA, or TS A) finds a better noise cancellation point for the first noise canceller. In block 3010, the control device uses one or more of the cancellation algorithms (eg, BSA, FBA, BCA, MSA, DSA, or TSA) to find a better cancellation point for the second noise canceller, while The settings for the first noise canceller remain unchanged at the preferred cancellation points found during execution of block 3005. In block 3015, the control device obtains the feedback values generated by the two noise cancellers in the event that the two cancellers use their respective preferred cancel point operations. In block 3020, the control device compares the obtained feedback value to a preset threshold. If the feedback value is better than the threshold or if the method 3_ has been sent more than a predetermined number of repetitions, then the method 3 continues to block 5 . Otherwise, the method returns to block 3〇〇5 where the current (I’Q) settings for the two cancellers are used as the starting values for the algorithm. In block simplification, the control device stores the settings for the two noise cancellers and uses these settings to control the noise canceller. FIG. 31 is a flow diagram s L depicting an alternative method 3100 for discovering one of the two noise annihilators for placement in a communication system, in accordance with certain exemplary embodiments, . L program map. This method 3100 solves the problem of using two noise cancellers to increase the cancellation bandwidth. See only gas implantation (in block 3105), controlling one of the dreams (such as 'two noise cancellers' _ ) based on the feedback value for the part below the bandwidth

Now used for two noise cancellers ^ X such as (!, Q) setting), at the same time: two elimination settings (cut the second noise canceller when the door is closed. In the block 3 ιι〇 65 201214999 'control device storage A preferred noise cancellation setting for the first-noise cancellation chirp cold removal. In block 3U5, the control device is found in the noise canceller based on the feedback value for the upper portion of the bandwidth. The second one's preferred cancellation setting (eg, U'Q) setting) 'cuts off the first-noise canceller at the same time. Eliminate the settings in the zone. ^ The first (4) core better noise/block 3125 'control device turns on two noise cancellers and applies respective better cancellation settings to each of the two noise cancellers. In block 3130, the control unit performs a step-by-step process on the first-to-noise cancellation for the portion below the bandwidth. In block η", the control device performs a one-step MSA on the second noise canceller for the upper portion of the bandwidth. In block 3140, the control device obtains a feedback value for the noise canceller and applies the feedback. The value is compared to a pre-set value. If the feedback value is greater than the pre-set value or if blocks 313 and 3135 have been executed more than a predetermined number of iterations, then method 3100 proceeds to block 3145. Otherwise, the method 3ι〇〇 returns to block 3^0. In block 3145, the control device stores the final settings in the memory and controls the noise canceller using the final settings, although the decision is made for two noise cancellers. Preferably, the methods 2900, 3000, and 3100 are depicted and described, but each of the methods 29, 3, 31 can also be used to determine a preferred cancellation point for any number of noise cancellers. For example, methods 2900, 3000, 3100 can be used to find a preferred cancellation point for three or more noise cancellers in a parallel configuration. Although depicted = 66 201214999 2 9 0 0 3 0 〇〇 and 3 1 〇〇 found for comparison of two noise cancellers Or improved elimination point' but each method 2900, 3000, 3 100 can also be used to find better or improved elimination for more than two noise cancellers (eg, three or more noise cancellers) In summary, a communication system in accordance with some exemplary embodiments of the present invention may include a transmitter that transmits information at a first frequency, and a receiver that is the same as the first frequency or at a first frequency Receiving a communication signal at a nearby second frequency; and - an interference suppression device that cancels, corrects, addresses, or compensates for interference, noise, noise, clutter, or other imposed on the receiver by a signal transmitted by the transmitter. a poor spectral component. The interference suppression device can be coupled to the transmission path of the transmission thief (eg, at the output of the power amplifier of the transmitter) to obtain a sample of the transmitted signal. The interference compensation circuit can include a plurality of filters (4 &quot ; bandpass data), which blocks or suppresses the out-of-band signal of the receiver while transmitting noise or other interfering signals in the frequency band of the receiver. The interference compensation circuit can also A modulator that generates an interference compensation signal using a signal output by the filter. The interference compensation signal may have an amplitude that is the same as or close to the amplitude of the noise and a phase shift relative to the stiffness of the interference. These parameters are tuned during the feedback from the victim receiver's Received Signal Quality Indicator. The interference compensation signal generated by the (8) modulator is applied to the receiver's receive path to cancel or suppress interference imposed by the transmitted signal on the receiver. The communication system described herein can be embodied in a variety of communication devices, including cellular phones, mobile computers, PDAs, personal navigation devices 1 (eg, GPS devices), or containing two or more communication elements' Dantratong 67 201214999 Letter device. For example, the communication system can be embodied in a smart phone having an LTE/CDMA/GSM transceiver and a mobile τν tuner. Another example is a smart phone with a GSM/PCS/DCS/W-CDMA transceiver and a GPS receiver. Yet another example includes a notebook computer having a WLAN transceiver and a WiMAX or Bluetooth transceiver. In a mobile device embodiment, two or more communication elements can communicate via two or more antennas with minimal spatial separation. Therefore, signals transmitted by two or more communication elements can impose interference on each other. In order to suppress or eliminate this interference, the HIPCF canceller as described above can be used in each communication direction. φ, ie, the first-HIPCF canceller can eliminate or suppress interference imposed by a second one of two or more communication elements on the first of the two or more pass elements, and the second hIPCF The canceller eliminates or suppresses interference imposed by a first one of the two or more communication elements on the second of the two (four) two or more communication λ pieces. Some of the components of the two HIPCF cancellers can be fabricated on a single integrated circuit or on multiple integrated circuits (such as - or multiple CMOS circuits). Embodiments of the present invention are available to perform the methods and processing functions described above = hardware and software use. As will be appreciated by those skilled in the art, the systems, methods, and programs described herein can be embodied in a programmable computer computer software or digital circuit. The soft altar can be stored on a computer readable medium + example. The computer readable medium can include floppy disk, RAM, ROM, hard disk removable media 'flash memory, memory stick, optical media, magneto-optical media, CD-ROM, and the like. Digital circuits can include integrated circuits, gate arrays, block block logic, field programmable gate arrays ("FpGA"), and the like. 68 201214999 Although specific embodiments of the invention have been described in detail above, this description is for illustrative purposes only. Therefore, it is to be understood that the invention is not intended to be limited Various modifications of the illustrative aspects of the illustrative embodiments and corresponding to the exemplary embodiments, without departing from the spirit and scope of the invention as defined in the appended claims. The equivalents of the disclosed aspects can also be made by those skilled in the art having the benefit of the invention, and the scope of the claims should be construed in the broadest scope of the invention. BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a functional block diagram of a communication system in accordance with some exemplary embodiments. 2 is a block diagram of a high input power series chopper (HIPCF) canceller, in accordance with some demonstrative embodiments. 3 is a block diagram of certain components of the HIPCF canceller of FIG. 2, in accordance with some demonstrative embodiments. 4 depicts a frequency spectrum of a signal received at a victim receiver antenna, in accordance with certain exemplary embodiments. 5 depicts a frequency spectrum of a signal received at the input of a victim receiver after the cancellation of the in-band bad spectral component by the HIpCF canceller, in accordance with certain exemplary embodiments. 6 is a block diagram of a Q enhanced bandpass filter (Q enhanced BPF) in accordance with certain exemplary embodiments. A block diagram of the components. FIG. 8 depicts a functional block diagram of a communication system in accordance with some demonstrative embodiments. FIG. 9 depicts a lookup table in accordance with some demonstrative embodiments. 10 is a flow chart depicting a method for calibrating certain components of HIPCF 4 of FIG. 2 in addition to $, in accordance with certain exemplary embodiments. 11 is a flow diagram depicting a method for a filter for a HIPCF canceller of FIG. 2, in accordance with certain exemplary embodiments. 12 is a flow diagram depicting an input bandpass sputum (input BpF) & method for calibrating the HIPCF / XI (4) of FIG. 2, in accordance with certain exemplary embodiments. 13 is a flow chart depicting a method for calibrating a low noise amplifier bandpass filter (LNA_BpF) of the HIPCF canceller of FIG. 2, in accordance with some demonstrative embodiments. 14A and 14B (collectively Fig. 14) depict a flow chart of a method for calibrating a Q enhanced BPF of the HIPCF canceller of Fig. 2, in accordance with certain exemplary embodiments. FIG. 15 is a flow chart depicting a method for calibrating the input bpf of the HIPCF canceller of FIG. 2, in accordance with certain exemplary embodiments. 16 is a flow chart depicting a method for determining switch settings for a given frequency, in accordance with some demonstrative embodiments. 17 depicts noise and/or interference cancellations in accordance with certain exemplary embodiments. 201214999 „ Except for the implementation layer of the algorithm. FIG. 18 is a diagram depicting phase mismatch with respect to a power amplifier in accordance with certain exemplary embodiments. A plot of receiver sensitivity plotted. Figure 19 is a graph depicting the output signal to noise ratio (SNR) of a motion τν tuner plotted against the received mobile TV tuner signal strength, in accordance with certain exemplary embodiments. 20 is a flow diagram depicting a fast binary algorithm for eliminating noise or interference, in accordance with some demonstrative embodiments. Figure 21 depicts in-phase (I) adjustment using a binary algorithm, in accordance with certain exemplary embodiments. And a graph of orthogonal (Q) values. Figure 22 is a flow diagram depicting a minimum step size algorithm for eliminating noise and/or interference, in accordance with some demonstrative embodiments. Figure 23 depicts some exemplary An IQ plane with pseudo-random feedback values for an embodiment. Figure 24 is a graph depicting a reception quality indicator plotted against j or Q values generated from implementation of a dual slope algorithm (DSA), in accordance with certain exemplary embodiments. Figure 25 is a flow diagram depicting DSA for noise cancellation and/or interference, in accordance with some demonstrative embodiments. Figure 26 is a diagram depicting a dual slope algorithm relative to self Figure 24, in accordance with some demonstrative embodiments. The resulting I or Q value yields a graph of the quality of the inconsistency. Figure 27 depicts a tracking and search algorithm (TSA) for eliminating noise and/or interference, in accordance with certain exemplary embodiments. Flowchart. Figure 71 201214999 Figure 28 is a graph depicting the elimination points along the IQ plane evaluated in the implementation of the TSA of 圊27, in accordance with certain exemplary embodiments. Figure 29 is a depiction depiction in accordance with certain exemplary implementations. A flowchart of a method for discovering a preferred noise cancellation point for one of two noise cancellers disposed in a communication system. FIG. 30 depicts a depiction for discovery, in accordance with certain exemplary embodiments. A flowchart of a method for better noise cancellation points in one of two noise cancellers disposed in a communication system. FIG. 31 is a diagram for discovering for placement in a communication, in accordance with certain exemplary embodiments. One of the two noise cancellers in the system A flow chart of a method for eliminating noise points. A number of aspects of the present invention can be better understood with reference to the above drawings. These drawings illustrate only exemplary embodiments of the present invention and therefore should not be construed as limiting The scope of the present invention is to be understood as being limited by the description of the exemplary embodiments of the present invention. Some dimensions are used to help visually convey such principles. In the drawings, reference numerals indicate identical or corresponding (but not necessarily identical) elements. [Main element notation] 100: communication system 105: (violators) transmitter 107 : Transmission path 110 : Power amplifier 72 201214999 11 5 : Transmission antenna 120 : Receiving antenna 125 : Sampling device 127 : Electrical conductor 130 : High input power series filter (HIPCF ) canceller 133 : Receiving path 134 : Elimination point 1 3 5: Receiver 140: Optional Receive (RX) Filter 1 8 0: Feedback Routine 205: Bandpass Filter (Input BPF) 210: Low Noise Amplifier (LNA) 2 1 5: Low Noise Large Bandpass Filter (LNA-BPF), 220: Variable Gain Amplifier (VGA) 225: Q Enhanced Bandpass Filter (Q-Enhanced-BPF) 23 0 : I/Q Modulator 235: Controller 240: Auxiliary circuit 250: wafer boundary 290, 291, 292: Q enhancement circuit 3 00: block diagram of some components of the HIPCF canceller 3 05: first switched capacitor array (SCA)

310: Second SCA 315: SCA 73 201214999 400: Spectrogram 402: Signal Frequency 403: Amplitude 404: First Peak 405: Second Peak 406: Noise Side Band 5 0 0: Spectrogram 506: Noise Sideband 507 : Notch 610 : LC Slot

615: Switched Capacitor Array/SCA 620: Cross-Coupling Pair 650: Digital/Analog (D/A) Converter 655: Center Taps 670, 720, 725: Bypass Switch 745: Power Detector 750: Analog/Digital (A/D) converter 755: temperature sensor 760: memory (storage) device 770: buffer 800: communication system 805: communication device 8 10: transmitter 8 1 5: power amplifier 74 201214999 820: receiver 825: first antenna 833, 834: elimination point 850: communication device 855: transmitter 860: power amplifier 865: receiver 870: second antenna 880: first HIPCF canceller 885: second HIPCF canceller 890, 895: Sampling device 900: lookup table 910, 92 0, 93 0: center frequency setting 940: seed value 950: miscellaneous setting 1700: implementation layer 1 7 10: link control layer 1720: signal processing layer 1730: algorithm control layer 1740: Algorithm execution layer 1800: Fig. 1 805: first curve 1 8 1 0 · · second curve 1900: Fig. 75 201214999 1905: first curve 1910: second curve 1915: third curve 2000: first two Algorithm (FB A ) 2075 ' 2085 : Bit 2100 : Binary school Algorithm (BCA) 2200 : Minimum Step Size Algorithm (MS A ) 2300 : IQ Plane 2400 : Graph 2405 : Received Signal Quality Indicator 2500 : Double Slope Algorithm (DSA) 2600 : Graph 2605 : (Curve) Receive Signal Quality Indicator 2610: Straight Line 2615: Straight Line 2620: Point 2700: Tracking and Search Algorithm (TSA) 2701a, 2701b: I Value 2800: Graph 2801: IQ Plane 2801a, 2801b: Q Value 2805: Elimination Point C1-C4 : Switchable capacitor FR : Channel frequency 76 201214999 _ FT : Carrier frequency

Ref_C : Reference current VC1-VC2 : Voltage controlled capacitor 77

Claims (1)

201214999 VII. Patent application scope: 1 _ A method for determining a noise cancellation control setting for generating an interference compensation signal, the method comprising: (a) generating the interference compensation signal based on at least the control setting; (b) applying the interference compensation signal to an input signal path of a receiver; (c) receiving a feedback signal including a feedback value indicating a level of interference imposed on the input signal path; (d) storing the feedback value and the control setting; (e) adjusting the control setting by the controller; (f) repeating a plurality of repetitions from (a) to (e), thereby storing a plurality of feedbacks And a plurality of control settings; and (g) determining which of the plurality of feedback values is preferred. 2. The method of claim 2, wherein the generating the interference compensation signal comprises a phase, an amplitude, and a delay based on the control setting to impose a signal on one of the input signal paths. At least - adjust. Such as the scope of patent application! The method of claim, wherein the feedback signal is connected to the receiver, and wherein the feedback value includes a received signal strength mismatch, a bit error rate, a packet error rate, a noise floor, and a letter UlMfLU Error vector magnitude ' _ position accuracy... one of the adjacent channel leakage ratio, and an amplifier gain. 4. The method of claim 1 wherein the control setting includes an in-phase setting and an orthogonal setting. 78 201214999 • 5. The method of claim 4, wherein the in-phase setting comprises a first set of binary bits corresponding to one of the in-phase values, and the orthogonal setting comprises one of the corresponding orthogonal values a second set of binary bits, and wherein adjusting the control setting comprises inverting a binary bit of at least one of the in-phase setting and the orthogonal setting. 6. The method of claim 4, wherein the adjusting the control setting comprises alternately alternating between adjusting the in-phase setting and adjusting the orthogonal setting. 7. The method of claim 1, wherein the control setting is adjusted based on a previously received feedback value. 8. A method for determining a noise cancellation control setting for use in a noise cancellation device, the noise cancellation device being operative to generate an interference compensation signal based on the control setting. An in-phase parameter and a quadrature parameter, the method comprising: receiving a first compensation value indicating a first interference level in response to applying the interference compensation signal to an input L-number path of an electrical device; The improved control settings are searched for by adjusting the in-phase parameters and the orthogonal parameters of the control settings one by one in an alternating manner until the coincidence-stop condition; and each time the in-phase parameter is adjusted and each time Adjustment of orthogonal parameters: generating an updated interference compensation signal based on the adjusted control settings, and responsive to applying the updated interference compensation signal to the input signal path to receive an updated feedback value; 79 201214999 determining the update Whether the feedback value is better than a previous feedback value; and the feedback value in response to the update is better than one of the previous feedback values In a subsequent adjustment of the control setting adjustment; and in response to the stop line with the conditions, using the adjusted control setting of the noise eliminating device is operated. 9. The method of claim 8, wherein the generation of the interference compensation 彳 § includes a control setting based on the adjustment to impose an interference on the same phase of the signal imposed on the electrical device, At least one of amplitude, and delay is adjusted. 10. The method of claim 8, wherein the electrical device comprises a receiver, and wherein the feedback value is received from the receiver, and wherein the feedback value comprises a received signal strength indicator, one bit The elementary error rate, a packet error rate, a noise floor, a signal to noise ratio, a wrong • Wu Xiangli amplitude, a position accuracy, an adjacent channel leakage ratio, and an amplifier gain. 11. The method of claim 8, wherein the stopping condition comprises a plurality of repetitions. 12. The method of claim 8, further comprising: comparing each updated feedback value with a threshold feedback value; and responding to the updated feedback value in accordance with the threshold feedback value or the updated feedback The value exceeds the value of the threshold feedback value to determine that the stop condition has been reached. 13. If the method of applying for patent scope 帛8 is applied, the in-phase parameter and the orthogonal parameter are sequentially adjusted in an alternating manner by the following: 80 201214999 One or more bits of the in-phase parameter are inversed. Turning; and inverting one or more bits of the orthogonal parameter. M. The method of claim _ 13$, wherein after each adjustment, 'from one or more bits of the in-phase parameter to the least significant one or more bits of the in-phase parameter, and from the positive One or more bits of the intersection parameter are advanced to the least significant one or more bits of the orthogonal parameter. 15. The method of claim 8, wherein the in-phase parameter and the quadrature parameter are sequentially adjusted in an alternating manner by: incrementing the in-phase parameter; determining whether an in-phase parameter of the increment causes An improved feedback value; the in-phase parameter responsive to the increment does not cause one of the improved feedback values to be determined, the in-phase parameter is decremented; whether the in-phase parameter of the bonding decrement causes an improved feedback value; The parameter increment is determined; determining whether the orthogonal parameter of the increment causes an improved feedback value; and the orthogonal parameter is decremented in response to the increment of the intersection parameter that does not make an "one of the improved return value determinations"; And determine whether the orthogonal parameter of the decrement causes an improved feedback value. 16. The method of claim 8, wherein the in-phase parameter and the quadrature parameter are sequentially adjusted in an alternating manner by: decrementing the in-phase parameter; determining whether the in-phase parameter of the decrement causes an improvement The feedback value; the in-phase parameter responsive to the decrement does not cause one of the improved back-spread values, and the in-phase parameter increment 丨81 201214999 determines whether the in-phase parameter of the increment causes an improved feedback value; Parameter decrement; determining whether the orthogonal parameter of the decrement causes an improved feedback value; the orthogonal parameter responsive to the decrement does not cause one of the improved feedback values to be determined, the orthogonal parameter is incremented; and the incremental Whether the orthogonal parameter results in an improved feedback value. 1 7. A method for determining a control setting for use in a noise canceling device s. The noise canceling device is operative to generate an interference compensation signal based on the control setting. The interference compensation signal is responsive to being The interference compensation signal applied to an input signal path of one of the receivers operatively suppresses interference imposed on the receiver, the control setting including an in-phase parameter and an orthogonal parameter, the method comprising: Applying the interference compensation signal to the input signal path, receiving a first feedback value of one of the first interference levels; adjusting the in-phase parameter and the orthogonal parameter one by one in an alternating manner Searching for an improved control setting until a stop condition is met. 'Each adjustment includes: incrementing one of the in-phase parameter and the orthogonal parameter to generate a first adjusted control setting based on the first adjustment Controlling a setting to generate a first updated interference compensation signal and responsive to applying the first updated interference compensation signal to the input signal path Receiving a first updated feedback value; incrementing one of the in-phase parameter and the orthogonal parameter to generate a second adjusted control setting, based on the second adjusted control setting, 82 201214999 ' And a second updated interference compensation signal, and responsive to applying the second updated interference compensation signal to the input signal path to receive a second updated feedback value; determining the first updated feedback value, the second updated feedback Which of the value and a previous feedback value is preferred; and the use of a subsequent adjustment will result in a better feedback control setting. 18 _ The method of claim 17, wherein generating each of the interference supplements includes a phase based on the control setting to impose a phase, amplitude, or amplitude on a signal that causes interference on the receiver. Adjust with at least one of the delays. 19. The method of claim 17, wherein the feedback value is received from the receiver, and wherein the feedback value comprises a received signal strength indicator, a payout, a one-bit error rate, and a packet error rate. The noise floor, a signal-to-noise ratio, an error vector magnitude, a positional accuracy, an adjacent channel leakage ratio, and an amplifier gain. 20. The method of claim 7, wherein the one of the in-phase parameter and the orthogonal parameter comprises incrementing an in-phase parameter or a quadrature parameter by a predetermined step size. 21. The method of claim 17, wherein the deducting one of the in-phase parameter and the edge-orthogonal parameter comprises decrementing the _ in-phase parameter or the ---------- 22. The method of claim 17, wherein the cessation condition comprises a plurality of replies. 83 201214999 23. The method of claim 17, further comprising: comparing each feedback value with a threshold feedback value; and responsive to the feedback value meeting the threshold feedback value or the feedback value exceeding One of the threshold feedback values to determine that the stop condition has been reached. 24. A method for determining a noise cancellation control setting for generating an interference compensation signal for a cellular telephone application, the method comprising: (a) selecting a plurality of initial control settings; (b) from the plurality Identifying one of the initial control settings for interference suppression in the initial control settings; (c) selecting a plurality of subsequent control settings based on the performance of the identified control settings; (d) identifying the interference suppression from the subsequent control settings a subsequent control setting; and (e) repeating (c) through (d) until a stop condition is met. 25. The method of claim 24, wherein identifying from the initial control settings one of the initial control settings for interference suppression comprises: generating an initial interference compensation signal for each initial control setting; The initial interference compensation signal is applied to one of the input signal paths of the receiver; ° each of the initial interference compensation signals received from the signal path applied to the s-round is received. ################# An initial feedback value; and determining the initial feedback signal results in better interference compensation. The method of claim 24, wherein identifying one of the subsequent control settings that is better for interference suppression from the subsequent control 84 201214999 system includes: generating a subsequent interference compensation signal for each subsequent control setting; a subsequent interference compensation signal is applied to an input signal path of a receiver; for each subsequent interference compensation signal applied to the input signal path, receiving a subsequent feedback value indicating an interference suppression level from the subsequent interference compensation signal And determine which subsequent feedback signal causes better interference compensation. 27. The method of claim 24, wherein each control setting comprises an in-phase parameter and an orthogonal parameter, and wherein selecting a subsequent control setting comprises: generating a first subsequent control setting and an increment The in-phase value, the first subsequent control setting includes an orthogonal parameter of the identified control setting, and the same phase value of the increment includes the in-phase value increment of the identified control setting is one step long; generating a first performance Controlling a set and an in-phase value of the decrement, the second subsequent control setting includes an orthogonal parameter of the identified control setting, and the in-phase value of the decrement includes the step of reducing the in-phase value of the identified control setting; generating one a third subsequent control setting and an incremental orthogonal value, the third subsequent control setting includes an in-phase parameter of the identified control setting, and the incremental orthogonal value includes an orthogonal value of the identified control setting Incrementing the step size; generating a fourth subsequent control setting and an offsetting orthogonal value, the fourth 85 201214999 subsequent control setting including the identified control setting The phase parameter, and the orthogonal value of the decrement includes the step of subtracting the orthogonal value of the identified control setting; and decreasing the step size after each control setting block. 28. A method for determining a noise cancellation control setting for generating an interference compensation signal, the method comprising: (a) identifying, from a plurality of control settings, one of control settings for interference suppression, each The control setting includes an in-phase parameter and an orthogonal parameter; (b) storing the identified control setting; (c) generating a first updated control setting and a second updated control setting, the first updated control setting The in-phase value including the stored control set orthogonal parameter and the -incremental in-phase value 纟 includes an in-phase value increment of the stored control setting, and the second updated control setting includes the Orthogonal parameter of the stored control setting and an in-phase value of a decrement 'where the in-phase value of the decrement includes the step of decrementing the in-phase value of the stored control setting; (d) receiving one of the control settings corresponding to the first update a first feedback value and a second feedback value corresponding to the control setting of the second update; (e) a control setting based on the first update, a control setting of the second update, the first feedback value, and The second feedback value uses a double slope equation to generate an updated in-phase value; (f) generates a third updated control setting and - a fourth update control setting. The third updated control setting includes the updated in-phase The orthogonal value of the parameter and the increment, wherein the orthogonal value of the increment includes the stored control 86 201214999 system set the orthogonal value increment one step long, the 箆m trial is as long as the a younger four update The control setting includes an updated in-phase parameter of the stored control setting and an orthogonal value of the decrement, wherein the orthogonal value of the decrement includes a positive = value decrement of the stored control setting; & (g Receiving a third feedback value corresponding to the control setting of the third update and a fourth feedback value corresponding to the control setting of the fourth update; (h) control setting based on the third update, control of the fourth update Setting, the third feedback value, and the fourth feedback value use a double slope equation to generate an updated orthogonal value; (i) reducing the value of the step; and (j) repeating (b) to (i) until Meet a stop condition . 29. A method for determining-controlling each of a plurality of noise canceling devices, the plurality of noise canceling devices being configured to generate an interference compensation based on the control settings a signal that is operatively inhibited from being applied to the receiver in response to the interference compensation signal being applied to the input signal path of the receiver, each control setting being included An in-phase value and an orthogonal value, the method comprising: generating a control setting for the combination of one of the in-phase value and the combined orthogonal value of the noise cancellation device construction, the in-phase value of the combination being: The same as (4) for each control setting of the per-sequence configuration, the orthogonal value of the combination includes an orthogonal value of each control setting configured in a sequence, and a control for modifying the combination according to at least one computer program Set the control settings for the search-better combination until a stop condition is met; 87 201214999 Each modification of the control settings for this combination: will correspond to each noise cancellation The modified combination of the device 6 and the quadrature value and the quadrature value are applied to the noise canceling device; the noise canceling device is used to generate an interference compensation signal; and the interference compensation signal is applied in response to the interference compensation signal Up to the input signal path 'receiving one of the interference levels back to the town value; and storing the feedback value; and the alarm-stop conditional match, from the stored feedback value + identifying the feedback value pair imposed on the receiver The interference will result in better suppression. 30. A method for setting the = decision-control in a plurality of noise canceling devices, the plurality of noise canceling = configuration to generate - interference based on the corresponding control settings The compensation signal; the disturbance compensation signal is responsive to the interference compensation being applied to a receiver: the diameter of the interference compensation is said to be, and the operation is imposed on the silly supplement, and the method includes: (2):: The corresponding control setting is applied to the plurality of noise cancellations; the mother noise cancellation device is placed in the first month, (b) by performing on the plurality of noise cancellation devices - computer program To identify the improvement of the one-to-one improvement μ~ the solid-state elimination device j' while maintaining the control settings for other miscellaneous constants; for the device name (c) to operate for the one noise cancellation device The noise cancellation device; and. The control ti (4) repeats (b) to (c) for each of the plurality of noise canceling devices 88 201214999. 3 1. As claimed in claim 30, the method for identifying an improved system includes determining, for a noise cancellation device, a plurality of control settings from a plurality of control settings. Better suppression of interference imposed on the receiver. 2·^ The method of claim 3, wherein identifying an improved "X疋 system comprises from a plurality of controls for the one noise cancellation device::: determining which control setting will be for the receiver Causes a better signal quality indicator. (d) a method for tuning each noise cancellation " setting in a plurality of noise canceling devices, the plurality of noise canceling devices::: will be across a given bandwidth The noise is suppressed::: 籍: by the first eight of the bandwidth > to 嘈 钻 钻 咕 咕 op op op & & & & & & & & & & & & & 电脑 电脑 电脑 电脑 电脑 电脑 电脑 电脑 电脑 电脑 电脑 ° a first control setting. By performing a second portion of the bandwidth to r to identify a second control setting for the first 汛 汛 elimination device for the 筮 M M 1 LJ 1: 腩 %; The first control setting and the second control setting; and, operating the first control setting according to the first control setting to operate the second noise cancellation device, and 34. In the method of item 33, the method includes a higher (four) H" of the bandwidth, and a lower frequency portion of the first portion of the system width - 35. The method of claiming the patent scope further includes : 89 201214999 (a) in at least one repetition 'by performing a calculation on the first control setting To improve the first control setting; (b) in at least one of the repetitions 'by performing an algorithm on the second control setting to improve the second control setting; (c) repeating (a) to (b) until 36. A honeycomb telephone system comprising: an interference compensation circuit for generating an interference compensation signal based on a noise cancellation control setting. By selecting a plurality of initial control settings; (b) identifying one of the initial control settings for the interference suppression from the plurality of initial control settings; (c) selecting based on the effectiveness of the identified control settings a plurality of subsequent control settings'; (d) identifying, from the subsequent control settings, a subsequent control setting that is better for interference suppression; and (e) repeating (c) through (d) until a stop condition is met. Type: (such as the next page) 90