201203833 六、發明說明： 【發明所屬之技術領域】 本發明係有關一種驅動控制電路，用於對振子相對定 子直線狀往復振動的線性振動馬達進行驅動控制。 【先前技術】 以往，線性振動馬達係使用於電顙刀等特定用途中， 但近年來，其用途正不斷擴大。例如，使用在產生將按下 觸控板時的操作感覺回饋給使用者用的振動的元件中。隨 著這種觸覺（haptics)用途的擴大，今後，預計線性振動馬 達的出貨量將會不斷增加。 線性振動馬達較佳為以儘量接近固有振動數（以下酌 情，也稱作諧振頻率）的頻率來進行驅動，在其諳振頻率與 驅動頻率一致時會產生最強的振動。 專利文獻1 : JP特開2001-16892號公報 【發明内容】 (發明所欲解決之課題） 由於線性振動馬達的固有振動數係由振子的質量以及 彈簧常數來決定，故在製品之間固有振動數會存在偏差。 因此，在對線性振動馬達的驅動電路一律設定固定的驅動 頻率的習知方法中，在製品中，馬達的固有振動數與驅動 頻率也會產生大的偏離，而成為使良率降低的原因。另外， 即使最初馬達的固有振動數與驅動頻率一致，因經時變化 的緣故，兩者也會偏離，而有振動變弱的情況。 本發明是鑒於這種狀況而提出的，其目的在於提供一 4 322708 201203833 種無論線性振動馬達處於哪種狀態，都能夠以儘量接近其 固有振動數的頻率來進行驅動的技術。 (用以解決課題的手段） 本發明一樣態之線性振動馬達的驅動控制電路中，該 線性振動馬達具有定子和振子，兩者之至少一方由電磁鐵 構成，對該電磁鐵的線圈供給驅動電流，使振子相對於定 子進行振動，該線性振動馬達的驅動控制電路包括：驅動 信號生成部，係生成用於使正電流和負電流夾介著非通電 期間交替地流至線圈的驅動信號；驅動部，生成與驅動信 號生成部所生成的驅動信號相應的驅動電流，並供給到線 圈；感應電壓檢測部，在非通電期間，檢測出線圈中產生 的感應電壓；和零交叉檢測部，係檢測出由感應電壓檢測 部檢測出的感應電壓的零交叉。驅動信號生成部根據零交 叉的檢測位置來推定線性振動馬達的固有振動數，使驅動 信號的頻率接近該固有振動數。 此外，以上的構成要素的任意組合、將本發明的表現 形式在方法、裝置、系統等之間進行變換而得到的創新設 計，也作為本發明的形式而同樣有效。 根據本發明，無論線性振動馬達處於哪種狀態，都能 夠以儘量接近其固有振動數的頻率來進行驅動。 【實施方式】 (基本構成） 第1圖是表示本發明實施形態的線性振動馬達200的 驅動控制電路100的結構圖。首先，線性振動馬達200具 5 322708 201203833 有定子210和振子220，兩者中的至少一個係由電磁鐵構 成。在本實施形態中，定子21〇係由電磁鐵構成。定子21〇 是由線圈L1繞在磁性材料的芯件211上而形成，若對線圈 L1通電，則會作為磁鐵產發揮作用。振子220包含永久磁 鐵221 ’永久磁鐵221的兩端（s極侧和N極側）分別透過彈 簧222a、222b固定在樞架（frame)223上。定子210與振 子220是相隔規定的間隙並排配置。此外，也可以與第1 圖的例子相反，振子220由電磁鐵構成，定子210由永久 磁鐵構成。 驅動控制電路100對上述線圈L1供給驅動電流’使振 子220相對於定子210作直線狀往復振動。驅動控制電路 100包括驅動信號生成部10、驅動部20、感應電壓檢測部 30以及零交又檢測部40。 驅動信號生成部10生成用於使正電流和負電流夾介 非通電期間交替流至線圈L1的驅動信號。驅動部20生成 與驅動信號生成部10所生成的驅動信號相應的驅動電 流，並供給到線圈L1。感應電壓檢測部30與線圈L1的兩 端連接，檢測線圈L1的兩端電位差。主要是在非通電期 間，檢測線圈L1中產生的感應電壓。零交叉檢測部40係 檢測由感應電壓檢測部30檢測出的感應電壓的零交又。 驅動信號生成部10根據由零交又檢測部40檢測出的 感應電壓的零交叉的檢測位置，來推斷線性振動馬達200 的固有振動數，並使上述驅動信號的頻率儘量地接近該固 有振動數。亦即，使上述驅動信號的頻率適應性地變化， 322708 6 201203833 以使上述驅動信號的頻率與該固有振動數一致。 更具體而言，驅動信號生成部ίο算出上述驅動信號的 一周期的終止位置與應對應此終止位置的零交叉的檢測位 置之間的差分，並將此差分加到現在之驅動信號的周期寬 度中，從而適應性地控制上述驅動信號的周期寬度。當上 述驅動信號的一周期由通常的相位（零—正電壓—零—負 電壓—零）形成時，應對應上述終止位置的零交叉的檢測位 置即為從上述感應電壓的負電壓零交叉到正電壓的位置。 相反地，當上述驅動信號的一周期由反相位（零—負電壓— 零—正電壓—零）形成時，應對應上述終止位置的零交又的 檢測位置則為從上述感應電壓的正電壓零交叉到負電壓的 位置。 以下，對驅動控制電路100的結構更具體地說明。首 先，對驅動部20、感應電壓檢測部30、零交叉檢測部40 的結構進行說明。零交叉檢測部40包含比較器41以及邊 緣檢測部42。比較器41係比較由感應電壓檢測部30檢測 出的感應電壓和用於檢測零交叉的基準電壓。在該感應電 壓與該基準電壓交叉（cross)的時間點，比較器41會使輸 出反轉。例如，從低位準信號反轉到高位準信號。，邊緣 檢測部42係檢測出比較器41的輸出反轉的位置作為邊緣。 第2圖是表示驅動部20、感應電壓檢測部30以及比 較器41的結構例圖。在第2圖所示的例子中，用Η橋 (bridge)電路構成驅動部20 *用差動放大電路構成感應電 壓檢測部30。 7 322708 201203833 該Η橋電路包含第1電晶體Ml、第2電晶體M2、第3 電晶體M3以及第4電晶體M4。此外，在第2圖中，為了 說明方便，也將線性振動馬達200的線圈L1晝在驅動部 20的框内。由第1電晶體Ml和第3電晶體M3構成的第！ 串聯電路、以及由第2電晶體M2和第4電晶體M4構成的 第2串聯電路係分別連接在電源電位vdd與接地電位之 間。在第1電晶體Ml和第3電晶體M3的連接點（以下，稱 作A點）、與第2電晶體M2和第4電晶體M4的連接點（以 下，稱作B點）之間連接有線圈L1。 在第2圖中，第1電晶體Ml以及第2電晶體M2由P 通道MOSFET構成，在各者的源極-汲極之間，連接有第1 二極體D1以及第2二極體D2作為本體二極體（body d i ode)。第3電晶體M3以及第4電晶體M4由N通道MOSFET 構成’在各者的源極-沒極之間，連接有第3二極體j)3以 及第4二極體D4作為本體二極體。 從驅動信號生成部1〇(更嚴格而言，係後述的解碼器 14)向第1電晶體Ml、第2電晶體M2、第3電晶體M3以及 第4電晶體M4的閘極輸入上述驅動信號。若藉由該驅動信 號控制使得第1電晶體Ml和第4電晶體M4導通；使第2 電晶體M2和第3電晶體M3截止，則正電流會流至線圈L1; 若藉由控制使得第1電晶體Ml和第4電晶體M4截止，使 第2電晶體M2和第3電晶體M3導通，則負電流會流至線 圈L1 〇 上述差動放大電路包含運算放大器〇P1、第1電阻R卜 8 322708 201203833 第2電_、第3電阻R3以及第4電阻R4。運算放大器 0P1的反轉輸入端子經由第1電阻R1而與㈣連接，非反 轉輸入端子則經由第2電阻R2_A點連接。運算放大器 0P1的反轉輸入端子與輸出端子係經由第3電阻R3連接。 將基準電壓Vref作為偏置（咐_電壓，經由第4電阻 R4施加到運算放大器0P1的非反轉輸入端子。 設定第i電阻R1和第2電阻R2的電阻值為相同值， 並設定第3電阻R3和第4電阻R4的電阻值為相同值。在 此條件下，上述差減大電路的放A率成為R3/R1。例如， 設定第1電阻R1和第2電阻R2的電阻值為編，第3 電阻R3和第4電阻R4的電阻值為2_，則將線圈u的 兩端電壓（A-B間電壓）放大2倍。 .對比較器4U由開環的運算放大器構成）的反轉輸入端 子施加基準電壓Vref。比較器41的非反轉輸入端子與運 鼻放大器0P1的輸出端子連接，對該非反轉輸入端子施加 運算放大器0P1的輸出電壓。使基準電壓Verf作為偏置電 壓（例如，l/2Vdd)施加到上述差動放大電路的情況下，為 了使運算放大器0P1與比較器41的範圍（range)相配合， 乃使用基準電壓Vref作為比較器41的參照電壓。此外， 當不施加偏置電壓到上述差動放大電路時，使用接地電壓 作為比較器41的參照電壓。 如此’藉由上述差動放大電路來放大線圈L1的兩端電 壓（A-B間電壓）’然後輸入到比較器41，能夠提高在線圈 L1中產生的感應電壓的零交叉的檢測精度。 9 322708 201203833 第3圖是表示實施形態的驅動控制電路100的動作例 的時序圖。此動作例是以單相全波驅動線性振動馬達200 為例。此時，設定非通電期間。非通電期間設定在正電流 通電期間以及負電流通電期間的各者的前後。亦即，在全 周期中，第1半周期由非通電期間、正電流通電期間、以 及非通電期間構成，第2半周期由非通電期間、負電流通 電期間、以及非通電期間構成。在以下的例子中，在半周 期的180°中，分配40°給非通電期間，分配100°給正（負） 電流通電期間，再分配40°給非通電期間。因此，一周期 中的5/9分配給通電期間，4/9分配給非通電期間。以下， 在本說明書中，將依照此比率的驅動方式稱作100度通電。 在第3圖中，在上述Η橋電路的導通-1狀態（Ml、M4 導通，M2、M3截止）下，正電流係流至線圈L1。在上述Η 橋電路的關斷狀態（Ml至Μ4截止）下，驅動電流不流至線 圈L1。在上述Η橋電路的導通-2狀態（M卜M4截止、M2、 M3導通）下，負電流流至線圈L1。 正電流流至線圈L1中的狀態下，定子210被激磁為Ν 極，由於此磁力，振子220受到向永久磁鐵221的S極侧 的力。由於此力，振子220反抗彈簧222a向永久磁鐵221 的S極側移動，一直移動到彈簧222a的收縮界限。在線圈 L1中沒有驅動電流流通的狀態下，定子210不進行激磁， 不產生磁力。振子220由於彈簧222a的恢復力而向中心位 置移動。在線圈L1中流通負電流的狀態下，定子210被激 磁為S極，由於此磁力，振子220受到向永久磁鐵221的 10 322708 201203833 N極侧的力。由於此力，振子22〇反抗彈簧222b向永久磁 鐵221的N極侧移動，一直移動到彈簧222b的收縮界限。 如此’驅動信號生成部10以導通狀態―導通-1狀態 —截止狀態〜導通-2狀態—截止狀態的循環（cycle)來控 制上述H橋電路，能使線性振動馬達200往復運動。 若上述Η橋電路從導通< 狀態轉移到截止狀態，第1 電晶體Ml至第4電晶體μ全被切換為截止，則再生電流 透過上述本體—極體而流動。在上述Η橋電路從導通狀 ，轉移到截止狀態時也相同。藉由有效利用此再生電流， 月匕夠提尚能量效率，減少驅動控制電路1〇〇的消耗功率。 、上述再生電流以與此前流過線圈U的電流相同的方 向机動。若上述再生電流流動完畢，則因振子220的移動 而感應得到的感應電流會流到線圈L1中。在振子220停止 、狀態下，該感應電流並不流動。振子220停止的狀態係 在振子220到達振子22〇的振動範圍的兩端的瞬間產生。 感應電壓檢測部30藉由監視在非通電期間線圈η中 產生的反電動勢電壓，而能夠估計振子22()的位置。該反 電=勢電壓為零的狀態表示振子220處於停止（即，位於振 動祀圍的S極側最大到達地點或者N極侧最大到達地點）。 因此’零交又檢測部40乃檢測線圈u的兩端電壓（a_b 間電麗）零交又（除了由驅動電流以及再生電流引起的零交 又）的時刻，並測量檢測出的零交又的期間，而能夠求出線 性振動馬達200的固有振動數1外，連續的零交叉的期 間表示線性振動馬達的半振動周期寬度，跳過一個的 322708 11 201203833 零交叉的期間表示其全振動周期寬度。 在本實施形態中，零交叉檢測部40僅檢測在非通電期 間線圈L1的兩端電壓（A-B間電壓）從負零交叉到正的時間 點。在此情況下，第2圖所示的比較器41被設定成：在運 算放大器0P1的輸出電壓比基準電壓Vref低的期間，輸出 低位準信號；若運算放大器0P1的輸出電壓比基準電壓 Vref高時，則輸出高位準信號。 驅動信號生成部10利用與所測量的線性振動馬達200 的固有振動數對應的周期寬度，來調整下一驅動信號的周 期寬度。藉由重複此測量和調整，驅動控制電路100能夠 以其諧振頻率或者其附近的頻率來持續地驅動線性振動馬 達 200。 回到第1圖，更具體地說明驅動信號生成部10。驅動 信號生成部10包括：第1鎖存（latch)電路11、主計數器 12、迴圈（loop)計數器13、解碼器14、第2鎖存電路15、 差分計算電路16、第3鎖存電路17、加法電路18、以及 第4鎖存電路19。 第1鎖存電路11係鎖存應與上述驅動信號一周期的終 止位置對應的計數終止值，並在第3時脈信號CLK3指示的 時刻輸出到主計數器12以及解碼器14。此外，也能夠輸 出到差分計算電路16。在第1鎖存電路11中，在線性振 動馬達200的驅動開始時從未圖示的暫存器等設定上述計 數終止值的初始值。在驅動開始後，從第4鎖存電路19輸 入的值成為上述計數終止值。 12 322708 201203833 值=器12係從第1鎖存電路^設定上述計數終止 ΐ數:Γ始值起進行重複計數到該計數終止值為止。 Η#，t㈣樹止值設定 為199時，主計數器12成為從 的進制計數器。主計數薄12 進订重複向上計數 數器，器14以及第;鎖存 其二13每一次在主計數器12計數迴圈終止時 次數。並保存主計數器12的計數迴圈 計數初始值起^；Γ計數迴圈是指從主計數器12的上述 對應一個驅料數終止值為止。—次計數迴圈 解喝器14°/ ’故计數迴圈次數對應驅動周期次數。 上述計數終±值相11^^12提㈣計數值，生成與 詳細結構容德、+、％周期寬度的驅動信號。解碼器14的 12提供的計數信，第2鎖存電路15依次鎖存由主計數器 的位置所鎖广& ’並將在零交又檢測部40檢測出零交又 零交又的位】2數值輸出到差分計算電路16。該檢測出 知。若檢测出該檢測部42輸入的邊緣信號來通 產生，則第2 ^、 位置始終理想地在相同的時間點 差分計I雷子電路/5的輸出始終為相同的計數值。 值與現在16算峻第2鎖存電路15輸入的計數 1鎖存電路u於、止值之間的差分。第1圖中描述了從第 計算電路:入，在之計數終止值的例子。此外，差分 以是從第^保存現在之計數終止值的結構，也可 鎖存電路19提供計數終止值的結構。 322708 13 201203833 當檢測出零交叉的位置的計數值（=從第2鎖存電路15 輸入的計數值）比現在之計數終止值小時，差分計算電路 16從前者中減去後者。例如，檢測出零交叉的位置的計數 值為197，而現在之計數終止值為199時，差分計算電路 16輸出-2。 當檢測出零交叉的位置的計數值比現在之計數終止值 大時，從第2鎖存電路15輸入的計數值是相對於現在之計 數終止值的增加部分的值。在此情況下，差分計算電路16 直接輸出從第2鎖存電路15輸入的計數值。例如，檢測出 零交叉的位置的本來的計數值為201，而現在之計數終止 值為199時，從第2鎖存電路15輸入的計數值為2，差分 計算電路16直接輸出2。由於該計數值在199被重置 (reset)，故從第2鎖存電路15輸入的計數值不是201， 而是2。 第3鎖存電路17鎖存從差分計算電路16輸入的差 分，在由第1時脈信號CLK1指示的時間點，將此差分輸出 到加法電路18。加法電路18將第3鎖存電路17輸入的差 分加到第4鎖存電路19輸入的現在之計數終止值上。第4 鎖存電路19鎖存從加法電路18輸入的值，在由第2時脈 信號CLK2指示的時間點，輸出到第1鎖存電路11。在第4 鎖存電路19中，在線性振動馬達200的驅動開始時，也從 未圖示的暫存器等設定上述計數終止值的初始值。 由加法電路18生成的值作為新的計數終止值，經由第 4鎖存電路19以及第1鎖存電路11被設定給主計數器12 14 322708 201203833 以及解碼器14。因此，在主計數器12以及解碼器14中始 終設定反應了剛才的零交叉的檢測位置的計數終止值。 第4圖是表示邊緣信號、第1時脈信號CLK1、第2時 脈信號CLK2、以及第3時脈信號CLK3 —例的時序圖。邊 緣信號係從邊緣檢測部42設定至第2鎖存電路15。第1 時脈信號CLK1是使邊緣信號延遲半個時脈的信號。此半個 時脈的延遲係考慮到要藉由差分計算電路16進行運算處 理的緣故。第2時脈信號CLK2是使第丄時脈信號cui延 遲半個時脈的信號。此半個時脈的延遲是考慮到要藉由加 法電路18進行運算處理的緣故。 第3時脈信號CLK3是使第2時脈信號⑽延遲幾個 時脈的信號。該幾個時脈的延遲是為了要在現在之驅動周 期的計數終止前抑制現在之驅動周期的計數終止值被改 變。例如’若不設置第1鎖存電路u ’則在現在之驅動周 期中，在其終土位置之前檢測出零交又時，反應出其零交 又位置的新的計數終止值有可能從現在之驅動周期起被使 用，而不是從下一個驅動周期被使用。在這種情泥下，由 於以更新前的計數終止值為基準來決定通電期間，故通電 期間與非通電期間的比率將不能維持。在本實施形態中， 將導致100度通電不能維持。 藉由在第4鎖存電路19與主計數器12之間設置第i 鎖存電路11，即能夠使將主計數器12設定中的現在之計 數終止值更新成反應出零交又位置的新計數終止值的時間 點延遲。 15 322708 201203833 (解碼器結構） ，第5圖是表示解碼器14的結構例圖。解碼$ i4根據 上述計數終止值乘㈣數之後得到的值，來決定與上述驅 動信號的通電㈣對應的計數寬度，該係數是用於使通電 期間相對於上述驅動信號的—周期的比率為固定。如上所 述在上述驅動信號的一周期中包括正電流通電期間和負 電流通電期間。因此，在上述⑽度通電的情況下，各通 電：間相對於上述驅動信號的一周期的比率為1〇〇。/36〇。 (.〇· 28)。另外，各通電期間的半期間相對於上述驅動信 號的一周期的比率為50。/360。（％〇· 14)。 ° 此外，解碼器14根據上述計數終止值乘以用於決定上 述，動信號的通電期間的中心位置的係數而得到的值，來 決疋與上述驅動彳s號的通電期間的開始位置以及終止位置 對應的計數值。如上所述，上述職信號的—周期由在前 後設定了非通電期間的正電流通電期間以及在前後設定了 非通電期間的負電流通電期間形成。在此，設定正電流通 電期間的長度以及貞f流通電㈣的長度相等，並設定非 通電期間的長度全部相等。 因此，用於決定上述驅動信號的正電流通電期間的中 心位置的係數被設定為0.25,用於決定上述驅動信號的負 電流通電期間的中心位置的係數被設定為〇. 。此外，合 上述驅動信號的相位相反時，用於決定負電流通電期間的 中心位置的係數被設定為〇. 25，而用於決定正電流通電期 間的中心位置的係數被設定為〇. 75。 322708 16 201203833 如此，解碼器14能夠算出與各通電期間對應的計數寬 度、以及與各通電期間的中心位置對應的計數值。而且， 藉由從與該中心位置對應的計數值中減去上述計數寬度的 一半的值，能夠算出與各通電期間的開始位置對應的計數 值。另外，藉由在與該中心位置對應的計數值中加上上述 計數寬度的一半的值，能夠算出與各通電期間的終止位置 對應的計數值。 以下，進行更具體的說明。解碼器14包括：驅動寬度 計算部51、正驅動中心值計算部52、負驅動中心值計算部 53、正侧減法部54、正側加法部55、負侧減法部56、負 側加法部57、正驅動信號生成部58、以及負驅動信號生成 部59。 驅動寬度計算部51係將各通電期間（以下也適當地稱 作驅動期間）的半期間相對於上述驅動信號的—周期的比 率作為係數㈣保存。在上述⑽度通電的情況下，保存 0· 14。驅動寬度計算部51從第1鎖存電路丨1被提供計數 終止值。驅動寬度計算部51在其計數終止值上乘以該係 數。如此’能夠算出與各驅動期_半期間對應的計數寬 度。 。正驅動中心值計算部52係保存用於決定上述驅動信 號的正電流通電期間（以下也適當地稱作正驅動期間）的中 心位置的係數。在本實施形態中，保存G.25。正驅動令心 值計算部52從第i鎖存電路u被提供計數終止值。正驅 動中〜值4算部52在其計數終止值±乘以該係數。如此， 322708 17 201203833 能夠算出與各正驅動期間的中心位置對應的計數值。 負驅動中心值計算部53係保存用於決定上述驅動信 號的負電流通電期間（以下也適當地稱作負驅動期間）的中 心位置的係數。在本實施形態中，保存0. 75。負驅動中心 值計算部53從第1鎖存電路11被提供計數終止值。負驅 動中心值計算部53在其計數終止值上乘以該係數。如此， 能夠算出與各負驅動期間的中心位置對應的計數值。 正側減法部54藉由從與正驅動中心值計算部52供給 的正驅動期間的中心位置對應的計數值中減去由驅動寬度 計算部51供給的計數寬度，而算出與正驅動期間的開始位 置對應的計數值。正側加法部5 5藉由在與正驅動中心值計 算部52供給的正驅動期間的中心位置對應的計數值中加 上由驅動寬度計算部51供給的計數寬度，而算出與正驅動 期間的終止位置對應的計數值。 負側減法部56藉由從與負驅動中心值計算部53供給 的負驅動期間的中心位置對應的計數值中減去由驅動寬度 計算部51供給的計數寬度，而算出與負驅動期間的開始位 置對應的計數值。負側加法部57藉由在與負驅動中心值計 算部53供給的負驅動期間的中心位置對應的計數值中加 上由驅動寬度計算部51供給的計數寬度，而算出與負驅動 期間的終止位置對應的計數值。 正驅動信號生成部58係從主計數器12被供給作為同 步時脈的計數值，從正側減法部54被供給與正驅動期間的 開始位置對應的計數值，從正側加法部55被供給與正驅動 18 322708 201203833 =應的計數值。正驅動信號生成部58依照 的;數;:、3十數值，從與正驅動期間的開始位置對應 到與正驅動期間的終止位置對應的計數值為止 輸出有思義的信號（例如，高位準信號）作為正驅動信號。 在除此以外的期間，輸出非有意義的信號（例如，低位準作 號）。 ° 此外’正驅動信號生成部58能夠以所設定的工作比的 PWM仏號來生成該正驅動信號。由正驅動信號生成部58生 成的正驅動信號被輸入到驅動部2〇，具體而言係輸入到第 1電晶體Ml以及第4電晶體M4的閘極。此外，在第1電 晶體Ml的刖段設置有未圖示的反相器（inverter)，使該正 驅動信號反轉相位，並輸入到第1電晶體Ml的閘極。 負驅動信號生成部59係從主計數器12被供給作為同 步時脈的計數值，從負側減法部56被供給與負驅動期間的 開始位置對應的計數值’從負侧加法部57被供給與負驅動 期間的終止位置對應的計數值。負驅動信號生成部59依照 作為同步時脈的計數值，從與負驅動期間的開始位置對應 的計數值起到與負驅動期間的終止位置對應的計數值為止 輸出有意義的信號（例如，高位準信號）作為負驅動信號。 在除此以外的期間，輸出非有意義的信號（例如，低位準信 號）。 此外，負驅動信號生成部59能夠以所設定的工作比的 PM信號來生成該負驅動信號。由負驅動信號生成部59生 成的負驅動信號被輸入到驅動部20，具體而言’係輸入到 19 322708 201203833 第2電晶體M2以及第3電晶體M3的閘極。此外，在第2 電晶體M2的前段設置有未圖示的反相器，使該負驅動信號 反轉相位，並輸入到第2電晶體M2的閘極。 第6圖是表示驅動信號的一周期的波形圖。在第6圖 中，網點區域表示正驅動期間（前）以及負驅動期間（後）。 與正驅動開始值a對應的計數值由正側減法部54生成，與 i驅動中心值b對應的計數值由正驅動中心值計算部52生 成’與正驅動終止值c對應的計數值由正側加法部55生 成。同樣’與負驅動開始值d對應的計數值由負側減法部 56生成’與負驅動中心值e對應的計數值由負驅動中心值 計算部53生成，與負驅動終止值f對應的計數值由負側加 法部5 7生成。 如第5圖所示，藉由構成解碼器14，即使因上述驅動 信號的頻率改變而使其周期寬度改變，驅動信號生成部1〇 也能夠調整上述驅動信號，以便維持上述驅動信號的通電 期間與非通電期間的比。另外，即使上述驅動信號的周期 寬度改變’驅動信號生成部10也能調整上述驅動信號，以 維持在一周期中的通電期間的信號相位的相對位置關係。 第7圖是驅動信號的通電期間寬度的控制的說明圖。 第7圖（a)是表示在驅動周期為預設狀態下的線圈驅動電 壓的推移圖’第7圖（b)是表示使驅動周期調整得比預設狀 態長之後的線圈驅動電壓（通電期間寬度的無調整）的推移 圖’第7圖（c)是表示使驅動周期調整得比預設狀態長之後 的線圈驅動電壓（通電期間寬度的有調整）的推移圖。 20 322708 201203833 在第7圖（a)中，設定為上述1〇〇度通電。亦即，！個 驅動周期中的通電期間與非通電期間的比被設定為5 ·· 4。 在第7圖（b)中，表示使驅動周期調整成比預設狀態長以 後，仍維持預設狀態下的通電期間寬度的例子。該情況下， 存在對線性振動馬達2〇〇的驅動力降低，線性振動馬達2〇〇 的振動變弱的可能性。 在第7圖（c)中，使驅動周期調整得比預設狀態長以 後，進行控制，使得仍維持丨個驅動周期中的通電期間與 非通電期間的比。在本實施形態中，係進行控制，以便維 ，上述100度通電。該控制藉由解碼器14内的驅動寬度計 算部51的作用來實現。 在此，說明了使驅動周期調整得比預設狀態長的例 子，但調整成比預設狀態短的例子也相同。若使驅動周期 調整得比預設狀態短以後，仍維持預錄態中的通電期間 寬度，則存在對線性振動馬達2〇〇的驅動力上升，線性振 動馬達200的振動變強的可能性。關於這一點，在本實施 形態中’也進行控制使得在驅動周期從預設狀態被調整短 以後仍維持100度通電。 第8圖是用於說明驅動信號的相位控制的圖。第8圖 表示調整為線性振動馬達2GG的諧振頻率後的線圈u的兩 端電壓的推移。此外，為了簡化說明，再生電流省略描述。 第1段的波形表*在驅動信號的相位是最佳㈣下驅動線 性振動馬達200的狀態。 第2段的波形表示從其第2周期起驅動信號的相位為 322708 201203833 ^目位延遲後狀態下驅動線性振動馬達2〇〇的狀態。此狀態 疋驅動周期被調整得比之前短的情況，該調整之後，各通 電期間的開始位置以及終止位置在仍維持其調整前的位置 的情況下產生的。 第3段的波形表示從产第2周期起驅動信號的相位為 相位超前狀態下驅動線性_動馬達2〇〇的狀態。此狀態是 驅動周期被s周整得比之前丨养的情況，該調整之後，各通電 期間的開始位置以及終止位置也在維持其調整前的位置的 情況下產生的。 亦即’在各通電期間^開始位置以及終止位置固定的 情況下，若改變驅動周期寬度，則驅動信號的相位會產生 延遲或超前。與此相對地，在本實施形態中，若改變驅動 周期’則由於適應性地調整各通電期間的開始位置以及終 止位置’故能夠將驅動信號的相位保持在最佳。該開始位 置以及終止位置的調整，主要藉由解碼器14内的正驅動中 心值計算部52以及負驅動中心值計算部53的作用來實現。 如以上所說明，根據本實施形態的驅動控制電路1〇〇， 藉由利用與所測量的線性振動馬達200的固有振動數對應 的周期寬度’來調整下一驅動信號的周期寬度，從而無論 線性振動馬達2〇〇處於哪種狀態，都能夠以儘量接近其固 有振動數的頻率來持續地進行驅動。 因此’能夠吸收線性振動馬達2〇〇的製品間所產生的 固有振動數的偏差，能夠防止在量產馬達時的良率的降 低。另外，即使彈簧222a、222b等產生經時變化，由於以 22 322708 201203833 與經時變化後的固有振動數對 故也能夠抑制振動減弱。 、°動頻率來進行驅動， 另外，在適應性地控制驅 振動馬達咖的固有振動數與周期寬度使得線性 夠將因周期寬度改變造成的影響抑頻率一致時，能 即使驅動信號的周期寬度被改變，藉最小。具體而言， 以維持-周期中的通電期間與非通電整通電期間寬度 夠維持對祕振動料綱的_力’。，從而能 驅動力的變動而使線性振動馬達2QQ 能夠抑制因 此外，即使驅動信號的周期寬度被改變變^；，象: 電期間的開始位置以及終止位置調整成最佳位^以冬各通 -周期中通電期間的相對位置關係’也能夠= 的下降。亦即，若驅動信號的相位偏離，則振子22〇 ' ' 置和驅動力的供給位置會產生偏離，驅動效率會下降。 於這一點，能夠藉由將驅動信號的相位維持在最佳位置： 從而以相同的消耗功率得到最大限度的振動。 (上升控制） 以下，對本實施形態的驅動控制電路100進行的可追 加到上述驅動控制中的第1上升控制進行說明。如第6圖 所示，上述驅動信號的一周期係藉由在前後設定了非通電 期間的正電流通電期間以及在前後設定了非通電期間的負 電流通電期間形成。藉此，能夠如第3圖所示地以高精度 地檢測感應電壓的零交又，如第8圖所示也能夠提高驅動 效率。 322708 23 201203833 因此，原則上’在上述驅動信號中的最初周期的正電 流通電期間（反相位的情況下’為負電流通電期間）之前也 設定非通電期間。然而，此非通電期間卻朝線性振動馬達 200的上升時間延遲的方向作用。因此，為了改善這種情 況，驅動信號生成部10可以執行以下的上升控制。 亦即，驅動信號生成部10 ’在線性振動馬達200的驅 動開始後，將上述驅動信號的至少應設定在最初的通電期 間之前的非通電期間寬度設定得比在線性振動馬達200穩 定動作時應設定在各通電期間之前的非通電期間寬度短。 例如，驅動信號生成部10在線性驅動馬達200的驅動開始 後，亦玎將上述驅動信號的至少應設定在最初的通電期間 之前的#通電期間寬度設定為零。 應在前面設定有比在穩定動作時應設定在各通電期間 之前的非通電期間寬度還短的非通電期間寬度的通電期 間，可以僅有最初的通電期間’亦可為從最初的通電期間 起到第n(n為自然數）個通電期間。在後者的情況下，可以 依照從最初的通電期間向第η個通電期間接近的順序，而 加長應設定在各者前面的非通電期間寬度。 另外，在通電期間之前，在設定了比在穩定動作時應 設定在各通電期間前的非通電期間寬度更短的非通電期間 寬度的期間，驅動信號生成部1 〇可以停止上述驅動信號的 周期寬度的調整處理。該情況下，可以停止由感應電壓檢 測部30以及零交又檢測部4〇進行的上述感應電壓的零交 叉檢測處理。 322708 24 201203833 接下來，對本實施形態的驅動控制電路1〇〇進行的可 追加到上述驅動控制中的第2上升控制進行說明。如第5 圖所示’驅動信號生成部10能夠以PWM信號生成各通電期 間的信號。由此，能夠配合線性振動馬達2〇〇的性能，調 整驅動能力。 第2上升控制的前提在於以PWM信號生成各通電期間 的信號。驅動信號生成部1〇在線性振動馬達2〇〇的驅動開 始後，將上述驅動信號的在至少最初的通電期間生成的p w M 信號的工作比設定得比在線性振動馬達200穩定動作時在 各通電期間生成的PWM信號的工作比高。例如，驅動信號 生成部10在線性振動馬達200的驅動開始後，可以將上述 驅動信號的在至少最初的通電期間生成的PWM信號的工作 比設定為1。 生成比在穩定動作時各通電期間内所生成的p醫信號 的工作比更高工作比的PWM信號的通電期間可以僅是最初 的通電期間，亦可為從最初的通電期間到第mb為自然數） 個的通電期間。在後者的情況下，可以依照從最初的通電 期間向第m個通電期間接近的順序，降低在各通電期間所 生成的PWM信號的工作比。 另外’在生成比穩定動作時各通電期間内所生成的p题 4號的工作比更高工作比的PWM信號的期間，驅動信號生 成部10可以停止上述驅動信號的周期寬度的調整處理。該 情況下’可以停止由感應電壓檢測部3〇以及零交叉檢測部 40進行的上述感應電壓的零交叉檢測處理。 25 322708 201203833 第1上升控制以及第2上升控制既可以各自單獨使 用，也可以並用。以下，說明當採用第丨上升控制以及第 2上升控制中的至少一個時的解碣器14的結構例。 第9圖是表示追加了上升控制功能的解碼器14的結構 例的圖。如第9圖所示的解碼器14是在第5圖所示的解碼 器14中追加了上升控制部60的結構。當執行第1上升控 制時，上升控制部60修正從主計數器12輪入給正驅動信 被生成部58以及負驅動L就生成部59的計數值。 例如，將應設定在通電期間之前的非通電期間寬度設 定為零的情況下，上升控制部60將與穩定動作時應設定在 各通電期間之前的#通電期間寬度相對應的計數寬度加到 從主計數器12輸入的計數值中。由此，正驅動信號生成部 58以及負驅動信號生成部59可以省略應設定在正電流通 電期間以及負電流通電期間的各者之前的非通電期間。 此外，同樣的處理也能夠在將應設定於通電期間之前 的非通電期間寬度設定為零的期間，將主計數器12的計數 初始值設定為在穩定動作期間的計數初始值中加入上述計 數寬度的值。在本實施形態中，將主計數器12的計數初始 值设定為上述1〇〇度通電開始時的計數值。此處理可以由 解碼器14以外的未圖示的其他上升控制部執行。 當執行第2上升控制時，上升控制部6〇在正驅動信號 生成部58以及負驅動信號生成部的中，設定上述驅動信 號的至少最初的通電期間内生成的pWM信號的工作比。此 時’設^比穩定動作時各通電期間所生成的酬信號的工 26 322708 201203833 作比高的工作比。 第10圖是用於說明第i上升控制的圖。帛1〇圖⑷ 疋表不虽不執行第i上升控制時的線圈驅動電壓以及線性 f動馬達200的振動的推移圖’第10圖⑹是表示當執行 第1上升控制時的線圈驅動電壓以及線性振動馬達2〇 振動的推移圖。 :第10圖⑷、第10圖⑻中，描述了在驅動信號的 第2周麟性振動馬達2〇〇的振動到達期望水準（即，穩 動作時的水準）的例子。在第1Q圖（b)中，驅動信號生㈣ 10將應設定在上述驅動信號的最初的通電期間之 電期間寬度設定為零。 非通 "第10圖⑷内的期㈣表示不執行第！上升控制時的 從驅動開始辆到振動到達期望水準為止的期間，第^圖 ⑹内的期間t2表示執行了第1上升控制時的從驅動開始 時起到振㈣達㈣水料止的期I若比較朗tl與期 間t2，則可知期間t2較短，藉由執行第1上升控制，而 縮短了從驅_始時起到振關軸望水料止的期間。 Θ第11圖是用於說明第2上升控制的圖。第u圖⑷ 是表不當不執行第2上升控制時的線圈驅動㈣的推移 圖’第11圖⑸是表示當執行了第2上升控制時的線圈驅 動電壓的推移圖。在第11圖⑷卜驅動信號生成部1〇在 驅動開始後，根據最初的通電期間的信號以簡信號生成 各通電期間的信號。在第u圖⑻中，驅動信號生成部1〇 在驅動開始後’以非簡信號生成最初的通電期間的_ 322708 27 201203833 號，以PWM信號生成第2周期以後的通電期間的信號β 如以上說明，若採用第1上升控制，則能夠縮短從驅 動開始到線圈L1中通電的時間，能夠縮短從線性振動馬達 200的驅動開始時起至得到期望的振動為止的上升時間。 另外，若採用第2上升控制，則相較於穩定動作時的驅動 力能夠提高上升時的驅動力’能夠縮短該上升時間。 (停止控制） 以下，對本實施形態的驅動控制電路100進行的可追 加到上述驅動控制中的停止控制進行說明。驅動信號生成 部10在線性振動馬達200的驅動終止後，生成與該驅動時 所生成的驅動信號的相位相反的.驅動信號。驅動部2〇藉由 對線圈L1提供與驅動信號生成部1〇所生成的反相位的驅 動信號相應的反相位的驅動電流，而使線性振動馬達2〇〇 的停止加速《若對線圈L1梃給該反相位的驅動電流，則定 子210將發揮使振子220停止的制動作用。在本說明書中， 所謂線性振動馬達200的驅;動終止時，是指不包含用於停 止控制的反驅動期間在内的jL規驅動終止時。 驅動信號生成部10可以用PWM信號來生成在線性振動 馬達200驅動終止後生成的反相位驅動信號的各通電期間 的信號。藉由調整此ρ·信號的工作比，能夠靈活地調整 制動力。 如上所述，驅動信號生成部1〇能夠以PWM信號來生成 各通電期間的信號。在以用PWM信號來生成各通電期間的 信號為前提的情況下，驅動信號生成部10可以採用以下的 28 322708 201203833 停止控制。亦即，驅動信號生成部ίο可以將線性振動馬達 200驅動終止後的反相位驅動信號的通電期間生成的PWM 信號的工作比，設定得比線性振動馬達200驅動時的驅動 信號的各通電期間生成的P丽信號的工作比低。 另外，驅動信號生成部10可以根據線性振動馬達200 驅動時的驅動信號的供給期間，來調整線性振動馬達200 的驅動終止後的反相位驅動信號的供給期間。例如，上述 驅動時的驅動信號的供給期間越短，則驅動信號生成部10 將上述驅動終止後的反相位的驅動信號的供給期間設定得 越短。例如，使上述反相位的驅動信號的供給期間與上述 驅動時的驅動信號的供給期間成比例。此外，在上述驅動 時驅動信號的供給期間超過規定的基準期間的區域中，上 述反相位的驅動信號的供給期間可以是固定。上述驅動信 號的供給期間可以由驅動周期次數來確定。 另外，驅動信號生成部10可以根據線性振動馬達200 驅動時的驅動信號的供給期間，來調整在線性振動馬達200 驅動終止後的反相位驅動信號的通電期間生成的PWM信號 的工作比。例如，上述驅動時的驅動信號的供給期間越短， 則驅動信號生成部10將該PWM信號的工作比設定得越低。 例如，使該PWM信號的工作比與上述驅動時的驅動信號的 供給期間成比例。此外，在上述驅動時驅動信號的供給期 間超過規定的基準期間的區域，上述PWM信號的工作比可 以是固定的。 第12圖是表示追加了停止控制功能的解碼器14的結 29 322708 201203833 構例圖。第12圖所示的解碼器14是在第5圖所示的解碼 器14中追加了停止控制部61的結構。若線性振動馬達200 的驅動終止，則停止控制部61會指示正驅動信號生成部 58以及負驅動信號生成部59,使其生成與該驅動時所生成 的驅動信號的相位相反的驅動信號。此時，可以指示以PWM 信號生成該反相位驅動信號的通電期間的信號。 另外，當根據線性振動馬達200驅動時的驅動信號的 供給期間，來調整上述反相位驅動信號的供給期間時，停 止控制部61從迴圈計數器13接受計數迴圈次數（即驅動周 期次數）。停止控制部61係指示正驅動信號生成部58以及 負驅動信號生成部59，使其生成反應此驅動周期次數的上 述反相位驅動信號。根據線性驅動馬達200驅動時的驅動 信號的供給期間來調整上述PWM信號的工作比的情況也同 樣。 第13圖是用於說明上述停止控制的基本概念圖。第 13圖（a)是表示當不執行停止控制時的線圈驅動電壓的推 移圖，第13圖（b)是表示當執行了停止控制時的線圈驅動 電壓的推移圖，第13圖（c)是表示當藉由PWM信號執行停 止控制時的線圈驅動電壓的推移圖。 在第13圖（b)、第13圖（c)中描述了驅動終止後的反 相位驅動信號的周期為一次的例子，但也可以是多次。在 多次的情況下，當以PWM信號生成該驅動信號的通電期間 的信號時，隨著該反相位驅動信號的周期前進，可以降低 該PWM信號的工作比。 30 322708 201203833 ^ 圖疋用於說明在上述停止控制中反相位驅動信 躺周期次數固定的例子。帛14圖⑷是表示驅動時的驅 動七號的周U較多時的線圈轉電_及線性振動馬 達200的振動的推移圖，帛Η圖⑻是表示驅動時的驅動 ㈣的周期次數較少時的線圈驅動電壓以及線性振動馬達 200的振動的推移圖。 在第14圖中，描述了將驅動終止後所生成的反相位驅 動信號的周期次數固^為2的例子。第14圖⑷描述了驅 動時的驅動化號的周期次數為4的例子，第14圖⑹描述 了驅動時的驅動信號的周期次數為2的例子。在第14圖⑷ 中可知藉由對線圈Li供給兩周期份的反相位驅動信號， 在線!·生振動馬達200的驅動終止後，線性振動馬達2〇〇的 振動會快速收斂。 、另一方面，在第14圖（b)中，藉由對線圈L1供給兩周 期份的反相位驅動㈣，在線性振動馬達2⑽的驅動終止 後，線性縣馬達的難會快賴斂，但其後會產生 反相位的振動（參照橢圓區域的部分）。這意味著針對線性 振動馬達200驅動時的振動提供了過剩的制動力。 。第15圖是用於說明在上述停止控制中反相位驅動信 號的周期次數為可變的例子4 15圖⑷是表示驅動時的 驅動信號的㈣次數較多時的線圈驅動以及線性振動 馬達細的振動的推移圖，第15圖⑹是表示驅動時的驅 動信號的周期次數較少時的_軸錢以及線性振動馬 達200的振動的推移圖。 322708 31 201203833 第15圖（a)是與第14圖（a)相同的圖。第15圖（b)描 述了驅動時的驅動信號的周期次數為2，而在驅動終止後 所生成的反相位驅動信號的周期次數為丨的例子。由第 圖（b)中可知’藉由對線圈L1供給一周期份的反相位驅動 信號，在線性振動馬達2〇〇的驅動終止後，線性振動馬達 200的振動會快速收斂。與第14圖化）比較，可知在第15 圖（b)中線性振動馬達2〇〇不產生反相位的振動。 在第14圖中，並未考慮線性振動馬達2〇〇的驅動終止 則的線性振動馬達200的振動強度，就提供了固定的制動 力。因此，會發生其制動力過大或者過小的情況。對此， 在第15圖中，藉由提供反應線性振動馬達2〇〇的振動強度 的制動力，從而能夠實現最佳的停止控制。 如以上說明，若採用上述的停止控制，則能夠縮短線 性振動馬達200的驅動終止時的振動停止時間。另外，藉 由以PWM信號生成上述反相位驅動信號的通電期間的信 號，能夠靈活地設定制動力。此外，根據線性振動馬達2〇〇 驅動時的驅動信號的供給期間來調整上述反相位驅動信號 的供給期間’則無論該驅動時的驅動信號的供給期間的長 短’都能夠實現最佳的停止控制。在觸覺用途中，藉由使 振動急劇地變化，使用者容易親身感受因接觸產生的振 動°藉由採用上述的停止控制，能夠使振動急劇地變化。 (檢測窗設定） 接下來，對零交又檢測部40設定用於回避上述感應電 壓以外的電壓的零交又檢測的檢測窗的例子進行說明。零 32 322708 201203833 =又檢測部4G將在婦職内檢測出的零交叉認定為有 ^而將在該檢測窗外檢測出的零交叉認定為無效。在此， 所明上述感應電壓以外的電屋的零交又，主要是由驅動信 號生成σ|5 1〇進行通電的驅動電虔的零交又以及再生電麼 的零X又（參照第3 m。因此’該檢測窗原則上為使設定 在正（負）電流通電期間與負（正）電流通電期間之間的非通 電期間設定在往内側變窄的期間。 此時，有必要從該非通電期間至少除去再生電流流通 的期間H若將上述檢測窗蚊得過窄，助法檢測 正規的感應f㈣零交叉的可能性會變高。為此，考慮檢 測上述感應電壓以外的電壓零交叉的可能性和無法檢測正 規的感應電壓零交又的可能性之間的權衡（tradeoff)關 係’來決定上述檢測窗的期間。 接下來，對在上述檢測窗内未檢測出零交又的情況進 行說明。在上述情況下，零交叉檢測部40，在上述檢測窗 的開始位置，上述感應電壓的零交叉已經終止的情況下， 假設在上述檢測窗的開始位置附近檢測出了零交叉，並將 假S史的零交叉的檢測位置供給到驅動信號生成部1〇。所謂 在上述檢測窗的開始位置上述感應電壓的零交又已經終止 的情況’是指在上述檢測窗的開始位置線圈L1的兩端電壓 處於零交叉後的極性的情況。在第3圖所示的例子中，是 在上述檢測窗的開始位置線圈L1的兩端電壓為正的情況。 另外’零交叉檢測部40在上述檢測窗内未檢測出零交 又的情況下，亦即當在上述檢測窗的终止位置，上述感應 322708 33 201203833 電壓的零交又未終止時，假設在上述檢測窗的終止位置附 近檢測出了零交又，並將假設的零交叉的檢齡置供給到 驅動仏唬生成部10。所謂在上述檢測窗的終止位置時，上 述感應電壓的零交又未終止的情況，是指在上述檢測窗的 終止位置線圈L1的兩端電壓處於零交叉前的極性的情 況。以下，說明用於實現這些處理的零交叉檢測部40的结 構例。 第16圖是表示具有檢測窗設定功能的零交叉檢測部 40的結構圖。第16圖所示的零交叉檢測部40是在第1圖 所示的零交叉檢測部4〇中追加了檢測窗設定部Μ以及輸 出控制部44的構成。檢測窗設定部43將用於設定檢測窗 的信號供給到輸出控制部44。更具體而言，提供檢測窗信 號2以及檢測窗開始信號。 第17圖是用於說明檢測窗信號丨、檢測窗信號2以及 檢測窗開始信號的圖。檢測窗信號丨是依據上述見解所生 成的信號，也就是設定有將非通電期間向内側縮小的檢測 窗信號。檢測窗信號2與檢測窗信號丨相比，是檢測窗的 終止位置被延伸到包含後續的通電期間的開始位置的信 號。如此，比較器41不僅根據上述感應電壓的零交又，還 根據在該通電期間所供給的驅動電壓的零交又，使輸出反 轉。檢測窗開始信號是表示檢測窗的開始位置的信號。更 具體而言，是在該檢測窗的開始位置邊緣立起的信號。 回到第16圖，在上述檢測窗的開始位置處比較器41 的輸出未反轉的情況下，輸出控制部44將邊緣檢測部42 322708 34 201203833 檢測出的邊緣位置作為零交叉的檢測位置供給到驅動信號 生成部10(更嚴格地說，是第2鎖存器電路15)。在上述檢 測窗的開始位置，比較器41的輸出已反轉的情況下，輸出 控制部44將上述檢測窗的開始位置作為零交又的檢測位 置供給到驅動信號生成部10(更嚴格地說，是第2鎖存器 電路15)。以下，說明用於實現這些處理的輸出控制部44 的結構例。 第18圖是表示輸出控制部44的結構例圖。該輸出栌 制部44包括第1AND閘71、第2AND閘72以及〇R閘？3 = 上述檢測窗開始信號以及比較器41的輸出信號被輸入到 第1AND閘7卜第聰閉71在兩者為高位準信號時輸出 ，位準’在至少其中—方為低位準信號時輸出低位準信 號。更具體而t ’在上述檢測f的開始位置處比較器Μ的 輸出已反轉的情況下，第議閘71輸出高位準信號。 上述檢測窗信號2以及邊緣檢測部42的輸出信號被輸 2M，72。第_閘72在兩者為高位準信號時 〜间位準U纟至少其中_方為低位準信號時 =號=體而言’在上述檢測窗内，邊緣檢測部42 的輸出信號中邊緣立起時，第鳩閘？2輪 。第議閉71的輪出信號以及第_閘72的輸出作 娩被輸入到GR㈣。⑽閘73以兩者 出邊緣信號，閘73在兩者 二”為基礎輸 位準信號時輸出高位準㈣，在㈣2的任思一個為高 準信號時輸出低位準信均為低位 ，、媸而5，在上述檢測窗的 322708 35 201203833 開始位置處比較器41的輸出已反轉時，〇R閘73輸出高位 準信號。在上述檢測窗的開始位置處比較器41的輸出尚未 反轉時，在上述檢測窗内，在邊緣檢測部42的輸出信號中 邊緣立起時輸出高位準信號。 第19圖是用於說明使用檢測窗信號丨的零交又檢測部 40(未使用檢測窗開始信號）的動作圖。第19圖（幻表示在 檢測窗内發生了感應電壓的零交叉時線圈L1的兩端電壓 以及邊緣信號的推移，第19圖（b)表示在檢測窗内未發生 感應電壓的零交叉時（驅動頻率〈諧振頻率）線圈L1的兩端 電壓以及邊緣信號的推移，第19圖（c)表示在檢測窗内未 發生感應電壓的零交又時（驅動頻率〉諧振頻率）線圈的 兩端電壓以及邊緣信號的推移。 在使用檢測窗信號1的零交叉檢測部4〇(未使用檢測 窗開始信號）中，輸出控制部44僅由第18圖所示的第2AND 閘72構成。檢測窗信號1和邊緣檢測部42的輸出信號被 輸入到此第2AND閘72。 在第19圖（a)中，由於在由檢測窗信號1設定的檢測 窗内發生感應電壓的零交叉，因此在發生該零交叉的位 置’邊緣立起於邊緣信號中。此外，由於設定有該檢測窗， 故在發生再生電壓的零交又的位置，在該邊緣信號中邊緣 未立起。 在第19圖（b)中，表示了線性振動馬達200的諧振頻 率比上述驅動信號的頻率高且其差較大的狀態。因此，應 生成上述感應電壓的零交又的線性振動馬達200的停止狀 36 322708 201203833 態（即，位於振動範圍的§極側最大到達地點或者N極侧最 大到達地點）未產生在上述檢測窗内。在進入上述檢測窗的 時間點，該停止狀態已終止。在此情況下，在使用檢測窗 信號1的零交叉檢測部4〇(未使用檢測窗開始信號）中，在 邊緣信號中邊緣未立起（參照橢圓區域的部分）。 在第19圖（c)中，表示了線性振動馬達2〇〇的諧振頻 率比上述驅動信號的頻率低且其差較大的狀態。因此，應 生成上述感應電壓的零交叉的線性振動馬達2〇〇的停止狀 態不產生在上述檢測窗内。在離開上述檢測窗後此停止狀 態才產生。在此情況下，在使用檢測窗信號1的零交叉檢 測部40(未使用檢測窗開始信號）中，在邊緣信號中邊緣不 立起（參照橢圓區域的部分）。 第20圖是用於說明使用檢測窗信號2以及檢測窗開始 信號的零交又檢測部40的動作圖。第20圖（a)表示在檢測 窗内未發生感應電壓的零交叉時（驅動頻率〈諧振頻率）線 圈L1的兩端電壓以及邊緣信號的推移，第2〇圖（b)表示在 檢測窗内未發生感應電壓的零交叉時（驅動頻率〉諧振頻率） 線圈L1的兩端電壓以及邊緣信號的推移。 在使用檢測窗信號2以及檢測窗開始信號的零交又檢 測部40中’係利用第18圖所示的輸出控制部44。第20 圖（a)所示的線圈L1兩端電壓的推移與第19圖（b)所示的 線圈L1兩端電壓的推移相同。第20圖（b)所示的線圈L1 兩端電壓的推移與第19圖（c)所示的線圈L1兩端電壓的推 移相同。 37 322708 201203833 在第20圖（a)中*於第18圖所示的第⑽以 及㈣73的作用’在檢測窗的開始位置 在 邊緣信號中。在第20圖⑹中，由於將檢測窗的I土位置 延伸的作用’在正電流通電開始位置，邊緣立起在邊緣信 號中。 如以上說明上述，料設定上述檢㈣，當適應性地 控制驅動#號的周期寬度，使得線性振動馬達的固有振動 數與驅動信號的頻率一致時，能夠提高線圈u中產生的感 應電壓的零交叉的檢測精度。亦即，能夠抑制錯誤地檢測 驅動電壓和再生電流的零交叉。 在設疋了檢測由的情況下，若線性振動馬達2 〇 〇的譜 振頻率與驅動信號的頻率之間產生大幅偏離，則有時感應 電壓的零交叉會從檢測窗脫離。在本實施形態中，藉由在 檢測窗的開始位置附近或者終止位置附近使暫時的邊緣立 起，從而能夠不中途間斷地持續上述驅動信號的周期寬度 的適應性控制。即使線性振動馬達2〇〇的諧振頻率與驅動 信號的頻率偏離較大，也能夠藉由此暫時的邊緣使兩者逐 漸接近》 如此，藉由經常執行適應性控制，使得線性振動馬達 200的譜振頻率與驅動信號的頻率一致，即使驅動控制電 路1〇〇内的生成基本時脈的内置振盪子的精度下降，也不 需要調節（tri丽ing)内置振盪子的頻率，從而極大地有助 於驅動IC(驅動控制電路1〇〇)的製造成本的降低。 另外’作為在檢測窗的終止位置附近立起的暫時性邊 322708 38 201203833 緣，通過利用非通電期間後續的通電期間的起動，能夠簡 化信號控制。不需要❹上驗職開純置信號等檢測 窗信號以外的信號。 以上，以實施形態為基礎說明了本發明。此實施形態 僅是例示性，本領域的技術人員應理解；在這些各構成因 素與各處理過程的組合中可能產生各種變形例，該等變形 例也在本發明的範圍中。 上述第2上升控制也能夠藉由不包含非通電期間的驅 動信號，而應用於驅動線性振動馬達2〇〇的驅動控制電 路。該驅動信蚊在正電流通電期間與貞電流通電期間之 間不夾介非通電期間的情況下交替設定的信號。亦即，上 述第2上升控制也㈣應驗錢行上_動信號的周期 寬度的適應性㈣的驅動㈣電^上述停止控制同樣也 可以藉由不包含非通電期間的驅動信號，應用於驅動線性 振動馬達2GG的鶴㈣電路。亦即，也㈣應用於不執 行上述驅動信號的周期寬度的適應性控制的驅動控制電 路0 第21圖是表示第！圖所示的線性振動馬達2〇〇的驅動 控制電路100的結構變形例i的圖。帛21圖所示的驅動信 號生成部10的結構與第i圖所示的驅動信號生成部1〇相 比’省略了差分計算電路16、第3鎖存電路17、加法電路 18以及第4鎖存電路19。 取而代之的是，在主計數器12中採科數器可計數到 比對應上述驅動錄-職的計數值紅的賴值。若沿 322708 39 201203833 用上述例子進行說明，則使用250進制計數器或300進制 計數器，而不是200進制計數器。第2鎖存電路15依次鎖 存從主計數器12供給的計數值，並在零交叉檢測部4〇檢 測出零交叉的位置將鎖存的計數值輸出到第1鎖存電路 1卜 根據變形例1，由於主計數器12能夠計數到比對應上 述驅動信號一周期的計數值還大的計數值，故第2鎖存電 路15能夠將鎖存的計數值直接作為新的計數終止值來使 用。因此’能夠省略差分計算電路16、第3鎖存電路17、 加法電路18以及第4鎖存電路19而能夠簡化電路結構。 第22圖是表示第1圖所示的線性振動馬達2〇〇的驅動 控制電路100的結構變形例2的圖。在第22圖所示的零交 叉檢測部40中，不用比較器41，而使用類比/數位變換器 41a。類比/數位變換器41a將感應電壓檢測部3〇(在第22 圖的例子中為差動放大器）的輸出類比信號變換成數位信 號。邊緣檢測部42根據類比/數位變換器41a的輸出數位 信號’生成表示檢測出上述零交叉的位置的數位值，並輸 出到第2鎖存電路15。 例如，在不帶偏差地設計該差動放大器的情況下，邊 緣檢測部42係在類比/數位變換器41 a的輪出數位值為零 的時間點’將高位準信號輸出到第2鎖存電路15。 根據變形例2,由於從檢測感應電壓的零交叉的相位 (phase)來進行數位處理，故能夠使零交又時刻的檢測精度 提高。 322708 40 201203833 第23圖是表示第丨圖所示的線性振動馬達2〇〇的驅動 控制電路100的結構變形例3的圖示。變形例3是組合了 變形例1和變形例2之後的結構。 【圖式簡單說明】 第1圖是表示本發明實施形態的線性振動馬達的驅動 控制電路的結構圖。 第2圖是表示驅動部、感應電壓檢測部以及比較器的 結構例圖。 第3圖是表示實施形態的驅動控制電路的動作例的時 序圖。 第4圖是表示邊緣信號、第i時脈信號、第2時脈信 號以及第3時脈信號的一例的時序圖。 第5圖是表示解碼器的結構例圖。 第6圖是表示驅動信號的一周期的波形圖。 第7圖（a)至（c)是驅動信號的通電期間寬度的控制的 說明圖。 第8圖是驅動信號的相位控制的說明圖。 第9圖是表示追加了上升镡制功能的解碼器的結構例 圖。 第10圖是第1上升控制的說明圖。第10圖^)是表示 不執行第1上升控制時的線圈驅動電壓以及線性振動馬達 的振動的推移圖，第10圖（b)是表示執行了第i上升控制 時的線圈驅動電壓以及線性振動馬達的振動的推移圖。 第11圖是第2上升控制的說明圖。第u圖（3)是表示 322708 41 201203833 不執行第2上升控制時的線圈驅動電壓的推移圖，第11圖 (b)是表示執行了第2上升控制時的線圈驅動電壓的推移 圖。 第12圖是表示追加了停止控制功能的解碼器的結構 例圖。 第13圖是上述停止控制的基本概念的說明圖。第13 圖（a)是表示不執行停止控制時的線圈驅動電壓的推移 圖，第13圖（b)是表示執行了停止控制時的線圈驅動電壓 的推移圖，第13圖（c)是表示由PWM信號執行停止控制時 的線圈驅動電壓的推移圖。 第14圖是在上述停止控制中反相位的驅動信號的周 期次數固定的例子說明圖。第14圖（a)是表示驅動時的驅 動信號的周期次數較多時線圈驅動電壓以及線性振動馬達 的振動的推移圖，第14圖（b)是表示驅動時的驅動信號的 周期次數較少時線圈驅動電壓以及線性振動馬達的振動的 推移圖。 第15圖是在上述停止控制中反相位的驅動信號的周 期次數為可變的例子說明圖。第15圖（a)是表示驅動時的 驅動信號的周期次數較多時線圈驅動電壓以及線性振動馬 達的振動的推移圖，第15圖（b)是表示驅動時的驅動信號 的周期次數較少時線圈驅動電壓以及線性振動馬達的振動 的推移圖。 第16圖是表示具有檢測窗設定功能的零交叉檢測部 的結構圖。 42 322708 201203833 第17圖是檢測窗信號1、檢測窗信號2以及檢測窗開 始信號的說明圖。 第18圖是表示輸出控制部的結構例圖。 第19圖是使用檢測窗信號1的零交叉檢測部（未使用 檢测由開始彳§號）的動作說明圖。第19圖（a)是表示在檢測 窗内產生感應電壓的零交又時線圈的兩端電壓以及邊緣信 號的推移圖，第19圖（b)是表示在檢測窗内未產生感應電 壓的零交叉時（驅動頻率〈諧振頻率）線圈的兩端電壓以及 邊緣信號的推移圖，第19圖（c)是表示在檢測窗内未產生 感應電壓的零交又時（驅動頻率 >諧振頻率）線圈的兩端電 壓以及邊緣信號的推移圖。 第2 0圖是使用檢測窗信號£以及檢測窗開始信號的零 交叉檢測部的動作說明圖。第2〇圖（a)是表示在檢測窗内 未產生感應電壓的零交叉時（驅動頻率〈諧振頻率）線圈的 兩端電壓以及邊緣信號的推移圖，第2〇圖…）是表示在檢 測窗内未產生感應電壓的零交又時（驅動頻率〉諧振頻率） 線圈的兩端電壓以及邊緣信號的推移圖。 第21圖是表示帛！圖所示的線性振動馬達的驅動控制 電路的結構變形例1的圖。 第22圖是表示第！圖所示的線性振動馬達的驅動控制 電路的結構變形例2的圖。 第23圖是表示第丨圖所示的線性振動馬達的驅動控制 電路的結構變形例3的圖。 【主要元件符號說明】 322708 43 201203833 10 驅動信號生成部 11 第1鎖存電路 12 主計數器 13 迴圈計數器 14 解碼器 15 第2鎖存電路 16 差分計算電路 17 第3鎖存電路 18 加法電路 19 第4鎖存電路 20 驅動部 30 感應電壓檢測部 40 零交叉檢測部 41 比較器 41a 類比/數位變換器 42 邊緣檢測部 43 檢測窗設定部 44 輸出控制部 51 驅動寬度計算部 52 正驅動中心值計算部 53 負驅動中心值計算部 54 正側減法部 55 正侧加法部 56 負側減法部 57 負側加法部 58 正驅動信號生成部 59 負驅動信號生成部 60 上升控制部 61 停止控制部 71 第1 AND閘 72 第2AND-閘 73 OR閘 100 驅動控制電路 200 線性振動馬達 210 定子 211 芯件 220 振子 221 永久磁鐵 222a 、222b彈簧 223 框架 — LI 線圈 44 322708201203833 VI. Description of the Invention: [Technical Field] The present invention relates to a drive control circuit for driving control of a linear vibration motor in which a vibrator is linearly reciprocated with respect to a stator. [Prior Art] Conventionally, a linear vibration motor has been used in a specific application such as an electric trowel, but in recent years, its use has been expanding. For example, an element that generates a vibration that is returned to the user when the touch panel is pressed is generated. With the expansion of this haptics use, shipments of linear vibration motors are expected to increase in the future. The linear vibration motor is preferably driven at a frequency as close as possible to the natural vibration number (hereinafter, also referred to as a resonance frequency), and the strongest vibration is generated when the resonance frequency coincides with the drive frequency. [Problem to be Solved by the Invention] Since the natural vibration number of the linear vibration motor is determined by the mass of the vibrator and the spring constant, the natural vibration between the products is known. There will be deviations in the number. Therefore, in a conventional method of uniformly setting a fixed driving frequency to a driving circuit of a linear vibration motor, in the product, the natural vibration number of the motor and the driving frequency are also largely deviated, which causes a decrease in yield. Further, even if the number of natural vibrations of the first motor coincides with the drive frequency, the two may deviate due to the change over time, and the vibration may be weak. The present invention has been made in view of such circumstances, and an object thereof is to provide a technique for driving a frequency as close as possible to the natural vibration number regardless of the state of the linear vibration motor. (Means for Solving the Problem) In the drive control circuit of the linear vibration motor of the present invention, the linear vibration motor has a stator and a vibrator, and at least one of them is composed of an electromagnet, and a drive current is supplied to the coil of the electromagnet. a vibration of the vibrator relative to the stator, the drive control circuit of the linear vibration motor comprising: a drive signal generating unit that generates a drive signal for causing the positive current and the negative current to alternately flow to the coil during the non-energization period; a drive current corresponding to the drive signal generated by the drive signal generating unit is supplied to the coil, and the induced voltage detecting unit detects the induced voltage generated in the coil during the non-energization period; and the zero-cross detecting unit detects A zero crossing of the induced voltage detected by the induced voltage detecting unit is generated. The drive signal generating unit estimates the natural vibration number of the linear vibration motor based on the detected position of the zero cross, so that the frequency of the drive signal approaches the natural vibration number. Further, any combination of the above constituent elements and an innovative design obtained by converting the expression of the present invention between methods, apparatuses, systems, and the like are also effective as the form of the present invention. According to the present invention, regardless of the state of the linear vibration motor, it is possible to drive at a frequency as close as possible to the natural vibration number. [Embodiment] (Basic configuration) Fig. 1 is a configuration diagram showing a drive control circuit 100 of a linear vibration motor 200 according to an embodiment of the present invention. First, the linear vibration motor 200 has 5 322708 201203833 having a stator 210 and a vibrator 220, at least one of which is composed of an electromagnet. In the present embodiment, the stator 21 is made of an electromagnet. The stator 21 is formed by winding the coil L1 around the core member 211 of the magnetic material. When the coil L1 is energized, it acts as a magnet. The vibrator 220 includes permanent magnets 221'. Both ends (the s-side and the N-pole side) of the permanent magnet 221 are fixed to the frame 223 via springs 222a and 222b, respectively. The stator 210 and the vibrator 220 are arranged side by side with a predetermined gap therebetween. Further, contrary to the example of Fig. 1, the vibrator 220 may be composed of an electromagnet, and the stator 210 may be composed of a permanent magnet. The drive control circuit 100 supplies a drive current ' to the coil L1' to reciprocate the vibrator 220 linearly with respect to the stator 210. The drive control circuit 100 includes a drive signal generating unit 10, a drive unit 20, an induced voltage detecting unit 30, and a zero-crossing detecting unit 40. The drive signal generating unit 10 generates a drive signal for alternately flowing the positive current and the negative current to the coil L1 during the non-energization period. The drive unit 20 generates a drive current corresponding to the drive signal generated by the drive signal generating unit 10, and supplies it to the coil L1. The induced voltage detecting unit 30 is connected to both ends of the coil L1 to detect a potential difference between both ends of the coil L1. Mainly during the non-energization period, the induced voltage generated in the coil L1 is detected. The zero-cross detecting unit 40 detects the zero-crossing of the induced voltage detected by the induced voltage detecting unit 30. The drive signal generation unit 10 estimates the natural vibration number of the linear vibration motor 200 based on the detection position of the zero-crossing of the induced voltage detected by the zero-cross detection unit 40, and makes the frequency of the drive signal as close as possible to the natural vibration number. . That is, the frequency of the drive signal is adaptively changed, 322708 6 201203833 to match the frequency of the drive signal with the natural number of vibrations. More specifically, the drive signal generating unit calculates a difference between the end position of one cycle of the drive signal and the detection position of the zero crossing corresponding to the end position, and adds the difference to the cycle width of the current drive signal. Thereby, the period width of the above drive signal is adaptively controlled. When a period of the above driving signal is formed by a normal phase (zero-positive voltage-zero-negative voltage-zero), the detection position of the zero-crossing corresponding to the above-mentioned termination position is the negative voltage zero crossing from the induced voltage to The position of the positive voltage. Conversely, when a period of the above driving signal is formed by an inverse phase (zero-negative voltage-zero-positive voltage-zero), the detection position corresponding to the zero-crossing of the termination position is positive from the induced voltage The voltage zero crosses to the position of the negative voltage. Hereinafter, the configuration of the drive control circuit 100 will be more specifically described. First, the configuration of the drive unit 20, the induced voltage detecting unit 30, and the zero-cross detecting unit 40 will be described. The zero-cross detecting unit 40 includes a comparator 41 and an edge detecting unit 42. The comparator 41 compares the induced voltage detected by the induced voltage detecting portion 30 with a reference voltage for detecting zero crossing. At the point in time when the induced voltage crosses the reference voltage, the comparator 41 inverts the output. For example, reversing from a low level signal to a high level signal. The edge detecting unit 42 detects a position at which the output of the comparator 41 is inverted as an edge. Fig. 2 is a view showing an example of the configuration of the drive unit 20, the induced voltage detecting unit 30, and the comparator 41. In the example shown in Fig. 2, the drive unit 20 is constituted by a bridge circuit. * The inductive voltage detecting unit 30 is constituted by a differential amplifier circuit. 7 322708 201203833 The bridge circuit includes a first transistor M1, a second transistor M2, a third transistor M3, and a fourth transistor M4. Further, in Fig. 2, the coil L1 of the linear vibration motor 200 is also placed in the frame of the driving portion 20 for convenience of explanation. The first transistor M1 and the third transistor M3 are composed! The series circuit and the second series circuit including the second transistor M2 and the fourth transistor M4 are connected between the power supply potential vdd and the ground potential, respectively. The connection point between the first transistor M1 and the third transistor M3 (hereinafter referred to as point A) and the connection point between the second transistor M2 and the fourth transistor M4 (hereinafter referred to as point B) are connected. There is a coil L1. In Fig. 2, the first transistor M1 and the second transistor M2 are composed of a P-channel MOSFET, and a first diode D1 and a second diode D2 are connected between the source and the drain of each. As a body di ode. The third transistor M3 and the fourth transistor M4 are composed of an N-channel MOSFET 'between the source and the gate of each of the three diodes j) 3 and the fourth diode D4 as the body diode body. The drive signal generating unit 1 (more strictly, a decoder 14 to be described later) inputs the above-described driving to the gates of the first transistor M1, the second transistor M2, the third transistor M3, and the fourth transistor M4. signal. If the first transistor M1 and the fourth transistor M4 are turned on by the driving signal control, and the second transistor M2 and the third transistor M3 are turned off, a positive current flows to the coil L1; When the transistor M1 and the fourth transistor M4 are turned off, and the second transistor M2 and the third transistor M3 are turned on, a negative current flows to the coil L1. The differential amplifier circuit includes an operational amplifier 〇P1 and a first resistor R.卜 8 322708 201203833 The second electric_, the third electric resistance R3 and the fourth electric resistance R4. The inverting input terminal of the operational amplifier 0P1 is connected to (4) via the first resistor R1, and the non-inverting input terminal is connected via the second resistor R2_A. The inverting input terminal and the output terminal of the operational amplifier 0P1 are connected via the third resistor R3. The reference voltage Vref is biased (咐_voltage) is applied to the non-inverting input terminal of the operational amplifier OP1 via the fourth resistor R4. The resistance values of the ith resistor R1 and the second resistor R2 are set to the same value, and the third value is set. The resistance values of the resistor R3 and the fourth resistor R4 are the same value. Under this condition, the A ratio of the difference-reduction circuit is R3/R1. For example, the resistance values of the first resistor R1 and the second resistor R2 are set to be edited. When the resistance value of the third resistor R3 and the fourth resistor R4 is 2_, the voltage across the coil u (the voltage between the ABs is doubled. The reverse input of the comparator 4U is composed of an open-loop operational amplifier). The reference voltage Vref is applied to the terminal. The non-inverting input terminal of the comparator 41 is connected to the output terminal of the nose amplifier OP1, and the output voltage of the operational amplifier OP1 is applied to the non-inverting input terminal. When the reference voltage Verf is applied as a bias voltage (for example, 1 / 2 Vdd) to the differential amplifier circuit described above, in order to match the range of the operational amplifier OP1 and the comparator 41, the reference voltage Vref is used as a comparison. The reference voltage of the device 41. Further, when a bias voltage is not applied to the above differential amplifying circuit, the ground voltage is used as the reference voltage of the comparator 41. By the above-described differential amplifier circuit amplifying the voltage across the coil L1 (voltage between A and B) and then inputting it to the comparator 41, the detection accuracy of the zero crossing of the induced voltage generated in the coil L1 can be improved. 9 322708 201203833 Fig. 3 is a timing chart showing an operation example of the drive control circuit 100 according to the embodiment. This operation example is an example of a single-phase full-wave drive linear vibration motor 200. At this time, set the non-energization period. The non-energization period is set before and after each of the positive current energization period and the negative current energization period. That is, in the full cycle, the first half cycle is composed of a non-energization period, a positive current energization period, and a non-energization period, and the second half period is composed of a non-energization period, a negative current period, and a non-energization period. In the following example, in a half cycle of 180°, 40° is allocated to the non-energized period, 100° is allocated to the positive (negative) current during energization, and 40° is allocated to the non-energized period. Therefore, 5/9 in one cycle is allocated to the power-on period, and 4/9 is assigned to the non-power-on period. Hereinafter, in the present specification, the driving method according to this ratio is referred to as 100-degree energization. In Fig. 3, in the on-1 state of the bridge circuit (M1, M4 are turned on, M2, M3 are turned off), a positive current flows to the coil L1. In the off state (M1 to Μ4 off) of the above-described bridge circuit, the drive current does not flow to the coil L1. In the on-state of the above-mentioned bridge circuit (M Bu M4 off, M2, M3 on), a negative current flows to the coil L1. In a state where a positive current flows into the coil L1, the stator 210 is excited to be a drain, and due to this magnetic force, the vibrator 220 receives a force toward the S pole side of the permanent magnet 221. Due to this force, the vibrator 220 moves against the spring 222a toward the S pole side of the permanent magnet 221, and moves to the contraction limit of the spring 222a. In a state where no drive current flows in the coil L1, the stator 210 is not excited, and no magnetic force is generated. The vibrator 220 moves to the center position due to the restoring force of the spring 222a. In a state where a negative current flows through the coil L1, the stator 210 is excited to the S pole, and due to this magnetic force, the vibrator 220 receives a force toward the N pole side of the permanent magnet 221 10 032708 201203833. Due to this force, the vibrator 22 〇 moves against the spring 222b toward the N pole side of the permanent magnet 221, and moves to the contraction limit of the spring 222b. The drive signal generating unit 10 controls the H-bridge circuit in a state in which the drive signal generating unit 10 is turned on, the on-state, the off-state, and the on-off state, and the linear vibration motor 200 can reciprocate. If the above-mentioned bridge circuit is turned on < The state shifts to the OFF state, and all of the first transistor M1 to the fourth transistor μ are switched off, and the regenerative current flows through the body-pole. The same applies when the above-mentioned bridge circuit is switched from the conduction state to the OFF state. By effectively utilizing this regenerative current, the monthly efficiency is sufficient to reduce the power consumption of the drive control circuit. The regenerative current is maneuvered in the same direction as the current flowing through the coil U. When the regenerative current flows, the induced current induced by the movement of the vibrator 220 flows into the coil L1. When the vibrator 220 is stopped, the induced current does not flow. The state in which the vibrator 220 is stopped is generated at the instant when the vibrator 220 reaches both ends of the vibration range of the vibrator 22A. The induced voltage detecting unit 30 can estimate the position of the vibrator 22() by monitoring the counter electromotive voltage generated in the non-energized period coil η. The state in which the back power = potential voltage is zero indicates that the vibrator 220 is at a stop (i.e., at the S-pole side maximum arrival point or the N-pole side maximum arrival point of the vibration range). Therefore, the zero-crossing detection unit 40 detects the time at which the voltage across the coil u (a_b between the two) is zero-crossed (except for the zero-crossing caused by the drive current and the regenerative current), and measures the detected zero-crossing. In the period of time, the natural vibration number of the linear vibration motor 200 can be obtained, and the period of the continuous zero crossing indicates the half-vibration period width of the linear vibration motor. One of the 322708 is skipped. 201203833 The zero-crossing period indicates the full vibration period. width. In the present embodiment, the zero-cross detecting unit 40 detects only the time point at which the voltage across the coil L1 (the voltage between A and B) crosses from negative zero to positive during the non-energization period. In this case, the comparator 41 shown in FIG. 2 is set to output a low level signal while the output voltage of the operational amplifier OP1 is lower than the reference voltage Vref; if the output voltage of the operational amplifier OP1 is higher than the reference voltage Vref At the time, the high level signal is output. The drive signal generating unit 10 adjusts the cycle width of the next drive signal by the cycle width corresponding to the measured natural vibration number of the linear vibration motor 200. By repeating this measurement and adjustment, the drive control circuit 100 can continuously drive the linear vibration motor 200 at its resonant frequency or at a frequency in its vicinity. Returning to Fig. 1, the drive signal generating unit 10 will be described more specifically. The drive signal generation unit 10 includes a first latch circuit 11, a main counter 12, a loop counter 13, a decoder 14, a second latch circuit 15, a difference calculation circuit 16, and a third latch circuit. 17. Adding circuit 18 and fourth latch circuit 19. The first latch circuit 11 latches the count end value corresponding to the end position of one cycle of the drive signal, and outputs it to the main counter 12 and the decoder 14 at the timing indicated by the third clock signal CLK3. Further, it can also be output to the difference calculation circuit 16. In the first latch circuit 11, the initial value of the count termination value is set from a register or the like (not shown) when the drive of the linear vibration motor 200 is started. After the start of driving, the value input from the fourth latch circuit 19 becomes the above-described count end value. 12 322708 201203833 The value = 12 is set from the first latch circuit ^ to the above-mentioned count termination ΐ: the start value is counted until the count end value. When the Η#, t(4) tree stop value is set to 199, the main counter 12 becomes the slave counter. The master count thin 12 is ordered to repeat the count up counter, the 14th and the second latches, and the number of times the second 13 is counted each time the main counter 12 counts the loop end. And the count loop of the main counter 12 is saved from the initial value of the count; the count loop is the end value of the corresponding one of the feed counters of the main counter 12. —Second count loop The decompressor 14°/ ', so the number of count loops corresponds to the number of drive cycles. The above-mentioned count final value phase 11 ^ ^ 12 mentions (four) the count value, and generates a drive signal with a detailed structure of the width, +, and % cycle width. The count signal provided by the decoder 12 12, the second latch circuit 15 sequentially latches the position locked by the position of the main counter & ' and will detect the zero-crossing zero-crossing bit at the zero-crossing detection unit 40] The 2 value is output to the difference calculation circuit 16. This detection is known. When the edge signal input from the detecting unit 42 is detected to be generated, the second ^ position is always ideally at the same time point. The output of the differential meter I sub-circuit /5 is always the same count value. The value is equal to the current count of 16 counts of the second latch circuit 15 input 1 the difference between the latch circuit u and the stop value. Figure 1 depicts an example of the calculation of the termination value from the first calculation circuit: In. Further, the difference is a structure for storing the current count end value from the second, and the latch circuit 19 can also provide a structure for counting the end value. 322708 13 201203833 When the count value of the position where the zero crossing is detected (= the count value input from the second latch circuit 15) is smaller than the current count end value, the difference calculation circuit 16 subtracts the latter from the former. For example, the count value of the position where the zero crossing is detected is 197, and when the count end value is 199, the difference calculation circuit 16 outputs -2. When the count value of the position at which the zero crossing is detected is larger than the current count end value, the count value input from the second latch circuit 15 is a value relative to the increase portion of the current count end value. In this case, the difference calculation circuit 16 directly outputs the count value input from the second latch circuit 15. For example, the original count value of the position where the zero crossing is detected is 201, and when the count end value is 199, the count value input from the second latch circuit 15 is 2, and the difference calculation circuit 16 directly outputs 2. Since the count value is reset at 199, the count value input from the second latch circuit 15 is not 201 but 2. The third latch circuit 17 latches the difference input from the difference calculation circuit 16, and outputs the difference to the addition circuit 18 at the timing indicated by the first clock signal CLK1. The adding circuit 18 adds the difference input from the third latch circuit 17 to the current count end value input from the fourth latch circuit 19. The fourth latch circuit 19 latches the value input from the adding circuit 18, and outputs it to the first latch circuit 11 at the timing indicated by the second clock signal CLK2. In the fourth latch circuit 19, when the driving of the linear vibration motor 200 is started, the initial value of the count end value is also set from a register or the like (not shown). The value generated by the adding circuit 18 is set as a new count end value, and is set to the main counter 12 14 322708 201203833 and the decoder 14 via the fourth latch circuit 19 and the first latch circuit 11. Therefore, the count end value of the detection position in which the previous zero crossing is reflected is always set in the main counter 12 and the decoder 14. Fig. 4 is a timing chart showing an example of an edge signal, a first clock signal CLK1, a second clock signal CLK2, and a third clock signal CLK3. The edge signal is set from the edge detecting unit 42 to the second latch circuit 15. The first clock signal CLK1 is a signal that delays the edge signal by half a clock. The delay of this half clock is taken into account for the arithmetic processing by the difference calculation circuit 16. The second clock signal CLK2 is a signal for delaying the second clock signal cui by half a clock. The delay of this half clock is taken into consideration for the arithmetic processing to be performed by the addition circuit 18. The third clock signal CLK3 is a signal for delaying the second clock signal (10) by several clocks. The delay of the several clocks is to suppress the count end value of the current drive period from being changed before the end of the count of the current drive period. For example, if the first latch circuit u' is not set, then in the current driving cycle, when the zero crossing is detected before the final position, the new count termination value reflecting its zero crossing and position may be from now on. The drive cycle is used instead of being used from the next drive cycle. In this case, since the energization period is determined based on the count end value before the update, the ratio between the energization period and the non-energization period cannot be maintained. In the present embodiment, the energization of 100 degrees is not maintained. By providing the ith latch circuit 11 between the fourth latch circuit 19 and the main counter 12, it is possible to update the current count end value in the main counter 12 setting to a new count that reflects the zero-cross position. The time point of the value is delayed. 15 322708 201203833 (Decoder Configuration) FIG. 5 is a diagram showing an example of the configuration of the decoder 14. The decoding $i4 determines a count width corresponding to the energization (four) of the drive signal based on the value obtained by multiplying the number of (four) numbers by the count end value, and the coefficient is used to fix the ratio of the period of the energization period to the period of the drive signal. . The positive current energization period and the negative current energization period are included in one cycle of the above-described drive signal as described above. Therefore, in the case where the above (10) degree is energized, the ratio of each power supply to one cycle with respect to the above-described drive signal is 1 〇〇. /36〇. (. 〇· 28). Further, the ratio of the half period of each energization period to one period of the above-described drive signal is 50. /360. (%〇· 14). Further, the decoder 14 multiplies the above-described count end value by a value obtained by determining the coefficient of the center position of the energization period of the above-described dynamic signal, and determines the start position and termination of the energization period with the above-described drive 彳s number. The count value corresponding to the position. As described above, the period of the above-mentioned job signal is formed by the positive current energization period in which the non-energization period is set before and after, and the negative current energization period in which the non-energization period is set before and after. Here, the lengths of the positive current power-on periods and the lengths of the 流通f power-discharges (4) are set to be equal, and the lengths of the non-energization periods are all set to be equal. Therefore, the coefficient for determining the center position of the positive current during the energization of the above drive signal is set to 0. 25. The coefficient for determining the center position of the negative current during the energization of the drive signal is set to 〇. . Further, when the phases of the above-mentioned drive signals are opposite, the coefficient for determining the center position during the energization of the negative current is set to 〇. 25, and the coefficient used to determine the center position during the energization of the positive current is set to 〇. 75. 322708 16 201203833 In this manner, the decoder 14 can calculate the count width corresponding to each energization period and the count value corresponding to the center position of each energization period. Further, by subtracting the value of half of the count width from the count value corresponding to the center position, the count value corresponding to the start position of each energization period can be calculated. Further, by adding a value of half of the count width to the count value corresponding to the center position, the count value corresponding to the end position of each energization period can be calculated. Hereinafter, a more specific description will be made. The decoder 14 includes a drive width calculation unit 51, a positive drive center value calculation unit 52, a negative drive center value calculation unit 53, a positive side subtraction unit 54, a positive side addition unit 55, a negative side subtraction unit 56, and a negative side addition unit 57. The positive drive signal generating unit 58 and the negative drive signal generating unit 59. The drive width calculating unit 51 stores the half period of each energization period (hereinafter also referred to as a drive period as appropriate) with respect to the ratio of the period of the drive signal as a coefficient (four). In the case of the above (10) degree energization, 0·14 is stored. The drive width calculating unit 51 is supplied with the count end value from the first latch circuit 丨1. The drive width calculating portion 51 multiplies the count end value by the coefficient. Thus, the count width corresponding to each drive period_half period can be calculated. . The positive drive center value calculation unit 52 stores coefficients for determining the center position of the positive current energization period (hereinafter also referred to as the positive drive period as appropriate) of the drive signal. In this embodiment, G is saved. 25. The positive drive command core value calculation unit 52 is supplied with the count end value from the i-th latch circuit u. The positive drive-to-value 4 calculation unit 52 multiplies the count end value ± by the coefficient. Thus, 322708 17 201203833 can calculate the count value corresponding to the center position of each positive drive period. The negative drive center value calculation unit 53 stores coefficients for determining the center position of the negative current energization period (hereinafter also referred to as a negative drive period as appropriate) of the drive signal. In this embodiment, save 0. 75. The negative drive center value calculation unit 53 is supplied with the count end value from the first latch circuit 11. The negative drive center value calculating portion 53 multiplies the count end value by the coefficient. In this way, the count value corresponding to the center position of each negative driving period can be calculated. The positive side subtraction unit 54 calculates the start of the positive drive period by subtracting the count width supplied from the drive width calculation unit 51 from the count value corresponding to the center position of the positive drive period supplied from the positive drive center value calculation unit 52. The count value corresponding to the position. The positive side adder 55 calculates the count width supplied by the drive width calculating unit 51 from the count value corresponding to the center position of the positive drive period supplied from the positive drive center value calculation unit 52, and calculates the positive drive period. The count value corresponding to the end position. The negative side subtraction unit 56 calculates the start of the negative drive period by subtracting the count width supplied from the drive width calculation unit 51 from the count value corresponding to the center position of the negative drive period supplied from the negative drive center value calculation unit 53. The count value corresponding to the position. The negative side addition unit 57 calculates the termination of the negative driving period by adding the count width supplied from the driving width calculating unit 51 to the count value corresponding to the center position of the negative driving period supplied from the negative driving center value calculating unit 53. The count value corresponding to the position. The positive drive signal generating unit 58 supplies the count value as the synchronization clock from the main counter 12, and supplies the count value corresponding to the start position of the positive drive period from the positive side subtraction unit 54, and supplies it from the positive side adder 55. Positive drive 18 322708 201203833 = The count value should be. The positive drive signal generating unit 58 outputs a meaningful signal (for example, a high level) from the start position corresponding to the positive drive period to the count value corresponding to the end position of the positive drive period in accordance with the number; Signal) as a positive drive signal. During other periods, a non-significant signal (e.g., a low level reference) is output. Further, the positive drive signal generating unit 58 can generate the positive drive signal with the PWM number of the set duty ratio. The positive drive signal generated by the positive drive signal generating unit 58 is input to the drive unit 2, specifically to the gates of the first transistor M1 and the fourth transistor M4. Further, an inverter (not shown) is provided in a section of the first transistor M1, and the positive drive signal is inverted in phase and input to the gate of the first transistor M1. The negative drive signal generation unit 59 is supplied with the count value as the synchronization clock from the main counter 12, and the count value corresponding to the start position of the negative drive period is supplied from the negative side subtraction unit 56, and is supplied from the negative side addition unit 57. The count value corresponding to the end position of the negative drive period. The negative drive signal generating unit 59 outputs a meaningful signal (for example, a high level) from the count value corresponding to the start position of the negative drive period to the count value corresponding to the end position of the negative drive period in accordance with the count value as the synchronization clock. Signal) as a negative drive signal. During other periods, a non-significant signal (e.g., a low level signal) is output. Further, the negative drive signal generating unit 59 can generate the negative drive signal with the PM signal of the set duty ratio. The negative drive signal generated by the negative drive signal generating portion 59 is input to the drive unit 20, specifically, to the gates of the second transistor M2 and the third transistor M3 of 19 322708 201203833. Further, an inverter (not shown) is provided in the front stage of the second transistor M2, and the negative drive signal is inverted in phase and input to the gate of the second transistor M2. Fig. 6 is a waveform diagram showing one cycle of the drive signal. In Fig. 6, the dot area indicates the positive driving period (front) and the negative driving period (back). The count value corresponding to the positive drive start value a is generated by the positive side subtraction unit 54, and the count value corresponding to the i drive center value b is generated by the positive drive center value calculation unit 52. The count value corresponding to the positive drive end value c is positive. The side addition unit 55 is generated. Similarly, the count value corresponding to the negative drive start value d is generated by the negative side subtraction unit 56. The count value corresponding to the negative drive center value e is generated by the negative drive center value calculation unit 53, and the count value corresponding to the negative drive end value f is generated. It is generated by the negative side adder 57. As shown in Fig. 5, by configuring the decoder 14, even if the period width is changed by the frequency change of the drive signal, the drive signal generating unit 1 can adjust the drive signal to maintain the energization period of the drive signal. The ratio to the period of non-energization. Further, even if the period width of the drive signal is changed, the drive signal generating unit 10 can adjust the drive signal to maintain the relative positional relationship of the signal phases during the energization period in one cycle. Fig. 7 is an explanatory diagram of control of the width of the energization period of the drive signal. Fig. 7(a) is a transition diagram showing a coil driving voltage in a driving state in a preset state. Fig. 7(b) is a diagram showing a coil driving voltage after the driving period is adjusted longer than a preset state (power-on period) (Fig. 7(c) is a transition diagram showing the coil drive voltage (the adjustment of the width of the energization period) after the drive period is adjusted longer than the preset state. 20 322708 201203833 In Fig. 7(a), the above 1 degree power is set. that is,! The ratio of the energization period to the non-energization period in one drive cycle is set to 5 ··4. In Fig. 7(b), an example in which the driving period is adjusted to be longer than the preset state and the power-on period width in the preset state is maintained is shown. In this case, there is a possibility that the driving force of the linear vibration motor 2A is lowered, and the vibration of the linear vibration motor 2A is weak. In Fig. 7(c), after the drive period is adjusted longer than the preset state, control is performed so that the ratio of the energization period to the non-energization period in the one drive period is maintained. In the present embodiment, control is performed so that the above-described 100-degree energization is performed. This control is realized by the action of the drive width calculation unit 51 in the decoder 14. Here, an example in which the drive period is adjusted longer than the preset state has been described, but the example in which the adjustment is shorter than the preset state is also the same. When the drive period is adjusted to be shorter than the preset state and the energization period width in the pre-recorded state is maintained, the driving force to the linear vibration motor 2A increases, and the vibration of the linear vibration motor 200 becomes strong. In this regard, in the present embodiment, control is also performed so that the power supply is maintained at 100 degrees after the drive period is adjusted from the preset state to be short. Fig. 8 is a view for explaining phase control of a drive signal. Fig. 8 shows the transition of the voltages at both ends of the coil u after being adjusted to the resonance frequency of the linear vibration motor 2GG. Further, in order to simplify the explanation, the description of the regenerative current is omitted. The waveform table of the first stage * drives the state of the linear vibration motor 200 when the phase of the drive signal is optimum (four). The waveform of the second stage indicates the state in which the phase of the drive signal from the second cycle is 322708 201203833, and the state of the linear vibration motor 2〇〇 is driven in the state after the target position is delayed. In this state, the driving period is adjusted to be shorter than before, and after the adjustment, the start position and the end position of each power-on period are generated while maintaining the position before the adjustment. The waveform of the third stage indicates a state in which the phase of the drive signal is driven from the second cycle to the state in which the linear motor 2 is driven in the phase advance state. This state is a case where the driving cycle is maintained by the s week than before, and after the adjustment, the start position and the end position of each energization period are also generated while maintaining the position before the adjustment. That is, in the case where the start position and the end position are fixed during each energization period, if the drive period width is changed, the phase of the drive signal is delayed or advanced. On the other hand, in the present embodiment, when the drive period is changed, the phase of the drive signal can be optimally adjusted by adaptively adjusting the start position and the end position of each energization period. The adjustment of the start position and the end position is mainly realized by the action of the positive drive center value calculation unit 52 and the negative drive center value calculation unit 53 in the decoder 14. As described above, according to the drive control circuit 1 of the present embodiment, the period width of the next drive signal is adjusted by using the period width 'corresponding to the measured natural vibration number of the linear vibration motor 200, thereby linearly In which state the vibration motor 2 is in, it is possible to continuously drive at a frequency as close as possible to the natural vibration number. Therefore, it is possible to absorb the variation in the number of natural vibrations generated between the products of the linear vibration motor 2A, and it is possible to prevent a decrease in the yield at the time of mass production of the motor. Further, even if the springs 222a, 222b and the like change over time, the vibration is reduced by 22 322708 201203833 and the number of natural vibrations after the change with time. And the dynamic frequency is used for driving. In addition, when the natural vibration number and the period width of the drive vibration motor are adaptively controlled so that the linearity is sufficient to match the influence of the cycle width change, even if the cycle width of the drive signal is Change, borrow the least. Specifically, the width of the energization period and the period of the non-energization constant energization period in the sustain-cycle are maintained at the _force' of the secret vibration material. Therefore, the linear vibration motor 2QQ can be suppressed by the fluctuation of the driving force, and even if the period width of the driving signal is changed, the starting position and the ending position of the electric power are adjusted to the optimum position. - The relative positional relationship during the energization period in the cycle can also be reduced by =. That is, if the phase of the drive signal deviates, the vibrator 22'' and the supply position of the driving force are deviated, and the driving efficiency is lowered. At this point, the phase of the drive signal can be maintained at the optimum position: thereby maximizing the vibration with the same power consumption. (Rising Control) Hereinafter, the first rising control that can be added to the above-described drive control by the drive control circuit 100 of the present embodiment will be described. As shown in Fig. 6, one cycle of the above-described drive signal is formed by a positive current energization period in which the non-energization period is set before and after, and a negative current energization period in which the non-energization period is set before and after. Thereby, the zero crossing of the induced voltage can be detected with high precision as shown in Fig. 3, and the driving efficiency can be improved as shown in Fig. 8. 322708 23 201203833 Therefore, in principle, the non-energization period is also set before the positive current energization period of the first period of the above-described drive signal (in the case of the reverse phase is the negative current energization period). However, this non-energization period acts in the direction in which the rise time of the linear vibration motor 200 is delayed. Therefore, in order to improve this, the drive signal generating section 10 can perform the following rise control. In other words, after the driving of the linear vibration motor 200 is started, the drive signal generating unit 10' sets the non-energization period width of the drive signal at least before the first energization period to be higher than when the linear vibration motor 200 is stably operated. It is set that the width of the non-energization period before each energization period is short. For example, after the driving of the linear drive motor 200 is started, the drive signal generating unit 10 sets the width of the # energization period before the first energization period of the drive signal to zero. The energization period of the non-energization period width which is shorter than the width of the non-energization period before the energization period in the steady operation period may be set in the front, and only the first energization period may be set from the first energization period. To the nth (n is a natural number) power-on period. In the latter case, the non-energization period width to be set in front of each can be lengthened in accordance with the order from the first energization period to the n-th energization period. Further, before the energization period, the drive signal generating unit 1 can stop the period of the drive signal while the non-energization period width shorter than the non-energization period width before the energization period is set during the steady operation. Width adjustment processing. In this case, the zero-cross detection processing of the induced voltage by the induced voltage detecting unit 30 and the zero-crossing detecting unit 4 can be stopped. 322708 24 201203833 Next, the second rise control that can be added to the drive control performed by the drive control circuit 1A of the present embodiment will be described. As shown in Fig. 5, the drive signal generating unit 10 can generate a signal for each energization period with a PWM signal. Thereby, the driving ability can be adjusted in accordance with the performance of the linear vibration motor 2A. The premise of the second rise control is to generate a signal for each energization period by the PWM signal. The drive signal generating unit 1 sets the operation ratio of the pw M signal generated by the drive signal in at least the first energization period to be higher than when the linear vibration motor 200 is stably operated after the start of the drive of the linear vibration motor 2A. The duty ratio of the PWM signal generated during power-on is high. For example, after the driving of the linear vibration motor 200 is started, the drive signal generating unit 10 can set the duty ratio of the PWM signal generated by the drive signal in at least the first energization period to 1. The energization period of the PWM signal that generates a higher duty ratio than the operation ratio of the p medical signal generated during each energization period during the steady operation may be only the first energization period, or may be from the initial energization period to the mb. Number of power-up periods. In the latter case, the duty ratio of the PWM signal generated during each energization period can be lowered in accordance with the order from the first energization period to the mth energization period. Further, the drive signal generating unit 10 can stop the adjustment processing of the cycle width of the drive signal while the operation of the p-question No. 4 generated in each energization period is higher than the PWM signal of the higher duty ratio. In this case, the zero-cross detection processing of the induced voltage by the induced voltage detecting unit 3A and the zero-cross detecting unit 40 can be stopped. 25 322708 201203833 The 1st rise control and the 2nd rise control can be used individually or in combination. Hereinafter, a configuration example of the de-buffer 14 when at least one of the third rising control and the second rising control is employed will be described. Fig. 9 is a view showing an example of the configuration of the decoder 14 to which the rising control function is added. The decoder 14 shown in Fig. 9 has a configuration in which the rise control unit 60 is added to the decoder 14 shown in Fig. 5. When the first rising control is executed, the rising control unit 60 corrects the count value of the generating unit 59 from the main counter 12 to the positive drive signal generating unit 58 and the negative drive L. For example, when the non-energization period width before the energization period is set to be zero, the rise control unit 60 adds the count width corresponding to the # energization period width to be set before each energization period during the steady operation. The count value input by the main counter 12 is included. Thereby, the positive drive signal generating unit 58 and the negative drive signal generating unit 59 can omit the non-energizing period before each of the positive current power-on period and the negative current power-on period. Further, in the same process, the count initial value of the main counter 12 can be set to be the above-mentioned count width in the count initial value during the steady operation period while the non-energization period width to be set before the energization period is set to zero. value. In the present embodiment, the count initial value of the main counter 12 is set to the count value at the start of the above-described 1st power supply. This processing can be executed by other rising control units (not shown) other than the decoder 14. When the second rising control is executed, the rising control unit 6 turns the duty ratio of the pWM signal generated in at least the first energization period of the drive signal in the positive drive signal generating unit 58 and the negative drive signal generating unit. At this time, the ratio of the work generated by the power-on period during the steady-state operation is higher than the ratio of the work ratio of 26 322708 201203833. Fig. 10 is a view for explaining the i-th rise control. (4) The transition of the coil drive voltage and the vibration of the linear motor 200 when the i-th rise control is not performed is shown in FIG. 10 (6), which is the coil drive voltage when the first rise control is executed and Linear vibration motor 2 〇 vibration transition diagram. : In Fig. 10 (4) and Fig. 10 (8), an example in which the vibration of the second-stage vibration motor 2A of the drive signal reaches a desired level (i.e., the level at the time of steady operation) is described. In Fig. 1(b), the drive signal generation (4) 10 sets the width of the electrical period to be set to zero during the first energization period of the drive signal. Non-communication " Period (4) in Figure 10 (4) means not executing the first! In the period from the start of driving to the time when the vibration reaches the desired level during the ascent control, the period t2 in the (6) diagram indicates the period from the start of the drive to the vibration (four) to the fourth (4) water stop when the first rise control is executed. Comparing the latitude t1 with the period t2, it can be seen that the period t2 is short, and by performing the first rising control, the period from the start of the drive to the start of the oscillating axis is shortened. Fig. 11 is a view for explaining the second rise control. (4) is a transition diagram of the coil drive voltage when the second rise control is not performed. FIG. 11(5) is a transition diagram showing the coil drive voltage when the second rise control is executed. In the eleventh diagram (4), the drive signal generating unit 1A generates a signal for each energization period with a simple signal based on the signal of the first energization period after the start of the drive. In the first diagram (8), the drive signal generating unit 1 generates, after the start of the drive, the _ 322708 27 201203833, which is the first energization period, by the non-simplified signal, and generates the signal β of the energization period after the second period by the PWM signal. In the first rise control, the time from the start of driving to the energization of the coil L1 can be shortened, and the rise time from when the linear vibration motor 200 is started to when the desired vibration is obtained can be shortened. Further, when the second ascent control is employed, the driving force at the time of the rising can be increased as compared with the driving force during the steady operation, and the rising time can be shortened. (Stop Control) Hereinafter, the stop control that can be traced to the above-described drive control by the drive control circuit 100 of the present embodiment will be described. The drive signal generating portion 10 generates a phase opposite to the phase of the drive signal generated at the time of the drive after the drive of the linear vibration motor 200 is terminated. Drive signal. The drive unit 2 加速 accelerates the stop of the linear vibration motor 2 〇 by supplying the drive current of the opposite phase corresponding to the drive signal generated by the drive signal generating unit 1 对 to the coil L1. When the driving current of the opposite phase is given by L1, the stator 210 will exert a braking action for stopping the vibrator 220. In the present specification, the term "stopping of the linear vibration motor 200" refers to the termination of the jL gauge drive, which does not include the counter drive period for stopping the control. The drive signal generating unit 10 can generate a signal for each energization period of the inverse phase drive signal generated after the linear vibration motor 200 is terminated by the PWM signal. By adjusting the duty ratio of this ρ·signal, the braking force can be flexibly adjusted. As described above, the drive signal generating unit 1 can generate a signal for each energization period with the PWM signal. In the case where a signal for each energization period is generated by the PWM signal, the drive signal generation unit 10 can adopt the following 28 322708 201203833 stop control. In other words, the drive signal generating unit can set the duty ratio of the PWM signal generated during the energization period of the reverse phase drive signal after the linear vibration motor 200 is terminated to be higher than the energization period of the drive signal when the linear vibration motor 200 is driven. The generated P-signal has a lower work ratio. Further, the drive signal generating unit 10 can adjust the supply period of the reverse phase drive signal after the drive of the linear vibration motor 200 is terminated, based on the supply period of the drive signal when the linear vibration motor 200 is driven. For example, the shorter the supply period of the drive signal at the time of the drive, the shorter the supply period of the drive signal of the opposite phase after the end of the drive is set. For example, the supply period of the drive signal of the opposite phase is proportional to the supply period of the drive signal at the time of the drive. Further, in the region where the supply period of the drive signal during the driving is longer than the predetermined reference period, the supply period of the drive signal of the opposite phase may be fixed. The supply period of the above drive signal can be determined by the number of drive cycles. Further, the drive signal generating unit 10 can adjust the duty ratio of the PWM signal generated during the energization period of the reverse phase drive signal after the linear vibration motor 200 is driven to be driven, based on the supply period of the drive signal when the linear vibration motor 200 is driven. For example, the shorter the supply period of the drive signal at the time of the above driving, the lower the operation ratio of the PWM signal is set by the drive signal generation unit 10. For example, the operation ratio of the PWM signal is made proportional to the supply period of the drive signal at the time of the above-described driving. Further, in the region where the supply period of the drive signal during the above driving exceeds the predetermined reference period, the duty ratio of the PWM signal may be fixed. Fig. 12 is a diagram showing a configuration of a node 29 322708 201203833 of the decoder 14 to which the stop control function is added. The decoder 14 shown in Fig. 12 has a configuration in which the stop control unit 61 is added to the decoder 14 shown in Fig. 5. When the driving of the linear vibration motor 200 is terminated, the stop control unit 61 instructs the positive drive signal generating unit 58 and the negative drive signal generating unit 59 to generate a drive signal having a phase opposite to the phase of the drive signal generated at the time of the drive. At this time, it is possible to instruct to generate a signal during the energization period of the inverted phase drive signal with the PWM signal. In addition, when the supply period of the reverse phase drive signal is adjusted in accordance with the supply period of the drive signal when the linear vibration motor 200 is driven, the stop control unit 61 receives the number of count loops from the loop counter 13 (that is, the number of drive cycles). . The stop control unit 61 instructs the positive drive signal generating unit 58 and the negative drive signal generating unit 59 to generate the reverse phase drive signal that reflects the number of drive cycles. The same applies to the case where the duty ratio of the above-described PWM signal is adjusted in accordance with the supply period of the drive signal when the linear drive motor 200 is driven. Fig. 13 is a basic conceptual diagram for explaining the above stop control. Fig. 13(a) is a transition diagram showing a coil drive voltage when the stop control is not executed, and Fig. 13(b) is a transition diagram showing a coil drive voltage when the stop control is executed, Fig. 13(c) It is a transition diagram showing the coil drive voltage when the stop control is executed by the PWM signal. An example in which the period of the reverse phase drive signal after the termination of the drive is once is described in Figs. 13(b) and 13(c), but may be plural. In the case of a plurality of times, when the signal of the energization period of the drive signal is generated by the PWM signal, the duty ratio of the PWM signal can be lowered as the period of the reverse phase drive signal advances. 30 322708 201203833 ^ Figure 疋 is used to illustrate an example in which the number of reverse phase drive lie cycles is fixed in the above stop control. Fig. 14 (4) is a transition diagram showing the coil rotation _ and the vibration of the linear vibration motor 200 when the number U of the seventh drive is driven during driving, and Fig. 8 shows that the number of cycles of the drive (four) at the time of driving is small. A transition diagram of the coil drive voltage and the vibration of the linear vibration motor 200 at the time. In Fig. 14, an example in which the number of cycles of the inverse phase drive signal generated after the termination of driving is fixed to 2 is described. Fig. 14 (4) shows an example in which the number of cycles of the drive number at the time of driving is 4, and Fig. 14 (6) shows an example in which the number of cycles of the drive signal at the time of driving is 2. In Fig. 14 (4), it is understood that the vibration of the linear vibration motor 2〇〇 is quickly converged after the driving of the vibration motor 200 is terminated by the supply of the reverse phase drive signal of the two cycles to the coil Li. On the other hand, in Fig. 14(b), by supplying the two-phase reverse phase drive (4) to the coil L1, after the drive of the linear vibration motor 2 (10) is terminated, the difficulty of the linear county motor is fast, However, an anti-phase vibration is generated thereafter (refer to the portion of the elliptical region). This means that excessive vibration is supplied for the vibration when the linear vibration motor 200 is driven. . Fig. 15 is a diagram showing an example in which the number of cycles of the reverse phase drive signal is variable in the above-described stop control. Fig. 4 (4) is a coil drive and a linear vibration motor when the number of times of the drive signal at the time of driving is large. Fig. 15 (6) is a transition diagram showing the vibration of the linear vibration motor 200 when the number of cycles of the drive signal at the time of driving is small. 322708 31 201203833 Fig. 15(a) is the same diagram as Fig. 14(a). Fig. 15(b) shows an example in which the number of cycles of the drive signal at the time of driving is 2, and the number of cycles of the reverse phase drive signal generated after the termination of the drive is 丨. As is apparent from the figure (b), the vibration of the linear vibration motor 200 is quickly converged after the driving of the linear vibration motor 2A is terminated by the supply of the reverse phase drive signal for one cycle of the coil L1. In comparison with Fig. 14 (b), it is understood that the linear vibration motor 2 〇〇 does not generate the vibration of the opposite phase in Fig. 15 (b). In Fig. 14, the vibration strength of the linear vibration motor 200 in which the driving of the linear vibration motor 2A is terminated is not considered, and a fixed braking force is provided. Therefore, the braking force is too large or too small. On the other hand, in Fig. 15, by providing the braking force that reflects the vibration intensity of the linear vibration motor 2, it is possible to achieve optimum stop control. As described above, when the above-described stop control is employed, the vibration stop time at the time of termination of the drive of the linear vibration motor 200 can be shortened. Further, by generating a signal of the energization period of the inverted phase drive signal by the PWM signal, the braking force can be flexibly set. Further, by adjusting the supply period of the reverse phase drive signal during the supply period of the drive signal when the linear vibration motor 2 is driven, the optimum stop can be achieved regardless of the length of the supply period of the drive signal during the drive. control. In the tactile use, the vibration is abruptly changed, and the user can easily feel the vibration caused by the contact. By using the above-described stop control, the vibration can be abruptly changed. (Detection Window Setting) Next, an example in which the zero-crossing detection unit 40 sets a detection window for avoiding the zero-crossing detection of the voltage other than the induced voltage will be described. Zero 32 322708 201203833 = The detecting unit 4G recognizes the zero crossing detected in the gynecological office as having ^ and the zero crossing detected outside the detection window as invalid. Here, it is clear that the zero crossing of the electric house other than the induced voltage is mainly the zero crossing of the driving electric power that is energized by the driving signal σ|5 1 , and the zero X of the regenerative electric power (refer to the third Therefore, the detection window is in principle set to a period in which the non-energization period between the energization period of the positive (negative) current and the energization period of the negative (positive) current is set to be narrowed toward the inside. At this time, it is necessary to During the period during which the regenerative current flows during the energization period, if the mosquito is too narrow, the possibility of detecting the normal induced f(four) zero crossing is increased. Therefore, it is considered to detect the voltage zero crossing other than the induced voltage. The probability of not being able to detect the tradeoff relationship between the possibility of the normal induced voltage zero-crossing determines the period of the above detection window. Next, the case where the zero crossing is not detected in the above detection window is performed. In the above case, the zero-cross detecting unit 40 assumes that the zero crossing of the induced voltage has been terminated at the start position of the detection window, and assumes that the detection window is A zero crossing is detected in the vicinity of the start position, and the detection position of the zero crossing of the false S history is supplied to the drive signal generating unit 1 . The case where the zero crossing of the induced voltage is terminated at the start position of the detection window is The case where the voltage between the two ends of the coil L1 at the start position of the detection window is zero-crossed. In the example shown in Fig. 3, the voltage at both ends of the coil L1 is positive at the start position of the detection window. In addition, when the zero crossing detecting unit 40 does not detect the zero crossing in the above detection window, that is, when the zero crossing of the sensing 322708 33 201203833 is not terminated at the end position of the detection window, it is assumed A zero crossing is detected in the vicinity of the end position of the detection window, and the assumed zero crossing detection period is supplied to the driving chirp generating portion 10. The so-called zero crossing of the induced voltage is performed at the end position of the detecting window. The case where the voltage is not terminated is the case where the voltages at both ends of the coil L1 are at the polarity before the zero crossing at the end position of the detection window. A configuration example of the zero-cross detecting unit 40 having a detection window setting function is shown in Fig. 16. The zero-cross detecting unit 40 shown in Fig. 16 is shown in Fig. 1. The detection window setting unit Μ and the output control unit 44 are added to the zero-cross detecting unit 4A. The detection window setting unit 43 supplies a signal for setting the detection window to the output control unit 44. More specifically, a detection window is provided. Signal 2 and detection window start signal. Fig. 17 is a diagram for explaining the detection window signal 丨, the detection window signal 2, and the detection window start signal. The detection window signal 丨 is a signal generated based on the above knowledge, that is, the setting is The detection window signal that is reduced inward during the non-energization period. The detection window signal 2 is a signal that the termination position of the detection window is extended to the start position including the subsequent energization period, compared to the detection window signal 丨. Thus, the comparator 41 reverses the output based not only on the zero crossing of the induced voltage but also on the zero crossing of the driving voltage supplied during the energization. The detection window start signal is a signal indicating the start position of the detection window. More specifically, it is a signal that rises at the edge of the start position of the detection window. Referring back to Fig. 16, when the output of the comparator 41 is not reversed at the start position of the detection window, the output control unit 44 supplies the edge position detected by the edge detecting unit 42 322708 34 201203833 as the detection position of the zero crossing. The drive signal generation unit 10 (more strictly, the second latch circuit 15). When the output of the comparator 41 is reversed at the start position of the detection window, the output control unit 44 supplies the start position of the detection window to the detection signal generating unit 10 as a zero-crossing detection position (more strictly speaking Is the second latch circuit 15). Hereinafter, a configuration example of the output control unit 44 for realizing these processes will be described. Fig. 18 is a view showing an example of the configuration of the output control unit 44. The output control unit 44 includes a first AND gate 71, a second AND gate 72, and a 〇R gate. 3 = The detection window start signal and the output signal of the comparator 41 are input to the first AND gate 7 and the second output is output when the two are high level signals, and the level 'outputs a low level when at least one of the low level signals is low. signal. More specifically, in the case where the output of the comparator 已 has been inverted at the start position of the above-described detection f, the first shutter 71 outputs a high level signal. The detection window signal 2 and the output signal of the edge detecting unit 42 are input 2M, 72. When the _ gate 72 is a high level signal, the 间 准 纟 纟 纟 纟 纟 = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = = At the beginning, the first brake? 2 rounds. The turn-off signal of the first closed 71 and the output of the first _ gate 72 are input to the GR (four). (10) Gate 73 outputs the high level (4) when both sides are based on the two-level transmission signal. When the (4) 2 is a high-precision signal, the output low-level information is low. On the other hand, when the output of the comparator 41 has been inverted at the start position of the detection window 322708 35 201203833, the 〇R gate 73 outputs a high level signal. When the output of the comparator 41 has not been inverted at the start position of the above detection window In the above detection window, a high level signal is output when the edge of the output signal of the edge detecting unit 42 rises. Fig. 19 is a diagram showing the zero crossing detection unit 40 using the detection window signal ( (the detection window is not used) Action diagram of signal). Figure 19 (phantom shows the voltage across the coil L1 and the transition of the edge signal when the zero crossing of the induced voltage occurs in the detection window, and Fig. 19(b) shows that no induction occurs in the detection window. When the voltage is zero-crossing (drive frequency <resonance frequency), the voltage across the coil L1 and the edge signal are shifted, and FIG. 19(c) shows that the zero-crossing of the induced voltage does not occur in the detection window (drive frequency > resonance frequency) The voltage at both ends of the coil and the transition of the edge signal. In the zero-cross detecting unit 4 (using the detection window start signal) using the detection window signal 1, the output control unit 44 is only the second AND shown in Fig. 18. The gate 72 is formed. The detection window signal 1 and the output signal of the edge detecting unit 42 are input to the second AND gate 72. In Fig. 19(a), the induced voltage is generated in the detection window set by the detection window signal 1. Zero crossing, so the edge rises in the edge signal at the position where the zero crossing occurs. In addition, since the detection window is set, the edge is not raised in the edge signal at the position where the zero crossing of the regenerative voltage occurs. In Fig. 19(b), the resonance frequency of the linear vibration motor 200 is higher than the frequency of the above-described drive signal and the difference is large. Therefore, the linear vibration motor 200 of the above-mentioned induced voltage should be generated. The stop shape 36 322708 201203833 state (ie, the maximum arrival point on the § pole side of the vibration range or the maximum arrival point on the N pole side) is not generated in the above detection window. At the time of entering the above detection window The stop state has been terminated. In this case, in the zero-cross detecting portion 4 (using the detection window start signal) using the detection window signal 1, the edge is not raised in the edge signal (refer to the portion of the elliptical region) In Fig. 19(c), the resonance frequency of the linear vibration motor 2A is lower than the frequency of the above-described drive signal and the difference is large. Therefore, a zero-crossing linear vibration motor of the induced voltage should be generated. The stop state of 2〇〇 is not generated in the above detection window. This stop state is generated after leaving the above detection window. In this case, the zero-cross detection unit 40 of the detection window signal 1 is used (the detection window start signal is not used). In the edge signal, the edge does not rise (refer to the portion of the elliptical region). Fig. 20 is a view for explaining the operation of the zero-crossing detecting unit 40 using the detection window signal 2 and the detection window start signal. Fig. 20(a) shows the transition of the voltage at both ends of the coil L1 and the edge signal when the zero crossing of the induced voltage does not occur in the detection window (the driving frequency <resonance frequency), and the second graph (b) indicates the detection window. When zero crossing of the induced voltage does not occur (driving frequency > resonant frequency) The voltage across the coil L1 and the transition of the edge signal. In the zero-cross detection unit 40 that uses the detection window signal 2 and the detection window start signal, the output control unit 44 shown in Fig. 18 is used. The voltage transition across the coil L1 shown in Fig. 20(a) is the same as the voltage transition across the coil L1 shown in Fig. 19(b). The voltage transition across the coil L1 shown in Fig. 20(b) is the same as the voltage shift across the coil L1 shown in Fig. 19(c). 37 322708 201203833 In Fig. 20(a), the roles of (10) and (iv) 73 shown in Fig. 18 are in the edge signal at the start position of the detection window. In Fig. 20 (6), since the action of extending the position of the I soil of the detection window is "at the positive current energization start position, the edge rises in the edge signal. As described above, the above-described inspection (4) is set, and when the period width of the driving ## is adaptively controlled so that the natural vibration number of the linear vibration motor coincides with the frequency of the driving signal, the zero of the induced voltage generated in the coil u can be increased. Cross detection accuracy. That is, it is possible to suppress erroneous detection of the zero crossing of the driving voltage and the regenerative current. When the detection factor is set, if the spectral frequency of the linear vibration motor 2 〇 大幅 greatly deviates from the frequency of the drive signal, the zero crossing of the induced voltage may be detached from the detection window. In the present embodiment, by temporarily raising the temporary edge in the vicinity of the start position of the detection window or in the vicinity of the end position, the adaptive control of the cycle width of the drive signal can be continued without interruption. Even if the resonance frequency of the linear vibration motor 2〇〇 deviates greatly from the frequency of the drive signal, the two can be brought closer by the temporary edge. Thus, the spectrum of the linear vibration motor 200 is made by frequently performing adaptive control. The vibration frequency coincides with the frequency of the drive signal, and even if the accuracy of the built-in oscillator that generates the basic clock in the drive control circuit 1〇〇 is lowered, it is not necessary to adjust the frequency of the built-in oscillator, thereby greatly assisting The manufacturing cost of the drive IC (drive control circuit 1) is lowered. Further, as the temporary edge 322708 38 201203833 rising near the end position of the detection window, the signal control can be simplified by using the activation of the subsequent energization period in the non-energization period. It is not necessary to put a signal other than the detection window signal such as the pure signal on the inspection. The present invention has been described above based on the embodiments. This embodiment is merely illustrative, and it should be understood by those skilled in the art that various modifications may be made in combination of these constituent elements and the respective processes, and such modifications are also within the scope of the invention. The second rise control can be applied to the drive control circuit for driving the linear vibration motor 2A by not including the drive signal during the non-energization period. The drive letter mosquito alternately sets a signal when the non-energized period is not interposed between the positive current energization period and the neon current energization period. That is, the second rising control is also (4) the adaptability of the period width of the moving signal on the money line (4). (4) The above-described stop control can also be applied to the driving linearity by not including the driving signal during the non-energizing period. Crane (four) circuit of vibration motor 2GG. That is, also (d) applied to the drive control circuit 0 which does not perform the adaptive control of the cycle width of the above drive signal, Fig. 21 shows the first! A diagram showing a modification of the configuration of the drive control circuit 100 of the linear vibration motor 2A shown in the drawing. In the configuration of the drive signal generating unit 10 shown in FIG. 21, the difference calculation circuit 16, the third latch circuit 17, the adder circuit 18, and the fourth lock are omitted from the drive signal generation unit 1A shown in FIG. Memory circuit 19. Instead, the counter in the main counter 12 can count up to the value corresponding to the count value of the above-mentioned drive record. If you use the above example along 322708 39 201203833, use a 250-ary counter or a 300-ary counter instead of a 200-ary counter. The second latch circuit 15 sequentially latches the count value supplied from the main counter 12, and outputs the latched count value to the first latch circuit 1 at the position where the zero-cross detecting unit 4 detects the zero-crossing. 1. Since the main counter 12 can count a count value larger than the count value corresponding to one cycle of the drive signal, the second latch circuit 15 can directly use the latched count value as a new count end value. Therefore, the difference calculation circuit 16, the third latch circuit 17, the adder circuit 18, and the fourth latch circuit 19 can be omitted, and the circuit configuration can be simplified. Fig. 22 is a view showing a second modification of the configuration of the drive control circuit 100 of the linear vibration motor 2A shown in Fig. 1. In the zero-crossing detecting unit 40 shown in Fig. 22, the analog/digital converter 41a is used instead of the comparator 41. The analog/digital converter 41a converts the output analog signal of the induced voltage detecting unit 3 (the differential amplifier in the example of Fig. 22) into a digital signal. The edge detecting unit 42 generates a digit value indicating the position at which the zero crossing is detected based on the output digit signal ' of the analog/digital converter 41a, and outputs it to the second latch circuit 15. For example, when the differential amplifier is designed without deviation, the edge detecting unit 42 outputs a high level signal to the second latch at a time point when the round-trip digit value of the analog/digital converter 41a is zero. Circuit 15. According to the second modification, since the digital processing is performed from the phase of detecting the zero crossing of the induced voltage, the detection accuracy of the zero-crossing time can be improved. 322708 40 201203833 Fig. 23 is a view showing a modification 3 of the configuration of the drive control circuit 100 of the linear vibration motor 2A shown in Fig. 。. The third modification is a configuration in which the first modification and the second modification are combined. BRIEF DESCRIPTION OF THE DRAWINGS Fig. 1 is a view showing the configuration of a drive control circuit of a linear vibration motor according to an embodiment of the present invention. Fig. 2 is a view showing an example of the configuration of a drive unit, an induced voltage detecting unit, and a comparator. Fig. 3 is a timing chart showing an operation example of the drive control circuit of the embodiment. Fig. 4 is a timing chart showing an example of an edge signal, an i-th clock signal, a second clock signal, and a third clock signal. Fig. 5 is a view showing an example of the structure of a decoder. Fig. 6 is a waveform diagram showing one cycle of the drive signal. Fig. 7 (a) to (c) are explanatory diagrams of control of the width of the energization period of the drive signal. Fig. 8 is an explanatory diagram of phase control of a drive signal. Fig. 9 is a view showing an example of the configuration of a decoder to which a rising throttle function is added. Fig. 10 is an explanatory diagram of the first rising control. Fig. 10 is a transition diagram showing the coil drive voltage and the vibration of the linear vibration motor when the first rise control is not performed, and Fig. 10(b) is a view showing the coil drive voltage and the linear vibration when the ith rise control is executed. The transition diagram of the vibration of the motor. Fig. 11 is an explanatory diagram of the second rising control. (3) is a transition diagram showing the coil drive voltage when the second rise control is not performed, and FIG. 11B is a transition diagram showing the coil drive voltage when the second rise control is executed. Fig. 12 is a view showing an example of the configuration of a decoder to which a stop control function is added. Fig. 13 is an explanatory diagram of the basic concept of the above stop control. Fig. 13(a) is a transition diagram showing the coil drive voltage when the stop control is not executed, and Fig. 13(b) is a transition diagram showing the coil drive voltage when the stop control is executed, and Fig. 13(c) is a view showing the transition of the coil drive voltage when the stop control is executed. A transition diagram of the coil drive voltage when the stop control is executed by the PWM signal. Fig. 14 is an explanatory diagram showing an example in which the number of cycles of the drive signal in the reverse phase in the above-described stop control is fixed. Fig. 14 (a) is a transition diagram showing the coil drive voltage and the vibration of the linear vibration motor when the number of cycles of the drive signal at the time of driving is large, and Fig. 14 (b) shows that the number of cycles of the drive signal during driving is small. The transition diagram of the coil drive voltage and the vibration of the linear vibration motor. Fig. 15 is an explanatory diagram showing an example in which the number of cycles of the drive signal in the opposite phase in the above-described stop control is variable. Fig. 15(a) is a transition diagram showing the coil drive voltage and the vibration of the linear vibration motor when the number of cycles of the drive signal at the time of driving is large, and Fig. 15(b) shows that the number of cycles of the drive signal during driving is small. The transition diagram of the coil drive voltage and the vibration of the linear vibration motor. Fig. 16 is a view showing the configuration of a zero-cross detecting unit having a detection window setting function. 42 322708 201203833 Figure 17 is an explanatory diagram of the detection window signal 1, the detection window signal 2, and the detection window start signal. Fig. 18 is a view showing an example of the configuration of an output control unit; Fig. 19 is an explanatory diagram of the operation of the zero-cross detecting unit (the unused detection start number) using the detection window signal 1. Fig. 19(a) is a transition diagram showing the voltage across the coil and the edge signal of the zero-crossing coil in which the induced voltage is generated in the detection window, and Fig. 19(b) shows the zero in which the induced voltage is not generated in the detection window. At the time of crossing (drive frequency <resonance frequency), the voltage across the coil and the transition of the edge signal, Fig. 19(c) shows the zero-crossing time (drive frequency > resonance frequency) in which no induced voltage is generated in the detection window. The voltage across the coil and the transition of the edge signal. Fig. 20 is an operation explanatory diagram of the zero-cross detecting unit using the detection window signal £ and the detection window start signal. Fig. 2(a) is a transition diagram showing the voltage across the coil and the edge signal when the zero-crossing of the induced voltage is not generated in the detection window (the driving frequency <resonance frequency), and the second diagram is shown in the detection There is no zero-crossing of the induced voltage in the window (drive frequency > resonant frequency). The voltage across the coil and the transition of the edge signal. Figure 21 shows 帛! Fig. 1 is a view showing a modification of the configuration of the drive control circuit of the linear vibration motor shown in the drawing. Figure 22 shows the first! Fig. 2 is a view showing a modification of the configuration of the drive control circuit of the linear vibration motor shown in the drawing. Fig. 23 is a view showing a third modification of the configuration of the drive control circuit of the linear vibration motor shown in Fig. 。. [Description of main component symbols] 322708 43 201203833 10 Drive signal generation unit 11 First latch circuit 12 Main counter 13 Loop counter 14 Decoder 15 Second latch circuit 16 Difference calculation circuit 17 Third latch circuit 18 Addition circuit 19 Fourth latch circuit 20 Driving unit 30 Inductive voltage detecting unit 40 Zero crossing detecting unit 41 Comparator 41a Analog/digital converter 42 Edge detecting unit 43 Detection window setting unit 44 Output control unit 51 Driving width calculating unit 52 Positive driving center value Calculation unit 53 Negative drive center value calculation unit 54 Positive side subtraction unit 55 Positive side addition unit 56 Negative side subtraction unit 57 Negative side addition unit 58 Positive drive signal generation unit 59 Negative drive signal generation unit 60 Rise control unit 61 Stop control unit 71 1st AND gate 72 2nd-gate 73 OR gate 100 drive control circuit 200 linear vibration motor 210 stator 211 core member 220 vibrator 221 permanent magnet 222a, 222b spring 223 frame - LI coil 44 322708