201126421 六、發明說明： 【發明所屬之技術領域】 [0001] 本發明係有關於一種無線射頻辨Μ 1 ώ 戴天線，特別 係有關於一種具有電感性輸入阻抗且7 Ημ „ 且了貼附於金屬物體 之無線射頻辨識標籤之圓極化微帶天線。 【先前技術】 [0002] 習知無線射頻辨識（RFU))系統所使用的頻段主要有 低頻（LF，125ΚΗΖ)、高頻（HF，13.56ΜΗζ)、超高頻 (UHF，860〜960MHz)以及微波（2. 4〜2. 483GHz和 5. 725~5. 875GHz)等頻段。近年來，Rfid在超^頻頻段 的應用快速成長，因其工作於趕高頻時有較遠的讀取距 離、更大的資料傳輸率及較小的標錢尺寸，而標藏天線 是RFID系統的關鍵元件。目前在超高頻標藏天線的設計 中，偶極天線是最被廣泛使用的一種天，此種天線一 般製作於成本低、厚度很薄的PET材質上，並可執行良好 的讀取距離。然而，當偶極天線貼附於金屬物體上時， 會使天線的輸入阻抗及輻射場型造成很大改變，進而導 致天線無法正常工作。目前在公開文獻已提出微帶天線 或倒F天線（PIFA)的相關設計，來解決上述問題。 習知標籤天線都是線性極化的天線。在一般情況下 ，讀取機使用圓極化天線，標籤使用線性極化天線，讀 取功能不會受到標箴擺放方向的影響。然而，在一些特 殊應用場合，如機場的行李檢測，其讀取機可能需要使 用指向性高、波束寬較窄的線性極化天線’此時’若使 用市面上線性極化的標籤天線產品’可能會因為標籤天 線擺放的方向使得讀取機天線與標籤天線兩者之間極化 098140456 表單編號A0101 第4頁/共24頁 098 201126421 不匹配，造成讀取距離大幅下降。此時，標籤必須使用 具有圓極化輻射的天線，才能解決天線之間因極化不匹 配對讀取功能的影響。201126421 VI. Description of the Invention: [Technical Field] [0001] The present invention relates to a radio frequency identification antenna, in particular to an inductive input impedance and 7 Ημ „ Circularly polarized microstrip antenna for radio frequency identification tags of metal objects [Prior Art] [0002] The frequency bands used in conventional radio frequency identification (RFU) systems mainly include low frequency (LF, 125 ΚΗΖ), high frequency (HF, 13.56ΜΗζ), UHF (UHF, 860~960MHz) and microwave (2.4~2.483GHz and 5.725~5. 875GHz) and other frequency bands. In recent years, Rfid has grown rapidly in the ultra-frequency band. Because it works at high frequency, it has a long reading distance, a large data transmission rate and a small standard size. The standard antenna is a key component of the RFID system. Currently in the design of UHF standard antenna. The dipole antenna is one of the most widely used days. This type of antenna is usually fabricated on a low-cost, thin-thick PET material and can perform a good reading distance. However, when the dipole antenna is attached to a metal object, When it goes up, it will The input impedance of the line and the radiation field type cause a great change, which leads to the antenna not working properly. The related design of the microstrip antenna or inverted-F antenna (PIFA) has been proposed in the open literature to solve the above problems. It is a linearly polarized antenna. In general, the reader uses a circularly polarized antenna, and the tag uses a linearly polarized antenna. The reading function is not affected by the direction in which the mark is placed. However, in some special applications, For baggage detection at the airport, the reader may need to use a linearly polarized antenna with high directivity and narrow beamwidth. 'At this time, if you use a linearly polarized tag antenna product on the market', it may be placed because of the tag antenna. The direction is such that the polarization between the reader antenna and the tag antenna is 098140456. Form No. A0101 Page 4 of 24 098 201126421 Mismatch, resulting in a significant drop in reading distance. At this time, the tag must use circularly polarized radiation. The antenna can solve the influence of the polarization mismatch on the read function between the antennas.
[0003][0003]
098140456 傳統圓極化微帶天線為具有50 Ω輸入阻抗的天線結 構，一般而言，其很容易在輻射元件内部找到50 Ω的饋 入位置，並激發圓極化波輕射。然而，在RFID的系統中 ，標籤晶片通常具有電容性的阻抗，為了使晶片與天線 兩者之間有最大功率傳輸，天線必須設計為具有電感性 的輸入阻抗’使其與晶片阻抗匹邱:然而’要在微帶天 線上找到一饋入位置，使其具有一特定的電感性輸入阻 抗且能激發圓極化轉射並不容易β截至目前為止，在公 開的文獻還未看到可應用於RFID系统且具有電感性輸入 阻抗的圓極化標籤天線的相關設計被提出。 【發明内容】 本發明之主要目的係在於提供一種無線射頻辨識標 籤之圓極化微帶天線，其包^--基:板、1 一輕射金屬層、 一微帶線以及一择地層。該基掏係具有一上表面、—下 表面及一連通該上表面與該下表面之導通孔。該輻射金 屬層係形成於該基板之該上表面，且該輻射金屬層係具 有一槽孔。該微帶線係形成於該基板之該上表面且位於 該輻射金屬層之至少一側邊，該微帶線係具有一鄰近誃 導通孔之饋入端，且該微帶線與該輻射金屬層之間係且 有一第一間距及一第二間距。該接地層係形成於該基板 之該下表面，且電性連接該導通孔。本發明係有别於一 般習知輸入阻抗50歐姆的圓極化微帶天線，本發明天線 係利用電磁耦合饋入方法來設計具有電感性輸入阻抗 表單煸號Α0101 第5頁/共24頁 凡的 0982069427-0 201126421 圓極化微帶天線，使天線能與標蕺晶片阻抗匹配，得以 應用於無線射頻辨識系統。本發明天線的一實施例’設 计刼作於922〜928MHz的頻段，在此操作頻段内，改變天 線架設的方向具有穩定的讀取距離’而且天線貼附於金 屬有更佳的讀取距離’非常適合於超高頻無線射頻辨識 系統之應用。 【實施方式】 [0004] 相 > 閱第1及2圖，其係本發明之第一較佳實施例， 一種無線射頻辨識標籤之圓極化微帶天線係包含一基板 1〇、一輻射金屬層2〇、一微帶線30以及一接地層4〇。該 基板10係具有—上表面1〇έ、一下表面l〇b及一連通該上 表面1〇a與該下表面l〇b之導通孔11。該輻射金屬層2〇係 形成於該基板1〇之該上表面1〇a，且該輻射金屬層係具 有一槽孔21 ,在本實施例中，該輻射金屬層20係呈正方 形狀，或者，請參閱第12圖，在另一實施例中，該輻射 金屬廣20係可呈圓形狀’請再參閱第1及2圖，在本實施 例中，該輻射金屬層2〇係具有一第一側邊2〇a第二側 邊2〇b、一相對於該第一側邊20al第三側邊20c及一相 對於該第二侧邊2〇b之第四側邊2〇d。又該輕射金屬層 20的該槽孔21係呈不等長度的十字形狀，用以產生頻率 接近的低頻模態與高頻模態。 098140456 請再參閱第1及2圖’該微帶線30係形成於該基板10 之該上表面l〇a且位於該輻射金屬層2〇之至少一側邊。在 本實施例中，該微帶線30係具有一鄰近該導通孔1丨之饋 入端3〇a且具有—第一導線部31及一第二導線部32，該第 一導線部31係位於該輻射金屬層20之該第一側邊2〇a，而 09820ί 表單編號侧 第6頁/共24頁 一 201126421 該第二導線部32係位於該輻射金屬層2〇之該第二侧邊2〇b ，且該第一導線部31與該第二導線部32係連接成[形狀， 或者，請再參閱第12圖，在另一實施例中，該第一導線 部31與該第二導線部32係可連接成圓弧形狀。請再參閱 第1及2圖，在本實施例中，該第一導線部31係具有—第 一長度L1，該第二導線部32係具有一第二長度L2，在本 實施例中，該第一長度L1與該第二長度L2之計算公式係 Ο [0005]098140456 The traditional circularly polarized microstrip antenna is an antenna structure with a 50 Ω input impedance. In general, it is easy to find a 50 Ω feed position inside the radiating element and excite a circularly polarized wave. However, in RFID systems, the tag wafer typically has a capacitive impedance. In order to maximize the power transfer between the chip and the antenna, the antenna must be designed to have an inductive input impedance that makes it comparable to the chip impedance: However, it is not easy to find a feed position on the microstrip antenna to have a specific inductive input impedance and to stimulate circular polarization. As of now, it has not been seen in the published literature. A related design of a circularly polarized tag antenna with an inductive input impedance for an RFID system is proposed. SUMMARY OF THE INVENTION The main object of the present invention is to provide a circularly polarized microstrip antenna of a radio frequency identification tag, which comprises: a board, a light metal layer, a microstrip line and a ground layer. The base has an upper surface, a lower surface, and a via hole connecting the upper surface and the lower surface. The radiant metal layer is formed on the upper surface of the substrate, and the radiant metal layer has a slot. The microstrip line is formed on the upper surface of the substrate and on at least one side of the radiant metal layer, the microstrip line has a feeding end adjacent to the 誃 conductive via, and the microstrip line and the radiant metal The layers have a first pitch and a second pitch. The ground layer is formed on the lower surface of the substrate and electrically connected to the via. The present invention is different from the conventional circularly polarized microstrip antenna with an input impedance of 50 ohms. The antenna of the present invention is designed by using an electromagnetic coupling feeding method to have an inductive input impedance form. 1010101 Page 5 of 24 The 0982069427-0 201126421 circularly polarized microstrip antenna enables the antenna to be matched to the standard wafer impedance for use in a radio frequency identification system. An embodiment of the antenna of the present invention is designed to be used in a frequency band of 922 to 928 MHz. In this operating frequency band, the direction in which the antenna is erected has a stable reading distance and the antenna is attached to the metal to have a better reading distance. 'Very suitable for UHF RFID applications. [Embodiment] [0004] Phases 1 and 2, which are the first preferred embodiment of the present invention, a circularly polarized microstrip antenna of a radio frequency identification tag includes a substrate, a radiation The metal layer 2〇, a microstrip line 30, and a ground layer 4〇. The substrate 10 has an upper surface 1 〇έ, a lower surface 10b, and a via hole 11 connecting the upper surface 1a and the lower surface 10b. The radiant metal layer 2 is formed on the upper surface 1 〇 a of the substrate 1 , and the radiant metal layer has a slot 21 . In the embodiment, the radiant metal layer 20 has a square shape, or Referring to FIG. 12, in another embodiment, the radiant metal strip 20 may have a circular shape. Please refer to FIGS. 1 and 2 again. In this embodiment, the radiant metal layer 2 has a first The second side 2〇b of one side 2〇a, the third side 20c with respect to the first side 20al and the fourth side 2〇d of the second side 2〇b. Further, the slot 21 of the light-emitting metal layer 20 has a cross shape of unequal length for generating a low frequency mode and a high frequency mode which are close in frequency. 098140456 Please refer to FIGS. 1 and 2 again. The microstrip line 30 is formed on the upper surface 10a of the substrate 10 and on at least one side of the radiant metal layer 2〇. In the embodiment, the microstrip line 30 has a feeding end 3〇a adjacent to the through hole 1丨 and has a first lead portion 31 and a second lead portion 32. The first lead portion 31 is Located on the first side 2〇a of the radiant metal layer 20, and 09820ί, on the form number side, page 6/24 pages, 201126421, the second lead portion 32 is located on the second side of the radiant metal layer 2〇 2〇b, and the first lead portion 31 and the second lead portion 32 are connected in a [shape, or please refer to FIG. 12 again. In another embodiment, the first lead portion 31 and the second portion The lead portions 32 are connectable in a circular arc shape. Referring to FIGS. 1 and 2 again, in the embodiment, the first lead portion 31 has a first length L1, and the second lead portion 32 has a second length L2. In this embodiment, the second lead portion 32 has a second length L2. The calculation formula of the first length L1 and the second length L2 is [0005]
其中又係為天線工作中心頻率之波長、义丨係為天線輸入阻 抗之虛部值及ZQ係為該微帶線30之特性阻抗、較佳地， 該第一導線部31之該第一長度L1係與該第二導線部32之 該第二長度L2不相等，以使低頻模態與高頻模態的輻射 電場強度較接近，進而可獲得較佳的圓極化輻射。又， 098140456 該微帶線30與該輕射金屬層20.之間係具有一第一間距g 1 及一第二間距Gf，其中第一間距G1扁介於該微帶線3〇之 該第一導線部31與該輻射金屬層20之該第一側邊2〇a之間 ，而該第二間距G2係介於該微帶線30之該第二導線部32 與該輻射金屬層20之該第二侧邊20b之間。在本實施例中 ，該第一間距G1及第二間距G2係用以調整天線的實部輸 入阻抗，以使天線的實部輸入阻抗與一晶片50的實部阻 抗大約相等》較佳地，該第一間距G1係與該第二間距G2 相等，且其值係介於1. 5mm至3mm之間。請再參閱第1及2 圖，該接地層40係形成於該基板10之該下表面i〇b，且電 表單編號A0101 第7頁/共24頁 0982069427-0 201126421 性連接該導通孔11。 關於本發明之該圓極化微帶天線之阻抗匹配方法， 係詳細說明如下。在本實施例中’其所使用之標籤晶片 在925MHz的阻抗為13.5— ΠΙΟΩ，為了使該圓極化微帶 天線及標藏晶片兩者之間付到良好的功率傳輸，其必須 將該圓極化微帶天線在925MHz之輸入阻抗設計為 13.5+jllOQ ’以達成共轆匹配。本發明係透過選擇該 第一導線部31之該第一長度L1、該第二導線部32之該第 二長度L2 '該微帶線寬度w、該第一間距g 1及該第二間距 G2來達成阻抗匹配’其中該圓極化微帶天線之虛部輸入 阻抗主要來自該微帶線30之電抗值，而提供電抗j no Q 之該第一導線部31之該第一長度L1及該第二導線部32之 該第二長度L2係可由公式（1)決定，而由公式（1)可知， LI、L2及w有無窮多組解。在本實施例中，其係選擇 w=l. 5mm及Ll+L2 = 73. 2mm。至於該圓極化微帶天線之實 部輸入阻抗13. 5Ω，則可藉由調整該第一間距G1及該第 二間距G2來達成。 請參閱第3A及3B圖，其係顯示該圓極化微帶天線之 模擬與量測輸入阻抗結果圖，而第4圖係為其對應的返回 損失結果圖。另外，第3B圖係加入只有該微帶線30(不含 6玄輕射金屬層20)的電抗計算值作為比較。由第3B圖可發 現，在925MHz附近該圓極化微帶天線的模擬及量測電抗 值約為j 110 Ω，與只有微帶線的計算值非常吻合，證明 該圓極化微帶天線的輸入電抗主要由該微帶線3〇的參數 所控制。另外，由第3A、3B及4圖亦可看到該圓極化微帶 天線的模擬及量測結果相當接近。請參閱第5A、5B及6圖 098140456 表單編號A0101 第8頁/共24頁 0982069427-0 201126421 ，其係分別顯示改變該第一間距G1及該第二間距G2所得 之量測輸入阻抗及返回損失結果圖，由第5A圖可發現， 當該第一間距G1及該第二間距G2等於0. 5mm時，因為該 微帶線30與該輻射金屬層20的耦合強度較大，該圓極化 微帶天線的輸入電阻峰值達到57Ω，導致該圓極化微帶 天線不易完成阻抗匹配；反之，隨著該第一間距G1及該 第二間距G 2的增加，其輸入電阻會減小，當 G1=G2 = 1. 5~3mm時，可以達成良好的阻抗匹配。 請參閱第7A及7B圖，其係改變該第一導線部31之該 第一長度L1及該第二導線部32之該第二長度L2所得之量 測輸入阻抗結果圖，第8圖為其對應的返回撢失結果圖， 而第9圖為其對應的模擬轴比:結果圓，在本實::施例中，該 第一導線部31之該第一長度L1及該第二導線部32之該第 二長度L2之和皆固定為li+L2 = 73. 2mm。廣第7A圖可發 現在925MHz附近有兩個鄰近模態被激發《由於本發明該 ：,： | 圓極化微帶天線之低頻模態:是由x方向的:¾流所產生，而 高頻模態是由y方向的電流和產生。因此，從Ll=26nini及 L2 = 47. 2mm的曲線，可觀察到低頻模態的峰值電阻較小 ’而高頻模態的峰值電阻較大。這是因為L1較短，使得 低頻模態的耦合強度較小；而L2較長，使得高頻模態的 耦合強度較大。又，當L1增加為32mm及L2減小為41. 2mm 時’兩個鄰近模態的峰值電阻大約相等，預期兩個鄰近 模態的輻射電場強度較接近，可以得到較好的圓極化輻 射。另外’由第9圖亦可看到使用Ll=32mm及L2 = 41. 2mm 時’在925MHz的軸比值低於idB，其3dB軸比的圓極化頻 寬為92卜928MHz，符合超高頻RFID的工作頻帶 098140456 表單編號A0101 第9頁/共24頁 0982069427-0 201126421 922〜928MHz。請參閱第1〇A及1〇β圖，其係顯示本發明 該圓極化微帶天線貼附於4Q x4Qcm2金屬銅板時之輸入阻 抗結果圖’由®中可發現標籤貼附於金屬銅板時其天 線阻抗值幾乎沒有改變。 最後β參閱第11®，其係顯*轉動標籤天線的方 向（¢)所得到的量測讀取距離結果圖，由圖中可發現使 用本發明該圓極化微帶天線之標籤在所有角度的讀取距 離皆非常相近，大約&1.4m~1.5me此外，圖中亦可發 現’當標藏天線貼附於鋼板時’其讀取距離會隨著銅板 面積的增加而增加’在本實施例中，標蕺天線貼附在4〇 X40cm2銅板的讀取距離可違到'2. 45m，證明本發明之該 圓極化微帶天線具有良好圓極化輻射，且非常適合應用 在金屬物件。 本發明之保護範圍當視後附之申請嫁利範圍所界定 者為準，任何熟知此項技藝者，在不脫離_發明之精神 和範圍内所作之任何變化與修改，均屬於本發明之保護 範圍。 【圖式簡單說明】 [0006] 第1圖：依據本發明之第一較佳實施例，—種無線射頻辨 識標籤之圓極化微帶天線結構示意圖。 第2圖：該圓極化微帶天線結構側視圖。 第3 A至3 B圖.該圓極化微帶天線之量測與模擬輸入阻抗 結果圖。 第4圖：該圓極化微帶天線之量測與模擬返回損失結果圖 098140456 表單編號A0101 第10頁/共24頁 0982069427-0 201126421 第5A至5B圖：改變第一間距G1及第二間距G2所得之輸入 阻抗結果圖。 第6圖：改變第一間距G1及第二間距G2所得之返回損失結 果圖。 第7A至7B圖：改變第一導線部之第一長度L1及第二導線 部之第二長度L2所得之輸入阻抗結果圖。 第8圖：改變第一導線部之第一長度L1及第二導線部之第 二長度L2所得之返回損失結果圖。 第9圖：改變第一導線部之第一長度L1及龛二導線部之第 二長度L2所得之模擬轴比結果圖。The wavelength of the antenna working center frequency, the imaginary part of the antenna input impedance, and the ZQ is the characteristic impedance of the microstrip line 30. Preferably, the first length of the first lead portion 31. The L1 is not equal to the second length L2 of the second lead portion 32 such that the low frequency mode and the high frequency mode have relatively close radiant electric field strengths, thereby obtaining better circularly polarized radiation. 098140456 The microstrip line 30 and the light-emitting metal layer 20 have a first pitch g 1 and a second pitch Gf, wherein the first pitch G1 is flat between the microstrip lines 3 a wire portion 31 is interposed between the first side edge 2a of the radiant metal layer 20, and the second pitch G2 is between the second wire portion 32 of the microstrip line 30 and the radiant metal layer 20. Between the second side edges 20b. In this embodiment, the first pitch G1 and the second pitch G2 are used to adjust the real input impedance of the antenna such that the real input impedance of the antenna is approximately equal to the real impedance of a wafer 50. Preferably, 5毫米至3毫米之间。 The first pitch G1 is equal to the second pitch G2, and its value is between 1. 5mm to 3mm. Referring to FIGS. 1 and 2 again, the ground layer 40 is formed on the lower surface i〇b of the substrate 10, and the via form 11 is connected to the conductive form No. A0101, page 7 of 24, 0982069427-0, 201126421. The impedance matching method of the circularly polarized microstrip antenna of the present invention will be described in detail below. In the present embodiment, the tag wafer used therein has an impedance of 13.5 ΠΙΟ Ω at 925 MHz, and in order to provide good power transmission between the circularly polarized microstrip antenna and the standard wafer, the circle must be The input impedance of the polarized microstrip antenna at 925MHz is designed to be 13.5+jllOQ' to achieve a common match. In the present invention, the first length L1 of the first wire portion 31, the second length L2' of the second wire portion 32, the microstrip line width w, the first pitch g1, and the second pitch G2 are selected. To achieve impedance matching, wherein the imaginary input impedance of the circularly polarized microstrip antenna is mainly from the reactance value of the microstrip line 30, and the first length L1 of the first lead portion 31 that provides the reactance j no Q and the The second length L2 of the second lead portion 32 can be determined by the formula (1), and it is known from the formula (1) that LI, L2, and w have an infinite number of solutions. 2毫米。 In this embodiment, it is selected w = l. 5mm and Ll + L2 = 73. 2mm. The actual input impedance of the circularly polarized microstrip antenna is 13.5 Ω, which can be achieved by adjusting the first pitch G1 and the second pitch G2. Please refer to Figures 3A and 3B, which show the simulated and measured input impedance results of the circularly polarized microstrip antenna, and Figure 4 shows the corresponding return loss results. In addition, Fig. 3B is a comparison of the reactance calculation values of only the microstrip line 30 (excluding the 6 Xuan light metal layer 20). It can be found from Fig. 3B that the analog and measured reactance values of the circularly polarized microstrip antenna near 925 MHz are about j 110 Ω, which is in good agreement with the calculated value of only the microstrip line, which proves that the circularly polarized microstrip antenna The input reactance is mainly controlled by the parameters of the microstrip line 3〇. In addition, it can be seen from Figures 3A, 3B and 4 that the simulation and measurement results of the circularly polarized microstrip antenna are quite close. Please refer to FIGS. 5A, 5B and 6 098140456 Form No. A0101, page 8 / 24 pages 0992069427-0 201126421, which respectively show the measured input impedance and return loss obtained by changing the first spacing G1 and the second spacing G2. The result is shown in FIG. 5A. When the first pitch G1 and the second pitch G2 are equal to 0.5 mm, since the coupling strength of the microstrip line 30 and the radiation metal layer 20 is large, the circular polarization The input resistance of the microstrip antenna reaches 57 Ω, which makes the circularly polarized microstrip antenna difficult to complete impedance matching; conversely, as the first pitch G1 and the second pitch G 2 increase, the input resistance decreases. Good impedance matching can be achieved when G1=G2 = 1. 5~3mm. Please refer to FIGS. 7A and 7B , which are diagrams for measuring the input impedance obtained by changing the first length L1 of the first lead portion 31 and the second length L2 of the second lead portion 32. FIG. 8 is Corresponding return loss result graph, and FIG. 9 is its corresponding analog axis ratio: result circle, in the embodiment: the first length L1 of the first lead portion 31 and the second lead portion The sum of the second lengths L2 of 32 is fixed to li + L2 = 73. 2 mm. In Figure 7A, it can be found that there are two adjacent modes excited near 925 MHz. Due to the present invention::: | Low-frequency mode of a circularly polarized microstrip antenna: generated by the x-direction: 3⁄4 stream, and the high-frequency mode The state is generated by the current in the y direction. Therefore, from the curves of Ll = 26nini and L2 = 47. 2mm, it can be observed that the peak resistance of the low-frequency mode is small and the peak resistance of the high-frequency mode is large. This is because L1 is shorter, so that the coupling strength of the low-frequency mode is smaller; and L2 is longer, so that the coupling strength of the high-frequency mode is larger. Moreover, when L1 is increased to 32 mm and L2 is decreased to 41. 2 mm, the peak resistances of the two adjacent modes are approximately equal, and the intensity of the radiated electric fields of two adjacent modes is expected to be close, and a better circularly polarized radiation can be obtained. . In addition, from the 9th figure, it can also be seen that when Ll=32mm and L2=41. 2mm, the axial ratio at 925MHz is lower than idB, and the circular polarization width of the 3dB axial ratio is 92 928MHz, which is in line with UHF. RFID working frequency band 098140456 Form number A0101 Page 9 / Total 24 page 0982069427-0 201126421 922~928MHz. Please refer to FIGS. 1A and 1〇β, which show the input impedance results of the circularly polarized microstrip antenna of the present invention attached to a 4Q x4Qcm2 metal copper plate, which can be found when the label is attached to the metal copper plate by the ® Its antenna impedance value hardly changes. Finally, β refers to the 11th, which shows the measurement read distance result obtained by rotating the direction of the tag antenna (¢). It can be found from the figure that the circularly polarized microstrip antenna of the present invention is used at all angles. The reading distances are very similar, about &1.4m~1.5me In addition, the figure can also be found that 'when the standard antenna is attached to the steel plate' its reading distance will increase with the increase of the copper plate area. In the embodiment, the reading distance of the standard antenna attached to the 4〇X40cm2 copper plate may be in violation of '2.45m, which proves that the circularly polarized microstrip antenna of the present invention has good circularly polarized radiation and is very suitable for application to metal. object. The scope of the present invention is defined by the scope of the application for the benefit of the application, and any changes and modifications made by those skilled in the art without departing from the spirit and scope of the invention are protected by the present invention. range. BRIEF DESCRIPTION OF THE DRAWINGS [0006] FIG. 1 is a schematic view showing the structure of a circularly polarized microstrip antenna of a radio frequency identification tag according to a first preferred embodiment of the present invention. Figure 2: Side view of the circularly polarized microstrip antenna structure. Figure 3A to 3B. Measurement of the circularly polarized microstrip antenna and analog input impedance results. Figure 4: Measurement and simulation return loss results of the circularly polarized microstrip antenna 098140456 Form No. A0101 Page 10 / Total 24 pages 0992069427-0 201126421 Figure 5A to 5B: Change the first spacing G1 and the second spacing The input impedance result graph obtained by G2. Fig. 6 is a graph showing the return loss result obtained by changing the first pitch G1 and the second pitch G2. Figs. 7A to 7B are diagrams showing input impedance results obtained by changing the first length L1 of the first lead portion and the second length L2 of the second lead portion. Fig. 8 is a graph showing the return loss result obtained by changing the first length L1 of the first lead portion and the second length L2 of the second lead portion. Fig. 9 is a graph showing the results of the simulation of the axial ratio obtained by changing the first length L1 of the first lead portion and the second length L2 of the second lead portion.
第10A至10B圖：該圓極化微帶天線結構貼附於40 X 40cm 2金屬銅板時之輸入阻抗結果圖。 第11圖：轉動標籤天線的方向（ψ)所得到的量測讀取距 離結果圖。 第12圖：依據本發明之一具體實施例，另一種無線射頻 辨識標籤之圓極化微帶天線結構示意圖。 【主要元件符號說明】 [0007] 10 基板 10a上表面 11導通孔 20輻射金屬層 10b 下表面 20a第一侧邊 20b 第二側邊 20c第三側邊 21槽孔 30微帶線 20d 第四侧邊 表單編號A0101 第11頁/共24頁 098140456 0982069427-0 201126421 31第一導線部 30a饋入端 3 2第二導線部 4 0接地層 50晶片 G1第一間距 G2第二間距 L1第一長度 L2第二長度 w微帶線寬度 098140456 表單編號A0101 第12頁/共24頁 0982069427-0Figures 10A to 10B are diagrams showing input impedance results of the circularly polarized microstrip antenna structure attached to a 40 X 40 cm 2 metal copper plate. Figure 11: The measurement read distance result obtained by rotating the direction of the tag antenna (ψ). Figure 12 is a block diagram showing the structure of a circularly polarized microstrip antenna of another radio frequency identification tag according to an embodiment of the present invention. [Main component symbol description] [0007] 10 substrate 10a upper surface 11 via hole 20 radiating metal layer 10b lower surface 20a first side 20b second side 20c third side 21 slot 30 microstrip line 20d fourth side Side form number A0101 Page 11 / Total 24 page 098140456 0982069427-0 201126421 31 First lead portion 30a Feed end 3 2 Second lead portion 40 Ground layer 50 Wafer G1 First pitch G2 Second pitch L1 First length L2 Second length w microstrip line width 098140456 Form number A0101 Page 12 / Total 24 page 0992069427-0