TW201001964A - Channel estimation device and method for orthogonal frequency division multiplexing system - Google Patents
Channel estimation device and method for orthogonal frequency division multiplexing system Download PDFInfo
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201001964 九、發明說明: 【發明所屬之技術領域】 本發明係有關於一種在無線通信系統中的信號接收技 術’特別是有關於一種在正交頻分複用(orthogonal frequency division multiplexing ; OFDM)系統中通道估計的 方法及裝置。 【先前技#f】 f 近年來,正交頻分複用技術以其在高速資料傳輸中具 有的良好的抗多徑干擾的特性以及較高的頻譜利用率等優 點得到了廣泛的應用,並且可以支援多用戶的多址接入。 尤其是與頻分多址接入相結合的正交頻分多址接入技術可 以更有效地控制每個用戶的資料速率,從而提高了通道利 用率。例如在802. 16e標準即微波存取全球互通接入系統 (WiMAX » Worldwide Interoperability for Microwave ( Access)中’物理層引入了正交頻分複用與多入多出天線相 結合的技術,使傳輸速度成倍提高的同時能夠支援一定的 移動性。可以在載波帶寬為20MHz時提供頻譜範圍為2GHz 〜11GHz的非視距資料傳輸,其資料速率可以達到 75Mbps,覆蓋半徑為10公里,移動速度最高玎達120 km/h。 然而在無線通信系統中’資料的傳輸往往受到頻率選 擇性衰落通道以及通道時變特性的影響。在無線終端中需 要對以上因素對資料傳輸過程中可能的影響進行估計。尤 其是在信號接收端,需要獲得通道資訊用於量化和解碼。 VTU08-0002/ 0608A-A41642-TW/Final 201001964 因此二通^/古計的方法成為無線資料接收處理的關鍵 在目則主要使用的通道估計方法中, 過二維維_波的方法獲得通道資訊,並基於== =抽頭係數進行有限脈衝回臟。這種方法;;= 反映通道的時變特性以及頻率選擇性衰落的影響。 然而在求維納濾波器的濾波係數時 大。雖然可以降低二維維納據波器的複雜度二 上導頻子載波的數目仍需要較大的運算量便 :效地獲得維納渡波器的係數相關資訊成為一二 【發明内容】 本發:針對正交頻分複用系統中導頻 Symb〇1)符號的特性提出了-種快速簡便地進行通道估叶 的方法’很好地解決了财通道估計演算法中的問題。 本發明提出了-種用於正交頻分複用系統中通道估計 =法’該方法接收多個正交頻分複用符號,產生通道估 =貝訊’該方法包括根據多個正交頻分複用符號中的多個 導頻信號位置’將該多個導頻信號劃分為第一導頻华,盆 中該第-導頻集中的每個導頻信號間為第一相對位^·對 中的導頻信號進行第一方向的第一估值係數 ^ ^第I純純’對第—導趣中的導頻信號進 方向的第二估值係數計算,產生第二估值係數,根 一估值係數以及該第二估值係數,獲得導頻信號估 值、、、。果,以及根據該導頻信號估值結果獲得該通道估計資 VTU08-0002/ 0608A-A41642-TW/Final 201001964 訊0 的裝t發^出了—種用於正交頻分複用系統中通道估計 許^置接收多個正交頻分複用符號,產生通道估 交該裂置包括:導頻信號選擇器’用於根據多個正 書/分中的多個導頻信號位置’將多個導頻信號 」刀為導頻集’其中該第—導頻集中的每 間為第一相對位置;導頻 ’。唬 梦撰㈣m ㈣W估倾組’錢到該導頻信 二第對該第一導頻集中的導頻信號進行第-方 的導頻信號進行第二方向的第二估值係數計算,=員 數'^彳^^’根據該第—估值係數以及該第二估值係 ^獲仔導頻信號估值結果;以及通道 果组,—頻信號二 置,談7 —種用於正交頻分複用系統的接收裝 資料·接收信號前端處理模組,用於將接收 多個正交頻分複用,通道估計裝置;接收 號魏Ρ JU 生通道估計資訊,·以及接收信 於根據該通道估計資訊以及多個正交 頻刀稷用符號進行資料解碼。 高運估計裝置以及方法,可以大幅提 的缺陷。了現有正交頻分複用系統通道估計技術 :’、、=本發月之特徵和優點能更明顯易懂,下文特舉出201001964 IX. Description of the Invention: [Technical Field] The present invention relates to a signal receiving technique in a wireless communication system, particularly relating to an orthogonal frequency division multiplexing (OFDM) system. Method and device for estimating the middle channel. [Previous technology #f] f In recent years, Orthogonal Frequency Division Multiplexing (OFDM) technology has been widely used for its advantages of high multipath interference resistance and high spectrum utilization in high-speed data transmission. Can support multiple access for multiple users. In particular, the Orthogonal Frequency Division Multiple Access (OFDM) technology combined with frequency division multiple access can more effectively control the data rate of each user, thereby improving channel utilization. For example, in the 802.16e standard, the Worldwide Interoperability for Microwave (Access) system, the physical layer introduces a combination of orthogonal frequency division multiplexing and multiple input and multiple output antennas to enable transmission. The speed can be doubled while supporting a certain mobility. It can provide non-line-of-sight data transmission with a spectrum range of 2GHz to 11GHz when the carrier bandwidth is 20MHz. The data rate can reach 75Mbps, the coverage radius is 10km, and the moving speed is the highest.玎120 km/h. However, in wireless communication systems, the transmission of data is often affected by the frequency selective fading channel and the time-varying characteristics of the channel. In the wireless terminal, it is necessary to estimate the possible influence of the above factors on the data transmission process. Especially at the signal receiving end, channel information needs to be obtained for quantization and decoding. VTU08-0002/ 0608A-A41642-TW/Final 201001964 Therefore, the method of two-way ^/古计 becomes the key to wireless data receiving processing. In the channel estimation method used, the method of obtaining the channel information by the two-dimensional dimension-wave method is based on the === tap system Perform a finite pulse back to the dirty. This method;; = reflects the time-varying characteristics of the channel and the effect of frequency selective fading. However, the filter coefficient of the Wiener filter is large. Although it can reduce the two-dimensional Wiener The number of pilot subcarriers on the complexity two still needs a large amount of computation: it is effective to obtain the coefficient related information of the Wiener wave transformer. [Inventive content] This is a guide for Orthogonal Frequency Division Multiplexing (OFDM) systems. The characteristics of the frequency Symb〇1) symbol proposes a fast and simple method for channel estimation, which solves the problem in the estimation algorithm of the financial channel. The present invention proposes a channel estimation method for an Orthogonal Frequency Division Multiplexing (OFDM) system. The method receives a plurality of orthogonal frequency division multiplexing symbols, and generates a channel estimate=beta. The method includes multiple orthogonal frequencies according to the method. And dividing a plurality of pilot signal positions in the multiplexed symbols into the first pilot, and each pilot signal in the first pilot set in the basin is a first relative bit ^· The first pilot coefficient of the first direction is calculated by the first estimation coefficient of the first pilot signal in the first direction, and the second estimation coefficient of the pilot signal in the first direction is calculated to generate a second estimation coefficient. The root-valued coefficient and the second coefficient of estimation obtain a pilot signal estimate, , , . And, according to the estimation result of the pilot signal, the channel estimation VTU08-0002/ 0608A-A41642-TW/Final 201001964 0 is sent out - the channel used in the orthogonal frequency division multiplexing system Estimating that a plurality of orthogonal frequency division multiplexing symbols are received, and generating a channel estimate for the split includes: the pilot signal selector 'for multiple pilot signal positions according to multiple books/minutes' will be The pilot signal "knife is a pilot set" where each of the first pilot sets is a first relative position; pilot '.唬 撰 ( 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' ' '^彳^^' according to the first-estimation coefficient and the second evaluation system, the pilot signal estimation result; and the channel fruit group, the frequency signal two sets, and the seventh type is used for orthogonal frequency The receiving and receiving data and receiving signal front-end processing module of the sub-multiplexing system is configured to receive a plurality of orthogonal frequency division multiplexing, channel estimating devices, receiving the Wei Wei JU channel estimation information, and receiving the signal according to the Channel estimation information and multiple orthogonal frequency masks are used for data decoding. The high-speed estimation device and method can greatly reduce the defects. The channel estimation technology of the existing orthogonal frequency division multiplexing system: ',, = the characteristics and advantages of this month can be more obvious and easy to understand, the following is a special
VTU0S-0002/0608A-A4J642-TW/FinaI 8 201001964 較佳實施例,並配合所附圖式,作詳細說明如下: 【實施方式】 在以下附圖以及實施例中,本發明以在微波存取全球 互通接入系統802.16d或802.16e版本中的應用為一個實施 例’並以母個符號包括1〇24個子载波的微波存取全球互通 接入系統為例。可以理解到,子载波的數目以及其他設定 並非用以限定於該系統,僅為使得本領域技術人員可以更 好地理解本發明,並且本發明也不局限為微波存取全球互 通接入系統,其他任何與本發明所需條件相符合的系統或 與本文所述的通道特性相似的系統並使用本文所述的方法 均可以作為本發明的一種實施方式。VTU0S-0002/0608A-A4J642-TW/FinaI 8 201001964 The preferred embodiment, with reference to the accompanying drawings, is described in detail below: [Embodiment] In the following drawings and embodiments, the present invention is used in microwave access. The application in the 802.16d or 802.16e version of the Global Interworking Access System is an embodiment 'and a microwave access global interworking access system in which the parent symbol includes 1 〇 24 subcarriers is taken as an example. It is to be understood that the number of subcarriers and other settings are not intended to be limited to the system, only to enable those skilled in the art to better understand the present invention, and the present invention is not limited to the microwave access global interworking access system. Any other system consistent with the conditions required by the present invention or systems having similar channel characteristics as described herein and using the methods described herein can be considered as an embodiment of the present invention.
第1圖為無線通信系統尤其是微波存取全球互通接入 接收系統的方塊圖。微波存取全球互通接入接收系統ι〇〇 包括天線接輯組11G、正交頻分複⑽調模組⑽、通道 估計模組130以及倾後歧理她⑽。在微波存取全 球互通接人接㈣統中’天線接收模組㈣可以包括 2發2收的天線陣列或者包括單發單收或多發單收等天線 妾收模組110還可以包括將射頻信號轉換為基 =说的裝置,例如模/數轉換器、濾波器等裝置。在微波 子取全球互通接人接收系統中’天線接收模組⑽輸出正 交=用符號以及天線判別資訊,正交頻分複用符號可 二個I载波以及多個用戶資料,其中子載波所傳輸 括二、关::為導頻信號以及資料信號。天線判別資訊包 田考X k天線為單天線時,單天線標誌位元為是,當發送 VTU08-0002/0608A.A41642.TW/Final 9 201001964 天線為多天線時’單天線標諸位it為否。正交頻分複用解 調益12G連接到天線接收模組iiQ,在正交頻分複用解調 器120中可以使用快速傅立葉變換(FFT)實現解調’其解調 所付到的導頻資訊送至通道估計模組13G,並將資料資訊 运至貧料後端處理模組M〇。後文將對通道估計模組13〇 做詳細描述。資料後端處理模組140接收通道估計模組130 產生的結果以及正交頻分複用解調模組12〇的解調結果。 需要注意的是’當天線接收模組11G使關如2發2收等 多發多收天線陣列時’資料後端處理模組14G需要對所接 收到資料信號進行空時解碼(STC,space_time c〇ding)。當 天線接收模組11G使用例如丨發2收等單發多收天線陣列 時料後端處理模組14〇需要對所接收到資料信號進行 合併操作。當天線接收模組u〇使用例如丨發丨收天線陣 列時,=貝料後端處理模組14〇無需針對天線陣列進行資料 整理或空時解碼。當然資料後端處理模組14〇還可以對根 據通道估計資訊進行解碼後㈣料進行正交相移鍵控或正 父幅度调製與解調、混合自動重發請求(HARQ,HybridFigure 1 is a block diagram of a wireless communication system, particularly a microwave access global interworking access receiving system. The microwave access global intercom access receiving system ι includes an antenna set 11G, an orthogonal frequency division (10) modulation module (10), a channel estimation module 130, and a post-discrimination her (10). In the microwave access global intercommunication (4) system, the antenna receiving module (4) may include an antenna array of 2 transmissions and 2 receptions, or an antenna receiving module 110 including single-issue single-receiving or multiple-receiving single-receiving, and may also include a radio frequency signal. A device that converts to a base = say, such as an analog-to-digital converter, a filter, and the like. In the microwave sub-global intercommunication receiving system, the antenna receiving module (10) outputs orthogonal = symbol and antenna discriminating information, and the orthogonal frequency division multiplexing symbol can have two I carriers and multiple user data, wherein the subcarriers Transmission includes two, off:: for the pilot signal and the data signal. When the antenna identification information packet Tianke X k antenna is a single antenna, the single antenna flag bit is YES. When transmitting VTU08-0002/0608A.A41642.TW/Final 9 201001964 antenna is multi-antenna, 'single antenna target is it no . The Orthogonal Frequency Division Multiplexing (OFDM) demodulation 12G is connected to the antenna receiving module iiQ, and the Orthogonal Frequency Division Multiplexing Demodulator 120 can be used to perform demodulation using the Fast Fourier Transform (FFT). The frequency information is sent to the channel estimation module 13G, and the data information is sent to the poor material back-end processing module M〇. The channel estimation module 13A will be described in detail later. The data backend processing module 140 receives the result generated by the channel estimation module 130 and the demodulation result of the orthogonal frequency division multiplexing demodulation module 12A. It should be noted that when the antenna receiving module 11G turns off the multi-transmitting antenna array such as 2 transmissions and 2 receptions, the data back-end processing module 14G needs to perform space-time decoding on the received data signals (STC, space_time c〇). Ding). When the antenna receiving module 11G uses a single-shot multi-receiver antenna array such as burst 2, the material back-end processing module 14 needs to perform a merge operation on the received data signals. When the antenna receiving module 〇 uses, for example, a burst antenna array, the hopper back-end processing module 14 does not need to perform data sorting or space-time decoding for the antenna array. Of course, the data backend processing module 14 can also perform quadrature phase shift keying or positive parent amplitude modulation and demodulation, hybrid automatic repeat request (HARQ, Hybrid) after decoding the channel estimation information.
Automatic Repeat request)、解交織以及通道解碼等操作。 第2圖為第1圖中通道估計模組的示意圖。通道估計 模、、且130包括V頻L號選擇器21 〇、初始值計算器220、導 頻b號估值模組230以及通道估計資訊計算器24〇。導頻 信號選擇ϋ 210根據接㈣正交頻分複料頻信號的位置 以及天線的判別結果將導頻信號分為可變導頻集和固定導 頻集分別進行相關係數的計算,或者將導頻信號看成一個 VTU08-0002/ 0608A-A41642-TW/Final 201001964 整體進行相關係數的計算。對於多個發送天線的情況,導 頻信號選擇器21〇將與同一發送天線相對應的導頰信號劃 分為一個導頻集。 初始值計算器220接收來自導頻信號選擇器21〇的導 頻#號並連接到導頻信號估值模組23〇,初始值計算器 為導頻信號估值模組230提供了進行信道估計的初始值。 在一個實施例中’初始值錯誤!物件無法用編輯功能變數 代碼來建立。,其中Y為接收到的導頻信號,X為已知發 送導頻信號。 " 導頻信號估值模組23〇根據導頻信號選擇器21〇產生 的導頻集的劃分結果以及初始值計算器220產生的信道估 計的初始值對導頻信號進行時域以及頻域上的相關係數估 值計算。其計算公式為: — ~ k')Trmx Af) ....................................(1) rt{f- V) = ··,·;·.·”.-------------(2) 其中錯誤!物件無法用編輯功能變數代碼來建立。,錯 誤!物件無法用編輯功能變數代碼來建立。分別為頻域以 及時域方向的導頻信號,錯誤丨物件無法用編輯功能變數 代瑪來建立。為最大時延擴展,錯誤丨物件無法用編輯功 能變數代碼來建立。為最大多普勒頻移,錯誤!物件無法 用編輯功能變數代碼來建立。為正交頻分複用符號間隔 (Spacing)’錯誤!物件無法用編輯功能變數代碼來建立。 為正交頻分複用符號長度。 通道估計資訊計算器24〇連接到導頻信號估值模組 VTU08-0002/ 0608A-A41642-TW/Final 201001964 230’根據導頻信W = R(R+盖1)_1號估計模組230求得的濾 波係數,透過生成通道估計資訊,完成通道估計運算。其 中β為與調製方式相關的係數,錯誤!物件無法用編輯功能 變數代碼來建立。’SNR為導頻信噪比,I為單位矩陣,R 為相關係數矩陣,錯誤!物件無法用編輯功能變數代碼來 建立。,即R受到時域以及頻域二維變數的影響。 以下將以微波存取全球互通接入系統的完全使用子通 道模式(FUSC,Fully Usable Shared Channels)、部分使用子 通道模式(PUSC,Partially Usable Shared Channels)以及使 用空時碼的部分使用子通道模式(STC PUSC)等工作模式 結合具體示例對本發明做進一步的描述。 完全使用子通道模式 在微波存取全球互通接入的完全使用子通道模式下, 每個子通道包含的子載波擴散在整個頻帶範圍内,發射端 可以使用全部子通道,獲得最大的頻率分集。 下行鏈路的完全使用子通道模式調用所有子通道,首 先在可用子載波中指定導頻子載波,然後將剩下的資料子 載波分成子通道。導頻信號可以分為兩個可變導頻集 VariableSet#0、VariableSet#l 、以及固定導頻集 ConstantSet#0、ConstantSet#l ’ 表 1 中注明 了在完全使用 子通道模式下導頻集的個數和位置。 表1 1024個子載波,下行鏈路完全使用子通道模式 導頻位置 __ 導頻信號 值 註解 VTU08-0002/ 0608A-A41642-TW/Final 12 201001964Automatic Repeat request), deinterleaving, and channel decoding. Figure 2 is a schematic diagram of the channel estimation module in Figure 1. The channel estimation module, and 130 includes a V-frequency L-number selector 21 〇, an initial value calculator 220, a pilot b-number evaluation module 230, and a channel estimation information calculator 24〇. The pilot signal selection ϋ 210 divides the pilot signal into a variable pilot set and a fixed pilot set according to the position of the (four) orthogonal frequency division complex frequency signal and the discrimination result of the antenna, respectively, or performs correlation coefficient calculation, or guides The frequency signal is regarded as a VTU08-0002/ 0608A-A41642-TW/Final 201001964. The correlation coefficient is calculated as a whole. In the case of a plurality of transmitting antennas, the pilot signal selector 21 划 divides the guide signal corresponding to the same transmitting antenna into one pilot set. The initial value calculator 220 receives the pilot # number from the pilot signal selector 21A and is connected to the pilot signal evaluation module 23, and the initial value calculator provides channel estimation for the pilot signal evaluation module 230. The initial value. In one embodiment 'the initial value is wrong! Objects cannot be created with the edit function variable code. Where Y is the received pilot signal and X is the known transmitted pilot signal. " The pilot signal evaluation module 23 进行 performs time domain and frequency domain on the pilot signal according to the division result of the pilot set generated by the pilot signal selector 21 以及 and the initial value of the channel estimation generated by the initial value calculator 220 The correlation coefficient is calculated on the estimate. Its calculation formula is: — ~ k')Trmx Af) .................................... 1) rt{f- V) = ····;···..-------------(2) where the error! The object cannot be created with the edit function variable code. Objects cannot be created with the edit function variable code. The pilot signals in the frequency domain and the time domain direction respectively, the error object cannot be created with the edit function variable. For the maximum delay spread, the error object cannot be edited. Variable code to establish. For maximum Doppler shift, error! Object cannot be created with the edit function variable code. For Orthogonal Frequency Division Multiplexing (Spacing) 'Error! Object cannot be created with the edit function variable code. The symbol length is orthogonally frequency division multiplexed. The channel estimation information calculator 24〇 is connected to the pilot signal evaluation module VTU08-0002/ 0608A-A41642-TW/Final 201001964 230' according to the pilot letter W = R (R+ cover 1) The filter coefficient obtained by the estimation module 230 is calculated by generating channel estimation information, wherein β is a coefficient related to the modulation mode, and the error! Created with the edit function variable code. 'SNR is the pilot signal-to-noise ratio, I is the unit matrix, R is the correlation coefficient matrix, error! The object cannot be created with the edit function variable code. That is, R is affected by the time domain and the frequency domain. The effect of dimension variables. The following will use the microwave access global interoperable access system's Fully Sub-Channel (FUSC), Partially Usable Shared Channels (PUSC) and Space Time Codes. The present invention is further described in part by using a working mode such as sub-channel mode (STC PUSC) in conjunction with a specific example. Fully using the sub-channel mode in the fully used sub-channel mode of the microwave access global interworking access, the sub-carriers included in each sub-channel Diffusion is spread over the entire frequency band, and all the sub-channels can be used by the transmitter to obtain the maximum frequency diversity. The full-subchannel mode of the downlink calls all sub-channels, first the pilot sub-carriers are specified in the available sub-carriers, and then the remaining The following subcarriers are divided into subchannels. The pilot signal can be divided into two variables. The frequency set VariableSet#0, VariableSet#l, and the fixed pilot set ConstantSet#0, ConstantSet#l' indicate the number and position of the pilot set in the fully used subchannel mode. Table 1 1024 subcarriers , the downlink uses the subchannel mode pilot position completely __ pilot signal value annotation VTU08-0002/ 0608A-A41642-TW/Final 12 201001964
VariableSet#0 36 0、24、48、72、96、120、144、168、 192、216、240、264、288、312、336、 360、384、408、432、456、480、504、 528、552、576、600、624、648、672、 696、720、744、768、792、816、840 ConstantSet#0 6 72*(2*n+k)+9 ;其中 k=0,n=0〜5,當 導頻子載波索引經過上式計算後,將 可包含DC子載波 VariableSet#l 35 0、24、48、72、96、120、144、168、 192、216、240、264、288、312、336、 360、384、408、432、456、480、504、 528、552、576、600、624、648、672、 696、720、744、768、792、816、840 ConstantSet#l 5 72*(2*n+k)+9 ;其中 k=l , n=0〜4,當導頻子載波索引經過上式 計算後,將可包含DC子載波 可以看到, —jtb 導頻的位置保持不變即固定導頻集, 而另一些導頻的位置滿足以下公式’即可變導頻集: 讎顯辦灘代51^®^ 〇 根據表1中的導頻信號,可以根據錯誤!物件無法用 編輯功能變數代碼來建立。獲得可變位置導頻信號以及固 定位置導頻信號的初始估計值。其中γ為接收到的導頻信 號,X為已知的發送導頻信號。 第3A圖為在微波存取全球互通接收系統中單發單收 VTU08-0002/ 0608A-A41642-TW/Final 13 201001964 天線完全使用子通道模式下©定位置導頻以及可變位置導 f的Γ意圖。可以看到,在可變位置導頻集中,每兩個相 = 導頻^號位置間隔6個子載波’而同—個符號的 相隔12個子載波。即第3A圖中偶數符號與奇數 帛最小間隔為6,偶數符號與奇數符號中的相鄰 2別Η隔為12。在其他的實施例中,其導頻間隔也可 以為其他數目。 一立第3Β圖為在完全使用子通道模式下固定位置導頻的 ^f可以看到,圖中偶數符號與奇數符號中的導頻位 為在微波存取全球互通接㈣統中的完全使用 、首下,該系統對接收的正交頻分複用符號進行通 Ϊ = Γ流程實施例。在步驟410中,將正交頻分複 =據^現位置劃分為固定位置導頻集以及 導頻集。其導頻信號位置可以參見第Μ圖以及VariableSet#0 36 0, 24, 48, 72, 96, 120, 144, 168, 192, 216, 240, 264, 288, 312, 336, 360, 384, 408, 432, 456, 480, 504, 528, 552, 576, 600, 624, 648, 672, 696, 720, 744, 768, 792, 816, 840 ConstantSet#0 6 72*(2*n+k)+9; where k=0, n=0~ 5. When the pilot subcarrier index is calculated by the above formula, it may include DC subcarriers VariableSet#l 35 0, 24, 48, 72, 96, 120, 144, 168, 192, 216, 240, 264, 288, 312, 336, 360, 384, 408, 432, 456, 480, 504, 528, 552, 576, 600, 624, 648, 672, 696, 720, 744, 768, 792, 816, 840 ConstantSet#l 5 72 *(2*n+k)+9; where k=l, n=0~4, when the pilot subcarrier index is calculated by the above formula, it can contain DC subcarriers to see, the position of the —jtb pilot The constant set of pilots remains unchanged, while the positions of other pilots satisfy the following formula'. The pilot set can be changed: 雠 办 办 代 51 51^®^ 〇 According to the pilot signal in Table 1, it can be based on the error! Objects cannot be created with the edit function variable code. An initial estimate of the variable position pilot signal and the fixed position pilot signal is obtained. Where γ is the received pilot signal and X is the known transmitted pilot signal. Figure 3A shows the single-shot single-receipt VTU08-0002/ 0608A-A41642-TW/Final 13 201001964 in the microwave access global intercommunication receiving system. The antenna uses the sub-channel mode and the variable position guide f in the sub-channel mode. intention. It can be seen that in the variable position pilot set, every two phase = pilot position is separated by 6 subcarriers and the same symbol is separated by 12 subcarriers. That is, in Fig. 3A, the even-numbered symbols and the odd-numbered 帛 minimum interval are 6, and the even-numbered symbols and the odd-numbered symbols in the odd-numbered symbols are separated by 12. In other embodiments, the pilot spacing may also be other numbers. A third figure shows that the fixed position pilot in the fully used subchannel mode can be seen. The pilot bits in the even and odd symbols in the figure are fully used in the global interworking of microwave access (4). First, the system performs a pass-by-turn process instance for the received orthogonal frequency division multiplexing symbols. In step 410, the orthogonal frequency division = data location is divided into a fixed location pilot set and a pilot set. The location of the pilot signal can be seen in the figure and
法。對Si在—個實施财,剌區分·儲存位址的方 法〜^頻㈣的劃分可以採用纟1中所述_分方 導頻特性進行的其他割分也可以為本發明的 在步驟420中對於上述的所有導頻集, 導頻集以及可變位置導難進㈣始_,匕括固定位置 中’採用LS演算法’即根據錯誤,•在一個實施例 數代瑪來建立。獲得導頻信制初始料、用鵠轉功能變 收到的導頻信號’ X為已知的發送導頻信號’。其中Υ為接 VTU08-0002/ 0608A-A41642-TW/Final 14 201001964 可以理解到,劃分固定位置導頻集以及可變位置導頻 集後再進行初始估計為本發明的一種實施方式,在其他實 施例中也可以在步驟420後進行步驟410。 在步驟430中判斷所接收的正交頻分複用符號個數, 當僅收到一個正交頻分複用符號時,進行步驟440。由於 僅收到一個正交頻分複用符號,因此無需進行時域濾波, 對於一個正交頻分複用符號來說,每個導頻間隔為12個子 載波,因此根據公式(1)求得維納濾波係數,進而求得係數 資訊。 當步驟430判斷收到多個連續的正交頻分複用符號 時,進行步驟450。在步驟450中,首先根據可變導頻集 的導頻位置進行時財向的㈣錄料錢波,此處即 為一維維納濾波。law. For the division of Si, the method of dividing the storage address, and the division of the frequency (four) may be performed by using the _ splitter pilot characteristic described in 纟1, or may be the step 420 of the present invention. For all of the pilot sets described above, the pilot set and the variable position are difficult to start (4), including the 'LS algorithm' in the fixed position, ie according to the error, • in one embodiment, the generation is established. The pilot signal initial material is obtained, and the pilot signal 'X is a known transmission pilot signal' received by the twirling function. Wherein Υ is connected to VTU08-0002/ 0608A-A41642-TW/Final 14 201001964 It can be understood that initial estimation by dividing the fixed position pilot set and the variable position pilot set is an embodiment of the present invention, in other implementations. In step, step 410 may also be performed in step 410. In step 430, the number of received orthogonal frequency division multiplexing symbols is determined. When only one orthogonal frequency division multiplexing symbol is received, step 440 is performed. Since only one orthogonal frequency division multiplexing symbol is received, no time domain filtering is needed, and for one orthogonal frequency division multiplexing symbol, each pilot interval is 12 subcarriers, and thus is obtained according to formula (1). The Wiener filter coefficient is used to obtain the coefficient information. When it is determined in step 430 that a plurality of consecutive orthogonal frequency division multiplexing symbols are received, step 450 is performed. In step 450, the (four) recording money wave of the time-of-flight is first performed according to the pilot position of the variable pilot set, here a one-dimensional Wiener filter.
在步驟452中’在兩個可變導頻集的同一個子載波位 置進行頻域方向的相關係數計算並濾、波,可以使用盘一個 導頻相鄰的導頻㈣以及該導頻信號自身共三個導頻㈣In step 452, 'the correlation coefficient in the frequency domain direction is calculated and filtered in the same subcarrier position of the two variable pilot sets, and the pilot (four) adjacent to the pilot of the pilot and the pilot signal itself may be used. A total of three pilots (four)
St:也可以選用其他數目的導頻信號進行相關 糸數運具’此處頻域方向的導頻信號選擇為本發明的一個 在步驟454中 進行固定導頻集的計算,在固道 集中,即對每個位置的導頻進行時β 疋導頻 等貝進仃時域方向的維納濾波。 在步驟456中,使用線性插值函數求得盆他 ί = Ϊ結果。由於在本實施例中使用線性插值可以在Ϊ 到準確㈣紐下提“道估計較,可㈣解到在= VTU08-0002/ 〇608A-A41642-TW/Final 15 201001964 實施例中也可以繼續使用頻域方向的維納濾波獲得估計結 果。同時本領域技術人員可以理解到,先進行頻域方向的 估計在進行時域方向的估計亦屬於本發明的範圍,以上步 驟為說明起見。 在步驟460中,根據步驟440或步驟456的結果獲得 通道估計資訊W,其中對應不同導頻集的相關係數的合集 即為所有導頻信號的相關係數,使用所有導頻信號的相關 係數求得通道估計資訊W,從而獲得正確的發送信號。通 道估計資訊W的計算參見第2圖中通道估計資訊計算器 240的描述。 第5圖為在完全使用子通道模式下,當接收到8個連 續的正交頻分複用符號,採用正交相移鍵控編碼模式,1/4 編碼速率,移動速度3km/h時,普通的線性濾波以及改進 的維納濾波與理想情況下的通道估計的性能模擬情況,其 中橫軸為信噪比,單位為dB,縱軸為誤幀率。可以看到, 在極大的降低了運算量開銷並減少了運算時間的情況下, 改進後的維納濾波方法仍然獲得了較好的估計結果。 單天線的部分使用子通道模式 在部分使用子通道模式下,發射端使用一部分子通 道,能夠實現部分頻率重用。 本發明的另一個實施例為當正交頻分複用系統工作於 部分使用子通道模式下時的應用示例。在部分使用子通道 模式中將子載波劃分為若干個互相交織的群(cluster),每個 群包含14個相鄰的子載波。部分使用子通道模式可以應用 VTU08-0002/ 0608A-A41642-TW/Final 16 201001964 於多磁區的情況,但是限制了 一個蜂窩小區中每個磁區内 群的數目。 在本發明的這一實施例中,由於每個資料塊(block)均 包括時域連續的多個正交頻分複用符號,並且在頻域方向 上每14個子載波劃分為一個群。由於每個群在頻域方向上 的長度有限,因此對頻域方向上的改進有限,僅需進行時 域方向上的改進計算。而對於頻域方向,仍然使用線性插 值函數進行其他子載波的估計。因此可以犧牲較少的性能 f 而得到運算的複雜度的大幅降低。 第6圖為在單發單收天線部分使用子通道模式下一個 群的導頻位置示意圖。可以看到在部分使用子通道模式下 無需劃分可變導頻集和固定導頻集。 第7圖為在部分使用子通道模式下本發明的一個方法 流程示例。步驟710與完全使用子通道模式下步驟420完 成的功能相似,在此不作贅述。在步驟720中,對每個群 進行時域方向的相關係數計算。在步驟730中,由於在本 實施例中相鄰導頻間隔為4個子載波,因此認為線性插值 函數可以滿足頻域方向的精度要求。在這種情況下,維納 濾波公式可以簡化為錯誤!物件無法用編輯功能變數代碼 來建立。。當然,在其他實施例中導頻間隔也可以為其他 數目,並且當導頻間隔較少不能進行導頻間隔時,也可以 繼續進行頻域方向的相關係數估值。在步驟740中,根據 相關係數求得通道估計資訊,通道估計資訊的求法與圖3 所述完全使用子通道模式下相似,在此不作贅述。 VTU08-0002/ 0608A-A41642-TW/Final 17 201001964St: Other numbers of pilot signals may be selected for the associated number of carriers. Here, the pilot signal in the frequency domain direction is selected as a fixed pilot set calculation in step 454 of the present invention. That is, the pilot of each position is subjected to Wiener filtering in the time domain direction such as β 疋 pilot. In step 456, the potent ί = Ϊ result is obtained using a linear interpolation function. Since the linear interpolation can be used in the present embodiment, the "channel estimation" can be further improved, and the (four) solution can be continued in the embodiment of = VTU08-0002/〇608A-A41642-TW/Final 15 201001964. The Wiener filtering in the frequency domain direction obtains the estimation result. At the same time, those skilled in the art can understand that it is also within the scope of the present invention to perform the estimation of the frequency domain direction in the time domain direction, and the above steps are for the sake of explanation. 460, the channel estimation information W is obtained according to the result of step 440 or step 456, wherein a combination of correlation coefficients corresponding to different pilot sets is a correlation coefficient of all pilot signals, and channel estimation is obtained by using correlation coefficients of all pilot signals. The information W is obtained to obtain the correct transmission signal. The calculation of the channel estimation information W is described in the channel estimation information calculator 240 in Fig. 2. Fig. 5 shows that 8 consecutive positives are received in the fully used subchannel mode. Inter-frequency division multiplexing symbol, using quadrature phase shift keying coding mode, 1/4 coding rate, moving speed 3km/h, common linear filtering and improved dimension Filtering and performance simulation of channel estimation under ideal conditions, where the horizontal axis is the signal-to-noise ratio, the unit is dB, and the vertical axis is the frame error rate. It can be seen that the computational overhead is greatly reduced and the computation time is reduced. In the case, the improved Wiener filtering method still obtains better estimation results. Part of the single antenna uses the subchannel mode. In the partial subchannel mode, the transmitting end uses a part of the subchannel, which enables partial frequency reuse. Another embodiment of the present invention is an application example when the orthogonal frequency division multiplexing system operates in a partial use subchannel mode. In the partial use subchannel mode, the subcarriers are divided into a plurality of interlaced clusters, each The group contains 14 adjacent subcarriers. Partially using the subchannel mode can apply VTU08-0002/ 0608A-A41642-TW/Final 16 201001964 in the case of multiple magnetic regions, but limits each magnetic region in one cell. The number of groups. In this embodiment of the invention, each block includes a plurality of orthogonal frequency division multiplexing symbols in a time domain, and Each 14 subcarriers are divided into one group in the frequency domain direction. Since each group has a limited length in the frequency domain direction, the improvement in the frequency domain direction is limited, and only an improved calculation in the time domain direction is required. In the frequency domain direction, the linear interpolation function is still used to estimate other subcarriers, so the complexity of the operation can be greatly reduced at the expense of less performance f. Figure 6 shows the subchannel mode in the single-shot single-receiving antenna part. Schematic diagram of the pilot position of a group. It can be seen that there is no need to divide the variable pilot set and the fixed pilot set in the partial use subchannel mode. Fig. 7 is an example of a method flow of the present invention in the partial use subchannel mode. Step 710 is similar to the function performed in step 420 in the sub-channel mode, and will not be described herein. In step 720, a correlation coefficient calculation for the time domain direction is performed for each cluster. In step 730, since the adjacent pilot interval is 4 subcarriers in this embodiment, it is considered that the linear interpolation function can satisfy the accuracy requirement in the frequency domain direction. In this case, the Wiener filtering formula can be reduced to an error! Objects cannot be created with the edit function variable code. . Of course, in other embodiments, the pilot interval may also be other numbers, and when the pilot interval is small and the pilot interval cannot be performed, the correlation coefficient estimation in the frequency domain direction may also be continued. In step 740, the channel estimation information is obtained according to the correlation coefficient, and the channel estimation information is similar to that in the fully used subchannel mode described in FIG. 3, and details are not described herein. VTU08-0002/ 0608A-A41642-TW/Final 17 201001964
第8圖為在部分使用子通道模式下’當接收到8個連 續的正交頻分複用符號,採用正交相移鍵控編碼模式,1/4 編碼速率,移動速度3km/h時’普通的二維維納遽波以及 改進的維納濾波與理想情況下的通道估計的性能仿真情 況’其中橫軸為信噪比,單位為dB,縱軸為誤幀率。可以 看到,在極大的降低了運算量開銷並減少了運算時間的情 況下,改進後的維納濾波方案仍然獲得了較好的估計結果。 使用空時編碼的部分使用子通道模式 在使用空時編碼的部分使用子通道模式下,天線陣歹4 為多發多收天線。輸入資訊首先分成k個符號_会且 [C1,C2,…Ck]。經過空時分組編碼後,在兩個符號週期内 多個天線同時發射k個符號。第1週期’天線丨發cl,天 線2發c2......天線k發送ck;在第2週期,天線丨 天線2發c 1 *......天線k發送ck*,上標*表示取複北輪 、 常認為在一個週期内,時間方向上的通道的狀況不 f生變Figure 8 shows the use of quadrature phase-shift keying coding mode, 1/4 coding rate, and moving speed of 3km/h when partially receiving sub-channel mode. The performance simulation of ordinary two-dimensional Wiener chopping and improved Wiener filtering and channel estimation under ideal conditions' where the horizontal axis is the signal-to-noise ratio in dB and the vertical axis is the frame error rate. It can be seen that the improved Wiener filtering scheme still obtains a good estimation result under the condition that the computational overhead is greatly reduced and the computation time is reduced. The sub-channel mode is used for the part that uses space-time coding. In the sub-channel mode where the space-time coding is used, the antenna array 4 is a multi-shot antenna. The input information is first divided into k symbols _ and [C1, C2, ... Ck]. After space-time block coding, multiple antennas simultaneously transmit k symbols in two symbol periods. In the first cycle 'antenna burst cl, antenna 2 sends c2... antenna k sends ck; in the second cycle, antenna 丨 antenna 2 sends c 1 *... antenna k sends ck*, on The mark * indicates that the north wheel is taken, and it is often considered that the condition of the channel in the time direction does not change in one cycle.
在一個實施例中,採用兩根天線用於資料接收,、 下 採用這一天線陣列設置來闡述本發明的技術特徵。第 及9B圖闡述了使用空時編碼的部分使用子通道模式的 作模式下接收系統所收到的正交頻分複用符號中的道 、』等頻信 號的位置。在第9A及9B圖中,每個位置接收到兩個八 來自不同的接收天線的導頻901 ’ 902 。其中導頻別 自天線0,導頻902來自天線1,導頻變化的週期為4 ^ 號。因此在使用空時編碼的部分使用子通道模式中,< 、T,來自 VTU08-0002/ 0608A-A41642-TW/Final 18 201001964 同一天線的導頻間隔較大,採用二位元維納濾波的話需要 對較f的正_分複㈣號進行運算。 第⑺圖闡述了本發明的一個實施示例。在步驟910中 由第1圖中天線接收模組110判斷接收天線情況,包括發 送/接收天線數目、資料分配規則等參數,在本實施例中使 用了 2個發送天線,因此單天線判斷資訊為否。在步驟920 中提取對應其中—天線的發送資料。在步驟930中,對其 中一根天線的導頻信號採用二維濾波的方法同時求得相關 係數W1,當然在其他實施例中也可以採用第4圖與第7 圖所述的方法獲得相關係數。在步驟940中進行另一天線 的相關係數W2計算。步驟930與步驟940也可以同時進 们* °在步驟950中’根據求得的相關係數W卜W2,采用 二維渡波的彳法得到通道估計資訊。 第Π圖為在使用空時編碼的部分使用子通道模式下, 田接收到8個連續的正交頻分複用符號,採用正交相移鍵 控扁馬模式1/4編喝速率,移動速度3km/h時,普通的二 維維'm以及改進的維納it波與理想情況下的通道估計 的性能仿真情況’其中橫軸為信噪比,單位為dB,縱軸為 誤,1 °可以看到’在極大的降低了運算量開銷並減少了 運算日寸間的情況下,改進後的維納濾波方案仍然獲得了較 好的估計結果。 —雖然本發明已以較佳實施例揭露如上’然其並非用以 限疋本發明’任何所屬技術領域中具有通常知識者,在不 脫離本發明之精神和範圍内,當可作些許之更動與潤飾, VTU08-0002/ 0608A-A41642-TW/Final 19 201001964 '因此本發明之保護範圍當視後附之申請專利範圍所界定者 ‘ 為準。 【圖式簡單說明】 第1圖為一個方塊圖,其根據本發明的一個實施例, 闡述了本發明所述的一個微波存取全球互通接收系統示 例; 第2圖為一個方塊圖,其根據本發明的一個實施例, ( 闡述了本發明所述的微波存取全球互通接收系統中的通道 估計模組示例; 第3A圖為一個示意圖,其根據本發明的一個實施例, 闡述了本發明所述的微波存取全球互通接收系統中完全使 用子通道模式下可變位置導頻信號示例; 第3B圖為一個方塊圖,其根據本發明的一個實施例, 闡述了本發明所述的微波存取全球互通接收系統中完全使 用子通道模式下固定位置導頻信號示例; 第4圖為一個流程圖,其根據本發明的一個實施例, 闡述了本發明所述通道估計方法流程示例; 第5圖為一個示意圖,其根據本發明的一個實施例, 闡述了本發明所述的通道估計方法與傳統通道估計方法的 性能比較示例; 第6圖為一個示意圖,其根據本發明的一個實施例, 闡述了本發明所述的微波存取全球互通接收系統中部分使 用子通道模式的導頻信號位置示例; 第7圖為一個流程圖,其根據本發明的一個實施例, VTU08-0002/ 0608A-A41642-TW/Final 20 201001964 闡述了本發明所述通道估計方法流程示例; 第8圖為一個示意圖,其根據本發明的一個實施例, 闡述了本發明所述的通道估計方法與傳統通道估計方法的 性能比較示例; 第9圖為一個示意圖,其根據本發明的一個實施例, 闡述了本發明所述的微波存取全球互通接收系統中使用空 時編碼的部分使用子通道模式的導頻信號位置示例; 第10圖為一個流程圖,其根據本發明的一個實施例, 闡述了本發明所述的微波存取全球互通接收系統中使用空 時編碼的部分使用子通道模式的通道估計方法流程示例; 以及 第11圖為一個示意圖,其根據本發明的一個實施例, 闡述了本發明所述的通道估計方法與傳統通道估計方法的 性能比較示例。 【主要元件符號說明】 100 :微波存取全球互通接入接收系統; 110 :天線接收模組; 120 :正交頻分複用解調模組; 130 :通道估計模組; 140 :資料後端處理模組; 210 :導頻信號選擇器; 220 :初始值計算器; 230 :導頻信號估值模組; 240 :通道估計資訊計算器; VTU08-0002/ 0608A-A41642-TW/Final 21 201001964 410〜460、710〜740、910〜950 :步驟; 901、902 :導頻。In one embodiment, two antennas are used for data reception, and this antenna array arrangement is used to illustrate the technical features of the present invention. Sections 9 and 9B illustrate the positions of the equal-frequency signals in the orthogonal frequency division multiplexing symbols received by the receiving system using the space-time coded partial subchannel mode. In Figures 9A and 9B, two eight pilots 901 '902 from different receive antennas are received at each location. The pilot is independent of antenna 0, pilot 902 is from antenna 1, and the period of pilot change is 4^. Therefore, in the sub-channel mode in which the space-time coded portion is used, < , T, from VTU08-0002/ 0608A-A41642-TW/Final 18 201001964, the pilot interval of the same antenna is large, and 2-bit Wiener filtering is used. It is necessary to operate on the positive_divided (four) number of f. Figure (7) illustrates an embodiment of the present invention. In step 910, the antenna receiving module 110 in FIG. 1 determines the receiving antenna condition, including the number of transmitting/receiving antennas, data allocation rules, and the like. In this embodiment, two transmitting antennas are used, so the single antenna determining information is no. In step 920, the transmission data corresponding to the antenna is extracted. In step 930, the correlation coefficient W1 is simultaneously obtained by using a two-dimensional filtering method on the pilot signal of one of the antennas. Of course, in other embodiments, the correlation coefficient may be obtained by using the methods described in FIG. 4 and FIG. . The correlation coefficient W2 calculation of the other antenna is performed in step 940. Step 930 and step 940 may also be performed at the same time in step 950. According to the obtained correlation coefficient WBu, the channel estimation information is obtained by using the two-dimensional wave method. The first diagram shows that in the sub-channel mode where the space-time coding is used, the field receives 8 consecutive orthogonal frequency division multiplexing symbols, and uses quadrature phase shift keying to control the flat horse mode 1/4 brewing rate, moving At a speed of 3km/h, the ordinary two-dimensional dimension 'm and the improved Wiener it wave and the performance simulation of the channel estimation under ideal conditions' where the horizontal axis is the signal-to-noise ratio, the unit is dB, and the vertical axis is error, 1 ° It can be seen that the improved Wiener filtering scheme still obtains a good estimation result in the case of greatly reducing the computational overhead and reducing the computational time. </ RTI> <RTIgt; </ RTI> <RTIgt; </ RTI> <RTIgt; </ RTI> <RTIgt; </ RTI> <RTIgt; </ RTI> </ RTI> <RTIgt; And the retouching, VTU08-0002/ 0608A-A41642-TW/Final 19 201001964 'The scope of the invention is therefore defined by the scope of the appended claims. BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a block diagram illustrating an example of a microwave access global intercommunication receiving system according to an embodiment of the present invention; FIG. 2 is a block diagram according to a block diagram An embodiment of the present invention, (an example of a channel estimation module in the microwave access global intercommunication receiving system of the present invention is illustrated; FIG. 3A is a schematic diagram illustrating the present invention according to an embodiment of the present invention; An example of a variable position pilot signal in a subchannel mode is fully used in the microwave access global interworking receiving system; FIG. 3B is a block diagram illustrating the microwave according to the present invention according to an embodiment of the present invention. Example for accessing a fixed-position pilot signal in a sub-channel mode in a global interworking receiving system; FIG. 4 is a flowchart illustrating an example of a channel estimation method according to the present invention according to an embodiment of the present invention; 5 is a schematic diagram illustrating a channel estimation method and a conventional channel estimation according to an embodiment of the present invention. Example of a performance comparison of the method; FIG. 6 is a schematic diagram illustrating an example of a pilot signal position partially using a subchannel mode in the microwave access global interworking receiving system according to an embodiment of the present invention; 7 is a flow chart illustrating an example of the channel estimation method flow of the present invention in accordance with an embodiment of the present invention, VTU 08-0002/0608A-A41642-TW/Final 20 201001964; FIG. 8 is a schematic diagram based on An embodiment of the present invention describes an example of performance comparison between the channel estimation method and the conventional channel estimation method according to the present invention; FIG. 9 is a schematic diagram illustrating the present invention according to an embodiment of the present invention. Example of a pilot signal position using a sub-channel mode in a portion of a microwave access global interworking receiving system using space time coding; FIG. 10 is a flow chart illustrating the microwave according to the present invention, in accordance with an embodiment of the present invention An example of a channel estimation method flow using a sub-channel mode in a portion of a global interworking receiving system that uses space-time coding; And FIG. 11 is a schematic diagram illustrating an example of performance comparison between the channel estimation method and the conventional channel estimation method according to the present invention according to an embodiment of the present invention. [Main Component Symbol Description] 100: Worldwide Interoperability for Microwave Access Access receiving system; 110: antenna receiving module; 120: orthogonal frequency division multiplexing demodulation module; 130: channel estimation module; 140: data back-end processing module; 210: pilot signal selector; : initial value calculator; 230: pilot signal evaluation module; 240: channel estimation information calculator; VTU08-0002/ 0608A-A41642-TW/Final 21 201001964 410~460, 710~740, 910~950: steps ; 901, 902: Pilot.
VTU08-0002/ 0608A-A41642-TW/Final 22VTU08-0002/ 0608A-A41642-TW/Final 22
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