TW200915658A - Antenna and resonant frequency tuning method thereof - Google Patents
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- TW200915658A TW200915658A TW96136122A TW96136122A TW200915658A TW 200915658 A TW200915658 A TW 200915658A TW 96136122 A TW96136122 A TW 96136122A TW 96136122 A TW96136122 A TW 96136122A TW 200915658 A TW200915658 A TW 200915658A
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200915658 九、發明說明: 【發明所屬之技術領域】 調頻方法。 本發明涉及-觀線,及增加傳輪頻寬與 【先前技術】 因為習知錢質紐器讀的败 習知介電質共振器天線的_,通“== 加以結合。例如將多塊 门开▲之U冓 辻相她士、▲ 也狀不同之共振結構相疊合,僅 其頻㈣而達到寬頻之效果。或是利用孔隙(㈣^吏 入至介電質共振器,結合介 、 貝 、/、振态天線與狹縫天線的頻帶 以延伸頻見。或在介電質共振器上 擾動原本之介電質轉〜立 金則此-金屬片將 輕模能,心 _叙料分佈,_赶額外之共 增加歧㈣寬。細,金則在義時會有顯 者的歐姆損耗,因而降低輕射效率。 ^曰有'”、員 雜度愈\了^^之啊’f知触料段纽高製程的複 以整:在曰益精:=’使得f知介電f共振器天線難 述f知介謝。故,本發縣針對上 貝、搌為天線之缺失進行改進。 200915658 【發明内容】 . 馨於上狀發明背景中,為了符合聽上料利益之需 求,=發明提供一種天線及其調頻方法可用以解決 天線未能達成之標的。 ’、、死之 ^發明之一目的係提供—種天線及其調頻方法,此一天 線包s -基板、-微帶線、—接地層與—共振結構,其 層與微帶線分卿成於基板相對之兩面,並且接地層包含一鏤 =。上述之共振結構配置於接地層之上,並且共振結構係藉 由-狹縫分喊-第-共振频―第二共振部, 部包含一第一底面與一第一側面 〃第“振 叫旬亚且弟二共振部包含一第二 底面與一第二側面。上述天線的 ^ +〜 、、1々111杈恶之共振頻率係藉由調 錢縫Lx鶴並增加,且其频頻寬亦隨狹縫寬度增加而 增加。 再者,一第-隨道配置於狹縫與第—底面之交接處,並 且-:二隧道配置於狹缝與第二底面之交接處,其中藉由調整 狹縫歧,以移動天線的私模態之共振頻率。糾,一第一 凹槽配置於第-側面’並且—第二凹槽配置於第二侧面,其中 藉由調整第-㈣與第二凹槽之尺寸與位置,以增加天線的 ⑽仏模態之頻寬。隨後即可利用微帶線當作訊號 線,並且訊號可透過鏤空部饋入共振結構。此—天線易於與其 他平面電路整合’並減少其他元件對天線的干擾。 200915658 【實施方式】 …本發明在此所顯財向為—種天線及其方法。為 了能徹底地瞭解本發明,將在下列的描述中提出詳盡的步驟及 其組成。_地,本發_贿縣限定於天線及其調頻方法 之技藝者所熟習的特殊細節。另—方面,料周知的組成或步 驟亚未描述於細節巾,_免造成本發料必要之_。样 明的較佳實施例會詳細描述如下,細除了這些詳細描述之 Γ本發明還可以廣泛地施行在其他的實施射,且本發明的 耗圍不受限定’其以之後的翻範圍為準。 本發明之—實蘭係將—直、_介電質共㈣均勾地分 ㈣形成—介電f共振11天線(胸_ Res_ Αη_ 驗)。因此’被分離之二介電f共翻間之狹縫巾的電場會 顯著地’並且m因子(Q_fa_)亦相此降低。 上述二介電質共振n分酬有二個凹槽,觀關整共振頻率 ^增加頻寬。本發鴨即揭露狹縫與凹槽對於共振頻率的影 響’其中私與呢模態之共振頻帶係可被調整以涵蓋丽仏 (3.4-3.7GHZ)與 WLAN (5.15-5.35GHZ)等頻帶。 立參考第一 A圖與第—B圖所示,其係為-天線励之結 構不意圖,其中天線卿係可為介電f共振器天線。此一天線 卿係由一第一共振部15〇與一第二共振部⑽所構成,並且 弟一共振部150與第二共振部17〇可為?全相同之二個矩形介 2〇〇915658 “、、振其中此一個共振結構150,丨70與一狹縫142寬之 總和長度為、此二個共振結構15〇, 17〇寬度為△、高度為d, 並且第-共振部15〇與第二共振部携係藉由寬度為p之狹縫 42所刀1W。上述二個共振部150,170之底部與側邊皆分別刻 有凹口,其中底部之一第一隧道156與一第二隧道176之長 又為A見度為&、尚度為沁,並且側邊之一第一凹槽I%與 一第二凹槽178之長度為卜寬度為&、高度為.上述二個 共振結構150, 170係配置於一接地層13〇上,其中接地層㈣ 之長度為%、寬度為A。此一接地層13〇係配置於一 基 板110上,其中此一 FR4基板11〇之厚度為卜介電常數為《4。 一微帶線120透過-鏤铸132以饋人至二個共振結構15〇, 17〇 ’其中此微帶線12〇延伸超過鏤空部132之長度為“,並 且上述鏤空部D2與第-共振結構15〇之間的輕疊長度為 ds ° 天線100之共振頻率主要係藉由二個共振結構15〇, 17〇 之三維參數e與介電常U決定。因為上述之凹口會 改變原始二個共振結構15〇, 170内部之電場分佈,因此共細 率亦會隨之改變。由於狹縫m垂直於完整無凹口之二個共振 結構150, 170 模態之電場的z分量,狹縫142内部之電 場即會因此增強。所以’和模態之共振頻率與輸人阻抗會被 顯著地影響,並且可藉_整上狀◊微帶線i2G的延伸長 200915658 度與鏤空部132 , 丨 長度乙以微調天線之輸入阻抗。 在位於工間^之二個共振結構150, 170中,電場與磁場 必須符合馬克士咸方程式(M_eii,s—Μ·广 ~~^χΕ〇 =ja〇Mfi〇 (i)200915658 IX. Description of the invention: [Technical field to which the invention pertains] Frequency modulation method. The present invention relates to - line-of-sight, and increases the bandwidth of the transmission wheel and [prior art] because the ignorance of the dielectric resonator is read by the _, "== is combined. For example, multiple blocks The door opening ▲ U 冓辻 她 her, ▲ also different resonant structure superimposed, only its frequency (four) to achieve the effect of broadband, or the use of pores ((4) ^ into the dielectric resonator, combined with , Bay, /, vibrating antenna and slot antenna frequency band to extend the frequency. Or disturb the original dielectric in the dielectric resonator ~ Li Jin then this - metal sheet will be light mode energy, heart _ Syria Material distribution, _ rush to add a total difference (four) wide. Fine, gold will have obvious ohmic loss in the sense of justice, thus reducing the efficiency of light shooting. ^ 曰 have '", the more the staff is more ^ ^ ^ 'f knowing the contact section of the New High process to complete: in the 曰益精:=' makes the f-known dielectric f-resonator antenna difficult to say f. Thanks, this is the county for the upper shell, the antenna for the antenna Missing for improvement. 200915658 [Summary of the Invention] In the background of the invention, in order to meet the needs of listening to the interests, the invention provides one The antenna and its frequency modulation method can be used to solve the problem that the antenna fails to achieve. ',, and the purpose of the invention is to provide an antenna and a frequency modulation method thereof, the antenna package s-substrate, the microstrip line, a grounding layer and a resonant structure, the layer and the microstrip line are formed on opposite sides of the substrate, and the ground layer comprises a 镂=. The above resonant structure is disposed on the ground layer, and the resonant structure is formed by the slit a second-resonant portion, a first bottom surface and a first side surface 〃 振 振 旬 旬 且 且 且 且 且 且 且 且 且 且 且 弟 弟 弟 弟 弟 弟 弟 弟 弟 弟 弟The resonance frequency of ^ +~ , 1々111 aversion is increased by the money-sewing Lx crane, and its frequency width also increases with the increase of the slit width. Furthermore, a first-channel is placed in the slit. a junction with the first bottom surface, and - two tunnels are disposed at the intersection of the slit and the second bottom surface, wherein the resonant frequency of the private mode of the antenna is moved by adjusting the slit difference. The groove is disposed on the first side 'and the second groove is disposed on the second side The width of the (10) 仏 mode of the antenna is increased by adjusting the size and position of the (-)th and the second groove. The microstrip line can then be used as the signal line, and the signal can be fed through the hollow portion. Resonant structure. This is an antenna that is easy to integrate with other planar circuits' and reduces the interference of other components on the antenna. 200915658 [Embodiment] The present invention is hereby shown as an antenna and a method thereof. The invention will be described in the following description with detailed steps and its composition. _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ The invention is described in detail below, and the preferred embodiment will be described in detail below, and the invention may be widely practiced in other embodiments, and The cost of the invention is not limited, which is based on the subsequent range. In the present invention, the solid-state, the straight-line, the _ dielectric, and the (four) are all divided into four (4) to form a dielectric f-resonance 11 antenna (thoracic_Res_ Αη_test). Therefore, the electric field of the slit towel in which the separated dielectrics f are turned over will be remarkably 'and the m factor (Q_fa_) is also lowered accordingly. The above two dielectric resonance n fractions have two grooves, and the entire resonance frequency is increased to increase the bandwidth. The hair duck reveals the effect of slits and grooves on the resonant frequency. The resonant frequency band of the private and modal modes can be adjusted to cover bands such as Radisson (3.4-3.7 GHz) and WLAN (5.15-5.35 GHz). Referring to the first A diagram and the -B diagram, it is not intended to be an antenna excitation structure, wherein the antenna system may be a dielectric f resonator antenna. The antenna is composed of a first resonating portion 15A and a second resonating portion (10), and the resonance portion 150 and the second resonating portion 17? The two identical rectangles are 2〇〇915658", and one of the resonant structures 150 is oscillated. The sum of the widths of the 丨70 and the slit 142 is the width of the two resonant structures 15〇, 17〇, the width is Δ, the height D, and the first-resonant portion 15〇 and the second resonating portion are carried by the slit 42 of the slit 42. The bottom and the sides of the two resonating portions 150, 170 are respectively inscribed with a notch, wherein the bottom portion One of the first tunnel 156 and the second tunnel 176 has a length of A and a degree of 沁, and the length of one of the first groove I% and the second groove 178 of the side is The two resonant structures 150 and 170 are disposed on a ground layer 13〇, wherein the ground layer (4) has a length of % and a width of A. The ground layer 13 is disposed on a substrate. 110, wherein the thickness of the FR4 substrate 11 为 is a dielectric constant of “4. A microstrip line 120 is transmitted through the sputum 132 to feed the two resonant structures 15 〇, 17 〇, wherein the microstrip line 12〇 extends beyond the length of the hollow portion 132, and the light stack length between the above-mentioned hollow portion D2 and the first-resonant structure 15〇 is ds ° The resonant frequency of the antenna 100 is mainly determined by the three-dimensional parameter e and the dielectric constant U of the two resonant structures 15〇, 17〇. Since the above-mentioned notch changes the electric field distribution inside the original two resonant structures 15〇, 170, the total fineness also changes. Since the slit m is perpendicular to the z component of the electric field of the two resonant structures 150, 170 modal, the electric field inside the slit 142 is thereby enhanced. Therefore, the resonant frequency and input impedance of the modality and the modality are significantly affected, and the input impedance of the antenna can be fine-tuned by the length of the extension of the microstrip line i2G and the length of the thin section 132, 丨 length B. In the two resonant structures 150, 170 located in the workplace, the electric and magnetic fields must conform to the Maxine's equation (M_eii, s-Μ·广~~^χΕ〇 =ja〇Mfi〇 (i)
VxHQ = jc〇QSEQ (2) 其中%即為共振頻率。 ^個共振結構150, 170的形狀藉由狹縫m2、隨道156, 或凹槽158, 178改變後,在空間^/中的共振頻率即會便成 人位置函數並且場的分佈與共振頻率亦會分微變為符 口馬克士威方程式之卜互與ω。由於原始場與擾動場之間的 作用,形狀變更後的二個共振結構15(U7G之共振頻率係可表 示如下: „、yFxg+$x互)‘ W = — ----:~~ 其中VxHQ = jc〇QSEQ (2) where % is the resonant frequency. After the shape of the resonant structures 150, 170 is changed by the slit m2, the track 156, or the grooves 158, 178, the resonant frequency in the space ^/ will be an adult position function and the field distribution and resonance frequency are also Will be divided into the mouth of the Maxwell's equation and ω. Due to the interaction between the original field and the disturbance field, the two resonance structures 15 after the shape change (the resonance frequency of the U7G can be expressed as follows: „, yFxg+$x mutually) ‘W = — ----:~~
Wm = HduWm = Hdu
VV
Wea = 200915658Wea = 200915658
Web = ll^K-EduWeb = ll^K-Edu
V 其中上述方程式麵共振辦係受絲與形變之介電質 共振器之間的反應所影響。上述方程式亦暗示了如果擾動= 被估抽更知確,則共振頻率亦可被更精確的預測。例如,一 介電質共振器上具有-小狹縫時,垂直於空氣之介電分界面的 .電場將會顯著地增強,其中此—現象可藉由模擬來觀察。 再者’配置於無限大之„接地層上並狀寸為之— 介電質共振器係等同於在自由空間中的單一矩形共振界,其中 此-矩形共振器之高為Μ ’如第二騎示。因為介電質絲 器的介電常數遠大於空氣的介電常數’所以在一階分析中,空 氣的介電分界面可魏為L導體(Perfect MagnetI Conch咖·,PMC)牆,滿足此邊界條件所求得的電磁場模能, 可被區分為喊態細鶴。_,近似完美磁導體之料 所求得細難之絲會比_狀結果更域確。再者,又 (Dielectric Waveguide Model, DWM) 以提供更精俩_,射驗結構倾視為在傳播方向中被 截斷之-介電質波導的部分。完美磁導體之近似職強加於波 導之表面,並且在截斷之截面中假設全反射存在,電磁場在被 鑛之兩截面間形成駐波。接著,具有奇數㈣模態之p 係可推導如下: # 200915658 E0x = -A:zAcos(A:JCJv:)cos(A:y>;)sin(A:zz) EOy =0 E0z = kxAsm(kxx)cos(kyy)cos(kzz) y Asin(kxx)sin(k y)cos(kzz)V wherein the above equation surface resonance system is affected by the reaction between the filament and the deformed dielectric resonator. The above equation also implies that if the disturbance = estimated is more accurate, the resonant frequency can also be predicted more accurately. For example, when a dielectric resonator has a small slit, the electric field perpendicular to the dielectric interface of the air will be significantly enhanced, and this phenomenon can be observed by simulation. Furthermore, 'disposed on the infinite „ground layer and the shape is the same—the dielectric resonator is equivalent to a single rectangular resonance in the free space, where the height of the rectangular resonator is Μ 'as the second Because the dielectric constant of the dielectric filament is much larger than the dielectric constant of air', in the first-order analysis, the dielectric interface of the air can be a L-conductor (Perfect MagnetI Conch, PMC) wall. The electromagnetic field modulus energy obtained by satisfying this boundary condition can be divided into shouting cranes. _, the material of the approximate perfect magnetic conductor can be more accurate than the _ shape result. Again, again ( Dielectric Waveguide Model, DWM) provides a more refined _, the structure of the structure is considered to be truncated in the direction of propagation - the part of the dielectric waveguide. The approximate strength of the perfect magnetic conductor is applied to the surface of the waveguide, and is cut off It is assumed in the section that total reflection exists, and the electromagnetic field forms a standing wave between the two sections of the ore. Next, the p-system with an odd (four) mode can be derived as follows: # 200915658 E0x = -A: zAcos(A:JCJv:)cos(A :y>;)sin(A:zz) EOy =0 E0z = kxAsm(kxx)cos(kyy)cos (kzz) y Asin(kxx)sin(k y)cos(kzz)
HH
OxOx
H J零kl+k2z 〇y A cos(kxx) cos(ky y) cos(kz z) ⑷H J zero kl+k2z 〇y A cos(kxx) cos(ky y) cos(kz z) (4)
Hr ](〇μ kzk -—A c〇s(A:丨)sin( A: y) sin(众 z) ](〇μ yHr ](〇μ kzk -—A c〇s(A:丨)sin( A: y) sin(众 z) ](〇μ y
其中為任思令數、灸^ = ;r / Μ、\ =册及/ 並且灸^係決定自 [Υ. Μ. Μ. Antar, D. Cheng, G. Seguin, B. Henry, and M. G. Keller, Modified waveguide model (MWGM) for rectangular resonator antenna (DRA)/5 Microwave 〇pt. Tech. Lett., vol. 19, no. 2pp. 158-160, Oct. 1998·],其中Among them, Ren Si Ling, moxibustion ^ = ; r / Μ, \ = book and / and moxibustion ^ decided from [Υ. Μ. Μ. Antar, D. Cheng, G. Seguin, B. Henry, and MG Keller, Modified Waveguide model (MWGM) for rectangular resonator antenna (DRA)/5 Microwave 〇pt. Tech. Lett., vol. 19, no. 2pp. 158-160, Oct. 1998·], where
(5) 共振頻率亦可同樣地計算如下: ⑹ 具有偶數”的%模態之場係可推導如下 200915658 E, Ολ: -kz B cos(kxx) cos(ky y) cos{kz z) % -0 ^〇z = kxB sm(kxx) cos{ky y) sin(A:2 z) kk H, H, .忍 sin(々 x)sin(A: y)cos«z) ]ωμ y z k2r + k7: -B cos{kxx) cos{ky y) sm{kz z) ⑺5 cos(々j) sin(b 3;) cos(l z) 頻率數、’並且〜與共振 J刀別由方程式(5)與⑹決定。 引的三弟 、第三B圖與第三C _科第三個下標所指 器内:=:其中上述之T標係表示介電質共振 .代表偶數槿〜=Μ Ζ軸的Α分量具有可變之偶數以 態中4分^ Γ可變之奇數以代表奇數模態。在偶數模 中4分量係為對稱。 狀對稱’並且在奇數模態 第四Α圖顯示了配置於—接地 , 携,其中此二矩形共振結構风振結構150, 並且狹縫M2係位於㈣。在^,扁由—狹縫⑷分隔, 會趨極大值,模態 之馬分量 振為之共振杈態會被激發。此—狹縫1 ^ "電貝八 . 之空氣介電分界面係(5) The resonance frequency can also be calculated as follows: (6) The field system with even-numbered % mode can be derived as follows: 200915658 E, Ολ: -kz B cos(kxx) cos(ky y) cos{kz z) % - 0 ^〇z = kxB sm(kxx) cos{ky y) sin(A:2 z) kk H, H, .bear sin(々x)sin(A: y)cos«z) ]ωμ yz k2r + k7 : -B cos{kxx) cos{ky y) sm{kz z) (7)5 cos(々j) sin(b 3;) cos(lz) frequency number, 'and ~ and resonance J knife not by equation (5) (6) Decision. The third sub-picture, the third B picture and the third C _ section of the third subscript are: /: wherein the above T mark system represents dielectric resonance. It represents even 槿~=Μ Ζ axis The Α component has a variable even number in the state of 4 minutes ^ Γ variable odd number to represent the odd mode. In the even mode, the 4 component system is symmetrical. The shape symmetry 'and the fourth Α diagram in the odd mode shows the configuration In the grounding, carrying, the two rectangular resonant structure wind vibration structure 150, and the slit M2 is located at (four). In the ^, the flat is separated by the slit (4), which will be maximal, and the modal horse component resonance The state will be excited. This - slit 1 ^ " electric shell eight. Department of surface
Hr ](〇μ Kky ΐωμ 12 200915658 垂直於z車由 地增強。 所以為了符合A上之連祕件,&分量會顯著 之較狹縫寬度順回伽效應,射天線励 寸芩數為:a=28mm、b=9mm、d=Wmm、e =20、 c〇a=2mm、^ r a mm、Ls=8mm、ds=7mm、Ws=Lg=70mm、 0.6mm ^ 〇) —i ic 與产所所。當 顯著地增加眸,t -珥趣之共振頻率 、β ^ 模態亦會輕微地受影響,值得注意 、1 '心會與72匕模態合併,然而兩模態在;平面具有不 同的輻射場形。 一 根據圖像論(imagethe〇ry),如果接地層係為無限大 . ___ 、'^ 圖與弟四Β圖的結構係為等效的。然而,被狹縫142 刀&之—個共振結構15〇, 17〇可被視為具有介電常數為之 門類J的一介電質共振器。因為狹縫142寬度/?被設定為遠 小於α,所以除了狹縫142内部之垂直電場尽被增強以符合空 氣介電之連續條件以外,在單一介電質共振器150, 170内部其 餘部位的電磁場之分佈則與沒有狹缝142之完整介電質共振 器内部的場之分佈大致相同。因此,在空氣狹缝142中的7^>u 與炫。模態可近似如下: & = m,屹 Asin«x)cos(心 _y)cos(l/?/2) Ex=Ey=0 H = H0 13 (8) 200915658 上述之Μ量係根據㈣子而增強。對和模態而今, ,縫则度很小時,趨♦。對%模態而言^分 里僅有u巾田地i曰加,並且產生—微小值〒其中叫大約在2 到3之間。所以,%模態之共振頻率微幅地增加。相對地, 空氣狹縫142中的⑽模態之場係可推導如下: Εχ ~Hr ](〇μ Kky ΐωμ 12 200915658 is perpendicular to the z-car to enhance the ground. Therefore, in order to meet the secrets on A, the & component will be significantly more smooth than the slit width, and the number of antennas is: a=28mm, b=9mm, d=Wmm, e=20, c〇a=2mm, ^ ra mm, Ls=8mm, ds=7mm, Ws=Lg=70mm, 0.6mm ^ 〇) —i ic What you are. When the enthalpy is significantly increased, the resonance frequency and β ^ mode of t - interest are also slightly affected. It is worth noting that the 1 'heart will merge with the 72 匕 mode, but the two modes are; the plane has different radiation Field shape. According to image theory (imagethe〇ry), if the ground plane is infinitely large, the structure of ___, '^ and the four diagrams is equivalent. However, a resonant structure 15 〇, 17 被 by the slit 142 knives can be regarded as a dielectric resonator having a dielectric constant of the kind J. Since the slit 142 width /? is set to be much smaller than α, the rest of the interior of the single dielectric resonator 150, 170 is excluded except that the vertical electric field inside the slit 142 is enhanced to conform to the continuous condition of air dielectric. The distribution of the electromagnetic field is substantially the same as the distribution of the field inside the complete dielectric resonator without the slits 142. Therefore, 7^>u in the air slit 142 is dazzled. The modalities can be approximated as follows: & = m, 屹Asin«x)cos(心_y)cos(l/?/2) Ex=Ey=0 H = H0 13 (8) 200915658 The above Μ quantity is based on (4) Enhance the child. For the modality and the modality, the seam is very small and tends to be ♦. For the % modality, only the u-zone field is added, and a small value is generated, which is called between about 2 and 3. Therefore, the resonance frequency of the % mode is slightly increased. In contrast, the field of the (10) modality in the air slit 142 can be derived as follows: Εχ ~
Ez=Ey ~0 X H = H0 ⑼ «,再將方程=:之:^ ’以及令方程式(9)之 具有狹縫之矩形介代入方程式⑶,即可推導出 ^“的㈤與私模態之共振_。 幸晏射圖形可由介 由於和模態之電場/佈上之切缘電場來決定。 沿z方向上,正負 每刀佈。咖(2_)在介電質共振器上表面 的輻射場形零值會出現在if::的電場方向,A在”面 態之共振頻率會逐漸靠近,並^"抑的士增加,和與⑽模 即會合併。然而,由於輻射時,上述之兩個頻帶 .頻帶分離係更好的。了⑽的差異,將冗與阿2模態之 根據方程式(3),—处严 之電場強、%彳工鱗道146係配置於⑹模態 4&%可忽略之處⑹模態之共接頻率 14 200915658 即可與鳴之共振頻率分 11-¾、若1 K /么 4号第/、A圖所不,一空氣 隧迢146係配置於—政 口吓丁 工札 應係顧示於第七圖,其中上;;2_146之半長度谢 ’L天線100之較佳三維尺寸參數 馬· a-28mm、b=9mm、扣 1 η mm、P=〇mm ' d^mm、sr=2Q、 ωα 二 2mm、、,聰、 一〜 ,8mm > d=7mm ^ W=L=70mm > t=0.6mm、 rnm~\\<mni ^Ez=Ey ~0 XH = H0 (9) «, and then equation =:: ^ ' and the rectangle with the slit of equation (9) is substituted into equation (3), then we can derive the "five" and private mode Resonance _. Fortunately, the radiation pattern can be determined by the electric field of the modal and the electric field on the cloth. In the z direction, positive and negative knives. Coffee (2_) radiation field on the upper surface of the dielectric resonator The shape zero value will appear in the direction of the electric field of if::, the resonance frequency of A in the "surface state will gradually approach, and ^" suppresses the taxi increase, and the (10) mode will merge. However, due to the above two bands, the band separation is better. The difference of (10) is that the redundancy and the A2 mode are based on the equation (3), the strict electric field is strong, and the % work scale 146 is arranged in (6) the modal 4&% negligible (6) mode The frequency of the connection 14 200915658 can be divided into 11-3⁄4 of the resonant frequency of the sound, if 1 K / 4 No. 4, and A is not, an air tunnel 146 is placed in the - Zhengkou scared work should be shown In the seventh figure, the upper; 2_146 half length of the 'L antenna 100's preferred three-dimensional size parameters Ma · a-28mm, b = 9mm, buckle 1 η mm, P = 〇 mm ' d ^ mm, sr = 2Q, ωα 2mm, ,, Cong, Yi~, 8mm > d=7mm ^ W=L=70mm > t=0.6mm, rnm~\\<mni ^
” i;=G·5〜2麵。隨著心與沁的增加, Γ£7ΐ2桓悲之共振頻率亦合隨$ ^ 4 、 千力θ I通之增加,並且由於此時隧道140附 近的T£7U與7¾模態之電場齡彳呀& & 电野孕乂U弱的,所以炫二與炫%模態之共 振頻率幾乎不受影響。 、:考第、B圖戶斤示,當共振結構⑽與隧道μ6的高度 皆為原先的兩倍時’第六B圖與第六A圖之共振結構係為等 放的因為reui與7¾模態之電場隨著y軸旋轉,所以電場會 正切於隧道I46之空氣介電之分界面。因此,e與&必須假設 e~e,$l'h = h, ° 至於7^12模態,因為隧道丨46係位於電場接近極大值之 處,所以在隧道146中之民分量即會增加,並且可近似如下:"i;=G·5~2 faces. With the increase of heart and sputum, the resonance frequency of ΐ£ΐ2桓2 亦 is also increased with $^4, 千力θI, and because of the vicinity of tunnel 140 at this time T£7U and 73⁄4 mode electric field age 彳 && electric field pregnancy 乂 U weak, so the resonance frequency of Hyun 2 and Hyun modal is almost unaffected. When the heights of the resonant structure (10) and the tunnel μ6 are both twice as large, the resonant structures of the sixth and sixth graphs are equal. Since the electric fields of the reui and 73⁄4 modes rotate with the y-axis, The electric field will be tangent to the interface of the air dielectric of tunnel I46. Therefore, e and & must assume e~e, $l'h = h, ° as for the 7^12 mode, because the tunnel 丨46 is located in the electric field is very close Where the value is, so the weight of the people in the tunnel 146 will increase, and can be approximated as follows:
Ex = kzaB cosC^) cos(^j) cos(/?z)Ex = kzaB cosC^) cos(^j) cos(/?z)
Ez=Ey=0 ~ ~ (10) H = H0 15 200915658 令欠=k/a之方程式(7)與(10)代入(3),即可預測坪^模態 共振頻率的偏移。隧道146對Γ^2模態之共振頻率的影響大於 對珲U與7¾%模態之共振頻率的影響,並且尽係隨著隧道146 的薄度呈α倍地增加。rw12模態之較佳共振頻率力係為 3.646GHz。 在^±«/2時,、7^12與7^3模態之尽會趨近極大值, 亚且上述模態之共振頻率亦會受到鄰近之凹槽⑸, 178的影響。第人a圖顯示—接地之共彳辰結構⑽,其中二個 凹槽158, 178係分別配置於此一共振結構14〇之兩側邊。上述 之凹槽158, 178將會扭曲電場的分佈,並且天線品㈣子亦合 降低,以產生較寬的頻帶。第九圖係為顯示藉由增加凹槽⑸, 178深度以增加上述三種模態之共振頻率,其中上述天線腦 之較佳三維尺寸參數為七烈_、㈣咖、和槪心、 〇)a=2mm、La = lQmm、ι=8 η s 8_ ds一7mm ' w~7〇瞻、 t=0.6mm ' (〇m^l.\5mm ^ s2=〇^2mm 〇Ez=Ey=0 ~ ~ (10) H = H0 15 200915658 Let equations (7) and (10) of minus=k/a be substituted into (3) to predict the offset of the modulo resonance frequency. The effect of the tunnel 146 on the resonant frequency of the 模^2 mode is greater than the effect on the resonant frequency of the 珲U and 73⁄4% modalities, and increases as a function of the thinness of the tunnel 146 by a factor of a. The preferred resonant frequency of the rw12 mode is 3.646 GHz. At ^±«/2, the 7^12 and 7^3 modes will approach the maximum value, and the resonant frequency of the above modes will also be affected by the adjacent grooves (5), 178. The first figure a shows a grounded common structure (10) in which two recesses 158, 178 are respectively disposed on both sides of the resonant structure 14A. The grooves 158, 178 described above will distort the distribution of the electric field and the antenna (4) will also be reduced to produce a wider frequency band. The ninth figure shows the increase of the resonance frequency of the above three modes by increasing the depth of the grooves (5), 178, wherein the preferred three-dimensional size parameters of the antenna brain are seven-strong, (four) coffee, and heart, 〇) a =2mm, La = lQmm, ι=8 η s 8_ ds - 7mm ' w~7〇, t=0.6mm ' (〇m^l.\5mm ^ s2=〇^2mm 〇
"⑽,兩凹槽158, 178之接地共振結構14( 係等效於側邊具有四個簡·立·結構。根縣八B 所示,纽考m凹槽178,其中第二凹_ _ 於自由空間中之共振結構⑽,並且第二凹槽178之 為"、。糊:凹槽178㈣心分量:: 在,所以弟-凹槽m内之電場即會更加的複雜,而藉由駒 16 200915658 得知&分量大於尽分量。垂直於第二凹槽178 ;|嘵空氣分 界面的&分量係會增加,其中民分量&分量必 ” ^ 、付合連續條 件’並且可近似如下: 在7¾與7¾模態:"(10), the grounding resonance structure 14 of the two grooves 158, 178 (equivalent to the side having four Jane-Lee structures. The root county is shown as BB, the Newcastle m groove 178, wherein the second concave_ _ in the free space of the resonant structure (10), and the second groove 178 is ", paste: groove 178 (four) heart component::, so the electric field in the brother-groove m will be more complicated, and borrow It is known from 驹16 200915658 that the & component is greater than the full component. The perpendicular to the second groove 178; | 哓 air interface & component will increase, where the folk component & component must " ^, pay continuous condition ' and Can be approximated as follows: In the 73⁄4 and 73⁄4 modes:
Ex = -^aficosC^J^cos^ v)cos(^) (11) 在re/12模態··Ex = -^aficosC^J^cos^ v)cos(^) (11) In re/12 mode··
Ex = m2kzBcos{kxdx)co^{k y)c〇s(k z) z (12) 其中較佳之4值係為4mm,並且較佳之%值大約為丄$。 將方程式⑷與(11)代人方程式⑶,即可求得具有凹槽之介電質 共振器三個模態的共振頻率。 、 上述之介電質共振器之較佳實施範例為㈣ ㈣麵古wmm令7mm、Ls=8mm、% = 2麵队_麵。 %、呵2與网3模態之共振頻率分別為2 92GHz、3 58版與 4.62GHz。為了調整私與私模態之共振頻率可涵4 肅AX (3.4-3.7GHz)與 WLAN (5 i5_5 35GHz)之頻帶, 共振結構⑽之形貌被設計如第—A圖所示,其中内麵、 ㈣=如m以及㈣=2_。藉由調整《、微帶線m之延 伸長度L,與鏤空部之長度4,共振結構上料配合加微帶 線120的饋入’並且共振頻率會因饋入結構而產生微幅影響。 第十圖顯示測量與模擬之返回她,其㈣、呢與㈣模態 200915658 之共振頻率分別涵蓋至3 375_3 93GHz (15%)、46_479GHz (4%)與515415GHz (6%),並且實線與虛線分別為量測 值與模擬數值。第十圖中之第一個頻帶191即涵蓋了 (3.4-3.7GHZ),並且第三個頻帶193即涵蓋了 wlan (5-15-5.35GHz)’其中上述天線1〇〇之較佳三維尺寸參數為: β=2細m、㈣卿、扣胸w、产;麵、义=4咖、麵、 d2 4mm、s2-2mm、sr =20、h=4mm、(〇a =2mm、l 、Ex = m2kzBcos{kxdx)co^{k y)c〇s(k z) z (12) wherein the preferred value of 4 is 4 mm, and the preferred % value is approximately 丄$. By equations (4) and (11) and equation (3), the resonant frequencies of the three modes of the dielectric resonator with grooves can be obtained. A preferred embodiment of the above-described dielectric resonator is (4) (4) face wmm 7mm, Ls=8mm, % = 2 face _ face. The resonance frequencies of %, 呵2, and NET 3 modes are 2 92 GHz, 3 58 and 4.62 GHz, respectively. In order to adjust the resonant frequency of private and private modes, the frequency band of AX (3.4-3.7GHz) and WLAN (5 i5_5 35GHz) can be used. The shape of the resonant structure (10) is designed as shown in Figure A, where the inner surface , (4) = such as m and (four) = 2_. By adjusting "the elongation L of the microstrip line m, and the length 4 of the hollow portion, the resonance structure is loaded with the feeding of the microstrip line 120" and the resonance frequency is slightly affected by the feeding structure. The tenth graph shows the return of measurement and simulation. The resonance frequencies of (4), (4) and modal 200915658 cover 3 375_3 93 GHz (15%), 46_479 GHz (4%) and 515415 GHz (6%), respectively, and the solid line The dotted lines are measured values and analog values. The first frequency band 191 in the tenth figure covers (3.4-3.7 GHz), and the third frequency band 193 covers the wlan (5-15-5.35 GHz) 'the preferred three-dimensional size of the above antenna 1 〇〇 The parameters are: β = 2 fine m, (four) Qing, buckle chest w, production; face, meaning = 4 coffee, face, d2 4mm, s2-2mm, sr = 20, h = 4mm, (〇a = 2mm, l,
Ls=2.5_、d,4mm、Wg=Lg=7〇mm、t=〇 6jnm 輿 % =U5mm。 第十- A圖與第十一 B圖分別顯示了第—個頻帶⑼與 第二個頻帶193之電場分佈,其中約為/=5篇邮之第三個 共振頻帶咖卩為上述之;模態。此—分離之共振結構风 170可被視為沿著z軸緊鄰配置之雙天線。 上述僅為本發明之較佳實施例’並非狀限定本發明之 申請專利範圍。因此’本發明更提出—_天線卿,其中此一 天線100包含-基板110、一微帶線120、一接地層⑽與一 共振結構H0。上述之接地層⑽與微帶線m分別形成於基 板no相對之兩面’並且接地層13〇包含一鎮空部132。上述 ^共振結構M0配置於接地層13〇之上,並且共振結構⑷係 错由-狹缝142分隔成-第一共振部15〇與一第二共振部⑽ 參考第A圖所不,上述之第一共振部15〇包含一第一 底面152與-第—侧面154,並且第二共振部携包含一第二 18 200915658 底㈣與-第二側面174,其中第—底面i52覆蓋於接地層 ⑽之上,並且與鏤空部132重疊。再者,上述之第一共振部 150射一共振部170皆可為平行六面體,例如長方體,而共 振結構140可為一介電質共振結構,更可包含低溫共燒陶究。 當無線訊號自微帶線120輸入時,無線訊號將透過鏤空 ^麵口至“振結構140,其中當無線訊號之電力線通過狹 縫142時’由於介電質共振結構與空氣間的分界面之電通密度 义,頁連續’並且共振結構之介電係數遠大於线,所以狹縫 142中之電場會因此增強數倍,藉此使得電磁波更有效率地韓 射’並且降低共振結構之品質因子,亦可有效地增加訊號傳輸 的頻寬。因此,藉_整狹縫m寬度即可移細成態之共 4 w員率JL i曰加7¾杈態之頻寬’藉此以涵i別說又 (3.3-3.7GHz)之頻帶,如第五圖所示。 同理,亦可配置—第一隧道156於狹縫142 一底面 ⑸之交接處,並配置—第二隧道176於狹縫142與第二底面 ⑺之交接處’如第六圖A所示。當電力線通過上述之第一隧 道156與第二隨道176時,藉由調整第一隨道與第二隨道 m之尺寸與位置,即可移動㈤模態之共振解,並增加% 與呢13_之傳輸頻寬,藉此以涵蓋WLAN (515_5·35·) 之頻帶,如第七圖所示。 多考弟C圖所示,上述之第一隧道156可沿著一第一 19 200915658 底軸160穿透於第—共振部150,並且第二随道176可沿著一 ^姉⑽穿透於第二共振部17G,其中第—絲160可正 父於弟-絲152之法線162與狹縫142之絲14 二底軸180可正交於第弟 於弟一底面172之法線182與狹縫142之法 線 144 ^ 芩考弟^圖所示,一第一凹槽158可配置於第一侧面 、並且帛—凹槽178可配置於第二側面174。當電力線 通過上返之第一凹槽158與第二凹槽Μ時,藉由調整第一凹 曰…”第凹槽178之尺寸與位置,即可微調%與% 才魏之頻率與增加傳輸頻寬,如第九圖所示。Ls = 2.5_, d, 4 mm, Wg = Lg = 7 〇 mm, t = 〇 6jnm 舆 % = U5 mm. The tenth-A diagram and the eleventh diagram B show the electric field distribution of the first frequency band (9) and the second frequency band 193, respectively, wherein the third resonance frequency band of about /=5 postal mails is the above; state. This—the separated resonant structure wind 170 can be viewed as a dual antenna that is placed next to the z-axis. The above is only the preferred embodiment of the present invention and is not intended to limit the scope of the invention. Therefore, the present invention further proposes an antenna, wherein the antenna 100 includes a substrate 110, a microstrip line 120, a ground layer (10) and a resonant structure H0. The above ground layer (10) and the microstrip line m are respectively formed on opposite sides of the substrate no and the ground layer 13A includes an empty portion 132. The above-mentioned resonance structure M0 is disposed on the ground layer 13A, and the resonance structure (4) is separated by the slit 142 into a first resonance portion 15A and a second resonance portion (10). Referring to FIG. The first resonant portion 15 includes a first bottom surface 152 and a - side surface 154, and the second resonant portion carries a second 18 200915658 bottom (four) and - second side 174, wherein the first bottom surface i52 covers the ground layer (10) Above and overlapping with the hollow portion 132. Furthermore, the first resonant portion 150 and the resonant portion 170 may each be a parallelepiped, such as a rectangular parallelepiped, and the resonant structure 140 may be a dielectric resonant structure, and may further comprise a low temperature co-fired ceramic. When the wireless signal is input from the microstrip line 120, the wireless signal will pass through the hollow interface to the "vibration structure 140, when the power line of the wireless signal passes through the slit 142" due to the interface between the dielectric resonance structure and the air. The electric flux density means that the page is continuous' and the dielectric constant of the resonant structure is much larger than the line, so the electric field in the slit 142 is thus enhanced by several times, thereby making the electromagnetic wave more efficient and reducing the quality factor of the resonant structure. It can also effectively increase the bandwidth of the signal transmission. Therefore, by using the width of the slit, the width of the slit can be adjusted to a total of 4 weeks, and the bandwidth of the JL i曰 plus 73⁄4 state is used. The frequency band of (3.3-3.7 GHz) is as shown in the fifth figure. Similarly, the first tunnel 156 is disposed at the intersection of the bottom surface (5) of the slit 142, and the second tunnel 176 is disposed at the slit 142. The intersection with the second bottom surface (7) is as shown in FIG. 6A. When the power line passes through the first tunnel 156 and the second satellite 176 described above, by adjusting the size of the first track and the second track m Position, you can move (five) the modal resonance solution, and increase the % with 13_ Transmission bandwidth, thereby covering the frequency band of WLAN (515_5·35·), as shown in the seventh figure. As shown in Figure C, the first tunnel 156 described above may be along a first 19 200915658 bottom axis 160 Passing through the first resonance portion 150, and the second satellite 176 can penetrate through the second resonance portion 17G along a line (10), wherein the first wire 160 can be normal to the normal line 162 and the narrow line of the wire-152 The second rib 158 of the slit 142 can be orthogonal to the normal 182 of the bottom surface 172 of the first brother and the normal 144 of the slit 142. As shown in FIG. The first side, and the 帛-groove 178 can be disposed on the second side 174. When the power line passes through the first groove 158 and the second groove 上, the first groove 曰... The size and position can be used to fine tune the % and % of the frequency and increase the transmission bandwidth, as shown in Figure 9.
餐考第- c _示,上述之第—側面154與狹縫⑷係 刀W於第共振部15〇之兩側,其中第一凹槽⑽沿著一第 側軸164牙透於第—共振部15〇。同樣地,上述之第二側面 174,、狹縫142係分別位於第二共振部㈣之兩侧,其中第二 凹槽178沿著一第二侧軸184穿透於第二共振部m。'上叙 第「側軸164可正交於第—侧面154之法線麻與接地㈣ 之法線134 ’並且第二側軸154可與第二側㈣之法線脱 與接地層130之法線134正交。 如上所述,狹縫142牡人筮—歧、苦t * 、’ 口弟一隧迢156與第二隧道176 即可移動7¾丨與τ^ΐ2模態之政择 、 ,、搌頒率並柘加7¾與7¾模態之 頻寬’或是狹縫142結合第一叫她t 技 凹槽158與第二凹槽178即可移 20 200915658 動这⑴輪悲、之共振頻率並增加%、私細u莫態之頻寬。狹 f ^42亦可結合第一隧道156、第二隧道176、第—凹槽158 與第二凹槽178即可移私細3模態之共振頻輕增 力l 7¾與7¾模態之頻寬。最後再藉由調整共振結構14〇 之尺寸,即可調整天線1〇〇之共振頻率。In the meal test - c _, the first side surface 154 and the slit (4) are knives W on both sides of the first resonance portion 15 , wherein the first groove ( 10 ) penetrates the first resonance along a first side axis 164 Department 15〇. Similarly, the second side surface 174 and the slit 142 are respectively located on two sides of the second resonance portion (four), and the second recess 178 penetrates the second resonance portion m along a second side shaft 184. The above-mentioned "the side shaft 164 may be orthogonal to the normal line 134 of the first side surface 154 and the normal line 134 of the ground (4) and the second side shaft 154 may be separated from the normal of the second side (4) from the ground layer 130. Line 134 is orthogonal. As described above, the slit 142 牡人筮-歧, bitter t*, 'the younger brother 156 and the second tunnel 176 can move the 73⁄4丨 and τ^ΐ2 modes,搌 率 柘 73 73 73 73 73 73 73 73 73 73 73 73 73 73 73 73 或是 或是 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 142 The frequency is increased by %, and the bandwidth of the private fine u state. The narrow f ^42 can also be combined with the first tunnel 156, the second tunnel 176, the first groove 158 and the second groove 178 to move the fine 3 mode. The resonant frequency is lightly increasing the bandwidth of the 73⁄4 and 73⁄4 modes. Finally, by adjusting the size of the resonant structure 14〇, the resonant frequency of the antenna 1〇〇 can be adjusted.
〃參考第- B圖所示’上述之第一随道156、第一凹槽说、 176與第二凹槽178皆可為矩形。上述之微帶線⑽ 沿者-弟—軸122延伸’並且鏤空部132沿著—第二轴⑽延 =’其中第-軸122投影至基板11〇之投影線與第二轴说投 影至基板110之投影線為正交。此外,上述之第-軸m投影 至基板m之投麟可通财二軸136投影至基板⑽之= 線的中心位置,亦通過第—底面152與第二絲172之中心位 置。上述之天線⑽更可包含—饋人點與—接地點,其中績入 點係位於微帶線12〇之—端,並且接地點係位於接地層伽上。 為了使上述天線100之呢與%模態的頻帶涵蓋 3.375-3·細z與5爆5.4l5GHz,㈣足聰从與區域 網路(WLAN)的需求,上述天線觀之較佳三維尺寸參數為, a=28mm、㈣mm、㈣_、p=lmm、di=4mm、的魏、 f2-一一,2Q 'w—10mm'Ls=25mm、 d=4mm、Wg=Lg=70mm、㈣·6mm 與叱= U5mm。 依據上述範例即可得知 ,不同的共振模態具有不同的電 21 200915658 ==此㈣槽贿道配置於共振結構之不同位置中,即 。不同杈悲之共振頻率,藉此以移 帶,或县腺ΓΒ . 振頻率至需要之頻 明更==開。故,根據第十-圖所示,本發 種天線調頻方法,可以分別調整共振結構之不同模 二宽=率垂直了加其頻寬,其中此-天線具有低金屬耗 究製程等雜大量=練娜之碰,处可透魏共燒陶 驟:=_示’上述之天線調頻方法包含下列步 述之天線卿,如步驟200。再如步驟210, =過結雜之尺寸,霄天線細之共振頻率。 =可糟由調整狹縫142寬度,以移動⑽模態之共振頻率 模態之頻寬,如步驟220。或是藉由調整第—隨道 /、紅_ m之尺寸與健,以軸呢鋪之共振頻 率’如步驟230。亦可藉由調整第„凹槽158與第二凹槽⑺ 寸”位置’以增加%、私與%模態之頻寬,如步驟 。其餘之相關細節部分皆如同上述之實施例所示,本說明 書於此不再加以贅述。 顯然地,依照上面實闕中的描述,本發明可能有許多 正舁差異。因此需要在其附加的權利要求項之範圍内加以 理解’除了上述詳細的描述外,本發明還可以廣泛地在其他的 實把例中把行。上述僅為本發明之較佳實施例而已,並非用以 22 200915658 限定本發明之申請專利範圍;凡其它未脫離本發明所揭示之精 神下所完成的等效改變或修飾,均應包含在下述申請專利範圍 内。 23 200915658 【圖式簡單說明】 第一A圖、第一B圖、第一C圖、第二圖、第三A圖、 第三B圖、第三C圖、第四A圖、第四B圖、第六A圖、第六 B圖、第八A圖、第八B圖、第十一A圖與第十一B圖係為一 天線之結構不意圖; 第五圖、第七圖、第九圖與第十圖係為係為一天線之返 回損耗與頻率之關係示意圖;以及 / · % 弟十一·圖係為·—天線调頻方法之流程不意圖。 【主要元件符號說明】 100天線 110基板 120微帶線 I 122第一軸 130接地層 • 132鏤空部 134接地層之法線 136第二軸 140共振結構 142狹縫 24 200915658 144狹缝之法線 150第一共振部 152第一底面 154第一侧面 156第一隧道 158第一凹槽 ' 160第一底軸 C · 162第一底面之法線 164第'一侧轴 170第二共振部 172第二底面 174第二側面 176第二隧道 178第二凹槽 I 180第二底軸 182第二底面之法線 184第二側軸 191第一個頻帶 193第三個頻帶 200〜240 步驟 25Referring to the first-B diagram, the first lane 156, the first groove, the 176, and the second groove 178 may each be rectangular. The microstrip line (10) extends along the body-shaft 122 and the hollow portion 132 extends along the second axis (10) = where the projection line of the first axis 122 is projected onto the substrate 11 and the second axis is projected onto the substrate The projection lines of 110 are orthogonal. In addition, the above-mentioned first axis m projection onto the substrate m is projected to the center position of the = line of the substrate (10), and also passes through the center positions of the first bottom surface 152 and the second wire 172. The antenna (10) may further include a feed point and a ground point, wherein the score point is located at the end of the microstrip line 12〇, and the ground point is located on the ground layer. In order to make the above-mentioned antenna 100 and the % modal frequency band cover 3.375-3·fine z and 5 blasting 5.4l5 GHz, (4) the requirements of the local network (WLAN), the preferred three-dimensional size parameter of the antenna view is , a=28mm, (four)mm, (four)_, p=lmm, di=4mm, Wei, f2-one, 2Q 'w-10mm'Ls=25mm, d=4mm, Wg=Lg=70mm, (four)·6mm and 叱= U5mm. According to the above example, it can be known that different resonance modes have different electric powers. 21 200915658 == This (4) channel bribe is arranged in different positions of the resonance structure, ie. The resonance frequency of different sadness, by means of the transfer band, or the county adenine. The frequency of the vibration to the frequency of need is more == on. Therefore, according to the tenth-figure, the antenna frequency modulation method of the present invention can separately adjust the different modes of the resonant structure, the second width = the ratio of the vertical direction and the bandwidth thereof, wherein the antenna has a low metal consumption process and the like. At the touch of the singer, the singer can pass through the Wei dynasty.: = _ The antenna tuning method described above includes the following antennas, as in step 200. Again, as in step 210, = the size of the junction is too large, and the antenna has a fine resonant frequency. = The width of the slit 142 can be adjusted to move the frequency of the (10) mode resonant frequency mode, as in step 220. Or by adjusting the size and health of the first-channel/red_m, the resonance frequency of the axis is as shown in step 230. The width of the %, private and % modalities can also be increased by adjusting the „groove 158 and the second recess (7)” position, as in the step. The rest of the relevant details are as shown in the above embodiments, and the description will not be repeated here. Obviously, the present invention may have many positive differences as described in the above. It is therefore to be understood that within the scope of the appended claims, the invention may be The above are only the preferred embodiments of the present invention, and are not intended to limit the scope of the invention as claimed in the claims of the present invention. All other equivalent changes or modifications which are not departing from the spirit of the invention are included in the following. Within the scope of the patent application. 23 200915658 [Simple description of the drawings] First A picture, first B picture, first C picture, second picture, third A picture, third B picture, third C picture, fourth A picture, fourth B Figure 6, sixth A, sixth B, eighth A, eighth B, eleventh and eleventh B are not intended to be an antenna structure; fifth, seventh, The ninth and tenth diagrams are diagrams showing the relationship between the return loss and the frequency of an antenna; and / · % XI 11. The diagram is · The flow of the antenna frequency modulation method is not intended. [Main component symbol description] 100 antenna 110 substrate 120 microstrip line I 122 first axis 130 ground layer • 132 hollow portion 134 ground layer normal line 136 second axis 140 resonance structure 142 slit 24 200915658 144 slit normal 150 first resonance portion 152 first bottom surface 154 first side surface 156 first tunnel 158 first groove '160 first bottom axis C · 162 first bottom surface normal line 164 first side axis 170 second resonance portion 172 Second bottom surface 174 second side 176 second tunnel 178 second recess I 180 second bottom shaft 182 second bottom line normal 184 second side axis 191 first frequency band 193 third frequency band 200~240 step 25
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TWI511381B (en) * | 2013-10-09 | 2015-12-01 | Wistron Corp | Antenna |
TWI661756B (en) * | 2017-10-27 | 2019-06-01 | 聯發科技股份有限公司 | Radar module |
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TWI511381B (en) * | 2013-10-09 | 2015-12-01 | Wistron Corp | Antenna |
TWI661756B (en) * | 2017-10-27 | 2019-06-01 | 聯發科技股份有限公司 | Radar module |
US11169250B2 (en) | 2017-10-27 | 2021-11-09 | Mediatek Inc. | Radar module incorporated with a pattern-shaping device |
US11821975B2 (en) | 2017-10-27 | 2023-11-21 | Mediatek Inc. | Radar module incorporated with a pattern-shaping device |
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