SE510033C2 - Method and apparatus for powering a magnetron from an AC power source. - Google Patents

Abstract

The present invention relates to a method and a device for providing power to a magnetron from an alternating voltage source. According to the invention, the power supply is accomplished by said alternating voltage being rectified into a rectified voltage which is then converted into a high direct current for powering said magnetron, wherein the power provided to said conversion is controlled based upon a detected actual effect in the power supply means.

Description

510 10 os 2 how the power supply is affected or controlled as a result of these changed conditions, the applied anode voltage and / or the applied anode current can vary greatly.

A known technique for regulating the supply to the magnetron is based on the principle that the anode current is regulated under the assumption that the anode voltage is or is kept relatively constant.

An example of this is shown in Fig. 2, where a magnetron comprising a first; rectifying stage and a second, high voltage generating stage supply a magnetron, the current being sensed just before the magnetron and the high voltage generation being regulated accordingly.

A lot of effort has been put into finding a good method for stabilizing the magnetron's effect, but due to the complexity of the problem, the success has been limited. Thus, there is a need for improved possibilities to control the power supply to a magnetron, especially in applications with difficult load conditions, such as microwave-powered plasma lamps. Object of the invention An object of the invention is to achieve more stable operation of magnetrons, especially in difficult load conditions.

A further object of the invention is to provide more accurate control of the power supply to a magnetron.

A further object is to achieve a power supply which shows very small losses.

SUMMARY OF THE INVENTION The above and other objects are achieved by means of a method and an apparatus having the features set forth in the appended claims. Thus, according to one aspect of the invention, power supply to a magnetron from an alternating voltage source is provided. The power supply comprises rectifying the alternating voltage from the alternating voltage source to a rectifying voltage which is then converted to a rectified high voltage for operation of said magnetron. Furthermore, the effect added to this conversion is regulated in dependence on a sensed prevailing effect.

The invention is thus based on the insight of the advantageousness of regulating the power in a step which precedes the conversion from rectified low voltage to rectified high voltage. Since the power control is performed before this conversion, the conversion itself does not have to include power regulating elements, which means that the efficiency of the conversion can be considerably improved.

The inventors have thus realized that the influence of load variations, mains supply disturbances, temperature influence and similar factors on the work of the magnetron can be reduced by constantly ensuring that the power obtained from an initial power supply stage is constant.

It is thus preferred that the sensed prevailing power is compared with a set desired power and that the power applied to said conversion is thus regulated so that the sensed prevailing power is kept substantially constant equal to the desired power.

Instead of using only a current value or voltage value as feedback for regulating the measurement, the inventors have realized that considerably increased control / control of the work of the magnetron is achieved only when the power regulation is based on a "true" power value.

Furthermore, the inventors have realized the advantage that this power regulation takes place in dependence on a power which is sensed in connection with the initial rectification and at least before the conversion to high voltage.

To avoid having to retrieve information about the current power level from the high voltage generating 510 stage or closer to the magnetron, where the high voltage levels make it difficult to sense a "true" power value, it is preferred that the power be sensed before said conversion.

Due to the fact that both power sensing and power regulation in dependence thereof for constant keeping of the power is performed before the conversion from rectified low voltage to rectified high voltage, this conversion can be designed without included regulating or feedback elements, ie the conversion can be given an optimal design with very low losses. This means that you actually have very good control over the power losses in the conversion step¿ By controlling the power before an optimized conversion step, you do not need to know the specific conditions closer to the microwave, as these in principle follow directly from the controlled / regulated power.

According to a preferred embodiment of the present which is emitted from said AC voltage source to be sinusoidal, or that and / or that invention is also regulated, the current having the same waveform as the AC voltage, is substantially in phase with the AC voltage. The power unit thus has a very high power factor and it is ensured that no harmonics to the mains frequency pass back out on the public electricity grid.

It has been found that this regulation of the current and the same According to and the regulation of the power can be integrated in a unit / function in a very advantageous manner. yet another embodiment of the invention thus comprises the control of high frequency switching of the rectified signal for outputting a regulated, constant and power factor controlled power.

Advantageously, a prevailing current and a prevailing voltage are sensed, which together give said prevailing effect. These prevailing current and voltage values can then be used to achieve both power control and current control as above. Thus, according to a particularly preferred embodiment of the invention, the measured current and voltage are used in a treatment operation, forming the ratio between the desired power and the average value of the prevailing voltage, which gives a value of the instantaneous desired average value of the current. This desired average value is then scaled by the ratio between the prevailing voltage and the average value of the prevailing voltage, which gives the phase factor of the voltage, in order to obtain a momentarily desired current value.

Based on the treatment above, or similar operations that provide a desired current value, the regulation takes place by relating the desired current value to the prevailing current. The current and voltage are then regulated so that the power is kept constant and the current is increased or decreased depending on this relationship.

As mentioned above, the invention has the great advantage that the converting step can be designed to have a high efficiency. According to a preferred embodiment, the conversion from rectified low voltage to rectified high voltage is effected by the rectified voltage being alternated at a relatively high frequency, after which the alternating voltage is transformed into a high voltage alternating voltage which is finally rectified to said rectified high voltage for magnetron operation.

Since these steps lack control, regulation, feedback or the like, input circuits and functions can be tuned in a very advantageous manner so that each alternating switching takes place at a current zero crossing and thus substantially without losses. Gear direction is advantageously performed by means of two transistors which are active alternately to produce the generally pulse-shaped, alternating voltage. Since the frequency or on-off ratio of the gear direction need not be varied to control the supply, since this control has already been achieved in a previous step, the directional work performed by the transistors can take place 510 us in 6 at an intermittent factor of 1, which enables said tuning, which in turn allows switching at current zero crossings. This is one of the main factors contributing to the high efficiency.

Further advantages, aspects and features of the present invention will become more apparent from the description below. Brief Description of the Drawings The invention will now be described in the form of exemplary embodiments, which are given with reference to the accompanying drawings, in which: Fig. 1 schematically shows a typical ratio of anode voltage to anode current for a magnetron; Fig. 2 shows a block diagram of a power unit according to which shows control according to prior art; Fig. 3 schematically shows a block diagram of a power unit according to a first embodiment of the present invention; Fig. 4 schematically shows a block diagram of a power unit according to a second embodiment of the present invention; Fig. 5 schematically shows a block diagram of a power supply according to a third embodiment of the present invention; Fig. 6 schematically shows a block diagram of a power unit according to a fourth embodiment of the present invention; Fig. 7 schematically shows a concretized embodiment of a device according to the present invention; Figs. 8a to 8e schematically show signal curves occurring in the circuit of Fig. 7; Fig. 8e schematically shows a simplified principle diagram of elements included in the circuit of Fig. 7; and Figs. 8f and 8g schematically show signal curves occurring in the circuit of Fig. 8e. Detailed Description of Preferred Embodiments Fig. 1 shows a schematic diagram of a typical relationship between the anode voltage U of the magnetron and the anode current I. With the exception of extremely small currents, the magnetron has a relatively low dynamic impedance.

Fig. 2 schematically shows a block diagram of an example of a known power supply for magnetrons, which comprises a rectifying stage 1, which rectifies the alternating voltage from an alternating voltage source to a rectified voltage, and a voltage converting stage 2, which converts the rectified voltage to a rectified high voltage for feeding a magnetron 3. In the known unit a feedback of the anode current takes place to a control which is incorporated in the voltage converting stage 2.

Fig. 3 schematically shows a block diagram of a power unit according to a first embodiment of the present invention. In Fig. 3, a power regulating stage 4, 5 is arranged between the rectifying stage 1 and the voltage converting stage 2. The power regulating stage 4, 5 senses the power of the voltage supplied to the converting stage 2, compares this with a set desired power Pset and regulates continuous effect as a result. Since there is no feedback or control in step 2, the converting step can be designed to have an optimally high efficiency, as will be described in more detail below.

Fig. 4 shows a block diagram of a power unit according to a second embodiment of the invention. The power unit in Fig. 4 comprises a first rectifying stage 11, a power regulating stage 12, frequency switched, alternating stage 14, a preferably high up-transformation stage 15 and a second rectifying stage 16, and the magnetron itself 17. A sensing / regulating stage 13 senses the power applied to the alternating stage 14 and controls the power regulating stage 12 so that this power is equal to the Pset, i.e. so that the power supplied to the subsequent stage 14 is kept constant.

Since steps 14, 15 and 16 lack control, regulation, feedback or the like, these steps are tuned so that each alternating switching takes place at a current zero crossing and thus substantially without losses. Gear direction in step 14 is achieved by means of two transistors operating alternately. Since power control has already been achieved in steps 12, 13, the transistors in step 14 can operate at an intermittent factor of 1.

The inductances and capacitances present in the three circuits are dimensioned to have a resonant frequency which has a ratio, 'usually equal to 1/1, to the switching frequency in step 14, which enables said tuning with switching in current zero crossings.

In the embodiment in Fig. 5, the block diagram in Fig. 4 has been supplemented with a power factor regulating step 18 which ensures that the current output from the mains AC follows the shape of the voltage curve, which is usually sinusoidal, and that no harmonics are emitted on the general mains.

With this arrangement, both power control and power factor control are achieved in the initial steps without affecting the tuned steps 14, 15 and 16.

Fig. 6 shows schematically how the power regulating the power factor regulating functions can be integrated in Fig. 6 retrieves the prevailing power at the rectifying one and the same control unit 19. The control unit 19 at the output of the stage, but could just as easily sense the power at the input to stage 14, in the embodiments above.

A more concrete embodiment of a power unit for a magnetron according to the invention will now be described with reference to Fig. 7.

The power supply in Fig. 7 consists of four function blocks: a power regulating block S1, a high voltage generating block S2, a control / monitoring block S4. Furthermore, the power supply is a glow current regulating block S3 and is arranged to drive a magnetron V1. In Fig. 7, the different blocks are schematically separated by dashed lines.

The structure of the power regulating block SL will now be described in detail.

The power regulating block S1 of the power unit has inputs 10 and 20 which are connected to a mains voltage AC, which is preferably obtained from the general mains supply which delivers an alternating voltage of 50 or 60 Hz.

Inputs 10 and 20 are connected to a full-wave rectifying diode bridge Br1.

The positive output of the diode bridge Br1 is connected to a first output 30 from the block S1 via an inductor L1 in series with a diode D1. The negative output of the diode bridge Br1 is connected to a second output 40 from the block S1 via a resistor Rs. A transistor T1 has its collector / drain connected to a point between the inductor L1 and the diode D1 and has its emitter / source connected to the second output 40 of that socket S1. Furthermore, capacitor C1 is connected between the two outputs of the block S1. Together, the inductor L1, the diode D1, the transistor T1 and the capacitor C1 form a high-frequency switched power regulator.

The on and off times of the switch transistor T1 are controlled by a regulator circuit R which in this example consists of a power factor control circuit, more particularly an integrated circuit of the type UC3854 from Unitrode Integrated Circuits, the connection of which is modified to achieve the object according to the invention. A more detailed description of the functions of this circuit is given in the "Unitrode Integrated Circuits Application Note", pages 303-310.

The controller circuit R comprises a squared X, a multiplier / divider M / D, a comparator C and a control circuit PWCI. A control signal Pæt (“Power Set”) from the control / control block S4 is connected to an input A of the multiplier / divider M / D. Furthermore, the positive output of the diode bridge Br1 is connected partly to an input B of the multiplier / divider M / D, partly via a low-pass filter LE and the squared X to an input C of the 510 '25 033 multiplier / divider M / D. The M / D output of the multiplier / divider is connected to a first input to the comparator C and to the negative output of the diode bridge Brl via a resistor R1. The second input of the comparator C is connected to the second output 40 of the power regulating block S1. Furthermore, the output of the comparator C is connected to an input of the control circuit PWC1, the output of which is connected to the control of the transistor T1.

The function of the power control block S1 will now be described.

The power regulating block S1 is arranged in accordance with the invention to ensure that a constant power, the magnitude of which is determined by the signal Ps fi from the control block 40, is applied to the subsequent block stages independently of mains voltage variations or variations in the magnetron load or internal properties.

The power regulating block S1 also ensures that the power outlet from the mains is sinusoidal and is in phase with the mains voltage.

The incoming mains voltage AC is rectified in a conventional manner in the diode bridge Br1 so that a pulsating voltage Vm is obtained across the positive and negative output of the diode bridge Br1. Examples of voltage curves for the mains voltage AC and the rectified voltage Vm from the diode bridge Br1 are shown in Figs. 8a and 8b, respectively.

The current from the rectifier bridge Br1 is then switched at a high frequency by the power regulator L1, D1, T1, E1 so that it flows alternately through the transistor T1 and the diode D1. the high-frequency, high-frequency-switched power regulator normally “chops” 100 or 120 Hz, the current with a frequency of the order of 100 pulse-kHz or higher.

In this specific embodiment, the high-frequency switched power regulator emits a rectified output voltage which is always greater than the input voltage. The subsequent block must therefore in this case be dimensioned so that it can always be supplied with a voltage which is higher than the peak value of the highest occurring mains voltage AC.

The work of the high frequency switched voltage converter will now be described in more detail. When the transistor T1 is switched on, the current grows through the inductor L1, which thereby stores energy. When the transistor T1 is switched off, the current through the inductor L1 will charge the capacitor C1 via the diode D1, whereby the current through the inductor L1 decreases. Before the current has reached zero, the transistor T1 is switched on again, which means that the converter operates with continuous current.

The larger the proportion of the on-off period of the transistor T1 that the transistor T1 is turned on, i.e. the larger the pulse width of the signal applied to the control of the transistor, the greater the difference between the output voltage across the capacitor C1, i.e. across the power regulating block S1 outputs. Br1, i.e. the transistor T1, controls the voltage obtained across the outputs 30, 40 of the power regulating stage.

The curves in Figs. 8c, 8d and 8e show the current through the energy storage inductor L1 for three different pulse width ratios but with the same period time. In Fig. 8c the switch-on time is 20% of the period, in Fig. 8d the switch-on time is 50% and in Fig. 8e 80% of the period. The derivative of the increasing current in the curves in Figs. 8c to 8e is proportional to the voltage from the rectifier bridge Br1 and the derivative of the decreasing current is proportional to the difference between the voltage across the capacitor C1 and the voltage from the diode bridge Br1.

The ratio between the time that the transistor T1 is switched on and the time that the transistor T1 is switched off thus controls the power supplied to the subsequent blocks. According to the embodiment in Fig. 7, this relationship is controlled by the regulator circuit R. The interaction of the control unit R with the transistor T1 thus ensures that the power emitted from the power regulating block S1 is constant and independent of load 510 "12 '25 or variations in the mains voltage AC , taking into account the desired power level indicated by the signal Put obtained from the control / monitoring block.

The selected power level, which is represented by the control voltage Put (“Power Set”) is supplied to the input A on the multiplier / divider M / D. The voltage Vm from the rectifier bridge Br1 is filtered in the low-pass filter LF, squared in the squared X and then fed to the multiplier / divider's M / D input C. The voltage Vm is also fed directly to the multiplier / divider's M / D input B. The voltage from the rectifier bridge Dbl is used as more or less normal for what the current curve shape should look like. In the equations below, for simplicity, it is assumed that the mains voltage is sinusoidal. The average value Vm of the rectified mains voltage Vm is obtained from the low-pass filter LP and is supplied via the square X to the input C.

The following applies to the voltage Vw and its mean value Vm: vin = Jï - vm -sin mt (1) The function of the multiplier / divider can be schematically described as follows. By dividing the desired effect Pæt by the prevailing average value Vm on the voltage, a value is obtained that corresponds to the average value that the current must have in order to give rise to the desired effect Pæt at the current time. By dividing the instantaneous value of the voltage VM by the prevailing average value Vm of the voltage, a factor is obtained which represents the waveform of the voltage. The desired average value of the current (which ensures the correct power) multiplied by the waveform of the voltage (which ensures that the current is in phase with the voltage) thus provides a measure of the instantaneous value that the current must have to meet these two requirements.

In the multiplier / divider M / D the quantity K is therefore formed, where oo: 10 13 516 033 A 'B Psez' Vin Psec CV 'VV mm IH Zsinmt (2) The output signal K from the multiplier / divider M / D is thus generated as the current " desired “current. This is compared by the comparator C with the current "prevailing" current Is. By means of the resistor R1, the output signal K is represented as a voltage K x R1, which is applied to one input of the comparator C and which has its reference point in the negative output of the rectifier bridge Br1. Correspondingly, the current Is is represented by the resistor Rs as a voltage Is x Rs, which is applied to the second input of the comparator C and which also has its reference point in the negative output of the rectifier bridge Br1.

The difference between the voltages K x R1 and Is x Rs applied to the comparator C forms an "error signal" which the power factor control circuit strives to eliminate by the comparator controlling the transistor T1 via the pulse width control circuit PWCI as above. As long as K is greater than Is, the transistor is kept turned on and when Is becomes greater than K, the transistor is turned off. In this way, the voltage drop across Rs is made to be equal to the voltage drop across R1, ie if one disregards the high-frequency ripple: s -Rs = K -R1 (3) which together with equation (2) gives R1 R1 P s = šš -K = šš ':? ¿' Jï -sin Apparently, the current Is follows the waveform of Vin. Furthermore, the power is obtained as the average value of the voltage VM multiplied by the current Is: 510 us in 14 l5 '25, which shows that the power is kept equal to Pat as close to the factor R1 / Rs.

According to the embodiment in Fig. 7, the power regulating block S1 thus performs several advantageous functions: on the one hand, the power from the diode bridge Br1 is regulated to be constant, depending on Psa, and on the other hand it is ensured that the current Is taken from the general mains has the same waveform as the mains voltage. AC and is in phase with the mains voltage. The power unit thus has a very high power factor. The power regulating block S1 further comprises a mains filter (not shown) which ensures that no high frequency variations in the current are applied to the general mains. The description of the embodiment in Fig. 7 will now focus on the high voltage converting block S2. block S2 a high-frequency switched half-bridge with capacitive voltage division, a transformer and a voltage-doubling rectifier, control circuit for reasons which will be described below.

In addition, there is a pulse width More specifically, the block S2 comprises the pulse width control circuit PWC2 which has an input which receives a control signal from the control / control block S4 and two outputs which are connected to the primary winding of a first transformer Tr1.

Two transistors T2 and T3 are connected in series between the outputs 30 and 40 of the block S1, furthermore two capacitors C2 and C3 connected in series are connected in parallel with the transistors T2 and T3. A first secondary winding of the transformer Tr1 is connected at one end to the control of the transistor T2 and is connected at its other end to the emitter / source of the transistor T2. A second secondary winding of the transformer Tr1 is connected at one end to the control of the transistor T3 and is connected at its other end to the emitter / source of the transistor T3.

Furthermore, the high voltage converting block S2 comprises a second transformer Tr2 having a primary winding one end of which is connected to a point between the two transistors T1 and T2 and the other end of which is connected to a point between the two capacitors C2 and C3 via an inductance L2.

The high voltage converting block S2 further has a first 50 and a second 60 output which are connected to the cathode V1 cathode and anode, respectively. In Fig. 7, the anode is connected to ground. An inductor L3 and two capacitors C4 and C5 are connected in series between the outputs 50 and 60. Furthermore, two diodes D2 and D3 are connected in series in parallel with the capacitors C4 and C5. The secondary winding of the transformer Tr2 is connected at one end to a point between the two diodes D2 and D3 and is connected at its other end to a point between the two capacitors C4 and C5.

The block S2 further comprises a third transformer Tr3, the winding of the transporter Tr2, with a secondary winding one end of which is connected to the other of the block S1 whose primary winding is in series with the primary output 40 and the other end of which is connected via a diode D4 to an input of the pulse width control circuit PWC2.

The function of the high voltage converting block S2 will now be described.

The output signal from the pulse width control circuit PWC2 controls, via the transformer Tr1, the transistors T2 and T3 to conduct alternately. The switching frequency of the transistors is, for example, 50 kHz. The frequency used depends, among other things, on the power level at which the magnetron is intended to operate.

During normal operation, the high-voltage converting block S2 is operated with an intermittent factor ("duty-cycle") of 100%, the sistors T2, T3 in the high-voltage generating block S2, ie the ratio between the time of any of the transformers and the total period time is equal to 1 The signal from the pulse width control circuit PWC2 causes each transistor T2, T3 to conduct 50% of the time. At start-up, the desired oscillation has been built up in this "quasi-resonant 510 055 in 16 nanta" converters, the transistors T2 and T3 are activated for a shorter time, ie with an intermittent factor lower than 1.

This type of start-up of the converter is usually called a “slow-start”. It will be appreciated that this initial control of the intermittent factor of the high voltage converting block S1 is for the purpose of providing a smooth start of the converter and thus is not intended to be used for continuous control of the power supplied to the magnetron during normal operation, as occurs. in prior art.

In normal operation, when the converter S2 operates at full duty cycle, half the output voltage from the power regulating block S1, which is at a relatively constant level, is applied with varying polarity on the primary winding of the transformer Tr2 which transforms the voltage sharply, usually to i of the order of a few kilovolts.

The leakage inductance of the transformer Tr2, in this case supplemented by a separate inductor L2, is approximately tuned to the switching frequency by means of the capacitors C2 and C3. The transistors T2 and T3 will therefore be switched on and off when the current through them is close to zero. The switch losses are consequently very small.

This is a very advantageous aspect of the invention. Since the converter S2 operates with intermittent factor 1, this tuning of the oscillation to the switching frequency is allowed, which in turn makes it possible to switch over the transistors T2 and T3 when the current is zero.

If the converter were to operate with a lower or alternating intermittent factor, this tuning would not be possible, ie reversing of transistors T2 and T3 would not take place at zero crossings and therefore lead to unnecessary reversal losses.

The utilization of a tuned oscillation with intermittent factor 1 consequently gives an increase in the efficiency of the high voltage converting stage S2.

The function of the quasi-resonant converter S2 can be partially illustrated by the schematic block diagram in Fig. 17 516 ”033 Fig. 8f. In Fig. 8f, the supply to the resonant circuit, which schematically consists of an inductor L and a capacitor C, is switched between the plus and minus half supply voltages E by means of the transistors Ta and Tb, and the load, which is schematically symbolized by the resistor r, is connected to ground. .

Figs. 8g and 8h show examples of waveforms for the voltage V and the current I in Fig. 8f, which clearly show that the current at the moment of switching or switching is very close to zero.

Referring again to Fig. 7, the series-connected diodes D2 and D3 and the capacitors C4 and C5 constitute a voltage-doubling rectifier ref, which on the one hand gives lower requirements on the transformer Tr2's turnover, and on the other hand contributes to overload overload in the microwave, which will be discussed below. The diodes and capacitors are connected in separate arms by a bridge, where the secondary line of the transformer Tr2 is connected to the diagonal which consists of the connection point for the two diodes and the two capacitors, respectively, and the output voltage is taken from the other diagonal, where the diodes are The inductor L3 ensures that the magnetron V1 is supplied with an inductive generator impeller connected to the capacitors. dance, which is normally required for stable operation.

When the magnetron is exposed to very difficult load conditions, flashover can occur in the microwave circuits.

In order not to cause damage to the magnetron or in connected circuits, the current is monitored by means of the transformer Tr3 which via the diode D4 gives a signal to the pulse width control circuit PWC2 to temporarily switch off the switching of transistors T2 and T3 when an override or a transient occurs in the microwave.

The energy that can be fed to the microwave in that position for half a period is limited to the energy stored in the capacitor C3 or C4 since the previous half period, depending on the half period of the switching that the interruption is made. 053 '18 The glow control unit S3 in Fig. 7 comprises an FCC control unit. The FCC control unit emits a glow current in a conventional manner which is fed through the V1 cathode of the magnetron. The control unit FCC receives its supply from the power regulating block S1, but can on the other hand be designed to use its own control function, independent of the power regulator's work, for regulating the size of the glow current. The FCC's operation can also be controlled by various parameters obtained from the control block S4.

The control / control block S4 comprises a microprocessor CPU which provides control and monitoring of the work of the power unit. The microprocessor provides control signals and monitors (not shown) the other function blocks. More specifically, the microprocessor CPU is connected to the input A of the multiplier / divider M / D in the control circuit R for supplying the signal Pwt thereto, which signal indicates the desired power level thereto. The microprocessor CPU is also connected to the pulse width control circuit PWC2 in the high voltage converting block S2 for controlling the above-mentioned "slow-start" function which is used in slow start or construction of a stable oscillation of the high-voltage converting stage S2. In addition, the microprocessor CPU is connected to the control unit FCC in the glow current controlling block S3 for supplying control signals thereto.

Furthermore, the microprocessor CPU can communicate with the outside world via an analog control voltage (for controlling the power of the magnetron) or via a digital, preferably serial interface (not shown) which can be bidirectional.

In a conceivable design, the interface can use the force measurement as a communication link. Although the invention has been described with reference to specific exemplary embodiments, it will be appreciated by those skilled in the art that several different modifications, variations and combinations of the various embodiments shown may be made within the scope of the invention, as defined by the appended claims.

It is realized, for example, that control / monitoring of the power unit's work can be realized in many different ways, the embodiment now described. and the invention is of course not limited to Although the invention is described herein as related to magnetron power supply, it will be appreciated that certain aspects of the invention may be applicable in other types of power supply contexts.

Claims (22)

510 us 20 lO 15 20 25 30 35 PATENT CLAIMS A method of power supply to a magnetron from an AC voltage source, comprising the steps of: a) converting said AC voltage to a rectified voltage; b) preferably converting said rectified voltage to a high frequency alternating voltage by means of high frequency switching; c) transforming said high frequency alternating voltage into a high frequency high voltage alternating voltage; and d) rectifying said high voltage AC voltage to a rectified high voltage supplied to said magnetron for operation thereof; comprises regulating the applied power in said step a) to said conversion in step b) depending on a sensed prevailing power level. The method of claim 1, wherein said power control step comprises: comparing the sensed current power level with a set desired power level; and controlling the power applied to said conversion in step b) so that the sensed prevailing power level is kept substantially constant equal to the desired power level. A method according to claim 1, wherein said prevailing power level is sensed before step b). A method according to any preceding claim, comprising regulating the current supplied from said AC voltage source to be sinusoidal, or to have the same waveform as the AC voltage, and / or to be substantially in phase with the AC voltage. 10 15 20 25 30 21 5104153 A method according to claim 4, wherein said current control step is performed integrated with said power control step. A method according to any preceding claim, wherein said control step comprises controlling the power applied to said conversion in step b) by means of high frequency switching. A method according to any preceding claim, wherein said current power level is sensed by sensing a current current and a current voltage giving said current power level. The method of claim 7, comprising the steps of: forming the ratio between said desired power level and the average of said prevailing voltage as a value of the instantaneous desired average of the current; scaling the instantaneous desired average value of the current with the ratio between said prevailing voltage and the average value of the prevailing voltage to obtain an instantaneous desired current value; and relating the desired current value to said prevailing current and performing said regulation in dependence on this relationship. A method according to any preceding claim, comprising tuning steps b) to c) so that each alternating switching in step b) takes place at current zero crossings and thus substantially without losses. A method according to any preceding claim, wherein said gear direction is performed at an intermittent factor of 1,510,053 in 22 10 15 20 25 30 35 An apparatus for supplying power to a magnetron from an AC voltage source, comprising: Brl) for rectifying said AC voltage to a DC voltage; (14: T2, T3) rectified voltage to an alternating voltage with relatively first rectifying means (l; ll; means for alternating direction of said high frequency; means (l5; Tr2) for transforming said alternating voltage into an alternating high voltage; and ( 16; D2, D3, C4, C5) rectifying said alternating high voltage to a second rectifying means for rectifying high voltage for operating said magnetron; means (5; 13; 19; R) power; means (4; 12; for sensing a prevailing and R, L1, T1, Cl, D1) for regulating the power supplied to said alternating means before said alternating means in dependence on the sensed prevailing effect. The device according to claim 11, wherein said control means are arranged to compare the sensed prevailing power with a set desired power and to control the power applied to said alternating means so that the sensed prevailing power is kept substantially constant with the desired power. Device according to claim 11 or 12, wherein said means for sensing a prevailing power are arranged to sense said power before said alternating means. Device according to any one of claims 11 to 13, (18: 19; R) regulating the current emitted from said AC voltage, said control means being arranged to source to be sinusoidal, or to have the same waveform as the AC voltage, and / or to be substantially in phase with the alternating voltage. 10 15 20 25 30 35 23 i 5:10 101313- Device according to claim 14, wherein said control means (19: R) control is integrated with said power control. are arranged to provide said current Device according to any one of claims 11 to 15, wherein said control means comprises a high-frequency switching converter (T1). Device according to any one of claims 11 to 16, wherein said sensing means are arranged to sense a prevailing current and a prevailing voltage which gives said prevailing effect. The apparatus of claim 17, wherein said control means comprises processing means (M / D) arranged to form the ratio between said desired power level and the average value of said current voltage as a value of the instantaneous desired average value of the current and to scale this value with the ratio between said prevailing voltage and the average value of the prevailing voltage to produce an instantaneous desired current value, said control means being arranged to relate the desired current value to the prevailing current and perform said regulating of the power depending on this relationship. The apparatus of claim 18, wherein said control means further comprises comparator means (C) for relating said desired current value to said current current and outputting a signal intended to form the basis for said control of the power in dependence on this relationship . Device according to any one of claims 11 to 19, comprising means (L2, C2, C3, C4, C5 Tr2) for tuning said alternating means so that each alternating switching takes place at current zero crossings and thus substantially without losses. 510 035 24 Device according to any one of claims 11 to 20, wherein said alternating means comprises two transistors (T2, T3) which are active alternately for generating said alternating voltage. Device according to any one of claims 11 to 21, wherein said alternating means (T2, T3) operates at an intermittent factor of 1. 10