RU2788974C1 - Aerosol delivery device - Google Patents

Abstract

FIELD: tobacco industry.

SUBSTANCE: invention relates to an aerosol delivery device and an aerosol delivery system. The aerosol delivery device contains several electrical components including a controller containing a processor and a data carrier containing a set of machine-readable instructions; the specified several electrical components are located in direct proximity to an outer surface of the specified device; a heating node containing an insulating element, an induction coil, and a heating component for heating, but not burning, of aerosol forming material by means of electromagnetic induction; and a battery for supply of power to the specified electrical components and the heating node under control of the specified controller. In this case, during a heating session, energy consumption by the heating node is from 15 to 25 W, and energy consumption by the specified electrical components is less than 0.1 W, so that a temperature of the outer surface of the specified device remains lower than 48°C. An aerosol delivery system containing such a device is also described.

EFFECT: energy saving.

13 cl, 18 dwg

Description

The technical field to which the invention belongs

The present invention relates to an aerosol supply device and an aerosol supply system.

State of the art

Smoking articles such as cigarettes, cigars, etc. during their use, tobacco is burned to produce tobacco smoke. Attempts have been made to provide alternatives to these products that burn tobacco by creating products that release compounds without burning. Examples of such an article are heating devices that release compounds by heating, but not burning, the material. The material may be, for example, tobacco or other non-tobacco products, which may or may not contain nicotine.

The essence of the invention

According to a first aspect of the present invention, an aerosol delivery device is provided, comprising:

several electrical components;

a heating unit containing a heating component for heating the aerosol-forming material; and

a battery for supplying power to the electrical components and the heating unit;

wherein, in use, the power consumption of the multiple electrical components is less than about 0.25 watts.

According to a second aspect of the present invention, an aerosol delivery device is provided, comprising:

several electrical components;

a heating unit containing a heating component for heating the aerosol-forming material; and

a battery for supplying power to the electrical components and the heating unit;

moreover, when using the invention, the energy consumption of these several electrical components is less than about 1% of the energy consumption of the heating unit.

According to a third aspect of the present invention, an aerosol delivery system is provided, comprising:

an aerosol supply device according to the first aspect; and

an article containing an aerosol-forming material.

Other features and advantages of the invention will become apparent from the following description of preferred embodiments of the invention given by way of example with reference to the accompanying drawings.

Brief description of the drawings

In FIG. 1 is a front view of an example of an aerosol generating device;

in fig. 2 is a front view of the aerosol generating device shown in FIG. 1, with outer cover removed;

in fig. 3 is a cross-sectional view of the aerosol generating device shown in FIG. 1;

in fig. 4 is a partial view of the aerosol generating device shown in FIG. 2;

in fig. 5A is a cross-sectional view of an example of a heating unit in an aerosol generating device;

in fig. 5B is an enlarged view of a portion of the heating assembly shown in FIG. 5A;

in fig. 6 is a schematic representation of an example of an induction heating circuit for the aerosol generating device shown in FIG. 1-5B;

in fig. 7A is a schematic representation of the current passing through an inductor for the example induction heating circuit shown in FIG. 6;

in fig. 7B is a schematic representation of the voltage across the current sensing resistor for the example induction heating circuit shown in FIG. 6;

in fig. 8 is a schematic representation of the voltage across the switching device of the circuit shown in FIG. 6;

in fig. 9 is another schematic representation of an example of an induction heating circuit for the apparatus shown in FIG. 1-5B;

in fig. 10-13 show various parts of an example control device for the example induction heating circuit shown in the previous figures;

in fig. 14 is a flow chart of an example method for controlling aspects of an induction heating circuit for an illustrative example of an induction heating circuit;

in fig. 15 is a flowchart of another example of a method for controlling aspects of an induction heating circuit for an illustrative example of an induction heating circuit; and

in fig. 16 is a schematic representation of the temperature of the inductor and the target power supplied to the inductor during operation of the induction heating circuit for an illustrative example of an induction heating circuit.

Detailed description of the invention

As used herein, "aerosol-forming material" includes materials that release volatile components when heated, typically in the form of an aerosol. The aerosol forming material includes any tobacco-containing materials and may, for example, include one or more of the following materials: tobacco, tobacco derivatives, ground tobacco, reconstituted tobacco, or tobacco substitutes. The aerosol forming material may also include other non-tobacco products, which may or may not contain nicotine, depending on the product. The aerosol-forming material may, for example, be in solid form, in liquid form, in the form of a gel or wax. The aerosol forming material can also, for example, be a combination or mixture of materials. An aerosol forming material may also be known as a "smoking material".

An apparatus is known that heats the aerosol-forming material to vaporize at least one component of the aerosol-forming material, typically to form an aerosol that can be inhaled without burning or igniting the aerosol-forming material. Such a device is sometimes described as an "aerosol generating device", "aerosol providing device", "non-combustion heating device", "tobacco product heating device" or "tobacco heating device", and the like. Similarly, there are also so-called electronic cigarettes, which typically vaporize the aerosol-forming material in liquid form, which may or may not contain nicotine. The aerosol-forming material may be provided in the form of or as part of a rod, cartridge, or cassette, or the like, which may be inserted into the device. A heater for heating and evaporating the aerosol-forming material may be provided as a "permanent" part of the apparatus.

The aerosol providing device may receive an article containing an aerosol-forming material for heating. "Article" in this context is a component that includes or contains, in use of the device, an aerosol-forming material that is heated to vaporize the aerosol-forming material, and optionally other components used. The user may insert the product into the aerosol delivery device before heating it to produce an aerosol which is then inhaled by the user. The product, for example, may be of a predetermined or strictly defined size, that is, it can be placed in a heating chamber of the device, which is sized to receive the product.

The present invention describes an aerosol providing device comprising a plurality of electrical components and a heating assembly comprising at least one heating component for heating an aerosol-forming material. The heating unit can use resistive or induction heating. In a particular example, the heating assembly includes at least one coil. The coil may be, for example, an inductor. Thus, the heating component may be a current collector and the coil may heat the current collector, which in turn heats the aerosol generating material. The coil may be an inductor or an inductor.

The device also contains a battery for supplying power to the electrical components and the heating assembly. In use, the power consumption of multiple electrical components is less than about 0.25 watts.

The device thus comprises a heating assembly configured to heat the aerosol-forming material and several other electrical components. Several electrical components may include a controller, such as a microprocessor, and a visual indicator containing, for example, several LEDs. Several electrical components do not include a heating assembly, such as a heater component and coil(s).

The power consumption of several electrical components is less than about 0.1 watts. It has been found that keeping the power consumption of electrical components below 0.1 W can reduce the surface temperature of the device as well as improve overall energy efficiency. For example, it may be desirable to maintain the surface temperature of the device below about 48°C or below about 43°C. Electrical components may have a greater effect on surface temperature than expected because they may be located closer to the surface than heating components and may be less well insulated. By using energy efficient components, the surface temperature of the device can be reduced. Also, if the electrical components consume less than 0.1W, then the device can be used for a longer period of time before the battery needs to be recharged.

The battery may be a DC source. The battery may be a battery module or a battery pack. The battery may be a 3-4V battery.

The power consumption of several electrical components may be less than about 0.05 W, less than about 0.04 W, or less than about 0.01 W.

In one example, multiple electrical components include a controller, and when the controller is active, the power consumption of the controller is from about 10 mW to about 20 mW, and when the controller is inactive, the power consumption of the plurality of electrical components is less than about 0.5 mW.

In one example, multiple electrical components comprise multiple LEDs, and each LED has a power consumption of less than about 0.01 watts at maximum brightness or intensity. In one example, there are four LEDs and the power consumption of the four LEDs is less than 0.04W at maximum brightness. For example, the power consumption of four LEDs may be about 0.03W.

Accordingly, several electrical components may include four LEDs and a controller, and the power consumption of the electrical components is less than about 0.5 mW when the controller is inactive and the LEDs are off. The power consumption of the electrical components is less than 0.05W when the controller is active and the LEDs are on. For example, the power consumption of electrical components may be from about 0.03 W to about 0.05 W when the controller is active and the LEDs are on.

During a heating session, the heating assembly may consume from about 15 watts to about 25 watts. For example, in use, the power consumption of the heating unit is from about 15 watts to about 25 watts. Preferably, the power consumption of the heating unit is from about 20 watts to about 25 watts, for example, from about 20 watts to about 23 watts.

In some examples, several electrical components include a WiFi interface and/or a Bluetooth interface and/or an NFC interface.

In some examples, the power consumption of several electrical components is less than about 1% of the energy consumption of the heating unit. In another example, the power consumption of several electrical components is less than about 0.5% of the power consumption of the heating assembly. Preferably, the power consumption of the electrical components is less than about 0.2% of the energy consumption of the heating unit. For example, the power consumption of several electrical components may be about 0.05 W, and the power consumption of a heating unit may be about 20 W.

The battery can have a capacity from 30,000 J to 35,000 J.

The heating unit may be configured to operate for a period of from about 3 minutes to about 5 minutes and may be configured to consume from about 1000 J to about 1400 J during this period. Thus, the battery can be configured to power the heating unit for about 20 to about 35 cycles. Each period may be known as a heating session. During this period, one or more inductors may be operated intermittently.

Preferably, the device is a tobacco heating device, also known as a non-burning heating device.

As briefly mentioned above, in some examples the coil(s) are(are) configured to, in use, cause at least one electrically conductive heating component/element (also known as a heater component/element) to be heated, such that thermal energy is transferred from the at least one electrically conductive heating component to the aerosol generating material, to thereby cause heating of the aerosol generating material.

In some examples, the coil(s) are(are) configured to generate, using an alternating magnetic field, to penetrate at least one heating component/element to thereby cause induction heating and/or magnetic hysteresis heating of at least one heating element. component. In such an arrangement, one or each heating component may be referred to as a "susceptor". A coil that is configured to generate by using an alternating magnetic field to penetrate at least one electrically conductive heating component to thereby cause inductive heating of at least one electrically conductive heating component may be referred to as an "induction coil" or "inductor".

The device may include a heating component(s), for example, an electrically conductive heating component(s), and the heating component(s) may be appropriately positioned or positioned relative to the coil(s) to provide such heating of the heating component(s). The heating component(s) may be in a fixed position relative to the coil(s). Alternatively, at least one heating component, such as at least one electrically conductive heating component, may be included in the article for insertion into the heating zone of the device, the article also containing an aerosol generating material and may be removed from the zone. heating after use. Alternatively, both the device and such article may comprise at least one suitable heating element, for example at least one electrically conductive heating element, and the coil(s) may cause heating of the heating element(s) of both device and article when the product is in the heating zone.

In some examples, the coil(s) is helical. In some examples, the coil(s) surrounds at least a portion of the heating zone of the device configured to receive the aerosol generating material. In some examples, the coil(s) is a helical coil that surrounds at least a portion of the heating zone. The heating zone may be a reservoir shaped to receive the aerosol generating material.

In some examples, the device comprises an electrically conductive heating component that at least partially surrounds the heating zone, and the coil(s) is a helical coil(s) that surrounds at least a portion of the electrically conductive heating component. In some examples, the electrically conductive heating element is tubular. In some examples, the coil is an inductor.

In FIG. 1 shows an example of an aerosol providing device 100 for generating an aerosol from an aerosol generating substance/material. In general terms, device 100 can be used to heat a replaceable article 110 containing an aerosol-forming agent to generate an aerosol or other inhalable substance that is inhaled by a user of device 100.

The device 100 includes a housing 102 (in the form of an outer cover) that surrounds and houses the various components of the device 100. The device 100 has an opening 104 at one end through which an article 110 can be inserted for heating by a heating assembly. In use, article 110 may be fully or partially inserted into a heating assembly where it may be heated by one or more components of the heating assembly.

The device 100 in this example includes a first end member 106 that includes a cover 108 that can be moved relative to the first end member 106 to close the opening 104 when the article 110 is not in place. In FIG. 1, lid 108 is shown in an open configuration, however, lid 108 may change to a closed configuration. For example, the user may slide cover 108 in the direction of arrow "A".

Device 100 may also include a user-operable control 112, such as a button or switch, that, when pressed, activates device 100. For example, a user may turn on device 100 using switch 112.

The device 100 may also include an electrical component, such as a socket/port 114, that can receive a cable to charge the battery of the device 100. For example, socket 114 may be a charging port, such as a USB charging port. In some examples, socket 114 may be used in addition to, or as an alternative to, communication between device 100 and another device, such as a computing device.

In FIG. 2 shows the device 100 shown in FIG. 1 with the outer cover 102 removed. The device 100 defines the longitudinal axis 134.

As shown in FIG. 2, a first end 106 is located at one end of the device 100 and a second end 116 is located at the opposite end of the device 100. The first and second ends 106, 116 together at least partially define the end surfaces of the device 100. For example, the bottom surface of the second end element 116 at least partially defines the bottom surface of the device 100. The edges of the outer cover 102 may also define part of the end surfaces. In this example, cover 108 also defines a portion of the top surface of device 100. FIG. 2 also shows a second printed circuit board 138 associated with the control element 112.

The end of the device closest to opening 104 may be known as the proximal end (or mouth end) of device 100 because it is closest to the user's mouth in use. In use, the user inserts article 110 into opening 104, operates user control 112 to begin heating the aerosol generating material, and draws in the aerosol generated in the device. This causes the aerosol to pass through the device 100 along a flow path to the proximal end of the device 100.

The other end of the device furthest from opening 104 may be known as the distal end of device 100 because it is furthest away from the user's mouth in use. When the user draws in the aerosol generated in the device, the aerosol passes from the distal end of the device 100.

The device 100 also contains a power source 118. The power source 118 may be, for example, a battery, such as a rechargeable battery or a non-rechargeable battery. Examples of suitable batteries include, for example, a lithium battery (such as a lithium ion battery), a nickel battery (such as a nickel cadmium battery), and an alkaline battery. The battery is electrically connected to the heating unit to supply electricity when needed and under the control of a controller (not shown) to heat the aerosol-forming material. In this example, the battery is connected to a central support 120 that holds the battery 118 in place. The power supply 118 may be a constant voltage source 118.

The device also includes at least one electronic module 122. The electronic module 122 may include, for example, a printed circuit board (PCB). Circuit board 122 may support at least one controller such as a processor and memory. Circuit board 122 may also include one or more electrical tracks for electrically interconnecting various electronic components of device 100. For example, battery terminals may be electrically connected to circuit board 122 so that power can be distributed throughout device 100. Socket 114 may also be electrically connected to the battery via electrical tracks.

In the exemplary device 100, the heating assembly is an induction heating assembly and includes various components for heating the aerosol-forming material of article 110 through an induction heating process. Induction heating is the process of heating an electrically conductive object (such as a current collector) using electromagnetic induction. The induction heating assembly may include an inductive element, such as one or more induction coils, and a device for passing a varying electrical current, such as an alternating electrical current, through the inductive element. The changing electric current in an inductive element creates a changing magnetic field. An alternating magnetic field penetrates through a pantograph appropriately positioned relative to the inductive element and creates eddy currents inside the pantograph. The current collector has an electrical resistance to eddy currents, and therefore the flow of eddy currents against this resistance causes heating of the current collector due to Joule heating. In cases where the current collector contains a ferromagnetic material such as iron, nickel or cobalt, heat can also be generated by magnetic hysteresis losses in the current collector, i.e. the changing orientation of the magnetic dipoles in the magnetic material as a result of their alignment with the changing magnetic field. With induction heating, compared to, for example, heating by heat transfer, heat is generated inside the current collector, which ensures rapid heating. In addition, there is no need for any physical contact between the induction heater and the current collector, allowing greater freedom in design and application.

The induction heating assembly of an exemplary device 100 includes a current collector structure 132 (referred to herein as a "susceptor"), a first inductor 124, and a second inductor 126. The first and second inductors 204, 206 are made of an electrically conductive material. In this example, the first and second inductors 124, 126 are made of winding wire/cable wound in a helix to form helical inductors 124, 126. The winding wire is made up of many individual wires that are individually insulated and twisted together to form a single wire. Winding wires are designed to reduce skin effect losses in the conductor. In the exemplary device 100, the first and second inductors 124, 126 are made from copper winding wire having a substantially circular cross section. In other examples, the winding wire may have a cross section of another shape, such as a rectangular one.

The first inductor 124 is configured to generate a first alternating magnetic field to heat the first section of the pantograph 132, and the second inductor 126 is configured to generate a second alternating magnetic field to heat the second section of the pantograph 132. Here, the first section of the pantograph 132 is called the first pantograph zone 132a , and the second pantograph section 132 is referred to as the second pantograph zone 132b. In this example, the first inductor 124 is adjacent to the second inductor 126 in the direction along the longitudinal axis 134 of the device 100 (ie, the first and second inductors 124, 126 do not overlap). In this example, current collector structure 132 includes one current collector containing two zones, however, in other examples, current collector structure 132 may include two or more current collectors. The ends 130 of the first and second inductors 124, 126 are connected to the circuit board 122.

It is understood that the first and second inductors 124, 126 may, in some instances, have at least one characteristic different from each other. For example, first inductor 124 may have at least one different characteristic than second inductor 126. More specifically, in one example, first inductor 124 may have a different inductance than second inductor 126. FIG. . 2, the first and second inductors 124, 126 have different lengths such that the first inductor 124 is wound on a smaller section of the current collector 132 than the second inductor 126. Thus, the first coil 124 may contain a different number of turns than the second coil 126 (provided that the distance between the individual turns is essentially the same). In another example, the first inductor 124 may be made from a different material than the second inductor 126. In some examples, the first and second inductors 124, 126 may be substantially identical.

In this example, the inductors 124, 126 are wound in the same direction. That is, both the first inductor 124 and the second inductor 126 are left-handed coils. In another example, both inductors 124. 126 may be right handed coils. In yet another example (not shown), the first inductor 124 and the second inductor 126 are wound in opposite directions. This can be useful when the inductors are turned on at different times. For example, the first inductor 124 may be operated first to heat the first section of product 110, and later the second inductor 126 may be operated to heat the second section of product 110. Winding the coils in opposite directions helps to reduce the current induced in the inactive coil when used. combined with a certain type of control scheme. In one example, the inductors 124, 126 are wound in different directions (not shown) where the first inductor 124 may be a right helix and the second inductor 126 may be a left helix. In another such embodiment, the first inductor 124 may be a left-handed helix and the second inductor 126 may be a right-handed helix.

The current collector 132 in this example is hollow and therefore limits the container in which the aerosol generating material is placed. For example, article 110 may be inserted into current collector 132. In this example, current collector 132 is tubular with a circular cross section.

Device 100 in FIG. 2 also includes an insulating element 128 which may be generally tubular and at least partially surround the current collector 132. The insulating element 128 may be made from any insulating material such as plastic. In this particular example, the insulating element is made of polyether ether ketone (PEEK). The insulating member 128 can help insulate the various components of the device 100 from the heat generated in the current collector 132.

The insulating member 128 may also fully or partially support the first and second inductors 124, 126. For example, as shown in FIG. 2, first and second inductors 124, 126 are located around insulating element 128 and are in contact with the radially outer surface of insulating element 128. In some examples, insulating element 128 does not abut against first and second inductors 124, 126. For example, there may be a small gap between the outer surface of the insulating member 128 and the inner surface of the first and second inductors 124, 126.

In a particular example, current collector 132, insulating element 128, and first and second inductors 124, 126 lie on the same central longitudinal axis of current collector 132.

In FIG. 3 shows a side view of the device 100 in partial section. In this example, the outer cover 102 is again missing. The circular cross-sectional shape of the first and second inductors 124, 126 is seen in FIG. 3 more clearly.

The device 100 also includes a support 136 that engages with one end of the current collector 132 holding the current collector 132 in place. The support 136 is connected to the second end element 116.

The device 100 also includes a second cover/cap 140 and a spring 142 located toward the distal end of the device 100. The spring 142 allows the second cover 140 to be opened to provide access to the current collector 132. For example, a user may open the second cover 140 to clean the current collector 132 and /or support 136.

The device 100 also includes an expansion chamber 144 that extends from the proximal end of the current collector 132 to the opening 104 of the device. At least partially within the expansion chamber 144 is a retaining clip 146 that abuts against the product 110 and holds it in the device 100. The expansion chamber 144 is connected to the end member 106.

In FIG. 4 is an exploded view of the device 100 shown in FIG. 1, again without outer cover 102.

In FIG. 5A shows a cross section of a portion of the device 100 shown in FIG. 1. In FIG. 5B is a close-up view of the area of FIG. 5A. In FIG. 5A and 5B show an article 110 placed in a current collector 132, with the article 110 sized such that the outer surface of the article 110 abuts the inner surface of the current collector 132. This provides the most efficient heating. The article 110 of this example contains an aerosol generating material 110a. Aerosol generating material 110a is disposed within current collector 132. Article 110 may also include other components such as a filter, wrapping materials, and/or a cooling structure.

In FIG. 5B shows that the outer surface of the current collector 132 is separated from the inner surface of the inductors 124, 126 by a distance 150, measured in a direction perpendicular to the longitudinal axis 158 of the current collector 132. In one particular example, the distance 150 is from about 3 mm to 4 mm, from about 3 mm to 3.5 mm or approximately 3.25 mm.

In FIG. 5B also shows that the outer surface of the insulating member 128 is separated from the inner surface of the inductors 124, 126 by a distance 152, measured in a direction perpendicular to the longitudinal axis 158 of the current collector 132. In one particular example, the distance 152 is approximately 0.05 mm. In another example, distance 152 is substantially 0 mm so that inductors 124, 126 abut against and touch insulating member 128.

In one example, current collector 132 has a wall thickness 154 of about 0.025 mm to 1 mm, or about 0.05 mm.

In one example, current collector 132 has a length of about 40 mm to 60 mm, about 40 to 45 mm, or about 44.5 mm.

In one example, the insulating member 128 has a wall thickness 156 of about 0.25 mm to 2 mm, 0.25 mm to 1 mm, or about 0.5 mm.

As discussed above, the heating assembly in the exemplary apparatus 100 is an induction heating assembly containing various components for heating the aerosol forming material of article 110 through an induction heating process. Specifically, the first inductor 124 and the second inductor 126 are used to heat the respective first 132a and second 132b zones of the current collector 132 to heat the aerosol-forming material and obtain an aerosol. Below with reference to FIG. 6-12 will describe in detail the operation of the device 100 using the first and second inductors 124, 126 to inductively heat the current-collecting structure 132.

The induction heating assembly of device 100 includes an LC circuit. An LC circuit has an inductance L provided by an inductive element and a capacitance C provided by a capacitor. In the device 100, the inductance L is provided by the first and second inductors 124, 126, and the capacitance C is provided by several capacitors, as will be described below. An induction heater circuit containing an inductance L and a capacitance C can in some cases be represented as an RLC circuit containing a resistance R provided by a resistor. In some cases, the resistance is provided by the ohmic resistance of the parts of the circuit connecting the inductive element and the capacitor, and therefore the circuit need not include a resistor as such. In such circuits, electrical resonance can be observed, which occurs at a certain resonant frequency, when the imaginary parts of the impedances or the admittance of the circuit elements cancel each other out.

One example of an LC circuit is a series circuit in which an inductive element and a capacitor are connected in series. Another example of an LC circuit is a parallel LC circuit in which an inductive element and a capacitor are connected in parallel. Resonance occurs in an LC circuit because the collapsing magnetic field of an inductive element generates an electric current in its windings, which charges the capacitor, while the discharging capacitor provides an electric current, which creates a magnetic field in the inductive element. When a parallel LC circuit is driven at the resonant frequency, the dynamic impedance of the circuit is maximum (since the reactance of the inductive element is equal to that of the capacitor) and the current in the circuit is minimum. However, for a parallel LC circuit, an inductive element and a capacitor connected in parallel act as a current multiplier (effectively multiplying the current in the loop and hence the current through the inductive element). Thus, if the RLC or LC circuit is allowed to operate at the resonant frequency for at least some of the time while the circuit is operating to heat the pantograph, efficient and/or efficient induction heating can be achieved by providing the highest value of the magnetic field penetrating the pantograph.

The LC circuit used in device 100 to heat current collector 132 may use one or more transistors to act as a switching device, as will be described below. A transistor is a semiconductor device for switching electronic signals. A transistor usually contains at least three leads for connection to an electronic circuit. A field effect transistor (FET) is a transistor in which the influence of an applied electric field can be used to change the effective conductance of the transistor. The FET may include a body, a source terminal S, a drain terminal D, and a gate terminal G. The FET contains an active channel containing a semiconductor through which charge carriers, electrons or holes, can flow between the source S and the drain D. The conductance of the channel, i.e. the conductance between the drain D and the source S, is a function of the potential difference between the gate terminals G and the source S, for example, generated by the potential applied to the terminal G of the gate. In enrichment mode FETs, the FET may be turned off (i.e. substantially prevent current from flowing through it) when there is essentially zero voltage between the gate G and source S, and may be turned on (i.e. essentially allow current to flow through it) when there is a substantially non-zero voltage between the gate G and the source S.

One type of transistor that may be used in the circuitry of device 100 is an n-channel (or n-type) field-effect transistor (n-FET). An n-channel FET is a FET whose channel consists of an n-type semiconductor in which electrons are majority carriers and holes are minority carriers. For example, n-type semiconductors may contain an intrinsic semiconductor (such as, for example, silicon) doped with donor impurities (such as, for example, phosphorus). In n-channel FETs, the drain terminal D is at a higher potential than the source terminal S (ie, there is a positive drain-to-source voltage, or in other words, a negative source-to-drain voltage). In order to turn on the n-channel FET (ie, allow current to flow through it), a switching potential is applied to the gate terminal G which is higher than the potential at the source terminal S.

Another type of transistor that can be used in device 100 is a p-channel (or p-type) field effect transistor (p-FET). A p-channel FET is a FET whose channel consists of a p-type semiconductor in which holes are majority carriers and electrons are minority carriers. For example, p-type semiconductors may contain an intrinsic semiconductor (such as, for example, silicon) doped with acceptor impurities (such as, for example, boron). In p-channel FETs, the source terminal S is at a higher potential than the drain terminal D (ie, there is a negative drain-to-source voltage, or in other words, a positive source-to-drain voltage). To turn on the p-channel FET (i.e., allow current to flow through it), a switching potential is applied to the gate terminal G which is lower than the potential at the source terminal S (and which may be, for example, higher than the potential at the drain terminal D ).

In examples, one or more FETs used in device 100 may be metal-oxide-semiconductor field-effect transistors (MOSFETs). The MOSFET is a field-effect transistor whose gate terminal G is electrically isolated from the semiconductor channel by an insulating layer. In some examples, the gate terminal G may be metallic and the insulating layer may be an oxide (such as, for example, silicon dioxide), hence the name "metal-oxide-semiconductor". However, in other examples, the gate may be made from materials other than metal, such as polysilicon, and/or the insulating layer may be made from materials other than oxide, such as other dielectric materials. Such devices are, however, commonly referred to as metal-oxide-semiconductor field-effect transistors (MOSFETs), and it is to be understood that the term field-effect transistors or MOSFETs as used herein should be interpreted to include such devices.

The MOSFET may be an n-channel MOSFET (or n-type transistor), where the semiconductor is n-type. An n-channel MOSFET (n-MOSFET, n-type MOSFET) can work in the same way as described above for an n-channel FET. As another example, the MOSFET may be a p-channel MOSFET (or p-type transistor), where the semiconductor is p-type. A p-channel MOSFET (p-MOSFET, p-type MOSFET) can work in the same way as described above for a p-channel FET. An n-type MOSFET typically has a lower source-to-drain resistance than a p-type MOSFET. Therefore, when on (i.e. when current is flowing through it), n-type MOSFETs generate less heat compared to p-type MOSFETs, and therefore can consume less power in operation than MOSFETs. p-type. In addition, n-type MOSFETs generally have a shorter switching time (i.e., the characteristic response time from a change in the switching potential applied to the gate G terminal to switching the current flow through the MOSFET) compared to p-type MOSFETs. -type. This can improve the switching speed and improve the switching control.

Now with reference to FIG. 6, a circuit for induction heating by device 100 will be described. In FIG. 6 shows a simplified schematic representation of a portion of the induction heating circuit 600 of the aerosol generating device 100. In FIG. 6 shows a portion of an inductive heating circuit 600 that includes a first inductor 124 for heating the first pantograph zone 132a when alternating current flows through the first inductor 124. The first pantograph zone 132a is shown in FIG. 6 as having an inductive element and a resistive element to show how the current collector 132 is inductively connected to the first inductor 124 and heated by the generation of eddy currents. It should be noted that device 100 also includes a second inductor 126, which is not shown in FIG. 6. The second inductor 126 is also part of the induction heating circuit 600 and is controlled to heat the second pantograph area 132b as will be described below. However, for the sake of clarity, circuit 600 will first be described with reference to features as shown in FIG. 6.

The circuit 600 includes a first resonator section 601, a constant voltage source 118 for supplying a constant voltage to the first resonator section 601, and a control device for driving the circuit 600. The first resonator section 601 includes a first inductor 124 and a switching device containing a first field-effect transistor 608, and the control device is configured to switch the FET 608 between the first state and the second state in response to the voltage detected in the circuit 600, as will be described in more detail below, to drive the first inductor 124. The circuit 600, except for the current collector 132, is located on the printed circuit board 122 of the device 100, while the inductor 124 is connected to the printed circuit board 122 at the first end 130a and the second end 130b.

The first resonator section 601 includes a first capacitor 606 and a second capacitor 610, both arranged in parallel with the first inductor 124 such that when the first resonator section 601 is allowed to resonate, an alternating current flows between the first capacitor 606 and the second capacitor 610 and through the inductor 124. As mentioned above, the first FET 608, in this example an n-channel MOSFET, is configured to operate as a switching device in the first resonator section 601.

It should be noted that in other examples, resonator section 601 may contain only one capacitor, such as in place of first capacitor 606 or in place of second capacitor 610. In other examples, resonator section 601 may contain any other number of capacitors, such as three or more capacitors. For example, the first capacitor 606 and/or the second capacitor 610 may be replaced by two or more capacitors in parallel with each other. As will be well understood, the resonator section 601 has a resonant frequency that depends on the inductance L and capacitance C of the resonator section 601. The number, type, and arrangement of capacitors in resonator section 601 may be selected based on the power levels to be used in circuit 600 and the desired frequency of circuit 600 operation. series resistance (ESR), as well as the limit of the ability of said capacitors to withstand current. Such features can be taken into account when determining the location of the capacitors to provide capacitance in the resonator section 601. For example, depending on the required power levels and frequency of operation, it may be advantageous to have multiple capacitors connected in parallel to provide higher capacitance or lower ESR. In this example, the first and second capacitors 606, 610 are C0G ceramic capacitors each having a capacitance of about 100 nF. In other examples, a capacitor and/or other types of capacitors with different capacitance values are used, for example, capacitors with different capacitance values may be used in accordance with the considerations set forth in this paragraph.

The first resonator section 601 is supplied with a constant voltage from a constant voltage source 118, which, for example, as described above, is the voltage supplied by the battery. As shown in FIG. 6, DC voltage source 118 includes a positive terminal 118a and a negative terminal 118b. In one example, a DC voltage source 118 supplies a DC voltage of about 4.2 V to the first resonator section 601. In other examples, DC voltage source 118 may supply, for example, 2 to 10 V, or about 3 to 5 V.

Controller 1001 is configured to control the operation of circuit 600. Controller 1001 may include a microcontroller, such as a microprocessor unit (MPU), having multiple inputs and outputs. In one example, controller 1001 is an MPU model STM32L051C8T6. In some examples, DC voltage source 118 provided in circuit 600 is output from controller 1001, which is itself powered by a battery or other power source.

The positive terminal 118a of the DC voltage source 118 is electrically connected to the first node 600A. In the example, a DC voltage source 118 is connected to a node 600A via a controller 1001 which receives power from the DC voltage source 118 and applies the voltage supplied by the DC voltage source to the components of the device, including circuit 600. The first node 600A is electrically connected to the first end 606a of the first capacitor. 606 and with the first end 130a of the first inductor 124. The second end 130b of the first inductor 124 is electrically connected to the second node 600B, which in FIG. 6 is represented at two electrically equivalent points in the circuit diagram. The second node 600B is electrically connected to the drain terminal 608D of the FET 608. In this example, the second node 600B is also electrically connected to the first end 610a of the second capacitor 610. Continuing to bypass the circuit, the source terminal 608S of the first FET 608 is electrically connected to the third node 600C. The third node 600C is electrically connected to ground 616 and, in this example, the second end 610b of the second capacitor 610. The third node 600C is electrically connected via a current-sensing resistor 615 to the fourth node 600D, and the fourth node 600D is electrically connected to the negative terminal 118b of the DC voltage source 118 current, which, like the positive terminal, is supplied through the controller 1001 in the example.

It should be noted that in examples where the second capacitor 610 is not present, the third node 600C may have only three electrical connections: to the FET's first source terminal 608S, to ground 616, and to the current-sensing resistor 615.

As mentioned above, the first FET 608 acts as a switching device in the first cavity section 601. The first FET 608 may be in a first state, ie, an "ON" state, and a second state, ie, an "OFF" state. As will be understood by those skilled in the art, if an n-channel FET is in the "OFF" state (ie, when an appropriate control voltage is not applied to its gate), then it is effectively acting as a diode. In FIG. 6, the diode functionality exhibited by the first FET 608 in the off state is represented by the first diode 608a. That is, when the FET 608 is in the off state, the first diode 608a operates, largely preventing current from flowing from the drain terminal 608D to the source terminal 608S, but allowing current to flow from the source terminal 608S to the drain terminal 608D, if the diode 608a is respectively forward-biased. An n-channel FET is in the on state when an appropriate control voltage is applied to its gate, so that there is a conductive path between its drain D and source S. Thus, when the first FET 608 is in the on state, it acts as a closed switch in the first resonator section 601.

As mentioned above, the circuit 600 can be considered to include the first resonator section 601 and an additional control device. The control device includes a comparator 618, a zero voltage detector 621, and a flip-flop 622 and is configured to detect voltage in the first resonator section 601 and control the first FET 608 in response to the detected voltage. This control of the first FET 608 by the control device will now be described in more detail.

At the second node 600B, a zero voltage detector 621 is electrically connected, which is configured to detect a voltage, that is, a voltage equal to or close to 0 V relative to the ground voltage, at a point in the circuit 600 to which the zero voltage detector 621 is connected. The zero voltage detector 621 is configured to output a signal to control the switching of the state of the FET 608. That is, the zero voltage detector 621 is configured to output a signal to the flip-flop 622. The flip-flop 622 is an electrical circuit that can be tuned between two stable states. The flip-flop 622 is electrically coupled to the first gate driver 623, which is configured to supply voltage to the first FET gate terminal 608G depending on the state of the flip-flop. That is, the first gate driver 623 is configured to apply an appropriate voltage to the first gate terminal 608G of the FET to turn the FET 608 ON when the flip-flop is in one state, but is configured not to apply an appropriate voltage to maintain the FET 608 in the on state when the flip-flop 622 is in another state. For example, the first gate driver 623 may be configured to supply an appropriate gate-to-source voltage to the first FET gate 608G to turn on the FET 608 when the flip-flop 622 is in a "1" state, and the first gate driver 623 may be configured to not apply a gate-source voltage when the flip-flop 622 is in the "0" state. Thus, the state of the trigger means 622 controls the on or off of the first FET 608.

In this example, the zero voltage detector 621 and the first controller gate driver 623 are configured to receive respective signals 1011, 1021 from the controller 1001, with which the controller 1001 can initiate and control the operation of the circuit 600, as will be described in more detail below.

In the fourth node 600D, the control voltage line 619 is electrically connected. The control voltage line 619 is electrically connected to the fifth node 600E through a resistor 617a, and the fifth node 600E is electrically connected to the voltage comparator 618, hereinafter referred to as the comparator 618. The fifth node 600E is electrically connected to the positive terminal of the comparator 618. The negative terminal of the comparator 618 is connected to ground 616 In this example, the comparator 618 is configured to output a signal based on a comparison of the voltage at the fifth node 600E with the ground voltage. The output of comparator 618 is sent to flip-flop 622. Control voltage 1031 is applied in this example from controller 1001 to control voltage line 619 through second resistor 617b.

As mentioned above, the comparator 618 is electrically connected to provide an output to the flip-flop 622. The flip-flop 622 is configured such that the output from the comparator 618 can change the state of the flip-flop 622 and thereby cause the first driver 623 to change the state of the first FET 608.

The operation of the exemplary circuit 600 will now be described in more detail in the context of the first resonator section 601 activated by the controller 1001 such that the first inductor 124 is actuated to heat the first pantograph zone 132a.

First, the first FET 608 is in the off state and thus acts as a diode 608a to prevent current from flowing through the inductor 124. Controller 1001 initiates circuit 600 to heat first pantograph zone 132a by causing FET 608 to switch from off to on. In this example, the controller initiates circuit 600 by applying a START signal 1011 to the zero voltage detector 621. Thus, the flip-flop 622 changes state and causes the first gate driver 623 to signal the gate terminal 608G of the FET to thereby switch the FET to an on state.

Once FET 608 is switched to the on state, what may be termed a self-oscillating heating cycle of circuit 600 begins. the first inductor 124 and return to the negative DC voltage terminal 118b through the current sensing resistor 615. The first inductor 124 counteracts this initial increase in current, as is well known, by creating a back EMF in accordance with the laws of Faraday and Lenz. In the on state, the voltage between drain terminal 608D and source terminal 608S is essentially zero.

In FIG. 7A shows a schematic graphical representation of the current flowing through the first inductor 124 as a function of time t since the FET 608 is turned on, at time t ₀ . From the time t ₀ in the inductor 124 begins to increase direct current from zero at a rate that depends on the inductance L1 of the inductor 124 and the resistance of the circuit 600 to direct current. In one example, current-sensing resistor 615 has a resistance of about 2 mΩ, while inductor 124 has a DC resistance of 2 to 15 mΩ, or 4 to 10 mΩ, or in this example, about 5.2 mΩ. This build-up of current in the inductor corresponds to the fact that the inductor 124 stores magnetic energy, and the amount of magnetic energy that the inductor 124 can store depends on its inductance L1, as will be understood.

In FIG. 7B shows a simplified representation of the voltage across current-sensing resistor 615 as a function of time t, again from time t ₀ at which FET 608 was turned on. , generated by the inductor 124, since the inductor counteracts the increase in current. Therefore, at this time, the voltage across the current-sensing resistor 615, as shown in FIG. 7B is small because almost all of the voltage difference provided by the DC source 118 falls across the inductor 124. Then, as the current through the inductor 124 increases and the back EMF of the inductor 124 decreases, the voltage across the current-sensing resistor 615 increases. This is seen as the occurrence of a negative voltage across the current-sensing resistor 615, as shown in FIG. 7B. That is, the voltage across the current-sensing resistor 615 becomes increasingly negative over the time that the FET 608 is in the on state.

Since an increasing negative voltage across current sensing resistor 615 corresponds to an increase in current flowing through inductor 124, the magnitude of voltage across current sensing resistor 615 is indicative of the current flowing through inductor 124. While FET 608 remains on, the current flowing through inductor 124 and the voltage across current-sensing resistor 615 increase substantially linearly to their respective maximum values I _max , V _max (which depend on the DC voltage supplied by DC source 118). current, and the DC resistance of the circuit 600) for a constant time depending on the inductance L1 and the DC resistance of the circuit 600. It should be noted that since the current flowing through the inductor 124 changes after time t ₀ , some inductive heating of the current collector 132 may occur while the DC current flowing through the first inductor 124 builds up.

The circuit 600 is configured such that the control device determines the amount of energy that is stored in the first inductor 124 during the time when the field effect transistor 608 is in the on state, and the controller 1001 can control it. That is, the controller 1001 controls the amount of DC current (and hence the amount of magnetic energy) that can build up in the inductor 124, as will be described below.

As described above, the control voltage 1031 is applied to the control voltage line 619. In this example, the control voltage 1031 is a positive voltage, and the voltage applied to the positive terminal of the comparator 618 (i.e., the voltage at the fifth node 600E) at any time depends on the value of the control voltage 1031 and the voltage at the fourth node 600D. When the negative voltage across the current sensing resistor 615 reaches a certain value, it cancels out the positive control voltage 1031 at the fifth node 600E and produces a voltage of 0 V (ie, ground voltage) at the fifth node 600E. In this example, resistor 617a has a resistance of 2 mΩ. Resistor 617b is an effective resistance for controller 1001 of 70 kΩ. The voltage at the fifth node 600E reaches 0 V when the negative voltage across the current-sensing resistor 615 has the same value as the control voltage 1031.

The comparator 618 is configured to compare the voltage at its positive terminal with the voltage of ground 616 connected to its negative terminal and output a signal as a result. In one example, the comparator is a standard FAN156 component that is available from On-Semiconductor. Accordingly, when the voltage at the fifth node 600E reaches 0 V, the comparator 618 receives a 0 V signal at its positive terminal, and the comparison result obtained by the comparator 618 is that the voltage at the positive terminal is equal to the voltage at the negative terminal. Therefore, comparator 618 outputs a signal to flip-flop 622 and causes FET 608 to turn off. Essentially, turning off FET 608 depends on the voltage detected in circuit 600. Namely, in this example, if comparator 618 detects, by comparing the voltage across its terminals, that the negative voltage across the current-sensing resistor 615 has reached the same value as the control voltage 1031 that occurs at time t ₁ , the FET 608 is turned off. In FIG. 7A, the DC current flowing through the inductor 124 at time t ₁ when the FET 608 is turned off is denoted by I ₁ .

When FET 608 is turned off at time t ₁ , then FET 608 switches from operation as a closed switch to operation as diode 608a in resonator section 601, and to effective operation as an open switch to supply power from DC source 118. At time t ₁ , the DC path through inductor 124 to ground 616 is interrupted by FET 608. This causes a drop in current flowing in first inductor 124 (this is not shown in FIG. 7A) and inductor 124 counteracts this change in current by generating induced voltage. Accordingly, the current begins to oscillate between the inductor 124 and the capacitors 606, 608 at the resonant frequency of the first resonator section 601.

Similarly, the voltage across the inductor 124, and thus between the drain terminal 608D and the source terminal 608S of the first FET, begins to oscillate at the resonant frequency of the first resonator section 601. When the current and voltage across the inductor 124 begin to oscillate, inductive heating of the current collector 132 occurs. Thus, turning the FET 608 off releases the magnetic energy stored in the inductor 124 at time t ₁ to heat the current collector 132.

In FIG. 8 shows a graph 800 of the voltage across the first FET 608 starting with the FET 608 in the on state from time t ₀ to t ₁ . During the time shown in Fig. 8, the first FET 608 is turned off and on twice.

The voltage curve 800 includes a first section 800a between times t ₀ and t ₁ when the first FET 608 is on and a second section 800b to 800d when the first FET 608 is off. At time 800e, the FET 608 is turned on again and the third section 800f begins, which is equivalent to the first section 800a, while the first FET 608 remains on and the above-described process of increasing the DC current flowing through the inductor 124 is repeated. In FIG. 8 also shows a fourth region 800g when the first FET 608 is turned off again to allow the voltage across the FET 608 to fluctuate, and a fifth region 800h when the first FET 608 is subsequently turned on again.

The voltage across the first FET 608 is zero when the first FET 608 is turned on in sections 800a, 800f, and 800h. When the first FET 608 is turned off, as shown by areas 800b-800d as well as area 800g, the first inductor 124 uses the energy stored in its magnetic field (the magnetic field resulting from the rise in DC current when the first FET 608 was turned on) to induce a voltage that resists the drop in current flowing through the first inductor 124 as a result of turning off the first FET 608. The voltage induced in the first inductor 124 causes a corresponding voltage change across the first FET 608. During this voltage change, the first the inductor 124 and the capacitors 606, 610 begin to resonate with each other in accordance with the sine wave. The voltage shown by the voltage line 800 first increases (see eg 800b), as the induced voltage in the first inductor 124 increases to counteract the current drop due to the first FET 608 turning off, reaches a peak (see eg , 800c), and then, as the energy stored in the magnetic field of the first inductor 124 decreases, decreases back to zero (see, for example, 800d).

The varying voltages 800b-800d and 800g produce a corresponding varying current (not shown), and since during the time that the first FET 608 is turned off, the capacitors 606, 610 and the first inductor 124 act as a resonant LC circuit, the total impedance the combination of the first inductor 124 and capacitors 606, 610 is minimal during this time. Therefore, it is understood that the maximum amount of changing current flowing through the first inductor 124 will be relatively large. This relatively large changing current causes a correspondingly large change in the magnetic field in the first inductor 124, which causes the current collector 132 to generate heat. The period of time during which the voltage across the first FET 608 changes as shown by region 800b-800d and region 800g depends in this example on the resonant frequency of the first resonant region 601.

Turning now to FIG. 6 and FIG. 8, the circuit 600 is configured such that when the first FET 608 is turned off and the voltage across the first FET 608 decreases back to 0 V, the zero voltage detector 621 detects this voltage condition and outputs a signal to the flip-flop 622 which causes the first FET 608 to switch back to the on state. That is, in response to this voltage condition detected in the first cavity section 601, the FET 608 switches from an off state to an on state. The zero voltage detector 621 may be considered to detect a voltage indicating that a predetermined portion of the current oscillation cycle between the inductive element and the capacitive element has ended since the FET 608 turned off. resonant frequency of the first resonator section 601 has been completed by detecting by the zero voltage detector 621 that the voltage across the FET 608 has returned to 0 V or nearly 0 V.

In some examples, the zero voltage detector 621 may detect when the voltage across the first FET 608 has returned to 801 or below, and as such may output a signal causing the state of the FET 608 to switch before the voltage across the FET 608 reaches exactly 0 V. As shown in FIG. 8, the operation of the zero voltage detector 621 limits the voltage fluctuations in the resonator section 601 after one half cycle and thus results in a substantially half-sinusoidal voltage profile across the first FET 608. Additional details of the operation of the zero voltage detector 621 will be described below with reference to FIG. nine.

When the first FET 608 is turned on again at point 800e, the DC current induced by the DC source 118 rises again in the first inductor 124. The first inductor 124 can then store energy again in the form of a magnetic field, which will be released when the first FET 608 next turns off to initiate resonance in the first resonator section 601. When the first FET 608 is cycled on and off in this manner, the above process is continuously repeated to heat the current collector 132.

It should be noted that the above-described rise in the current flowing through the inductor 124 described with reference to FIG. 7A and 7B occurs both when FET 608 is initially turned on in response to a RUN signal 1011 from controller 1001, and when FET 608 is subsequently turned on when zero voltage is detected by zero voltage detector 621. In the first case, in response to the RUN signal 1011, the current in the inductor 124 rises substantially linearly from 0. In the second case, when the FET 608 is turned on again in response to detecting zero voltage at point 800e, some excess current circulates in the circuit 600 ( for example, from previous turn-on and turn-off cycles of FET 608). When FET 608 is turned on again after detecting zero voltage, the recirculating current creates an initial negative current through FET 608. Then, while FET 608 remains on, current through FET 608 and inductor 124 rises substantially linearly from the initial negative current value, generated by the recirculating current. As the current through the inductor 124 increases, the voltage across the current-sensing resistor 615 correspondingly becomes more and more negative, as described above.

In the examples, FET 608 may turn on and off at a frequency of about 100 kHz to 2 MHz, or about 500 kHz to 1 MHz, or about 300 kHz. The frequency at which the FET 608 turns on and off depends on the inductance L, the capacitance C, the DC supply voltage supplied by the source 618, and in addition, the extent to which the current continues to recirculate through the resonator section 601, and the effect current collector load 132. For example, if the DC supply voltage is 3.6 V, the inductor 124 is 140 nH, and the capacitance of the resonator section 601 is 100 nF, then the time that the FET 608 remains on can be about 2700 ns, and the time to complete a half-cycle of oscillation when the FET 608 is turned off may be about 675 ns. These values correspond to a power of about 20 watts supplied from the DC voltage source 118 to the resonator section 601. The aforementioned amount of time FET 608 remains on depends on the amount of current that is recirculated in the circuit because, as described above, this recirculating current causes an initial negative current through the inductor when FET 608 is turned on. It should also be noted that the time it takes the current to reach a value that causes the FET 608 to turn off is also at least partly dependent on the resistance of the inductor 124, however, this has a relatively minor effect on time compared to the effect of the inductance of the resonator section 601. The time to complete the half-cycle of oscillation (675 ns in this example) depends on the resonant frequency of the resonator section 601, which is affected not only by the values of the inductance and capacitance of the inductor 124 and capacitors 606, 610, respectively, but also by the effective resistance provided by the loading of the inductor 124 by the current collector 132 .

So far, the circuit 600 has been described in terms of its operation to heat the current collector 132 with a single inductor, the first inductor 124, and thus only the part of the circuit 600 used by the device 100 has been described. However, as described above, the device 100 also includes a second inductor 126 for heating the second zone 132b of current collector 132. In FIG. 9 shows a simplified circuit diagram 600 including a second inductor 126 in addition to the first inductor 124.

As shown in FIG. 9, in addition to the features described with reference to FIG. 6-8, circuit 600 includes a second resonator section 701 including a second inductor 126, a third capacitor 706, a fourth capacitor 710, and a second FET 708 having a drain terminal 708D, a source terminal 708S, and a gate terminal 708G. In addition, the circuit 600 includes a second gate driver 723 configured to supply a gate-source voltage to the second FET gate terminal 708G. Controller 1001 is not shown in FIG. 9, but controller 1001 controls circuit 600 in the manner described with reference to FIG. 6-8 and is further configured to provide a control signal 1012 to the second gate driver 723. Some reference numerals for features of circuit 600 already described with reference to FIG. 6 have been omitted from FIG. 9 for clarity.

As described above, the first inductor 124 is for heating the first zone 132a of the current collector 132, and the second inductor 126 is for heating the second zone 132b of the current collector 132. The second inductor 126, the third and fourth capacitors 706, 710, and the second FET 708 are arranged to forming the second resonator section 701 in the same manner as the first inductor 124, the first and second capacitors 606, 610 and the first FET 608 are arranged to form the first resonator section 601. In one example, the third and fourth capacitors 706, 710 are also C0G capacitors and may have a capacitance of about 100 nF. The second inductor 126 in one example has a DC resistance of about 8 mΩ. In the active state, the second resonator section 701 operates to heat the current collector 132 in the same manner as described above for the first resonator section 601, and this description will not be repeated here.

It should be understood that the amount of DC resistance of the inductors 124, 126 will affect the efficiency of the circuit 600, since higher DC resistance will result in higher resistive losses in the inductor 124, 126 and thus it may be desirable to minimize the resistance of the inductor. direct current, for example, by changing the number of windings or the cross section of the coils 124, 126 inductors. In addition, it should be understood that the AC resistance of the inductor 124 increases compared to the DC resistance due to the skin effect. Thus, the use of winding wire in the examples provides a reduction in the skin effect and thus a reduction in AC resistance and corresponding resistive losses in the inductors 124, 126. As an example, where the first inductor 124 has a DC resistance of about 5 mΩ, and the second inductor 126 has a DC resistance of about 8 mΩ, and the circuit operates at a frequency of about 300 kHz, the particular arrangement of the winding wire forming the coils gives the effective resistance coils 124, 126 of about 1.14 times their DC resistance value.

The node 700A in the second cavity section 701 is equivalent to the first node 600A of the first cavity section 601 and is electrically connected to the first node 600A and thus to the positive terminal 118a of the DC source 118. Node 700C is in an equivalent position in second resonator section 701 as third node 600C of first resonator section 601, and node 700C is likewise connected to ground 616.

It is important to note that the circuit 600 is configured to operate under the control of the controller 1001 in such a way that only one of the resonator sections 601, 701 is active at any given time. Examples of this work will be described in more detail below.

During activation of one of the cavity sections 601, 701, the zero voltage detector 621 is configured to detect zero voltage in the active cavity section 601, 701 and thus control the switching of the respective FETs 608, 708 of the active cavity section 601, 701. The zero voltage detector 621 monitors when the corresponding FET 608, 708 of the active cavity section 601, 701 is turned on again (for example, at point 800e), and an example of this will now be described in more detail with reference to FIG. 8-10.

In circuit 600, the zero voltage detector 621 is configured to detect zero voltage at the second node 600B of the first resonator section 601 or at the equivalent node 700B of the second resonator section 701. When the first resonator section 601 or the second resonator section 701 is active, the zero voltage detector 621 detects that the voltage across that FET 608, 708 returns to zero each time the respective FET 608, 708 is turned off (e.g., point 800e in FIG. 8) or close to zero, such as below level 801. In response to this detection, zero voltage detector 621 outputs a signal to change the state of flip-flop 622. Then, the corresponding operating gate driver 623 outputs a gate-to-source voltage to switch the corresponding FET back to on. condition.

The first small signal diode 725 connects the zero voltage detector 621 to the second node 600B of the first cavity section, and the second small signal diode 726 connects the zero voltage detector 621 to the equivalent node 700B of the second cavity section 701. Specifically, the anodes of the first small signal diode 725 and the second small signal diode are connected to the input of the zero voltage detector 621 through the common node 701B, while the cathodes of the diodes 725, 726 are connected to the nodes 600B, 700B, respectively.

Now, in a specific example, the operation of the zero voltage detector 621 will be described with reference to FIG. 10 showing zero voltage detector 621 and trigger 622. FIG. 10, the components that make up the zero voltage detector 621 are enclosed in a dotted box. Assembly 701B is shown connected to the anodes of the first and second small signal diodes 725, 726. FIG. 13, you can also see the START signal 1011 from the controller 1001 to the zero voltage detector 621.

The zero voltage detector 621 in this example comprises an inverter gate U103 having input 2 from node 701B and output 4 connected to the trigger input 622. Inverter gate U103 is powered from pins 5 and 3, and capacitor C108 isolates pin 5 from ground. Logic power 621a, in this example, 2.5 V is applied to input 5 and through a pull-up resistor R111 to input 2 of the inverter gate U103. Logic power 621a in this example is supplied from controller 1001. Inverter gate U103 is configured to act as an OR gate for START signal 1011 and zero voltage detection signal from node 701B. That is, inverter gate U103 is configured to receive a logic low signal in the form of RUN signal 1011 from controller 1001 to initiate operation of circuit 600a. The START signal 1011 may be provided by the open drain signal contact of the controller 1001. Inverter gate U103 is also configured to receive a logic low signal from node 701B when the first or second signal diodes 725, 726 are forward biased due to one of nodes 600B, 700B have zero or near zero voltage, as will be explained below. If input 2 of the inverter gate is fed one or both of these logic low signals, then inverter gate U103 inverts the received signal and outputs a logic high signal to flip-flop 622.

When the first inductor 124 is operated to heat the current collector 132, the second FET 708 remains off. If the second FET 708 remains off, then the second small signal diode 726 is either unbiased or reverse biased depending on the voltages at the logic power supply, and the DC power 118, i.e., the voltage at the cathode end (closest to node 700B) the second small signal diode 726 is either substantially the same as or higher than the voltage at the anode end (closest to the zero voltage detector 621) of the second small signal diode 726.

During operation of the first resonator section 601, when the first FET 608 is turned off and the voltage across it changes, as shown at 800b-d in FIG. 8, the first small signal diode 725 is reverse biased. At the end of this voltage change, when the voltage reaches 0 V, as indicated by point 800e, or close to 0 V (eg, level 801 or lower), the first small signal diode 725 becomes positively biased. Accordingly, when the first small signal diode 725 is forward biased at point 800e, the signal applied to input 2 of inverter gate U103 becomes a logic low signal because a voltage drop is generated from the logic signal 621a across resistor R111. Thus, as soon as the inverter gate U103 inverts this low level signal, the output 4 of the inverter gate U103 is supplied with a high level signal.

Although in the above description, the operation of the zero voltage detector 621 has been described with respect to controlling the switching of the first FET 608, it will be understood that the zero voltage detector 621 functions the same when using the second small signal diode 726 instead of the first small signal diode 725 to drive the second FET 708.

Still with reference to FIG. 10, flip-flop 622 includes a clock input CLK, a reset/RST input, and a Q output. Flip-flop 622 also contains additional inputs D and VCC for power supply, in this example, the flip-flop receives the same power supply voltage 621a of 2.5V from the controller 1001 as inverter gate U103. The clock input CLK is connected to output 4 of inverter gate U103 and configured to receive a signal from it. When the output 4 of the inverter gate U103 is switched from a low logic level to a high logic level (due to the fact that the input 2 of the inverter gate U103 receives a detected zero voltage condition or a START signal 1011 as described above), the clock input CLK of the flip-flop 622 receives a rising signal. a logic high edge that “clocks” flip-flop 622 and makes flip-flop output Q high. Flip-flop 622 includes an additional input /RST, configured to receive a signal from the output of comparator 618, with which comparator 618 can change the state of flip-flop 621 so that the output of flip-flop Q is high. The Q output of the flip-flop is connected to the first and second gate drivers 623, 723, and upon receiving a high output signal from the flip-flop Q output, the active gate driver 623, 723 (due to receiving the signal 1021, 1022 as described above) supplies the gate driver signal to its corresponding FET 608, 708.

In one particular example, flip-flop 622 may be switched at half the voltage of logic power supply 621a, i.e., 1.25 V. This means that the positive bias voltage of the first small signal diode 725 and the voltage at the drain 608D of the first FET should add up to 1. 25 V to turn on the first FET 608. Therefore, in this example, the first FET 608 turns on when the voltage at its drain 608D is 0.55 V, not exactly 0 V. It should be noted that ideally, for maximum efficiency, the switching should occur at 0V across the FET 608. This zero voltage switching advantageously prevents the first FET 608 from discharging the capacitors 606, 610 and thus losing the energy stored in said capacitors 606, 610.

In FIG. 11 shows the first and second gate drivers 623, 723 and their connection to the gates 608G, 708G of their respective FETs 608, 708 in more detail. , 1022 to activate the heater supplied from the controller 1001. In addition, the signals received by the IN inputs of the gate drivers 623, 723 depend on whether the flip-flop output Q is high. The IN inputs are connected to the Q output of the flip-flop 622 through appropriate resistors R125, R128, each of which has a value of 2 kΩ in this example.

Each gate driver 623, 723 has two additional inputs VDD and XREF, with each VDD input receiving a 6V supply voltage from the controller 1001 and XREF receiving a 2.5V logic voltage which in this example is the same logic voltage supplied by the controller 1001 to flip-flop 622 and inverter gate U103. The VDD inputs of the first and second gate drivers 623, 723 are connected to a 6V supply voltage, and the VDD inputs are connected to ground via two buffer capacitors C120, C121. Each gate driver 623, 723 also has a GND pin connected to ground, with the VDD and GND pins acting to supply power to the gate drivers 623, 723. In this example, each capacitor C120, C121 has a value of 1uF. Gate drivers 623, 723 are configured to output gate drive voltages from respective OUT terminals. The OUT outputs of the gate drivers 623, 723 are connected respectively to the FET gates 608G, 708G via resistors R114, R115, each of which in this example has a resistance of 4.99 ohms.

Each gate driver 623, 723 is configured to receive a signal at its IN input to activate the gate driver only at the time when the trigger output Q is driven high and the heater activation signal 1021, 1022 is received from the controller 1001. At the controller 1001 an "open drain" signal output may be provided that is configured to provide signals 1021, 1022. In one example,

In the examples, turning on the circuit 600 to heat one of the resonator sections 601, 701 is performed by the controller 1001, which first initiates the desired one of the gate drivers 623, 723 with the appropriate heater on signal 1021, 1022. The controller 1001 then sends a START signal 1011 to the zero voltage detector 621 . The duration of the START signal 1011 should be shorter than the half cycle period of the active section 601, 701 of the resonator (this period may be referred to as the "resonant flyback period"). This allows the circuit to properly self-oscillate in response to a detected zero voltage condition. In another example, the order of the RUN signal 1011 and the corresponding heater on signal 1021, 1022 can be reversed so that the RUN signal 1011 is first applied to set the flip-flop output Q to logic high, and then one of the heater on signals 1021, 1022 is applied. to start self-oscillation of the resonator section 601, 701 corresponding to the heater being signaled 1021, 1022.

To continue a more detailed description of the control device for driving circuit 600, FIG. 12 shows a portion of the control device containing a comparator 618 and associated components. In FIG. 12, positive terminal 118a of DC power supply 118 is shown connected to node 1500A, which is connected to nodes 600A, 700A of first and second resonator sections 601, 701, respectively. The negative terminal 118b of the DC power supply is connected to node 1500B, which is equivalent to node 600D shown in FIG. 6. Node 1500B is connected to ground 616 through a current sense resistor 615. Capacitors C111, C112, C115, and C116 are connected in parallel between nodes 1500A and 1500B, each of which in this example has a capacitance of 100 uF, providing buffering between nodes 1500A and 1500B.

In FIG. 12 shows in more detail the components associated with the operation of the comparator 618 for detecting that the current flowing through the active inductor 124 or 126 has reached a predetermined level. As described with reference to the previous figures, the comparator 618 operates to compare the voltage indicating the amount of DC current flowing in the active inductor (124 or 126) with the control voltage 1031 coming from the controller 1001. The comparator 618 receives power through input 6, which connected through a 100 ohm resistor R116 to a 2.5 V logic power signal, in this example supplied by the 1001 controller, and the same logic signal as the 621a signal received by the flip-flop 622. Comparator power input 6 is connected to ground through a 10 nF capacitor C119 . The other end of comparator 618 is connected directly to ground.

In some examples, the controller 1001 is a microprocessor containing a timer (not shown) for generating a signal that generates a control voltage 1031. In this example, the control voltage 1031 is generated using a PWM_DAC signal generated by the controller 1001. The timer of the controller 1001 generates a pulse-width modulated square wave, for example, about 2.5 V, about 20 kHz, and having a specific duty cycle. The PWM_DAC signal is filtered with 10 nF capacitors C127 and C128, and two 49.9 kΩ resistors R121, R123, and a 10 kΩ resistor R124 to provide a substantially constant control voltage 1031 at the frequency at which the controller 1001 controls the control voltage of 1031 (for example, about 64 Hz in the examples). To adjust the control voltage 1031, the controller 1001 in the examples is configured to adjust the duty cycle of the PWM_DAC signal applied to the circuit 600. Thus, the components located between the PWM_DAC input and the positive terminal of the comparator 618 effectively provide the control voltage 1031 with using a PWM signal and adjusting the amount of control voltage 1031 by adjusting the duty cycle of this PWM signal. Thus, the control voltage line 619 shown in FIG. 6 and 9 may be replaced by these components. However, in other examples, the control voltage 1031 may be provided by a substantially constant voltage supplied, for example, by the controller 1001. In such examples, some or all of the components shown in FIG. 12 may be omitted to filter the PWM_DAC signal.

Node 1500B, which is connected to the positive input of comparator 618 as mentioned above, is equivalent to node 600D of circuit 600. In FIG. 12, it can be seen that, as described with reference to the simplified diagram shown in FIG. 6, node 1500B is connected through resistor 617a to the positive input of comparator 618. Essentially, the operation of comparator 618 is the same as described above: receive an input signal at its positive terminal, which depends on the control voltage 1031 and the voltage across the current-sensing resistor 615. When the voltage at When the positive terminal of comparator 618 reaches ground, the /FF RST signal is output through resistor R118 to the /RST input of flip-flop 622 to change the state of flip-flop 622 and thereby turn off the active FET 608/708.

In FIG. 13 shows additional components for a particular example control device for circuit 600. The components shown in FIG. 13, a current measuring device 1300 is defined to provide an I_SENSE signal indicating the amount of current drawn from the DC voltage source 118 during circuit 600 operation. From this signal, the controller 1001 can determine the current drawn from the voltage source 118 and can use it in conjunction with voltage supplied by DC voltage source 118 to determine the value of power supplied to circuit 600. In some examples, as will be described below, the determined power value may be used by controller 1001 to control circuit 600.

The input 1301 to the current measuring device 1300 is made through the resistor R120 shown in FIG. 12. Thus, the input is connected to node 1500B through resistor R120 and receives a voltage indicative of the voltage across the current-sensing resistor 615. Current measurement device 1300 operates as a low voltage side current measurement device for circuit 600. In this respect, current measurement device 1300 contains an op-amp U110 powered by 3.8V applied to input 5 of an op-amp U110 (component type TS507) configured to sense current on the low side using a current-sensing resistor 615, as will be understood. Transistor U109 with built-in bias resistor (part type RN4986) switches the 2.5V voltage supplied by the 1001 controller to a 3.8V supply for the op amp U110. The power line from transistor component U109 is connected to ground through a 10nF capacitor C132. In addition, a 1 kΩ resistor R130 is connected between the positive input of op-amp U110 and ground, and a 412 kΩ resistor R129 is connected between the 2.5 V input from the 1001 controller and the positive input of comparator U110. The negative terminal of op-amp U110 receives a voltage dependent on the voltage across the current-sensing resistor 615. Series resistor R131 and capacitor C133 filter the voltage signal received through input 1301. An additional resistor R133 (in this example with a resistance of 97.6 kΩ) and a 10nF capacitor C134 is connected in parallel between the input to the negative terminal of op-amp U110 and the output of op-amp U110, so that the op-amp operates in closed loop mode.

Operational amplifier U110 operates to output voltage signal I_SENSE to controller 1001 which indicates current through current sense resistor 615 and thus allows controller 1001 to determine the current drawn from DC voltage source 118 through circuit 600.

It should be noted that, taking into account the first and second field effect transistors 608, 708 and the topology of the circuit 600, the phasing of the first and second inductors 124 and 126 with respect to each other can be chosen such that if the first inductor 124 operates, then flow is prevented in the second inductor 126 of a current sufficient to significantly heat the current collector 132, and if the second inductor 126 is operating, then prevent the flow of current in the first inductor 124 sufficient to significantly heat the current collector 132.

As described above, the first 608 and second 708 FETs actually act as diodes 608a, 708a when they are off, and thus can conduct if they are positively biased (i.e., FETs are not ideal switches). Accordingly, in the examples, circuit 600 is configured such that when the first 124 or second 126 inductor is operated to heat the current collector 132, the voltage induced on the other non-operating inductor does not positively bias the intrinsic FET diode associated with that non-operating inductor. , and vice versa shifts it in the opposite direction.

The effect of the control device described above, configured to control the switching devices 608, 708 of the circuit 600 in response to detected voltage conditions, is that when the first resonator section 601 or the second resonator section 701 is active (that is, its gate driver 623, 723 is activated by the controller 1001), then this section of the resonator "auto-oscillates", while the other section remains inactive. That is, the switching of the respective FETs 608, 708 in the resonator section 601, 701 is repeated at a high frequency because the first voltage state (detected by the comparator 618) causes the FET to switch from on to off, and the second voltage state (detected by the zero voltage detector 621) causes the FET to switch from off to on.

The controller 1001 is configured to control the induction heating circuit 600 of the device 100 such that only the first inductor 124 or the second inductor 126 is active at any given time. The controller 1001 is configured to determine at a predetermined frequency which of the coils, the first 124 or the second 126, to activate.

In the examples, during use of the device 100, the controller 1001 determines at a predetermined frequency, that is, once for each of a plurality of predetermined time intervals, which resonator section, the first 601 or the second 701, to activate. In one example, each time controller 1001 determines which resonator section, first 601 or second 701, to activate, controller 1001 may decide to activate that resonator section to heat current collector 132 for the next predetermined time interval. That is, if the predetermined frequency (which may be referred to as the "chopping frequency") is, for example, 64 Hz, then the controller 1001 may determine at predetermined intervals of 1/64 s which resonator section 601, 701 should be activated for the next period, 1/64 s long, until the controller performs the next determination of the resonator section 601, 701 at the end of the next 1/64 s interval. In other examples, the interruption frequency may be, for example, 20 Hz to 80 Hz, or, respectively, the predetermined intervals may be 1/80 to 1/20 second long. To determine which inductor 124, 126 should be activated during a predetermined interval, the controller 1001 determines which pantograph zone 132a, 132b should be heated during this predetermined interval. In the examples, the controller 1001 determines which pantograph zone 132a, 132b is to be heated with reference to the measured temperature of the pantograph zones 132a, 132b, as will be described below.

In FIG. 14 shows a flowchart of an example of a method for determining which of the two resonator sections 601, 701 should be activated for a certain predetermined interval. In this example, the controller 1001 determines which of the resonator sections, the first 601 or the second 701, to activate during a predetermined interval based on the current temperature T1 of the first pantograph zone 132a heated by the first inductor 124 and the current temperature T2 of the second pantograph zone 132b heated the second coil 126 inductor. In one example, the current temperatures T1 and T2 of the first 132a and second 132b pantograph zones can be measured by respective thermocouples (not shown) attached to each pantograph zone 132. The thermocouples provide an input to the controller 1001, allowing the controller 1001 to determine the temperatures T1, T2. In other examples, another suitable means may be used to determine the respective temperatures of the pantograph zones 132a, 132b.

In block 1051, the controller 1001 determines the current temperature value T1 and compares it with the target temperature value target1 for the first zone 132a to be heated by the first inductor 124. The target temperature target1 of the first zone 132a has a value that may change during a session of use of the device activating circuit 600. For example, a temperature profile may be defined for the first zone that defines values for target1 throughout a session of use of the device 100.

In block 1052, the controller 1001 performs the same operation as for the first inductor 124 in block 1051 and determines whether the current temperature value T2 of the second zone 132b is below the target temperature target2 for the second zone 132b at that time or not. Again, the target temperature of the second zone 132b may be determined by a temperature profile defining target2 values during a usage session. The temperature of the second zone 132b can, like the first zone 132a, be measured by any suitable means such as a thermocouple.

If both responses at block 1051 and block 1052 are "no", that is, the temperature of both pantograph zones 132a, 132b is currently at or above their respective target temperatures target1, target2, then controller 1001 determines that neither first pantograph needs to be activated. , nor the second section 601, 701 of the resonator for the next predetermined interval.

If the answer at block 1051 is "no" and the answer at block 1052 is "yes", that is, the temperature of the first zone 132a is at or above its target temperature target1, but the temperature of the second zone 132b is below its target temperature target2, then the controller 1001 determines that the second cavity section 701 should be activated to heat the second zone 132b for the next predetermined interval.

If the answer at block 1051 is "yes" and the answer at block 1052 is "no", i.e. the temperature of the first zone 132a is below its target temperature target1 and the temperature of the second zone 132b is at or above its target temperature target2, then the controller 1001 determines that the first cavity section 601 should be activated to heat the first zone 132a for the next predetermined interval.

If the answer at block 1051 is "yes" and the answer at block 1052 is "yes", that is, the temperature of both the first zone 132a and the second zone 132b is below the respective target temperatures target1, target2, then the controller 1001 proceeds to block 1053. At block 1053, the controller 1001 is effectively operable to alternately turn on the first resonator section 601 and the second resonator section 701 for each predetermined interval during which both zones 132a, 132b remain below their respective target temperatures.

In one example, to alternately activate the first 601 and second 701 resonator sections, at block 1053, the controller 1001 in some examples determines whether an even number of predetermined intervals have passed since the start of the session. If an even number of predetermined intervals have passed since the start of the session, then the controller 1001 determines that the first resonator section 601 should be activated for the next interval. If an odd number of predetermined intervals have passed since the start of the session, then the controller 1001 determines that the second resonator section 701 should be activated for the next interval. It should be understood that in other examples, controller 1001 may instead activate second resonator section 701 if an even number of slots have passed, and first resonator section 601 if an odd number of slots have passed.

In some examples, the circuit 600 is configured such that once one of the resonator sections 601, 701 is activated by receiving a signal 1021 or 1022 on one of the gate drivers 623, 624, that resonator section 601/701 continues to operate, i. oscillate until it is deactivated by the controller 1001, for example, by applying another signal to the gate driver of this resonator section 601/701. Thus, after determining which of the resonator sections 601, 701 to activate during a given interval, the controller 1001 can deactivate one of the resonator sections 601, 701 that was active during the previous interval to initiate this activation.

To illustrate an example block 1053, where the method 1050 shown in FIG. 14 are performed at 1/64 second intervals if the controller 1001 determines that both zones 132a, 132b are below their respective target temperatures target1, target2 and an even number of 1/64 second intervals have elapsed since the start of the device 100 session. , the controller 1001 activates the first resonator section 601 for the next 1/64 second interval while the second resonator section 701 becomes inactive, which in the examples requires the controller 1001 to deactivate the second resonator section 701. If, after this next 1/64 second interval, the temperatures of both zones 132a, 132b remain below their respective target temperatures target1, taget2, then during the next 1/64 second interval, the controller 1001 activates the second resonator section 701 while the first section The resonator 601 becomes inactive, which in the examples requires the controller 1001 to deactivate the second resonator section 701. For each interval in which the temperatures of both zones 132a, 132b remain below their respective target temperatures, this alternation continues between activation of the first 601 and second 701 resonator sections.

In general, method 1050 results in the two inductors 124, 126 never being active at the same time. If it is determined that both inductors 124, 126 require activation to bring the temperature of the respective zones 132a, 132b to a predetermined temperature, then the controller 1001 alternately energizes the inductors 124, 126 at a predetermined frequency to bring the temperatures of both zones 132a, 132b to their respective target temperature. Thus, it can be seen that, for example, during a session period comprising a plurality of intervals during which the temperature of the first zone 132a is substantially below its target temperature and the temperature of the second zone 132b is at or above its target temperature, due to the method 1050 power may be supplied to the first resonator section 601 for almost 100% of this period. However, during a session period including multiple intervals during which both zones 132a, 132b are below target temperatures, each inductor may be energized for approximately 50% of that period.

As mentioned, if both zones 132a, 132b have their set temperatures, then neither of the inductors may be energized. In some examples, during a heating session that lasts for a predetermined period (eg, 3, 3.5, 4, or 4.5 minutes), none of the inductors can be energized more than 75% of the time or more than 80% of the time. In other words, both inductors can be at their desired temperature more than 75% of the time, or more than 80% of the time, so that a constant supply of energy is not required. Instead, power is only applied when the temperatures of zones 132a, 132b fall below their respective target temperatures. In a particular example, the heating session lasts for 260 seconds and the first inductor 124 is energized for a total of about 25 seconds during the heating session, and the second inductor 126 is energized for only about 19 seconds during the heating session. More generally, each inductor can be energized 7% to 15% of the time during a heating session, meaning that the inductors are energized about 14% to 30% of the time.

The power supplied during each 1/64 second interval can be determined as the product of the DC voltage at DC source 118 during that interval and the average DC current drawn from DC source 118 during that interval. In a specific example, when the inductor is operated for a certain interval, the power may be from about 15 watts to about 25 watts, or more preferably from about 20 watts to about 23 watts.

The energy consumed by the inductor during this interval is determined by the product of the power and the duration of the interval. Accordingly, the energy consumed by the inductor during this interval may be from about 0.31 J to about 0.36 J. inductors, and the energy consumed by the inductors can be negligible. For example, the energy consumed by the inductors may be less than about 0.01 J.

The first and second inductors 124, 126 may be referred to collectively as a heater/heater assembly. As mentioned above, assuming that the inductors only fire for about 25% of the time during a heating session, then the total energy consumed by the inductors during the session is between about 1200 J and about 1400 J. Assuming that the inductors are activated only for about 20% of the time during a heating session, the total energy consumed by the inductors is from about 1200 J to about 1400 J. Depending on the characteristics of the heating assembly, the environment and the product inserted into the device, the total energy, consumed by the inductors can be from about 1000 J to about 1400 J.

In one example, the DC source 118 (i.e., the battery) has a battery capacity of about 3000 mAh and a voltage of about 3-4 V, giving a capacity of about 30,000 J to about 35,000 J. In a specific example, the battery capacity is about 33,000 J. This allows DC source 118 to provide power for approximately 25 to 30 heating sessions.

In a specific example, when the device is inactive, the current drawn is about 0.1 mA. Most of this consumed current is the quiescent current consumed by the voltage regulator of the DC source 118 . For example, the regulator is a 2.5 volt regulator. Some of this current may be caused by small leakage currents on any additional integrated circuits present in the device, such as the integrated circuit associated with charging socket 114. Assuming the battery has a voltage of about 3 .7 V, the power consumed by the device in the inactive state can be about 0.370 mW. Preferably, the inactive device 100 consumes less than about 1 mW, or less than about 0.5 mW, or less than about 0.4 mW.

If the controller 1001 is active, then the current consumption can be about 3 mA. Thus, the controller 1001 consumes about 0.01 watts when in use. In one example, controller 1001 has a clock frequency of 16 MHz. Preferably, the controller 1001 consumes less than about 0.05 watts, or less than about 0.03 watts, or less than about 0.01 watts.

The device 100 may also include a visual indicator, such as a number of LEDs. As briefly mentioned above, in FIG. 2 shows the second circuit board 138. The second circuit board 138 has 4 LEDs. LEDs can light up to indicate certain events to the user, such as when the device is ready for use. Each LED can draw about 2 mA at maximum brightness or intensity. When all 4 LEDs are at maximum brightness, they consume a total power of about 0.03W. Preferably, the visual indicator consumes less than about 0.05 watts, or less than about 0.04 watts, or less than about 0.03 watts. The LEDs may be lit, for example, during a heating session.

Accordingly, during a heating session, the heating assembly may consume from about 15 watts to about 25 watts. Other electrical components within the device (i.e., all components except coils 124, 126) can consume a total power of about 0.04 W (about 0.03 W for LEDs and about 0.01 W for voltage regulator, controller 1001 and others). leakage currents). Preferably, the power consumed by the electrical components is less than about 0.1 W, or less than about 0.05 W.

In the examples above, it is understood that the power will vary with the voltage of battery 118. In the above examples, it is assumed that the voltage of battery 118 is about 3.7V.

In the example, the power drawn by the electrical components is less than about 0.5% of the power drawn by the heating assembly (ie, the inductors). Preferably, the power consumed by the electrical components is less than about 0.2% of the power consumption of the heating unit. For example, the power consumed by the heating assembly (ie, the inductors) may be about 20 watts, while the power consumed by other electrical components is about 0.04 watts. Accordingly, the power consumed by the electrical components is approximately 0.2% of the energy consumption of the heating unit.

In the examples, the controller 1001 is also configured at predetermined intervals, which in the examples coincide with the predetermined intervals at which the method 1050 is performed to determine the power supplied to one of the resonator sections 601, 701 from the DC source 118.

As described above with reference to FIG. 9-11, in particular, to control which of the resonator sections, the first 601 or the second 701, is active at any given time, the controller 1001, also transmitting the START signal 1001 to initiate the operation of the circuit 600, is configured to selectively transmit the first signal 1011 a heater operation signal to the first gate driver 623 to activate the first resonator section 601, or a second heater operation signal 1012 to the second gate driver 723 to activate the second resonator section 701.

For example, if controller 1001 initiates operation of circuit 600 and controller 1001 transmits a first heater operation signal 1011, then circuit 600 operates as described above to activate first inductor 124 to heat first pantograph zone 132a. If the controller 1001 transmits a second heater operation signal 1012, then the circuit 600 activates the second inductor 126 to heat the second pantograph zone 132b. If the controller 1001 transmits neither the first heater operation signal 1011 nor the second heater operation signal 1012, neither of the inductors 124, 126 is activated and the current collector 132 is not heated.

The controller 1001 is configured to control the power supplied from the DC voltage source 118 to the circuit 600 to inductively heat the current collector 132 based on a comparison of the measurement of the power supplied to the circuit 600 and the target power. The controller 1001 is configured to control the power supplied to the circuit 600 by controlling the switching device of the circuit 600, that is, by controlling the switching of the FETs 608, 708. The controller 1001 can control the switching of the FETs 608, 708 by setting the control voltage 1031, which determines a constant the current that can build up in the inductor 124, 126 corresponding to that FET 608, 708 before turning the FET 608, 708 off.

In FIG. 15 shows an example of a method 1100 performed by the controller 1001 to control the power supplied to the circuit 600. In step 1101, the controller 1001 determines the power P supplied from the DC source 118 to the circuit 600. For example, the controller 1001 may determine the average value of the power supplied to the circuit 600 during the previous predetermined interval. In the examples, the power P supplied to circuit 600 during an interval can be determined by measuring the voltage and DC current passing through a given one of the resonator sections 601, 701. The controller 1001 may then determine the product of the voltage and the DC current flowing through a given one of the resonator sections 601, 701 to determine the power P supplied to that resonator section 601, 701.

In the examples, the determined power P is the average power supplied from DC source 118 over a predetermined interval, which can be determined by performing the product of the average DC voltage at DC source 118 and the average DC current drawn from DC source 118 over the previous interval.

In the example device 100, the DC source 118 is a battery that is connected to the controller 1001, and the controller 1001 then outputs the voltage of the DC source 118 to the circuit 600. The controller 1001 is configured to determine the DC voltage supplied by the battery 118. The current drawn from battery 118 is determined by the operation of the device 1300 current measurement. Controller 1001 detects DC voltage and DC current once for each 1/64 second interval. The DC voltage can be considered almost constant during this short period of time. However, the current changes at a rate dependent on the speed at which the circuit is turned on and off. As described above, in some examples this frequency is about 300 kHz. The current measurement device 1300, as described above with reference to FIG. 13 outputs an I_SENSE signal which is filtered to remove this signal at about 300 kHz. Thus, the average DC current for the 1/64 s interval is obtained by measuring this filtered I_SENSE signal, and the I_SENSE measurement is made just before the end of the 1/64 s interval to allow the signal from the filter to settle. Thus, the controller 1001 obtains the DC voltage and DC current measurements for a 1/64 second interval and can calculate the product of these values to obtain a certain power P. This determined power P can be considered as the average value of the power supplied by the DC source 118 during the interval duration 1/64 s.

In block 1102, the delivered power P determined in block 1101 is compared with the target power. If the determined power P is the average power over a predetermined interval, then the target power is the target average power over the same interval. In one example, the target power is a target for the average power delivered over a predetermined interval and may be 10 to 25 watts, or 15 to 23 watts, or about 20 watts. In this example, the target power is in the range, for example, 20-21 watts or 15-25 watts. The controller 1001, respectively, in block 1102 may compare the value of the applied power P determined in block 1101 with the target range and determine whether the applied power is under the range, within the target range, or above the target range. For example, if the target range is 20-21 W, then at block 1102, the controller 1001 determines whether P < 20 W, or 20 W ≤ P ≤ 21 W, or P > 21 W.

Based on comparing the supplied power P with the target range, the controller 1001 determines whether and how to adjust the power for the next predetermined interval in order to bring the actual power supplied to the active inductor 124 or 126 during the next predetermined interval to the target power range. . That is, if the supplied power P is below the target range, then the controller 1001 decides to increase the power supplied to the circuit 600 during the next predetermined interval. If the supplied power P is above the target range, then the controller 1001 decides to reduce the power supplied to the circuit 600 during the next predetermined interval. If the supplied power P is below the target range, then the controller 1001 decides not to adjust the power supplied to circuit 600 for the next predetermined interval.

Due to the configuration of circuit 600 described above, the supplied power P for a given predetermined interval depends on the value of the control voltage 1031 for that interval. Taking the example of a single 1/64 s interval for which the first resonator section 601 is active, this 1/64 s interval contains a plurality of repeating cycles containing portions 800a to 800e of the voltage curve 800 and their repetition. For each cycle, during a period of time from t ₁ to t ₀ , resonator section 601 is allowed to resonate, and since FET 608 is turned off during this period, no power is drawn from DC source 118 through first resonator section 601. Thus, substantially all of the power drawn from DC source 118 during a predetermined 1/64 second interval to power resonator section 601 is consumed during the period between t ₀ and t ₁ while inductor 124 is "supplied" with current , that is, while the FET 608 is on. The time between t ₁ and t ₀ is determined by the resonant frequency of the first section 601 of the resonator. This resonant frequency may remain substantially constant for at least a predetermined 1/64 second interval (although it may vary over the duration of circuit 600 due to coil and pantograph temperature and battery voltage). The duration of time from t ₀ to t ₁ is determined by the value of the control voltage 1031, as well as the DC voltage supplied by the DC source 118, as well as the resistance and inductance of the first cavity section 601 (the same applies to the second cavity section 701). That is, for a given DC supply voltage, the control voltage 1031 sets the current I ₁ , which can build up in the inductor 124 between times t ₀ and t ₁ , but if the DC supply voltage decreases, then the time required for the rise of the set value I ₁ , increases. Thus, the average power supplied during the 1/64 s interval depends on the value of the control voltage 1031.

Therefore, in the examples, in order to control the power supplied to the circuit 600 during the next interval, the controller 1001 sets the value of the control voltage 1031 for the next interval. In the examples, for a given DC source 118, for a predetermined interval during which one of the resonator sections 601, 701 is active, a greater positive value of the control voltage 1031 results in a greater value of power P supplied to the circuit 600. Therefore, in such examples, if the controller 1001 determines that the applied power P in the last interval was above the target range, then the controller 1001 reduces the control voltage 1031 for the next interval. If the controller 1001 determines that the applied power P for the last interval was below the target range, then the controller 1001 increases the control voltage 1031 for the next interval. And if the controller 1001 determines that the applied power P in the last interval was above the target range, then the controller 1001 leaves the control voltage 1031 unchanged for the next interval.

It should be noted that in one example of the above method 1100, the supplied power P determined in block 1101 is the power supplied to a particular one of the resonator sections 601, 701. For example, the power P can be determined by measuring the voltage across the first resonator section 601 and the DC current flowing through the first resonator section 601. In such an example, the power P supplied to the first resonator section 601 is used to control the control voltage 1031. It should also be noted that for a given control voltage 1031, in some examples, the power supplied to each of the inductors 124, 126 when the respective sections 601, 701 resonators are active, may be different. This may be, for example, because the inductors 124, 126 have different values of inductance or DC resistance, or the capacitances of the two resonator sections 601, 701 are not equal. Therefore, in this example, the target power outside the target range can be supplied to the second resonator section 701 within a predetermined interval, but since the control voltage 1031 is adjusted based on the power P supplied to the first resonator section 601, in this example, the controller 1001 may not adjust the control voltage 1031.

For example, for a given control voltage value 1031, the controller 1001 may determine at block 1101 that an average power of 20 watts has been supplied to the first resonator section 601 for a given interval, with a target voltage of 20-21 watts in this example. At step 1102, the controller 1001 determines that the applied voltage was within the target range, and accordingly, the controller 1001 decides not to adjust the control voltage 1031. Considering that for the next predetermined interval, the controller 1001 determines (using the method example 1050) that it should be activated the second section 701 of the resonator, and not the first section 601 of the resonator. For a given value of the control voltage 1031, 22.5 watts is supplied in this example due to differences in the electrical properties of the first 601 and second 701 resonator sections. However, in this example, at block 1102, the controller 1001 compares the last measured value of the power P supplied to the first resonator section 601, and therefore, at block 1103, decides not to adjust the control voltage 1031. As such, in the example method 1100, the actual power supplied to the circuit 600 may be outside the target range. However, this may allow control of the power supplied to the inductors 124, 126 by measuring only the power P supplied to one of the resonator sections 601, 701. This can provide a simple and useful solution for keeping the power supplied to the circuit 600 within an acceptable range if, for example, the resonator sections 601, 701 and their components have approximately the same electrical properties.

As mentioned above, in some examples, the DC source 118 is a battery with a voltage of about 2 to 10 V, or 3 to 5 V, or, in one example, about 4.2 V. In some examples, the DC voltage generated DC source 118 may vary, such as decreasing over the duration of circuit 600. For example, if DC voltage source 118 is a battery, then the battery may initially supply 4.2 V, but the voltage supplied by the battery may decrease as the battery discharges. . Therefore, after a predetermined period, the DC voltage source 118 can supply, for example, 3.5 V instead of the initial 4.2 V.

As described above, at a given supply voltage, the value of the control voltage 1031 controls the amount of current that can build up in the active inductor 124/126 before turning off the corresponding FET 608/708. Power is supplied from a DC voltage source 118 to "energize" the active inductor 124/126, allowing the DC current to build up when the FET 608, 708 is turned on. As also described above, the time t ₁ for the current to rise to a value that causes the FET 608/708 to switch depends on the DC voltage source. Therefore, for example, if the voltage supplied by the DC source 118 decreases, the rate at which the current in the inductor 124 rises decreases, resulting in a decrease in the power P supplied to the circuit 600.

An example method 1100 can maintain the target power even when the voltage supplied from the DC source 118 changes. That is, since the actual supplied power P is determined and used to control the control voltage 1031, the controller 1001 can act to maintain the set power by adjusting the control voltage 1031. For example, if the battery level is depleted, then the controller 1001 measures that the power P, supplied to the circuit 600 at a given control voltage 1031 has decreased, and acts to increase the power P supplied to the circuit by increasing the control voltage 1031. As such, the target power level can be maintained while the battery used to power the circuit 600 is discharged . This is advantageous because maintaining the desired power level can provide optimal performance for the induction heating circuit 600. For example, maintaining a substantially constant applied power allows for constant heating of the aerosol material 110a regardless of the supply voltage. Similarly, exemplary method 1100 provides for substantially constant power regardless of other changing factors in the circuit that may affect the amount of power delivered, such as varying load on circuit 600 by pantograph 132 as the temperature of pantograph 132 increases. This provides a consistently good consumer experience, for example by providing a constant time to first puff, that is, a constant time between activation of the device 100 and its readiness to dispense an inhalation aerosol by the user.

In another example, the measured power value P, on which the control voltage 1031 is based, changes during a usage session. For example, during a particular usage session, for the first portion of the usage session (eg, for the first ~60 seconds of usage session), the temperature profile may be such that the first inductor 124 is mostly active and the second inductor 126 is inactive. For this first part of the session, it may be appropriate to base the control of the control voltage 1031 on measurements of the power supplied to the first resonator section 601. However, later in the session, again, for example, due to the temperature profile for the session, it may be that the second inductor 126 is mainly active, while the first inductor 124 is active for a shorter time. Thus, for the second part of the session (eg, after ~60 s) it may be useful to control the control voltage 1031 based on measurements of the power supplied to the second resonator section 701. The controller 1001 may accordingly switch from controlling the control voltage 1031 based on measurements of the power supplied to the first cavity section 601 to controlling the control voltage 1031 based on the measurements of the power supplied to the second cavity section 701. In this way, it is possible to more closely adhere to the target power throughout the entire session, since, for example, the control voltage 1031 is set based on a comparison of the actual power supplied to the active inductor 124, 126 with the target power range.

In some instances, where controller 1001 determines at block 1103 that power needs to be adjusted, controller 1001 may adjust control voltage 1031 in predetermined steps. For example, the controller 1001 may be configured to adjust the control voltage 1031 by a predetermined amount over a predetermined amount of time. If, at block 1102, controller 1001 determines that the supplied power P is below the target power range, then controller 1001 may increase control voltage 1031 by a predetermined number of volts for the next predetermined interval. Conversely, if at block 1102 the controller 1001 determines that the supplied power is above the target power range, then the controller 1001 may increase the control voltage 1031 by a predetermined amount for the next predetermined interval.

In the example described above with reference in particular to FIG. 12, the control voltage 1031 is generated by a pulse modulated PWM_DAC signal. The PWM_DAC signal as described above is a square wave at 2.5V. The duty cycle of the PWM_DAC signal is controlled by the controller 1001, which sets a value from 0 to 800 for the PWM_DAC duty cycle, this value corresponds to a duty cycle of 0% at 0 and 100% at 800 The PWM_DAC signal, after filtering, provides a substantially constant control voltage 1031, and therefore the duty cycle settings of 0 to 800 of the PWM_DAC signal ensure that the control voltage 1031 has a value of 0 to 2.5 V. In this example, the controller 1031 can adjust the duty cycle setting PWM_DAC signal by a set value, such as 8 out of 800, or leave the setting unchanged for each predetermined interval. In another example, the controller 1001 may provide adjustment of the control voltage 1031 by some other means, and if the controller 1001 determines that the control voltage 1031 should be adjusted, the controller 1001 may adjust the control voltage 1031, for example, to a level of 1%, or 2%, or 5 % of the maximum value of the control voltage 1031 for the next preset interval.

In some instances, when controller 1001 initiates circuit 600, such as to start a session of device 100 containing circuit 600, control voltage 1031 is set to a predetermined initial value. In one example, a control voltage value 1031 (eg, setting the duty cycle of the PWM_DAC signal that produces this control voltage value 1031) that corresponds to the target power level is determined during circuit 600 tuning. That is, the power supplied to circuit 600 can be determined (for example, measured or determined theoretically) for a number of values of the control voltage 1031, for example, to build a calibration curve. The control voltage value 1031 corresponding to the target power can then be determined. In one example, DC source 118 may supply 4.2 volts, and to achieve a target power of 20 watts, controller 1001 may determine during calibration a PWM_DAC signal duty cycle setting value of approximately 344 out of 800.

In one example, the controller 1001 is configured to set the control voltage 1031 to an initial value that is based on that determined value of the control voltage 1031. power. For example, if the duty cycle setting for the control voltage 1031, which is set to match the target power, is 344 out of 800, then the controller 1001 may start the session with the setting set to 152 out of 800 and increase the setting by a predetermined amount with each predetermined set interval until the measured power P is within the target range. This can result in the applied power being well below the target power at the beginning of a usage session, and then the applied power may be ramped up (by increasing the control voltage 1031 by the controller 1001) until it reaches the target power range. This initial increase in power input can provide increased safety for circuit 600 by preventing the pantograph from overheating at the start of a session and allowing circuit 600 to respond to the actual power input as determined by controller 1001.

In one example, the predetermined interval is the same predetermined interval that the controller 1001 uses in the method 1050 to determine which of the first 124 and second 126 inductors to activate. In one such example, as mentioned above, the predetermined intervals are 1/64 s long. The duration of the predetermined interval (or, equivalently, the frequency of interruptions) can be chosen to provide a favorable time interval during which the controller can monitor the circuit and adjust the parameters accordingly. For example, a chopping frequency of 64 Hz, or in the range of approximately 10-100 Hz, may be used. With these examples of interruption frequencies, the controller 1001 can measure the temperature rise of the pantograph zones at a fast enough rate that it can decide to stop heating a particular inductor 124, 126 before the temperature of the pantograph 132 zone 132a, 132b can also rise well above the target temperature. . Likewise, the examples given for the interrupt frequency may provide a preferred frequency at which the control voltage 1031 can be adjusted to provide adequate control of the power supplied to the inductors 124, 126 within a safe target range.

In an exemplary way of operating circuit 600, the target power to be used by controller 1001 in controlling the power supplied to circuit 600 is predetermined based on the characteristics of the scheduled usage session. For example, the target power range may be adjusted during a session of use.

In FIG. 16 shows a schematic example of a target temperature profile target1 for a portion of a usage session, which in this example is the target temperature for a single pantograph zone 132a. In this example, initially in the first part 1201 of the session, the temperature of the first zone 132a is well below its target temperature target1. In this first part 1201, the circuit 600 operates to bring the temperature of the first zone 132a to the target temperature target1. In such an example of a portion of a usage session, the target power P1 may have a range of values, eg 20-21 watts. The target power during the first part 1201 of the session may be relatively high in order to rapidly bring the temperature of the current collector 132, and hence the aerosolized material 110a, to a temperature suitable for generating an aerosol for inhalation by the user.

As the usage session continues, the first zone 132a substantially reaches its target temperature target1. The second part 1202 of the session may be set to begin shortly after the first zone 132a reaches its target temperature target1. For example, for this portion of the session 1202, the first zone 132a may substantially have a target temperature target1, such as 250°C, and may be held at the target temperature target1 in accordance with method 1050. Similarly, although not shown in FIG. 16, the temperature of the second zone 132b may be maintained equal to its own target temperature setpoint target2 by method 1050 (and the temperature target target2 of the second zone 132b may define a different temperature profile than that determined by target1).

The part 1202 in the session after the first zone 132a has substantially reached the set temperature target1 can be characterized in that the controller 1001 operates to maintain the temperature of the first zone 132a (or both zones 132a, 132b) and not to bring the temperature of the first zone 132a to the target value target1, as in the first part 1201. Thus, during the part 1202 of the session, relatively little power may be required to be supplied to the current collector zone 132a to maintain the target temperature target1, compared to the power required to bring the temperature of the zone 132a of the pantograph to the target temperature target1. In the second part 1202 of the session, it may be beneficial to reduce the value of the target power P1 compared to its value in part 1201. In one example, the target power level P1 can be reduced from 20-21 W in part 1201 to about 15 W in part 1202 of the session . Reducing the target power P1 in this manner may be advantageous in some instances because by using a lower power level, the energy losses in the circuit can be reduced and thus efficiency can be increased.

For the third part 1203 of the session, the value of the target temperature target1 is 0, that is, the first coil 124 is not activated. At this point, the target power P1 may also be reduced to 0 if the session has ended, or if the second inductor 126 is still active, then the target power P1 may remain at a non-zero value as long as the second inductor 126 is active. Accordingly, the target power may take into account the temperature profile of both zones 132a, 132b at any point in the session. If, for example, part of a session of use requires a significant increase in temperature in one of the zones, then a relatively high target power may be required. Conversely, for portions of the session where significant heating is not required for either zone 132a, 132b, a relatively low target power may be used.

As mentioned above, using lower power levels during certain periods of a usage session can provide the advantage that energy savings can be achieved over the duration of the session. For example, if the target power level is reduced from 20-21 watts in the first period to about 15 watts in the second period, then in some examples, about 5-10% energy savings can be achieved by reducing the energy loss in circuit 600 when operating at lower power. . In one example, during a typical session of about 260 seconds, maintaining the target power at about 20 watts for the duration of the session can result in an energy consumption of about 1000 joules. However, reducing the target power to about 15 watts after the first zone 132a first reaches set temperature, and maintaining the target power level at 15 W for the remainder of a session of substantially the same duration, could result in power consumption between 900 and 950 J. In the examples, almost all of the energy used by the device is due to the energy supplied to heating of the pantograph 132. The power consumption of electrical components other than the heating circuit, such as LED indicators and microcontroller, can be less than 0.1 W, and in some examples less than 0.01 W.

Certain methods described herein may be implemented by computer program code that may be stored on a non-volatile storage medium. For example, in some examples, controller 1001 may include a non-volatile computer-readable storage medium containing a set of computer-readable instructions stored thereon, and a processor for controller 1001 to execute the method described herein. Controller 1001 may include one or more processors. For example, in some examples, as described above, the controller 1001 is a programmable microprocessor. The controller 1001 may include a storage medium containing a set of machine-readable instructions, such as computer code, which, when executed by the controller 1001, causes the method described herein to be executed.

It should be noted that although a circuit containing two inductors has been described above, the aspects described above, for example, for controlling the power supplied to an induction heating circuit, can be applied to circuits with a different number of coils, for example, with one or more with two coils. In addition, although this document describes an inductive circuit containing inductors, the aspects described in this document can be equally applied to an inductive circuit using other types of inductive elements having inductance and suitable for creating an alternating magnetic field for heating current-carrying structure. In addition, although the circuits above have been described with respect to induction heating, features relating to the power consumption of electrical components and heating components are equally applicable to embodiments using resistive heating.

The above embodiments are to be understood as illustrative examples of the invention. Other embodiments of the invention are contemplated. It should be understood that any feature described with respect to any one embodiment may be used alone or in combination with other features described, and may also be used in combination with one or more features of any other embodiment, or any combination of any other embodiments. . In addition, equivalents and modifications not described above may also be used without departing from the scope of the invention as defined in the appended claims.

Claims (19)

1. An aerosol delivery device, comprising: several electrical components, including a controller containing a processor and a storage medium containing a set of machine-readable instructions; these several electrical components are located in close proximity to the outer surface of the specified device; a heating unit comprising an insulating element, an inductor, and a heating component for heating, but not burning, the aerosol-forming material by electromagnetic induction; and a battery for supplying power to said electrical components and a heating unit under the control of said controller; moreover, during the heating session, the energy consumption of the heating unit is from 15 to 25 W, and the energy consumption of said electrical components is less than 0.1 W, so that the temperature of the outer surface of said device remains below 4°C. 2. The aerosol delivery device according to claim 1, wherein the power consumption of said electrical components is less than 50 mW. 3. The aerosol delivery device according to claim 2, wherein the power consumption of said electrical components is less than 40 mW. 4. An aerosol delivery device according to any one of the preceding claims, wherein when the controller is active, the power consumption of the controller is between 10 mW and 20 mW, and when the controller is inactive, the power consumption of said electrical components is less than 0.5 mW. 5. An aerosol dispenser according to any one of the preceding claims, wherein said electrical components comprise a plurality of LEDs and each LED has a power consumption of less than 10 mW at maximum intensity. 6. An aerosol delivery device according to any one of the preceding claims, wherein said electrical components also comprise a WiFi interface and/or a Bluetooth interface and/or an NFC interface. 7. An aerosol supply device according to any one of the preceding claims, wherein, when using the device, the power consumption of the heating unit is between 15W and 25W. 8. An aerosol delivery device according to any one of the preceding claims, wherein the battery capacity is from 30,000 J to 35,000 J. 9. An aerosol delivery device according to any one of the preceding claims, wherein the heating unit is configured to operate for a period of 3 minutes to 5 minutes and configured to consume 1000 J to 1400 J during said period. 10. An aerosol delivery device according to any one of the preceding claims, wherein the heating unit also comprises at least one coil configured to heat said heating component. 11. The aerosol delivery device of claim 10, wherein the heating assembly comprises a first coil and a second coil configured to heat said heating component. 12. An aerosol delivery device according to claim 10 or 11, wherein the heating component comprises at least one electrically conductive heating component included in an article for insertion into a heating zone of said device, said article also containing an aerosol-forming material and being removable from the heating zone after use. 13. An aerosol delivery system, comprising: an aerosol supply device according to any one of paragraphs. 1-11; and an article containing an aerosol-forming material.

Images