RU2777046C1 - Class g amplifier - Google Patents

Abstract

FIELD: amplifying and generator technology.

SUBSTANCE: invention relates to the field of amplifying and generator technology and can be used in acoustic and hydroacoustic broadband power amplifiers. The class G amplifier additionally contains a current sensor and an interface circuit. The comparison circuit is made four-channel. The controlled power supply is additionally provided with the first, second, third and fourth control inputs connected through the interface circuit with the first, second, third and fourth outputs of the four-channel comparison circuit. The output of the class AB half-bridge amplifier is connected via a current sensor to the load input. The output of the current sensor is connected to the input of the feedback circuit. At the output voltage U>+E_v-ΔE, +E_v-ΔE>U>-E_v-ΔE and U<-E_v-ΔE, the power supply voltage of the forward conduction transistor is set at +E_o, +E_v, -E_v, and the power supply voltage of the reverse conductivity transistor is set at the level of -E_o, -E_v, +E_v at the output voltage U<-E_v+ΔE, - E_v+ΔE<U<+E_v-ΔE and U>+E_v-ΔE. The value of ΔE is (0.005…0.2) E_v.

EFFECT: increasing the energy efficiency of a class G linear power amplifier while ensuring its use as part of hydroacoustic transmitting equipment.

1 cl, 5 dwg

Description

The invention relates to the field of amplifying and generator technology and can be used in acoustic and hydroacoustic broadband power amplifiers.

Known technical solutions for amplifying analog signals [Kibakin V.M. Fundamentals of the theory and calculation of transistor low-frequency amplifiers M .: Radio and communication 1988, 240 s], based on the methods of linear amplification by transistor end stages of class AB.

Along with well-known advantages, such as low non-linear distortion (less than 0.2%) and a wide dynamic range (more than 60 dB), such amplifiers are characterized by low energy efficiency. When operating on a matched active load, the efficiency of amplifiers of this class does not exceed 75% with a relative energy loss of at least 30%. Moreover, the energy losses related to the total output power of the class AB amplifier increase with a decrease in the load active power factor cos ϕ _n (where ϕ _n is the phase shift between the current and voltage of the amplifier output signal). For small values of cos φ _n = 0.1-0.3, the energy loss can be commensurate with the rated output power of the amplifier. Heat dissipation power of a linear amplifier p _T in the most efficient mode of class B when operating on a complex load (Article Alexandrov V.A., Kazakov Yu.V., Markova L.V., Chuprov O.A. Amplifying devices for the study of hydroacoustic emitters. Sensors system No. 1/ 2019 (67) pp. 50-55) is determined by the expression:

where cos φ _n \u003d Z _n / R _n - with a parallel load equivalent circuit;

P _{Z max} \u003d (U _{n max} ) ² / Z _n

where P _{Z max} is the maximum output power of the amplifier at the maximum output voltage for the relative signal amplitude m=1,

U _{n max} - the maximum output voltage of the amplifier,

Z _n - load impedance.

The above ratio confirms the significant relative level of energy losses in a linear power amplifier, especially when operating on a complex load. The identified circumstances prevent the introduction of high-quality class AB amplifiers into hydrocommunication transmission equipment, which provides excitation of hydroacoustic transducers (HAP) with complex broadband signals. Also, the transducers are made mainly on piezoactive materials with a pronounced capacitive component of conductivity with an active power factor of not more than 0.1 ... 0.5 in the operating frequency band of 1 ... 2 octaves.

From an energy point of view, in hydroacoustic broadband power amplifiers, it is preferable to use the key amplification method based on pulse-width modulation, implemented in class D amplifiers (Artym A.D. Class D amplifiers and key generators of radio communications and broadcasting. M .: Svyaz, 1980, p. 207). Low-frequency amplifiers (ULF), which implement the key amplification method, are based on pulsed energy transfer through the filter inductor to the load and reverse regeneration from the load capacitance to the power supply capacitance. In this case, losses in semiconductor devices and filter elements are very small and, as a rule, do not exceed 10% of the total output power. At the same time, the non-linearity of the pulse conversion and the non-ideality of transistors and diodes in the key mode cause significant non-linear distortion (more than 1-3%) and dynamic range limitation (less than 40 dB). As a result, the identified factors associated with the deterioration of the quality indicators of the output signals limit the possibility of generating complex signals for excitation of broadband HAPs, which does not meet modern requirements for transmitting digital messages over a hydroacoustic channel. Accordingly, the new tasks of providing hydrocommunication modes force the developers of hydroacoustic radiating paths to turn to linear amplification methods with elements of key control of the power supply voltage of linear amplifiers to reduce energy losses.

This method of linear amplification by analog-discrete power amplifiers (N.B. Dogadin Analog-discrete amplifiers / Volgograd - St. Petersburg: Change, 2003. - 216 s) is based on a step change in voltage for each of the half-waves of the amplified signal. Amplifying devices with a discrete change in the power supply voltage according to the accepted classification belong to class G amplifiers. The most effective is the well-known class G amplifier in a two-stage version, described in US patent No. 8072266 (US No. 8072266 class G amplifier with improved supply rail transition control. Date of patent Dec. 6, 2011). This class G amplifier is based on a class AB half-bridge amplifier containing forward and reverse conduction transistors by discretely changing the positive and negative polarity power supply voltage from low to high with a corresponding change in control signals. The introduction of power supply voltage steps of a class AB high-frequency amplifier covered by deep feedback makes it possible to reduce energy losses in the amplifier without deteriorating the output signal quality, which favorably distinguishes the technical solution according to US patent No. 8072266 from known technical analogues and is the closest in terms of the number of common features analogue of the present invention.

Class G amplifier - the prototype is based on the use of a class AB half-bridge transistor amplifier covered by output voltage feedback, with a discrete change in the power supply voltage of positive and negative polarity from a low level ±E _n to a high level ±E _in when the excitation signals of the transistors are direct, or reverse conductivity of the set level. Its feature is the simultaneous change in the positive and negative voltage of the power supply, which achieves power supply at a reduced or increased voltage at the corresponding output voltage of the amplifier. The introduction of a reduced power supply stage at a low output voltage level makes it possible to reduce the residual voltage on amplifying devices and significantly reduce (by half when operating on a resistive load) energy losses in the rated power mode at maximum load voltage.

Class G amplifier - prototype (US patent No. 8072266), (Fig. 1) contains a controlled power source 1 (UIP 1), an analog signal driver 2, a half-bridge circuit 3 of a class AB amplifier, a comparison circuit 4 with a logic element "OR", a circuit 6 negative feedback (NFB 6), adder 7, load 8

Timing diagrams of signals explaining the principle of operation of the prototype amplifier are shown in Fig. 2. (2 a - output voltage U with limits of change U ₁ and U ₂ , corresponding to discrete changes in voltages E ₁ (ϕ) and E ₂ (ϕ) of positive and negative power supply, 2 b) - signal diagrams for active load, 2 c ) - signal diagrams for capacitive load).

The prototype device works as follows. The input signal U _inx is fed to the input of the adder 7, where it is compared with the negative feedback signal for the output voltage U, supplied to the second input of the adder from the output of the OOS 6. As a result, the differential signal from the output of the adder 7 is fed to the input of the analog signal driver 2, under the influence of which forms the control signals U ₁ , U ₂ transistors direct and reverse conduction half-bridge circuit 3 with the appropriate level offset to implement the operation of the amplifier in class AB. Control signals U ₁ , U ₂ transistors are also transmitted to the inputs of the comparison circuit 4, where they are compared with the thresholds corresponding to low levels of power supply voltage. At low levels of the output voltage U, the control signals are below the thresholds, which corresponds to low signal levels at the input of the logical adder of the comparison circuit 4. In turn, the low output voltage of the logical adder of the comparison circuit 4 corresponds to the formation of low voltages of positive and negative polarity at the inputs of UIP 1. When the output voltage increases, one of the control signals from the output of driver 3 (in accordance with the polarity of the output voltage) exceeds the response threshold, which leads to to the formation of a high level at the input and output of the logic adder of the comparison circuit 4, and ensures the inclusion of a high voltage of positive and negative polarity at the outputs _of the UIP ₁ . low voltage power supply (Fig. 2a).

As shown in FIG. 2b for an active load, such a step change in the power supply voltage Ε, taking into account the level of the output voltage, can significantly reduce the residual voltage on the conductive transistors of the half-bridge circuit 3 of the class AB amplifier. In this case, the current i is in phase with the output voltage U, which achieves a symmetrical change in the residual voltages relative to the maximum value of the output signal U _max .

However, the presence of a phase shift ϕ _n , due to the complex load parameter, leads to a sharp increase in residual stresses U ₀ . As shown in FIG. 2b, c c for the case of a capacitive load (ϕ _n = π/2) with different polarities of current i and voltage U, the residual voltage on the conductive transistor increases to the level [E _in +|U|]=U ₀ , which causes a significant increase in energy losses in an amplifying device. Clarification of known data [Aleksandrov V.A., Maiorov V.A., Markova L.V. Modules of power electronics of the transmitting paths of the SAC. Marine radio electronics №2(64) 2018, p. 18-23] allows you to determine the relative heat dissipation power of the prototype amplifier when operating on a complex load.

For a high level of output voltage, we get:

where e ₁ \u003d E _n / E _in - the relative level of low voltage power supply;

ϕ ₁ \u003d arcsin e ₁ /m - phase of switching on the high voltage of the power supply;

1 > m > e ₁ - relative signal amplitude.

Analyzing the influence of the diagram (Fig. 2 c) of the residual voltage and output current of amplification transistors 3, neglecting the very small joint conduction zone of transistors characteristic of the AB mode, we can compare the known relation (1) for a class B amplifier with expression (2) for a known amplifier class G. It can be stated that when working on a complex load, the gain from using the prototype amplifier is significantly reduced. Moreover, in the most energy-consuming mode m = 1 when operating on a capacitive load for ϕ _n = π/2, the decrease in relative energy losses does not exceed 25%.

This circumstance is explained by the absence in the prototype amplifier of regeneration of reactive energy from the load capacitance to the power source capacitance, and, as a result, additional losses, which prevents the use of the prototype device in hydroacoustic transmitting equipment.

The objective of the present invention is to improve the energy efficiency of a class G linear power amplifier while ensuring its use as part of hydroacoustic transmitting equipment.

To solve the problem, in a class G amplifier containing a comparison circuit, an analog signal driver, the input of which is connected to the output of the adder, and the first and second outputs are connected to the control inputs of direct and reverse conduction transistors as part of a half-bridge class AB amplifier circuit with a discrete change in the power supply voltage positive and negative polarity of transistors of direct and reverse conductivity from a low level + E _n and -E _n to a high level + E _in and -E _in with a corresponding change in the output voltage U, while the positive and negative power supply outputs of the half-bridge class AB amplifier circuit are connected to to the corresponding power supply outputs of the analog signal driver, as well as to the positive and negative voltage outputs of the controlled power supply, the input of which is connected to the power supply bus, the first input of the adder is connected to the input signal bus, and the second input is connected to the output of the negative feedback circuit, entering there are additional features, namely, it includes a current sensor, an interface circuit, and the comparison circuit is made four-channel, moreover, the controlled power source is additionally equipped with the first, second and third, fourth control inputs connected through the interface circuit with the first, second and third, the fourth output of the four-channel comparison circuit, the output of the half-bridge circuit of the class AB amplifier is connected through a current sensor to the load input, and the output of the current sensor is connected to the input of the feedback circuit, and at the output voltage U > + E _n - ΔΕ, + E _n - ΔΕ > U > -E _n - ΔE and U < -E _n - ΔΕ, and the power supply voltage of the reverse conductivity transistor is set at the level -E _in , -E _n , + E _H is at the output voltage U < -E _n + ΔE, -E _n + ΔЕ < U < +Е _n - ΔЕ and U > +Е _n - ΔЕ while the value of ΔЕ is (0.005…0.2) Е _n .

The technical result from the use of the claimed class G amplifier is to ensure the regeneration of reactive energy from the load capacitance through the linear amplification link to a controlled power source at a high level of the output voltage of the power supply of transistors of positive and negative conductivity in three stages + E _in , + E _n , -E _n and -E _in , -E _n , + E _n . moreover, the inclusion of feedback on the output current provides the maximum depth of compensation for the nonlinearity of the amplitude characteristic of the linear amplifier (the depth of the OOS is more than 40 dB) to prevent distortion of the output signal in the forward and reverse power transmission cycles. Accordingly, an increase in energy efficiency is achieved under conditions of high quality of the excitation signals of hydroacoustic transducers with a pronounced capacitive component of conductivity.

The controlled power supply in the claimed device provides switching of high E _in and low E _n voltages according to the result of the control circuit for comparing the voltage level U of the output signal, taking into account the specified offset ΔE. As illustrated by the timing diagrams of the signals (Fig. 5), explaining the principle of operation of the proposed device, for the case U > E _n at a very low active power factor of the load cos ϕ _n = 0 ... 0.2, a significant decrease in the residual voltage U ₀ is provided compared to diagram (Fig. 2.c), characteristic of the prototype device

Excluding segments of energy losses in accordance with the diagram U ₀ (ϕ), corresponding to the use of the proposed device (Fig. ), for the most energy-intensive mode under the conditions m = e ₁ , with ϕ _n > 2ϕ ₁ , an expression can be obtained for estimating the relative power of heat release.

Comparing expressions (2) and (3) for the case of the maximum amplitude of the output voltage (m = 1) when operating on a capacitive load (ϕ _n = π/2), we obtain the relative values of energy losses: for the prototype device, 95%, and for the proposed device 63% of maximum output power.

The evaluation confirms the increase in the energy efficiency of hydroacoustic transmitting equipment when using the proposed class G amplifier in conditions of reducing energy losses by more than 30%.

The essence of the invention is illustrated in Fig. 1-5, where in Fig. 1 shows a block diagram of the prototype amplifier, Fig. 2 - timing diagrams of signals explaining the principle of operation of the prototype amplifier, in Fig. 3 is a block diagram of the proposed amplifier, in Fig. 4 - detailing the structure of building a controlled power source, an interface circuit and a comparison circuit, in FIG. 5 - timing diagrams of signals that determine the principle of operation of the claimed class G amplifier, while in FIG. 5a) - output voltage U with limits of change U ₁ and U ₂ corresponding to discrete changes in the voltages E ₁ (ϕ) and E ₂ (ϕ) of the positive and negative power supply of the class AB amplifier, in Fig. 5 b) shows the signals V ₁ (ϕ), V ₂ (ϕ), V ₃ (ϕ), V ₄ (ϕ) received at the corresponding inputs of the controlled power source, in Fig. 5 c) shows the positive and negative voltages of the high and low levels + E _in , + E _n , -E _n , -E _in , the voltage at the load U and the load current i, as well as the residual voltage U ₀ for various half-waves of the load current.

The proposed class G amplifier (Fig. 3) contains a controlled power supply 1, an analog signal driver 2, a half-bridge circuit 3 of a class AB amplifier, a comparison circuit 4, an interface circuit 5, a negative feedback circuit 6, an adder 7, a load 8 (GAP), current sensor 9. Comparison circuit 4 can be performed on operational amplifiers 4.1-4.4 and attenuator 4.5. In turn, the interface circuit 5 is performed on the drivers of pulse signals 5.1-5.4, the outputs of which are adapted to the control outputs of the UIP 1, and the UIP 1 may contain two diode-transistor switches 1.2, 1.3 and a four-channel secondary voltage converter 1.1.

The implementation of the structural nodes of the proposed device is determined by the functionality and corresponds to the principle of realizability according to known rules.

The controlled power source 1 must provide a step change in the power supply voltage Ε ₁ (ϕ) and Ε ₂ (ϕ) in steps + E _in , + E _n , -E _n and -E _in , -E _n , + E _n depending on the level of the output voltage U (Fig. 5) by commands received at the control inputs from the comparison circuit 4 through the interface circuit 5. The principal feature of the functioning of the controlled source 1 in the proposed device is a separate change in the power supply voltage for the transistors of the half-bridge circuit of the amplifier 3, including the transition to voltages of reverse polarity. In this case, conditions are formed for energy recovery from the capacitive component of the load to the capacitive filters of the secondary voltages of the power supply.

Controlled power source 1 can be made on a multi-channel voltage converter and two switches to form a three-stage voltage of positive and negative polarity.

A block diagram of a controlled power source 1 is shown in Fig. 4, which also shows the functional implementation of the comparison circuit 4 and the conjugation circuit 5.

The proposed power source 1 contains a four-channel secondary voltage converter 1.1 (VPN 1.1), and two three-channel switches 1.2 and 1.3 of direct and reverse conduction. To implement VPN 1.1, known circuits of multichannel voltage converters can be used, for example, those described in patents (RU 2267218 DC voltage transformer, publ. 27.12.2005). With a VPN of this type, secondary voltages of a given range of different polarity can be formed in the presence of significant capacitive filters for each individual output.

Implementation of the required algorithm for connecting secondary voltages generated by VPN 1.1 in switches 1.2 and 1.3 of positive and negative conductivity, transistors and diodes 1.2.1, 1.2.2, 1.2.3, 1.2.4, 1.2.5 and 1.3.1, 1.3 are used respectively .2, 1.3.3, 1.3.4, 1.3.5 (Fig. 4). Moreover, to connect the lowest stage of power supply, only diode switching can be used, which does not require additional control.

Connecting higher power levels of positive +E _in , +E _n and negative -E _in , -E _n polarity is carried out by turning on the corresponding transistors 1.2.2, 1.2.1 and 1.3.2, 1.3.1.

The transistor control commands are generated by a comparison circuit 4, which (Fig. 4) includes an attenuator 4.5 of the input voltage U and four comparison circuits (operational amplifiers) 4.1, 4.2, 4.3, 4.4, the direct inputs of which are connected to the output of the attenuator, and the inverse ones are connected to reference voltages +U + Δ, -U + Δ, +U - Δ, -U - Δ, where U = E _n K _a (K _a - attenuator gain). The discriminator 4 can be made on typical comparators when generating reference voltages using the appropriate resistive dividers of secondary power supply voltages.

The interface device 5 serves to transmit the output signals of the discriminator through the drivers to the control inputs of the power source 1 to turn on the corresponding transistors as part of the switches 1.2 and 1.3. Taking into account the different levels of power supply voltages E (+) and E (-), the drivers in the interface circuit can be made on floating point chips, for example IRFS4227PBF.

The analog signal driver 2 is designed to ensure the linear operation of the class AB amplifier 3 circuit. As a driver 2, an APEX module can be used, for example, of the PB63 type for a power supply voltage E V _up to 60 V, or a similar module (including those used in the prototype device), designed for the required maximum power supply voltage.

Circuit 3 of a class AB linear amplifier must contain field-effect or bipolar transistors of various conductivity, designed for the required output power and adapted to the selected power supply voltage. It is possible to use a similar technical solution used in the prototype device.

The OOS circuit 6 and the adder 7 provide the formation of a differential signal at the input of the analog signal driver 2, based on the condition of maintaining the specified value i of the output current of the circuit 3, corresponding to the value of the output signal U. The OOS circuit can be performed on a resistive divider with a given transfer coefficient of the output signal of the sensor 9 current. Adder 7 can also be in the form of a resistive summation circuit, or a typical operational amplifier.

Load 8, for the case of using the proposed technical solution as part of hydroacoustic equipment, is a hydroacoustic emitter with a pronounced capacitive component of conductivity. The active power factor of such a load, as a rule, does not exceed 0.3-0.5 and can be reduced to 0.1-0.2 in an extended frequency range with a band of up to 2-3 octaves.

- пропорциональный выходному току схемы 3 усилителя класса АВ.The current sensor 9 can be made on a current transformer, or on a Hall sensor chip, for example, type FHS40-P/SP600. In this case, the current sensor must generate a galvanically isolated signal

- proportional to the output current of circuit 3 of the class AB amplifier.

The choice of output current feedback in the proposed technical solution is due to the possibility of providing the maximum depth of compensation for non-linear distortions in a class AB amplifier, especially when operating on a capacitive load.

The above description of the functional units included in the proposed device corresponds to the conditions of technical feasibility, and the combination of their application ensures the achievement of a technical result from the implementation of the claimed invention.

The operation of the claimed amplifier is as follows.

и в виде разностного сигнала U_p поступает на вход драйвера 2 аналогового сигнала:The input signal U _inx is fed to the input of the adder 7, where it is compared with the feedback signal for the output current

and in the form of a difference signal U _p is fed to the input of the analog signal driver 2:

where β is the transmission coefficient of the circuit 6 OOS;

- выходной сигнал датчика 9 тока с коэффициентом преобразования

- output signal of current sensor 9 with conversion factor

K _c - the transfer coefficient of the adder 7.

The signal U _p is converted by the driver 2 into analog signals U ₁ , U ₂ excitation of the transistors of direct and reverse conduction circuit 3 of the class AB amplifier with appropriate biases that ensure linear operation.

U ₁ \u003d U _p K _Δ +U _cΔ ; U ₂ \u003d-U _p K _Δ - U _cΔ ,

where Κ _Δ , U _c - the transfer coefficient and the necessary offsets of the output signals of the driver 2. In accordance with the excitation signals, the current is closed through the forward and reverse conduction transistors:

where S [A / B] is the slope of the transfer characteristic of transistors of circuit 3 of a class AB amplifier.

As a result, the gain of the claimed amplifier can be written as:

Соответственно, коэффициент передачи по выходному току соответствует равенству:The feedback depth of this amplifier can be provided by 30-40 dB without violating the stability of the operation, which corresponds to the condition

Accordingly, the output current transfer coefficient corresponds to the equation:

- текущие значения выходного сигнала и выходного тока, в том числе для гармонического входного сигнала с относительной амплитудой U_вx = m sin ϕ (где m = U_вхМ/Мах|U_вхМ|).where U _in (ϕ),

- current values of the output signal and output current, including for a harmonic input signal with relative amplitude U _inx = m sin ϕ (where m = U _inM /Max|U _inM |).

In this case, the phase of the output current coincides with the phase of the input signal, and the phase of the output voltage is shifted by ϕ _n (ϕ _n = 0 ... - π/2 - for a capacitive load).

схемы 3 усилителя класса АВ, напряжение на нагрузке U определяется ее комплексным импедансом

At a given value of the output current

class AB amplifier circuit 3, the load voltage U is determined by its complex impedance

Depending on the voltage level U(ϕ), the comparison circuit 4 generates control signals V ₁ , V ₂ , V ₃ , V ₄ (Fig. 5 b) coming through the interface circuit to the control inputs of the power source 1. The high level Η of the control signals is determined from the operating conditions of the comparators 4.1, 4.2, 4.3, 4.4 as part of the comparison circuit 4:

where K _a - attenuator gain 4.5,

U = E _n ⋅K _a , Δ = (0.05…0.2) Е _n ⋅K _a

This ensures the inclusion of the corresponding transistors VT as part of the switches 1.2, 1.3 direct and reverse conduction of the controlled power source 1. Moreover, when the transistors of the switches 1.2 and 1.3 are turned off, the voltage at the outputs is formed through diodes 1.2.5 and 1.3.5 at levels - E _n and + E _n , respectively.

в условиях обеспечения уровня напряжения электропитания проводящего транзистора, необходимого для его линейной работы.As a result of the described algorithm for controlling the switches of the power source 1, determined by the proposed device of a class G linear amplifier, stepwise varying voltages Ε1(ϕ) and E2(ϕ), shown in Fig. 5 a). This achieves a decrease in residual voltages U ₀ (Fig. 5 c) for the positive and negative half-wave current

in terms of providing the voltage level of the power supply of the conductive transistor, necessary for its linear operation.

Thus, the implementation of the proposed class G amplifier makes it possible to exclude energy loss segments when operating on a capacitive load under conditions of high quality of the hydroacoustic transducer excitation signal. When operating on a load with a very low active power factor, typical for broadband hydroacoustic radiating antennas, the use of the present invention makes it possible to reduce energy losses in the nominal operating mode by more than 30%, which makes it possible to recommend the use of the proposed technical solution in hydroacoustic transmitters.

The proposed circuit solution of the class G amplifier allows to significantly improve the energy efficiency characteristics with high quality of the emitted hydroacoustic signals.

At present, the enterprise has developed samples of class G amplifiers based on the proposed technical solution, which are undergoing the stage of experimental testing, the results of which confirm the technical effect of the present invention. Relative energy losses in the nominal operating mode do not exceed 50%, which is half the losses in the class B amplifier and one and a half times lower than in the class G amplifier, implemented on the basis of the prototype device. The highlighted advantages of the proposed implementation of power amplifiers make it possible to recommend the implementation of the present invention in new developments of broadband radiating paths for hydrocommunication modes.

Claims (1)

Class G amplifier containing a comparison circuit, an analog signal driver, the input of which is connected to the output of the adder, and the first and second outputs are connected to the control inputs of direct and reverse conduction transistors as part of a half-bridge class AB amplifier circuit with a discrete change in the power supply voltage of positive and negative polarity of transistors direct and reverse conduction from a low level + E _n and -E _n to a high level + E _in and -E _in with a corresponding change in the output voltage U, while the positive and negative power supply outputs of the half-bridge class AB amplifier circuit are connected to the corresponding power supply outputs of the analog driver signals, as well as to the positive and negative voltage outputs of the controlled power source, the input of which is connected to the power supply bus, and the common output is connected to the common load bus, the first input of the adder is connected to the input signal bus, and the second input is connected to the output of the negative feedback circuit, characterized in that it includes a current sensor, an interface circuit, and the comparison circuit is made four-channel, moreover, the controlled power source is additionally equipped with the first, second, third and fourth control inputs connected through the interface circuit with the first, second, third and fourth outputs of the four-channel comparison circuits, the output of the half-bridge circuit of the class AB amplifier is connected through a current sensor to the load input, and the output of the current sensor is connected to the input of the feedback circuit, and at the output voltage U>+E _n -ΔE, +E _n -ΔE>U>-E _n -ΔE and U<-E _n -ΔE, the power supply voltage of the forward conduction transistor is set at +E _in , +E _n , -E _n , and the power supply voltage of the reverse conduction transistor is set at -E _in , -E _n , + Е _n at the output voltage U<-Е _n +ΔЕ, -Е _n +ΔЕ<U<+Е _n -ΔЕ and U>+Е _n -ΔЕ, while the value of ΔЕ is (0.005…0.2) Е _n .

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