RU2675789C1 - Adaptive compensator of packet radiation interference - Google Patents

Abstract

FIELD: radio engineering.SUBSTANCE: invention relates to the field of radio engineering and can be used to provide reception in conditions of powerful interference, occupying the entire frequency band of the desired signal when the radio station is operating in batch mode or in the mode with pseudorandom hopping (PRH). Device consists of two blocks of decomposition of a real signal into quadrature components 1, 1, the amplitude equalization unit of the signals 2, the unit calculating phase difference 3…3, constant delay filter 4, filters of a constant delay of the multi-channel unit for calculating the phase difference 4…4, phase rotation blocks 5…5, subtraction blocks 6…6, delay filters 7…7, filter with compensating delay 7, squaring blocks 8–8, totalizers 9…9, low pass filters 10…10, block select the minimum value of 11.EFFECT: technical result is an increase in the depth of interference compensation.1 cl, 4 dwg

Description

The invention relates to the field of radio engineering and can be used to provide reception under conditions of powerful interference, occupying the entire frequency band of a useful signal, when the radio is in burst mode or in a mode with pseudo-random frequency tuning (PFC).

Currently, there is an increasingly acute problem of ensuring the reception of signals in the presence of interference. One of the known approaches to solving this problem is the use of adaptive interference cancellers. A device is known that performs the function of adaptive interference compensation [RU 2115233 C1, Н04В 1/10, publ. 07/10/1998]. In this device, the signals from two inputs are aligned in amplitude in two normalizing amplifiers. Next, the signal orthogonalization procedure is performed, as a result, two signals with the same amplitude and orthogonal to each other are obtained. To compensate for the interference, both signals are again rotated in phase with respect to the output signal and added together. However, when a radio link is operating with broadband signals, the current interference may cover the entire spectrum of the signal. Moreover, the known adaptive interference cancellers correct the amplitudes and phases of the received signals in such a way that, after their subsequent summation, the interference signal is completely compensated. However, when constructing a receiving system with several antenna inputs, the received signals will differ not only in the values of the amplitudes and initial phases, but also in the amount of time delays. As you know [Honorovsky I.S. Radio engineering circuits and signals: a textbook for universities. - 4th ed., Revised. and add. - M .: Radio and communications, 1986. - 512 p. p. 31], the time delay of the signal in the spectral region is equivalent to multiplying the signal spectrum by the value:

where ω is the frequency;

t ₀ - time delay;

e is the Euler number.

Thus, for a large value of t ₀ in the exponent (1), a different phase rotation of the spectral components with different frequencies ω can occur. As a consequence of this phenomenon, deep compensation of interference only by correcting the amplitude and phase of the signal becomes impossible.

Known adaptive interference canceller [RU 2282939 C1, H04B 1/10, G01S 7/36, publ. 08/27/2006 Bull. No. 24], providing effective compensation of continuous narrowband signals. In this device, the signals from two inputs are aligned in amplitude in two normalizing amplifiers. Next, the signal orthogonalization procedure is performed, as a result, two signals with the same amplitude and orthogonal to each other are obtained. To compensate for the interference, both signals are again rotated in phase with respect to the output signal and added together. In addition, the device uses an amplitude detector that disables the compensation circuit when the tracking circuit approaches the “alignment”, which allows reception without interference compensation when the amplitudes and phases of the useful signal in both channels are equal. In this invention, the interference compensation is carried out due to the feedback signal and as a consequence, this system has an inertia, which is a disadvantage of this device. As a consequence of the inertia of the device, at the beginning of receiving a short information packet, incomplete interference compensation occurs, which in turn does not allow reliable reception of a packet of information or frequency hopping signals.

The closest in technical essence to the proposed one is the device described in the article [Makovy V.A., Chupeev S.A. Adaptive jammer for a packet radio station. Theory and technique of radio communications, 2017 No. 2, pp. 115-118], taken as a prototype. The prototype device diagram is shown in FIG. 1, where indicated:

1 ₁ , 1 ₂ - blocks the decomposition of the material signal into quadrature components;

2 - block alignment of the amplitude of the signals;

3 - block calculating the phase difference;

4 ₁ , 4 ₂ - constant delay filters;

5 - phase rotation unit;

6 - block subtraction.

The prototype device contains blocks for decomposing the material signal into quadrature components 1 ₁ and 1 ₂ , the outputs of which are connected to the first and second input of the amplitude equalization block 2, the first output of which is connected to the first input of the phase difference calculation unit 3 and the input of the constant delay filter 4 ₁ , whose output is connected to a first input of the phase rotation unit 5, the second output signal amplitude equalizing unit 2 is connected to the second input of the phase difference calculation unit 3 and a filter constant delay February _4, whose output is connected to a the next input of the subtraction block 6. The output of the phase difference calculation block 3 is connected to the second input of the phase 5 rotation block, the output of which is connected to the first input of the subtraction block 6. As a result, the amplitude correction signal is generated from the output of the amplitude equalization block 2 from the output of the difference calculation block phase 3 - phase difference signal.

The prototype device works as follows.

The inputs of the device receives two signals S ₁ (t) and S ₂ (t):

where A ₁ , A ₂ are the amplitudes of the input signals;

ƒ ₁ - input signal frequency;

ϕ ₁ , ϕ ₂ are the initial phases of the input signals;

τ _1, τ ₂ - the delay time of signals.

The signals S ₁ (t) and S ₂ (t) differ from each other only in the magnitude of the amplitudes of the initial phases and the propagation delay time of the signal. This is due to the non-identity of the amplitude-frequency characteristics (AFC) and phase-frequency characteristics (PFC) of the receiving channels, as well as the delay of the signal in one channel relative to the other. Further, each signal is decomposed into quadratures, as a result, we obtain two quadrature signals I ₁ , Q ₁ and I ₂ , Q ₂ . To equalize the amplitudes of the interference signals, in each quadrature channel, the power of the received interference signal is estimated by the formula: P = I ² + Q ² . Then, the ratio of the power of the interference signal received on both channels is calculated. After that, the resulting ratio is filtered in a low-pass filter in order to reduce the influence of additive Gaussian noise. Next, the square root is extracted, resulting in a coefficient that relates the ratio of the amplitudes of the interference signals in both reception channels. To correct the amplitudes, the quadrature components of the signal of the second channel are multiplied by the obtained amplitude correction coefficient. As a result of the signal processing, both quadrature signals will have the same amplitude equal to the value of A ₁ . The above operations are performed in the block correction amplitude 2. This block has two quadrature inputs and two quadrature outputs.

To compensate for the interference, it is also necessary to generate signals coming from both receiving paths in antiphase. To solve this problem, in the phase difference calculation unit 3, the cosine and sine of the phase difference between the two quadrature reception channels are estimated. For this, the signal from the second channel is multiplied by one of the quadrature components of the signal of the first channel, after which the high-frequency components are filtered by low-pass filters. In fact, the cosine and sine of the phase difference of the received interference signal are estimated by the following formula:

where h is the impulse response of the low-pass filter;

* - convolution operation;

Δϕ is the difference between the initial phases;

I ₁ , I ₂ - the real part of the signals of the first and second channel;

Q ₂ - the imaginary part of the signal of the second channel.

The calculations carried out by f. (3) are made in the phase difference calculation unit 3. This unit has two quadrature inputs and one quadrature output.

Next, we rotate the quadrature components in phase, for which we multiply the quadrature components of the first signal delayed in the delay filter 4 ₁ by the phase difference obtained in block 3. Multiplication is performed according to the rules for multiplying complex numbers by the following formula:

where I ₁ is the real part of the signal of the first channel;

Q ₁ is the imaginary part of the signal of the first channel.

The calculations carried out by f. (4), are produced in a phase 5 rotation block. This block has two quadrature inputs and one quadrature output.

As a result of the operations, we obtain two interference signal vectors, presented in the form of quadrature components having the same amplitude and the same phase. To perform interference compensation, we subtract one vector from another. For this, the signal from the second output of block 2 is delayed in the delay filter 4 ₂ and subtracted from the signal rotated in phase in the phase 5 rotation block.

The disadvantage of the prototype device when working with a broadband signal: in the case of exposure to broadband interference, the interference signal is not fully compensated. Let's consider this question in more detail. As an example, we will analyze the operation of the prototype device in the VHF frequency range (30 ... 108 MHz) with a signal with a spectrum width of 1000 kHz. Assume that the receiving antennas are located at a distance of 10 meters from each other. The maximum delay of the front during propagation at a distance of 10 meters will be τ = 3.3⋅10 ^-8 sec. In accordance with formula (1), the phase deviation due to the delay of the signal during tuning at a frequency of 1000 kHz will be 1.91 °. Consequently, the level of uncompensated interference during frequency tuning at 1000 kHz will correspond to the difference of vectors with the same amplitude and rotated in phase by an angle ϕ. Using the rules of geometry, we get:

where L is the difference in distances that the received signal passes at the outputs of the antenna inputs;

∂F is the frequency offset;

c is the speed of light.

To determine the interference / signal ratio at which information is received reliably with interference cancellation, it is necessary to set the signal-to-noise ratio at which information is received reliably. Thus, we obtain the following relationship, which determines the maximum signal to noise ratio:

where R is the depth of noise suppression in dB;

SNR - signal-to-noise ratio at the input of the receiving device, which ensures stable reception of information.

Setting SNR = 15 dB and performing calculations in accordance with (6), we obtain in our case the maximum value of interference / signal, which ensures reliable reception of 14.5 dB, which in some cases is insufficient to ensure reliable operation of the communication system.

The claimed invention solves the problem of improving the quality of signal reception under the influence of interference by providing deep compensation for broadband interference.

The technical result of the invention is the achievement of the desired interference / signal ratio, which ensures the reception of a useful signal at the sensitivity level of the receiver.

To achieve a technical result, an adaptive interference compensator of a packet radio station contains blocks for decomposing a material signal into quadrature components, the inputs of which are device inputs, and the outputs are connected to the inputs of the signal amplitude equalization unit, a phase difference calculation unit, constant delay filters connected to the unit located in series phase rotation and subtraction unit, according to the invention, n delay filters are inserted into it, a multi-channel phase difference calculation unit, including It contains n constant delay filters, n phase difference calculation blocks and n phase rotation blocks, a compensating delay filter, n-1 subtraction block, a multi-channel power calculation block, including 2n squaring blocks and n summing devices connected to n low-pass filters and a minimum value selection unit, the inputs of the delay filters being connected to the first output of the signal amplitude equalization unit, and the outputs being connected to the inputs of the constant delay filters of the multi-channel phase difference calculation unit and by the first inputs of the phase difference calculation blocks, the compensating delay filter input is connected to the second output of the signal amplitude equalization block, and the output is connected to the second inputs of n phase difference calculation blocks, with the constant delay filter inputs of the multi-channel phase difference calculation block and the constant delay filter input connected the output with the second inputs of the subtraction blocks, the first inputs of which are connected to the outputs of the phase rotation blocks, the inputs of which are connected to the outputs of the blocks of the calculation of the phase difference and output the constant delay filters of a multi-channel phase difference calculation unit, wherein the output of each subtraction unit is connected to the inputs of two squaring units corresponding to the first and second quadrature of the signal, and the outputs of the squaring units are paired to the inputs of n summing devices, the outputs of which are connected to inputs of low-pass filters connected by outputs to the inputs of the minimum value selection block.

The invention is illustrated by drawings: in FIG. 1 shows a block diagram of a prototype device; in FIG. 2 is a functional diagram of the inventive device; in FIG. 3 schematically illustrates a signal reception process; in FIG. 4 - phase response signals.

On the functional diagram of the inventive device (Fig. 2) is indicated:

1 ₁ , 1 ₂ - blocks the decomposition of the material signal into quadrature components;

2 - block alignment of the amplitude of the signals;

3 ₁ ... 3 _n - n phase difference calculation blocks;

4 ₀ - constant delay filter;

4 ₁ ... 4 _n - n constant delay filters of a multi-channel phase difference calculation unit;

5 ₁ ... 5 _n - n blocks of phase rotation;

6 ₁ ... 6 _n - n blocks of subtraction;

7 ₁ ... 7 _n - n delay filters;

7 ₀ - filter with compensating delay;

8 ₁ ... 8 _2n - 2n squaring blocks;

9 ₁ ... 9 _n - n summing devices;

10 ₁ ... 10 _n - n low-pass filters;

11 is a block for selecting the minimum value.

The inventive device comprises blocks for decomposing the material signal into quadrature components 1 ₁ and 1 ₂ , the outputs of which are connected to the first and second input of the amplitude equalization block 2, the first output of which is connected to the inputs of the delay filters 7 ₁ ... 7 _n , the second output of the signal amplitude equalization block 2 connected to the input of the filter with a compensating delay of 7 ₀ . The outputs of the delay filters 7 ₁ ... 7 _{n are} connected to a multi-channel phase difference calculation unit, consisting of constant delay filters 4 ₁ ... 4 _n , phase difference calculation units and phase rotation units. In particular, the outputs of the delay filters 7 ₁ ... 7 _{n are} connected to the first inputs of the phase difference calculation blocks 3 ₁ ... 3 _n , and the inputs of the filters of constant delay 4 ₁ ... 4 _n , the outputs of which are connected to the first inputs of the phase rotation blocks 5 ₁ ... 5 _n . The filter output with a compensating delay of 7 _{0 is} connected to the input of the constant delay filter 4 ₀ , the output of which is connected to the second inputs of the subtraction blocks 6 ₁ ... 6 _n . The outputs of the phase difference calculation blocks 3 ₁ ... 3 _{n are} connected to the second inputs of the phase rotation blocks 5 ₁ ... 5 _n , the outputs of which are connected to the first inputs of the subtraction blocks 6 ₁ ... 6 _n , the outputs of which are connected to the multi-channel power calculation block, consisting of 2n blocks squaring and n summing devices. In particular, each of the quadrature outputs is connected to the input of the squaring block 8 ₁ ... 8 _2n , the outputs of which are paired with the corresponding first and second inputs of the summing devices 9 ₁ ... 9 _n . Their outputs are connected to the inputs of the low-pass filters 10 ₁ ... 10 _n , connected to the inputs of the block for selecting the minimum value of 11.

The inventive device operates as follows.

Before receiving a packet, the device measures the interference parameters in order to obtain the correction coefficients of the amplitude, cosine and sine of the phase difference, as well as the necessary signal delay time in the absence of a useful signal. A timing diagram of the operation of the device is shown in FIG. 3. The process of obtaining the correction coefficients of the amplitude, cosine and sine of the phase difference, as well as the necessary signal delay time, is performed by the device shown in FIG. 2 and consists in the execution by the device blocks of the following actions. Further, as in the prototype, the signals S ₁ (t) and S ₂ (t) from two receiving paths are fed to blocks for decomposing the material signal into quadrature components, from the outputs of which we obtain two quadrature signals I ₁ , Q ₁ and I ₂ , Q ₂ . Further, by analogy with the prototype, in block 2, the amplitudes of the quadrature components are aligned. As a result, we obtain two quadrature signals with the same amplitudes. From the output of the signal amplitude equalization block 2, the signal enters n delay filters. Each of them delays the signal for a while. The number of delay filters n and the delay values are selected so that the sum of the delay difference modules in the delay filters 7 ₁ ... 7 _n and in the filter with compensating delay 7 ₀ are equal to two signal delay values between the antennas.

where τ _i is the signal delay in the i-th filter 7 _i ;

τ ₀ - signal delay in the filter with a compensating delay;

L is the distance between the antennas;

c is the speed of light.

The value of n - is selected based on the necessary depth of compensation and the width of the spectrum of the useful signal. Given that when the signal passes through the delay filters 7 ₁ ... 7 _n , in one of the filters the maximum compensation of the delays between the antennas and the reception paths will take place, and also taking into account relation (6), having performed elementary transformations, we obtain the following relation for the number of delay filters n :

where the values of L, s, SNR are described above.

Next, the signals from the delay filters 7 ₁ ... 7 _n are fed to the first inputs of the phase difference calculation blocks 3 ₁ ... 3 _n , the second inputs of which receive signals from the output of the filter with a compensating delay. In the blocks of the calculation of the phase difference 3 ₁ ... 3 _{n the} calculation of the phase difference between the signals having a different delay in time. Each phase difference value is supplied to the corresponding phase rotation unit 5 ₁ ... 5 _n . In phase rotation blocks 5 ₁ ... 5 _n , phase rotation of the signal of the first channel S ₁ (t) is performed. For this, the signals from the outputs of the delay filters 7 ₁ ... 7 _{n are} additionally delayed in the constant delay filters 4 ₁ ... 4 _{n of the} multi-channel phase difference calculation unit in order to compensate for the signal delay in the phase difference calculation units 3 ₁ ... 3 _n . After that, the delayed signals are fed to the first inputs of the phase rotation units, the second inputs of which receive signals from the phase difference calculation unit. As a result, at the outputs of phase rotation blocks 5 ₁ ... 5 _n, we obtain n quadrature signals I ₁ , Q ₁ , rotated in phase in the same way as signals I ₂ , Q ₂ . Further, in the subtraction blocks 6 ₁ ... 6 _n , the difference of the signals of both channels is calculated. But taking into account that the signals have different time delays, the outputs of the subtraction blocks 6 ₁ ... 6 _n will have incomplete interference compensation. At the same time, at the output of each of the subtraction blocks 6 ₁ ... 6 _n, it will be different, since when delaying the signal in the delay filters 7 ₁ ... 7 _n , the delay in the arrival of the signal of one channel relative to the other is actually compensated. With an increase in the number of delay filters n, it is possible to achieve the necessary value of the interference suppression coefficient in the adaptive interference compensator. Further, the received signal differences are fed to the squaring blocks 8 ₁ ... 8 _2n . At the same time, one of the quadrature components of the signal enters the input of each squaring block. In the squaring blocks 8 ₁ ... 8 _2n, these signals are squared, after which they are summed in pairs in n summing devices 9 ₁ ... 9 _n . In this case, the summation of the signals of the corresponding quadrature components is performed in pairs. Next, the signals are fed to low-pass filters 10 ₁ ... 10 _n , in which high-frequency components are filtered. As a result of the operations, we obtain n signals, each of which corresponds to the power of an incompletely compensated noise with a corresponding compensating delay. Further, these signals are fed to the inputs of the minimum value selection unit, in which the number of the output delay filter is determined, which achieves the highest signal compensation value. Thus, we obtain the values of the correction coefficient of the amplitude, cosine and sine of the phase difference, as well as the signal delay value, providing the required depth of interference compensation. Next, the sync preamble, the header and the information part of the packet are received, while the interference is compensated by the correction factors obtained at the measurement phase.

The implementation of the blocks 1 ₁ , 1 ₂ , 2, 3 ₁ ... 3 _n , 4 ₀ ... 4 _n , 5 ₁ ... 5 _n , 6 ₁ ... 6 _n in the inventive device is similar to the implementation of the corresponding blocks 1 ₁ , 1 ₂ , 2, 3, 4 ₁ , 4 ₂ , 5, 6 of the prototype device.

The implementation of blocks 7 ₁ ... 7 _n and 7 ₀ , depending on how the device will be implemented in practice, is different. For implementation in the form of a digital device that works with signal samples at a certain sampling frequency, delay filters 7 ₁ ... 7 _n are digital filters with a finite impulse response (FIR) and must provide a signal delay by a rational number of samples. Examples of the implementation of such filters are known and described [Sergienko AB Digital signal processing: study guide. - 3rd ed. - SPb .: BVH-Petersburg, 2011 .-- 768 p. With silt. on pages 282-285]. The implementation of the filter 7 ₀ is possible, as a filter, and is described in the book [Sergienko AB Digital signal processing: study guide. - 3rd ed. - SPb .: BVH-Petersburg, 2011 .-- 768 p. With silt. on page 282].

The implementation of blocks 9 ₁ ... 9 _n and blocks 8 ₁ ... 8 _{n is} known and actually represents the performance of arithmetic operations. Depending on the use of digital number formats, examples of the implementation of these operations are given in the book [Solonina A.I., Ulahovich D.A., Yakovlev L.A., Algorithms and processors for digital signal processing. - SPb .: BHV-Petersburg, 2002 .-- 464 p.: Ill. on pages 175-192].

The implementation of low-pass filters 10 ₁ ... 10 _{n is} known and described in the book [Sergienko AB Digital signal processing: study guide. - 3rd ed. - SPb .: BVH-Petersburg, 2011 .-- 768 p. With silt. on pages 389-390].

The implementation of the block for selecting the minimum value of 11 is known and is given in the book [W. Titz, C. Schenk, Semiconductor Circuitry: A Reference Guide. Per. with him. - M .: Mir, 1982. - 512 p., Ill. on pages 329-331].

и, как следствие, меняется наклон ФЧХ, и имеет место набег фазы по частоте. При этом, чем более широкополосный будет сигнал и большая будет разница задержек сигнала в трактах приема, тем больший будет набег фазы. Коррекция фазы принимаемого сигнала при работе прототипа позволяет перемещать ФЧХ вдоль оси ординат, не меняя наклона. Таким образом, устройство-прототип не позволяет полностью скомпенсировать сигнал помехи. Как было показано выше, в случае для сигнала шириной спектра 1000 кГц, расстояния между приемными антеннами 10 метров, глубины компенсации 50 дБ и необходимым отношении сигнал/шум 15 дБ отношение помеха/сигнал составляет всего 14.5 дБ. Для устранения данного недостатка в заявляемое устройство добавлены фильтры задержек, которые осуществляют задержку сигнала одного из каналов приема на различное количество величин, при этом задержка сигнала другого канала задерживается на некоторое фиксированное значение. Далее производится компенсация по фазе так же, как и в прототипе, и ищется случай с наибольшим значением компенсации сигнала. Таким образом, глубина компенсации увеличивается. Задавшись необходимым значением глубины компенсации, по формуле (8) можно определить необходимое количество фильтров задержек n. Для вышеприведенного примера n>30.To prove the effectiveness of the inventive device, we consider in more detail its work. Consider the phase response of the signals shown in FIG. 4. The solid line represents the phase response of the original signal, the dashed line represents the phase response of the signal rotated in phase by the input circuits of the radio receiving device (RPU), and the dashed line represents the delayed signal. When the signal passes through the input circuits of the RPU, the input filters rotate the signal in phase. Given that the selective characteristic of the input circuits of the RPU is more broadband, then all spectral components of the signal will be rotated by approximately the same angle. As a result of parallel reception at two antenna inputs, we obtain almost parallel lines of the phase spectrum of the signal. Also, various time delays will act on the input signals. The time delay is due to the distance between the antennas, the signal delay in the feeder paths and RPU filters. As shown earlier, the time delay of the signal is equivalent to multiplying the signal by a factor

and, as a consequence, the slope of the phase response changes, and there is a phase incursion in frequency. In this case, the more broadband the signal and the greater the difference in signal delays in the reception paths, the greater the phase shift will be. The correction of the phase of the received signal during the operation of the prototype allows you to move the phase response along the ordinate, without changing the slope. Thus, the prototype device does not fully compensate for the interference signal. As shown above, in the case of a signal with a spectral width of 1000 kHz, the distance between the receiving antennas is 10 meters, the compensation depth is 50 dB and the required signal / noise ratio is 15 dB, the interference / signal ratio is only 14.5 dB. To eliminate this drawback, delay filters are added to the inventive device, which delay the signal of one of the receiving channels by a different number of values, while the delay of the signal of the other channel is delayed by some fixed value. Next, phase compensation is performed in the same way as in the prototype, and the case with the highest signal compensation value is searched. Thus, the depth of compensation increases. Given the necessary value of the compensation depth, using the formula (8), one can determine the required number of delay filters n. For the above example, n> 30.

Claims (1)

An adaptive jammer of a packet radio station containing blocks for decomposing a material signal into quadrature components, the inputs of which are the inputs of the device, and the outputs are connected to the inputs of the signal amplitude equalization block, a phase difference calculation unit, constant delay filters connected to a phase rotation unit and a subtraction unit characterized in that n delay filters are introduced into it, a multi-channel phase difference calculation unit including n constant delay filters, n blocks for calculating the phase difference and n phase rotation blocks, a compensating delay filter, n-1 subtraction block, a multi-channel power calculation block, including 2n squaring blocks and n summing devices, connected to n low-pass filters, and a minimum value selection block, moreover, the inputs of the delay filters are connected to the first output of the signal amplitude equalization unit, and the outputs are connected to the inputs of the constant delay filters of the multi-channel phase difference calculation unit and the first inputs of the difference calculation units phase, the compensating delay filter input is connected to the second output of the signal amplitude equalization block, and the output is connected to the second inputs of n phase difference calculation blocks, with the constant delay filter inputs of the multi-channel phase difference calculation block and the constant delay filter input connected to the output with the second inputs of the blocks subtractions, the first inputs of which are connected to the outputs of the phase rotation blocks, the inputs of which are connected to the outputs of the phase difference calculation blocks and the outputs of the constant delay filters of the phase difference calculation unit, wherein the output of each subtraction unit is connected to the inputs of two squaring blocks corresponding to the first and second quadrature of the signal, and the outputs of the squaring blocks are paired with the inputs of n summing devices, the outputs of which are connected to the inputs of the low-pass filters connected by the outputs to the inputs of the block for selecting the minimum value.

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