RU2618522C1 - Phase direction finder - Google Patents

Abstract

FIELD: radio engineering.

SUBSTANCE: invention relates to radio engineering and can be used in radio monitoring when searching for radio-frequency radiation sources within confined territory and in premises, for example, special electronic information interceptors. Said result is achieved due to introduction of the fourth adder, connected via the first direct input with output of the second phase changer, and via the second direct input with output of the third controlled phase changer, fifth adder, connected via the first direct input with output of the second phase changer, and via the second inverting input with output of the third controlled phase changer, second subtractor, connected via input of subtrahend through the third device of calculation module with output of the fourth adder, and via input of subtrahend through the fourth device of calculation module with output of the fifth adder, sixth adder connected via the first and second uncomplemented inputs with outputs of the third and fourth calculation devices of module, respectively, second amplifier, connected via input with the output of the second subtractor and output of the second base limiter at zero level simultaneously, second comparator, connected via the first input with output of the sixth adder, and via the second input with output of the second amplifier. Output of the second comparator is the second output of the phase direction finder.

EFFECT: achieved technical result – possibility of using direction finder to determine angular position of radio-frequency sources simultaneously in two orthogonal planes.

1 cl, 24 dwg

Description

The invention relates to the field of radio engineering and can be used in radio monitoring when searching for sources of radio emission (IRI) in a limited area and in rooms, for example, special electronic devices for intercepting information.

There are various schemes of phase direction finders, for example, US patent No. 4383301 IPC G01S 5/02, 7/04, as well as a direction finder according to the application No. 1333546 (Great Britain) IPC G01S 3/48, 3/10.

The classical diagram of the phase direction finder (analogue) is shown in Figure 8.1 c. 195 in the book: V.A. Cherdyntsev "Radio Engineering Systems". Minsk, “Higher School”, 1988, 369 pp. The disadvantage of the analogue is the fact that the determination of the bearing to the source of radio emission is made only in one plane of the spatial angles of azimuthal or elevation.

Of the known devices, the closest in technical essence to the claimed (prototype) is the phase direction finder given in the patent of the Russian Federation No. 2282872 for the application No. 2005107832/09 with priority dated March 21, 2005, registered in the GRI of the Russian Federation on August 27, 2006, the authors: Kamashev B.V., Kamashev A.B., Podluzhny V.I., Podduzhny A.V., Ryumshin R.I.

The known phase direction finder contains the first and second antennas spaced a certain distance, the first and second receiving paths connected by inputs to the first and second antennas, respectively, the first and second phase shifters having fixed and mutually opposite phase shifts and connected by inputs to the output of the first receiving path, a phase detector connected by one input to the output of the first phase shifter, and the second input through the third controlled phase shifter with the output of the second receiving path, the first and second facets lower numbers at the zero level, the first adder connected by the first direct input through the first limiter from the bottom at zero level with the output of the phase detector, and the second direct input with the output of the second limiter from the bottom at zero level, the second adder connected by the first direct input with the output of the first limiter from the bottom at the zero level, and the second inverse input with the output of the second limiter from the bottom at zero level, the first and second device computing module connected by the first input to the output of the first adder, the second input ohm with the output of the second adder, the first subtractor connected to the input of the module calculating unit being reduced with the output of the first device, and the input of the module subtracting output from the second unit, the third adder connected to the first and second direct inputs with the outputs of the first and second module calculating devices, respectively, the first an amplifier connected by an input to the output of the first subtractor, a first comparator connected by a first input to the output of the first amplifier, and a second input with the output of the third ora, a key circuit, a first input coupled with the output of the first comparator, the oscillator control voltage is coupled to second inputs of the output third controllable phase shifter circuit and the key simultaneously. The output of the key circuit is the first output of the direction finder.

The principle of measuring the angular coordinate (bearing) of a radiation source of radio waves in a known direction finder is implemented by comparing the phases of the signals received by antennas spaced apart in the measurement plane. The direction finder circuit is constructed in a two-channel version providing this comparison. A feature of the scheme is to increase the accuracy of determining the bearing. The solution to this problem is carried out by the joint application of the functional procedures of intersection and union.

For this, from two voltages shifted along the angular coordinate and intersecting in the equal-signal direction, the formation by the circuit elements of the “intersection” and “union” signals is ensured. These voltages are rigidly interconnected by a common point located at an angle in the equal-signal direction, and according to the level determined by phase shifters with fixed phase shifts. Then, from the “intersection and“ association ”signals, a narrow region is formed in the vicinity of the equal-signal direction, which determines the bearing reference time and is the direction-finding characteristic that determines the direction-finding accuracy for a given signal-to-noise ratio at the input of the receiving channels. In this case, due to the adjustment of the controlled phase shifter, it is possible to scan with the same signal direction within a certain sector.

The disadvantages of the known phase direction finder include limited functionality, which consists in the fact that the determination of the bearing to the source of radio emission is made only in one plane of the spatial angles of azimuthal or elevation. This is due to the analysis and use of only phase information contained in the received signals.

However, at present, when searching for IRI, the urgent task is to determine its angular positions simultaneously in two orthogonal planes. The analysis shows that this is possible by using not only phase, but also amplitude information in the received signals of the IRI.

The problem to which the claimed device is directed is to form a narrow bearing reference area based on the amplitude information in the received signals of the IRI and to apply the crossing and combining procedures when processing this information.

The technical result, the achievement of which the present invention is directed, consists in providing the possibility of determining the angular position of the IRI by the direction finder simultaneously in two orthogonal planes.

The technical result is achieved by the fact that in the known phase direction finder containing the first and second antennas spaced a certain distance, the first and second receiving paths connected by inputs to the first and second antennas, respectively, the first and second phase shifters having fixed and mutually opposite phase shifts and connected by inputs to the output of the first receiving path, a phase detector connected by one input to the output of the first phase shifter, and the second input through the third controlled phase shifter with the output the second receiving path, the first and second lower limiters at zero level, the first adder connected by the first direct input through the first lower limiter at zero level to the output of the phase detector, and the second direct input with the second lower limiter output at zero level, the second adder connected by the first direct input with the output of the first limiter from the bottom at zero level, and the second inverse input with the output of the second limiter from the bottom at zero level, the first and second module calculating devices connected to the first the input with the output of the first adder, the second input with the output of the second adder, the first subtractor connected to the input of the module computation being reduced with the output of the first device, and the input of the module subtracted with the output of the second calculator, the third adder connected to the first and second direct inputs with the outputs of the first and the second device for calculating the module, respectively, the first amplifier connected to the input of the output of the first subtracting device, the first comparator connected to the first input to the output of the first amplifier Itel, and the second input with the output of the third adder, the key circuit connected to the first input with the output of the first comparator, the control voltage generator connected to the output with the second inputs of the third controlled phase shifter and the key circuit at the same time, and the output of the key circuit is the first output of the direction finder, the fourth adder is introduced connected by the first direct input to the output of the second phase shifter, and the second direct input to the output of the third controlled phase shifter, the fifth adder connected by the first direct input with the output of the second phase shifter, and the second inverse input with the output of the third controlled phase shifter, the second subtractor connected to the input of the module being reduced through the third device and output of the fourth adder, and the input of the module subtracted through the fourth device to output the fifth adder, the sixth adder connected by the first and second direct inputs with the outputs of the third and fourth devices for calculating the module, respectively, the second amplifier connected to the input with the output of the second subtracting of the first device and the input of the second limiter from below at the zero level at the same time, the second comparator, connected by the first input to the output of the sixth adder, and the second input with the output of the second amplifier, the output of the second comparator are the second output of the phase direction finder.

The invention is illustrated by drawings. In FIG. 1 is a structural diagram of a phase direction finder. In FIG. 2 ... 7 show the dependences calculated theoretically. In FIG. 8 ... 15, 18 signals are presented at various points of the circuit according to the simulation results. In FIG. 16, 17, 19 ... 24 presents the results of a statistical evaluation of the effectiveness of the proposed direction finder.

The phase direction finder (Fig. 1) contains the first 1 and second 2 antennas spaced a certain distance d, the first 3 and second 4 receiving paths, the first 5 and second 6 phase shifters having fixed and mutually opposite phase shifts, the third controlled phase shifter 7, phase detector 8, first 15 and second 16 bottom limiters at zero, the first 17, second 18, third 24, fourth 9, fifth 10 and sixth 14 adders, the first 19, second 20, third 11 and fourth 12 of the module calculation device, first 21 and the second 13 subtracting devices, the first 25 and watts Ora 23 amplifiers, first 26 and second 22 comparators, key circuit 27, control voltage generator 28.

Before explaining the purpose of the elements and the principle of operation of the circuit, we justify the general approach to the construction of a direction finder.

When constructing direction finders for radio monitoring, especially manual ones, they strive to provide a combination of conflicting requirements such as compact equipment, speed of space viewing and high accuracy in determining the angular positions of IRI.

The implementation of these requirements is possible by using the principle of building complex monopulse systems [Leonov A.I., Fomichev K.I. Monopulse radar. M .: Radio and communication, 1984. - 312 p.], Electronic scanning of the investigated space at least in one plane and special measures to reduce direction finding errors. As these measures, the joint application of functional procedures of intersection and union is possible [Gordienko V.I., Dubrovsky S.E., Ryumshin R.I., Fenev D.V. Universal multifunctional structural element of information processing systems. / Electronics / Izv. Universities, No. 3, 1998, - from 13-17], allowing to obtain narrow resulting direction-finding characteristics (HR). The proposed direction finder scheme combines these requirements.

In order to manage only two interconnected channels during direction finding in two planes, antennas 1 and 2 in the microwave range can be irradiators of one mirror, which is a notch from a parabolic cylinder. For this, the irradiators providing the formation of two rays I and II are tilted to each other at an angle of 2ε _cm in the horizontal plane, and spaced apart by a distance d in the vertical plane. This ensures the direction finding of the IRI in the horizontal plane by the amplitude method for the mechanical movement of the antennas, and in the vertical - by the phase method in electronic scanning.

To create a tilt of the rays relative to each other in the horizontal plane (Fig. 2, a) the irradiators are located close to, but on different sides of the focal plane (one on the left, the other on the right).

In the formation of beams in the vertical plane, antennas are spaced apart by a distance d, and the upper and lower halves of the parabolic cylinder. Since the phase method is used in this plane, these rays can be considered as one, which is conventionally shown in FIG. 2, b.

Similarly, it is possible to form the corresponding radiation patterns in the longer wavelength part of the VHF range without using a mirror, for example, using antennas such as a vibrator-reflector or horn antennas.

The first 3 and second 4 receiving paths carry out standard processing operations at high and intermediate frequencies.

Phase shifters 5 and 6 with fixed and opposite phase shifts ± β form two phase-shifted voltages for the elevation channel from the signal of the first antenna, the angular dependences of which intersect at a constant and high level in the equal signal direction (RSN), providing further this direction.

The control voltage generator 28 and the third controlled phase shifter 7 provide scanning with an equal signal direction in the elevation plane in a given sector by changing the phase of the reference voltage generated from the signal from the second antenna.

Phase detector 8 forms the first PX elevation channel shifted by an angle β relative to the RSN.

The fourth 9, fifth 10 and sixth 14 adders, the third 11 and fourth 12 of the module calculating device and the second subtracting device 13 generate the intersection signals (from the output of the subtracting device 13) and the combination (from the output of the adder 14). In this case, the intersection signal for the elevation plane is equivalent to the signal from the output of the phase detector 8, but shifted by an angle of β relative to the RSN, that is, this is the second PX of the elevation channel, which is used further to obtain the RSN and the resulting PX of this channel. Thus, phase information regarding the bearing in the elevation plane is removed from the phase detector 8 and the second subtractor 13.

Further, the signals of the first and second PX pass the lower limiters at the zero level of 15 and 16, which exclude the negative branches of the shifted direction-finding characteristics of the elevation channel, eliminating false bearings in this channel.

The first 17, second 18 and third 24 adders, the first 19 and second 20 of the module calculating device and the first subtracting device 21 form from the signals of the first and second PX signals of intersection (from the output of the first subtractor 21) and combining (from the output of the adder 24). These signals have the same extrema on the equal signal direction and are used to obtain a narrow bearing reference area or the resulting direction finding characteristic of the elevation channel, which is formed at the output of the first comparator 26 after comparing the intersection signal amplified in the first amplifier 25 and the combining signal from the output of the third adder 24.

Further, the key circuit 27 provides the passage to the first output of the proposed direction finder voltage of the resulting PX from the output of the comparator 26 at the time of the enable voltage from the control voltage generator 28, which determines the position of the elevation bearing.

The use of amplitude information about the IRI bearing in the azimuthal plane is realized when the mechanical (for example, manual) scanning by the antenna system in this plane is slow (in comparison with the electronic in the elevation plane and simultaneously with it). This is ensured by the elements and connections introduced into the known direction finder. The difference in scanning speed allows us to consider processes in the elevation and azimuth channels independently of each other. In addition, elevation search can be enabled after azimuthal.

The intersection signal from the output of the second subtractor 13 in addition to the phase signal contains amplitude information about the position of the IRI in the azimuthal plane. This information is enclosed in envelopes I and II (Fig. 2, a), angularly shifted in the azimuthal plane and intersecting on the equal-signal direction radiation patterns of antennas 1 and 2, modulating the received signals. The same applies to the combining signal received from the output of the sixth adder 14. This is explained by the fact that the intersection and combining signals are generated by blocks 9, 10, 11, 12, 13, and 14 from the signals of the receiving channels containing this modulation. Crossing and combining signals are constructed based on the selection and specific conversion of envelope modulation. As in the elevation channel, the extrema of the intersection and union signals are in the direction that is equal to the signal in azimuth and is rigidly connected to each other. They are used to obtain a narrow bearing reference area or the resulting bearing characteristic of the azimuth channel, which is formed at the output of the second comparator 22 after comparing the intersection signal amplified in the second amplifier 23 and the combining signal from the output of the sixth adder 14. Thus, the technical result of the invention is achieved.

At a qualitative level, the principle of the formation of direction-finding characteristics in the azimuthal and elevation channels is shown in FIG. 3, 4, 5. Here in FIG. 3 shows the characteristic signals of the elevation channel: the intersection signal ∩ at the output of the first subtractor 21; the intersection signal at the output of the first amplifier 25, which is a product of the gain C ₁ ≥1 and the intersection C ₁ ⋅∩; combining signal ∪ at the output of the third adder 24; direction finding characteristic of the elevation channel at the output of the first comparator 26 as a result of the inequality C ₁ ⋅∩≥∪.

In FIG. 4 shows the characteristic signals of the azimuth channel: the intersection signal ∩ at the output of the second subtractor 13; the intersection signal at the output of the second amplifier 23, as a result of the product of the gain C ₂ ≥1 and the intersection C ₂ ⋅∩; combining signal ∪ at the output of the sixth adder 14; direction finding characteristic of the azimuth channel at the output of the second comparator 22 as a result of the inequality C ₂ ⋅∩≥∪.

Finally, in FIG. 5 shows the resulting direction-finding characteristic of the proposed direction finder in three-dimensional representation, if, for example, the outputs of the channels are connected to the corresponding indicator.

Let us explain the principle of operation of the inventive device using analytical relationships that describe the physical processes in the elements of the circuit, and the results of simulation.

Let the IRI be at an angle ε relative to the normal to the aperture plane of the antenna in the azimuthal (horizontal) plane and at an angle in θ of the elevation (vertical) plane. The antenna system forms in the horizontal plane two radiation patterns (BPS) due to the displacement of the irradiators from the focus F ₁ (ε) and F ₂ (ε) of FIG. 6. The type of antenna used as an example in the proposed direction finder allows, with sufficient accuracy for assessments, to accept in the horizontal plane the widespread bell approximation of the DND in the form of:

where ε _0,5р is the _bottom width at half power, and ε is the current angle.

Charts intersect in the same direction. The offset angle of each of the diagrams relative to the RSN is determined by the ratio

where K _cm is the displacement coefficient, the values of which are selected from the condition of the intersection of the diagrams at a given level.

In the theory and practice of direction finding, the level of intersection is taken, as a rule, no more than 0.5 to ensure the greatest steepness of the HR. However, this leads to significant energy losses.

The use of intersection and union procedures for the formation of HRP, as will be shown below, allows you to choose the level of intersection of the diagrams 0.7-0.9 from the maximum. Then for the bell approximation of the DND, this will correspond to K _cm = 0.19-0.28.

It should also be noted that in the horizontal plane it is necessary to have a narrower DND, since the amplitude information about the position of the IRI is extracted.

, а рефлектором будет являться соответствующая область зеркала с расстоянием до вибратора ~λ/4. Здесь λ - длина волны. Диаграмма направленности такой системы приближенно может быть представлена следующей зависимостью:In the vertical plane, bearing information is extracted from the phase differences of the signals received by each channel. Therefore, the amplitude diagram should not significantly affect the processing process, as a result of which it should be wide enough in the scanning sector. This requirement is ensured by the lack of focusing of the field in the aperture in the vertical plane and the formation of the field of each of the antennas by a system of two elements, one of which is active, the other is passive. Therefore, it seems quite acceptable for the vertical plane to take an approximation of each of the antennas in the form of an active weakly directed element, for example, a symmetric vibrator of length

, and the reflector will be the corresponding region of the mirror with a distance to the vibrator ~ λ / 4. Here λ is the wavelength. The radiation pattern of such a system can be approximately represented by the following relationship:

where α is the current value of the angle in the vertical plane; A, B, C - some constants determined by the parameters of the vibrator-reflector system adopted during modeling. An exemplary view of the bottom in the vertical plane corresponding to (3) is shown in FIG. 7. The radiation patterns of channels I and II in the vertical plane coincide, that is, F ₁ (α) = F ₂ (α) = F (α).

Since the analysis of signals in the azimuthal and elevation plane is carried out independently, the spatial diagram of the channel can be represented as F (ε, α) = F (ε) ⋅F (α).

The signals received by the antenna from the IRI at the output of irradiators 1 and 2 can be represented as

where ω ₀ is the angular frequency, ϕ ₁ , ϕ ₂ are the initial phases of the signals at the outputs of the irradiators; U _m is the amplitude of the signals; σ ₁ (t), σ ₂ (t) are the intrinsic noise of the channels, counted to the inputs, we assume uncorrelated and approximately the same σ ₁ (t) ~ σ ₂ (t) ~ σ (t).

Since the signals come from a single source, their initial phases in the elevation plane differ by a shift Δϕ due to the difference in the wave path, i.e.

Here d is the spacing of the irradiators in the elevation plane; θ is the angle between the direction to the IRI in the vertical plane and the normal to the aperture plane.

Next, the signals (4) and (5) are subjected to frequency conversion and amplification at an intermediate frequency in the receiving paths 3 and 4 and are fed to uncontrolled phase shifters 5 and 6 from the receiving path 3 and to the controlled phase shifter 7 from the receiving path 4.

At the output of the phase shifters, we will accordingly have:

Here ω _CR - the angular intermediate frequency; K ₁ , K ₂ - gain of the receiving paths; ± β — fixed phase shifts in phase shifters 5 and 6; α (t) is a relatively slowly varying component of the phase of the reference voltage, providing a single or periodic scan of the equal-signal direction in the elevation plane within a certain sector.

In the general case, α (t) = Ru (t), where R is the proportionality coefficient, and u (t) is the law of variation of the voltage, for example, sawtooth, of the control voltage generator 28.

.Simulation used to verify the operability of the proposed direction finder, conducted for the frequency of the IRI signal ƒ = 2.4⋅10 ⁶ Hz. A harmonic unmodulated signal was adopted as a signal, and a random process with a normal distribution of instantaneous values of zero mean and a given dispersion was adopted as additive noise. The sampling frequency is selected from the condition of minimizing the error in the representation of the signal and amounted to

.

The type of signals at the output of the phase shifters is shown in FIG. 8 in amplitude-time coordinates. Here u ₅ , u ₆ , u ₇ are the signals at the output of 5, 6, 7 phase shifters, respectively, for fixed angles and phase shifts. In the notation of the voltage diagrams at the outputs of the blocks shown in FIG. 8-14, the subscript corresponds to the block number of the direction finder structural diagram.

Next, the signal (7) is supplied to one input of the phase detector 8. At its other input, a signal (9) is output from the output of the controlled phase shifter 7 as a reference.

The signal (8) is simultaneously fed to the first direct inputs of the fourth 9 and fifth 10 adders. The signal (9), in addition to the second input of the phase detector 8, is simultaneously fed to the second direct input of the fourth adder 9 and to the second inverse input of the fifth adder 10.

To simplify the analysis, we set U _m >> σ (t), U _m = 1, K ₁ = K ₂ = K = 1. It can be shown that at the outputs of adders 9 and 10 we will accordingly have signals in the form

Where

Further, the signals (10) and (11) are subjected to the capture of the module in the third 11 and fourth 12 device calculation module, respectively.

подается одновременно на вход уменьшаемого второго вычитающего устройства 13 и на первый прямой вход шестого сумматора 14. Сигнал

с выхода вычислителя модуля 12 одновременно поступает на вход вычитаемого второго вычитающего устройства 13 и на второй прямой вход шестого сумматора 14.From the output of the calculator module 11 signal

fed simultaneously to the input of the reduced second subtracting device 13 and to the first direct input of the sixth adder 14. The signal

from the output of the calculator module 12 simultaneously enters the input of the subtracted second subtractor 13 and the second direct input of the sixth adder 14.

Blocks 9, 10, 11, 12, 13 are a functional node that implements the intersection operation on the input signals u ₆ (t) and u ₇ (t), and blocks 9, 10, 11, 12, 14 are a node that implements the union operation . Therefore, at the output of blocks 13 and 14, respectively, we will have

After the low-pass filters, excluding the high-frequency component, the intersection and combining signals, up to a constant factor, take the form

It can be shown, based on the principle of operation of a balanced phase detector with comparable amplitudes of the input signals, that the output signal of the phase detector 8 after excluding the high-frequency component has the form:

Expressions (15), (16) and (17) describe the physical signals at the output of the corresponding blocks and contain all the amplitude and phase information sufficient for the formation of the HRP, detection and determination of bearings in the azimuth and elevation plane as a result of subsequent processing.

The signals of intersection (15) and the first association (16) are used in the azimuth channel by analyzing only the amplitude parameters.

On the contrary, in the elevation channel only phase parameters of signals (15) and (17) are evaluated.

At the same time, from the analysis of the obtained expressions it follows that in each case there is a mutual influence of the parameters, which affects the quality indicators of detection and measurement. However, based on the operating principle of the direction finder, the formation of the HRP in this plane can be carried out independently provided that the IRI is located on the RSN in the orthogonal plane, which simplifies the analysis without significantly affecting the results. The same is confirmed by the simulation results.

With this in mind, we consider the process of determining the bearing in the azimuthal plane using the results of modeling the operation of the direction finder circuit.

, так как антенна в вертикальной плоскости слабонаправленная. Кроме того, поскольку ДНА в горизонтальной плоскости одинаковы, но смещены на угол ε_см относительно РСН, представим их в виде F₁(ε)=F(ε_см+ε), F₂(ε)=F(ε_см-ε).We simplify first the expressions (15) and (16) by accepting

, since the antenna in the vertical plane is slightly directional. Furthermore, since the same beam in the horizontal plane, but offset by an angle ε with respect _cm PCH, present them in the form of F ₁ (ε) = F (ε + ε _cm), F ₂ (ε) = F _(see ε -ε) .

We also assume that bearing estimation in azimuth occurs at the moment α (t) = - (Δϕ + β). This condition is realized automatically during the restructuring of the phase shifter 7.

Given these assumptions, expressions (15) and (16) are transformed to

The signals at the outputs of the azimuth channel blocks are shown in FIG. 9, 10 in coordinates the amplitude (relative units) is the azimuthal angle (degrees). Here in FIG. 9 shows: U ₁₁ — envelope module of the total signal at the output of block 11; U ₁₂ is the envelope module of the difference signal at the output of block 12. In FIG. 10 are shown: U ₁₃ —the envelope of the intersection signal at the output of the second subtractor 13; U ₁₄ is the envelope of the combining signal at the output of the sixth adder 14. The RSN value in azimuth is taken equal to ε = 0 °.

For further analysis, we expand the function F (ε _cm ± ε) in a Taylor series in the vicinity of the RSN and discard terms whose order of smallness is two or more; as a result, we obtain

в окрестности РСН, так как именно в этой узкой области значений ε формируется ПХ и справедливо принятое разложение.where F '(ε _cm ) is the first derivative of the normalized DND at the PCN point. Here, the analysis interval is selected within

in the vicinity of RSN, since it is in this narrow range of ε that the HRP and the correctly accepted decomposition are formed.

After the signal U ₁₃ (ε) passes through the second amplifier 23 with a threshold gain of C ₂ ≥1, we obtain U ₂₃ (ε) = 2⋅С ₂ ⋅U ₁₃ (ε).

Finally, in comparator 22, signals (20) and (21) (intersection and union) are compared with the moment of their equality being recorded U ₂₃ (ε) ≥U ₁₄ (ε).

The output signal of the comparator 22 and will be the resulting PX direction finder in the azimuthal plane.

It can be shown that the width of the HRP is determined by the relation

As follows from the obtained expression (22), the width of the HRP depends on the shape of the radiation pattern and the value of the threshold coefficient C ₂ . Moreover, the dependence on the latter is much stronger.

The choice of the operating value of the threshold coefficient C ₂ is determined by the level of intrinsic noise, the required indicators of the quality of detection and measurement of bearings will be discussed later.

When C _2> 1 extrema crossing signals U ₁₃ (ε) and ₁₄ U of association (ε) overlap. The degree of overlap is determined by the value of the threshold coefficient, and the resulting PX U ₂₂ (ε) at the output of the comparator 22 (second output of the phase direction finder) has a rectangular shape with a triangular top (Fig. 11). Here in FIG. 11 shows the coordinates obtained as a result of modeling: relative amplitude — azimuthal angle ε (deg) signals: U ₁₀ — envelope at the output of the fifth adder 10; U ₁₃ - at the output of the second subtracting device 13; U ₁₄ - at the output of the sixth adder 14; U ₂₃ - at the output of the second amplifier 23; U ₂₂ - at the output of the second comparator 22. Note that the envelope of the differential signal U ₁₀ can be used to determine the direction of motion of the direction finder antenna in azimuth to RSN.

It can be shown that the estimation of the accuracy of bearing measurement in the azimuthal plane in the general case can be carried out by determining the standard error in accordance with the ratio

where K _f - coefficient determined by the form of HRP, K _f = 0.5 ... 1.2 - for most practical cases; q is the signal-to-noise ratio in power.

The average value of the width of the HRP in the azimuthal plane for the adopted approximation of the DND is determined by the ratio:

Based on the analysis of (23) and (24), it can be concluded that obtaining the azimuth channel HRP based on the intersection and union procedures increases the direction finding accuracy. The dependence of the root mean square error of direction finding will be given below.

To form the bearing reference area in the elevation plane, phase information is used, consisting of signals (15) and (17).

We transform these expressions, taking as a condition that the formation of the PX by elevation is at the moment when the antenna system is in the same direction in azimuth. Physically, this is realized, for example, by pre-setting using a difference signal in the azimuth channel.

Given this condition, we can assume that F ₁ (ε) = F ₂ (ε) = F (ε). In addition, the condition F (α) ≈const remains valid within the scanning sector by elevation. Then expressions (15) and (17 are converted to the form

Recall that the angular displacement θ (bearing) in elevation is contained in the phase difference Δϕ (6) between the signals supplied to the antenna inputs, α (t) is the variable phase component due to the reference voltage, ± β are the fixed phase shifts in the phase shifters 5 and 6.

Signals (26) and (25) from the output of the phase detector 8 and the second subtractor 13 are fed to the input of the first 15 and second 16 limiters from the bottom at zero level, respectively. The form of these signals U ₈ and U ₁₃ according to the simulation results is shown in FIG. 12 in coordinates, the relative amplitude is the elevation angle α in degrees. The given dependences essentially represent direction-finding characteristics of subchannels in the elevation plane, shifted relative to each other by 2β. Limiters eliminate the negative branches of the dependencies U ₈ and U ₁₃ shown in FIG. 12, providing unambiguous assessment of the bearing.

To form the resulting PX in the elevation plane, the signals from the outputs of the limiters undergo crossing and combining procedures.

To this end, from the output of the limiter 15, the signal U ₁₅ (α) is simultaneously supplied to the first direct inputs of the first 17 and second 18 adders, and the signal U ₁₆ (α) from the output of the limiter 16 to the second direct and second inverse inputs of the first and second adders, respectively. As a result, the output of block 17 and block 18 will be the total and difference voltages, respectively: U ₁₇ (α) = U ₁₅ (α) + U ₁₆ (α) and U ₁₈ (α) = U ₁₅ (α) -U ₁₆ (α )

.These voltages are subjected to the operation of taking the module in the first 19 and second 20 device calculation module:

.

The voltages at the outputs of these blocks are shown in FIG. 13 in coordinates, the relative amplitude is the elevation angle α in degrees. These stresses were obtained as a result of simulation taking into account the effect of noise, the IRI is located at an angle θ = 10 degrees in the elevation plane.

Then, the sum module is simultaneously fed to the input of the reduced first subtractor 21 and the first direct input of the third adder 24. The difference module is simultaneously fed to the input of the subtracted first subtractor 21 and the second direct input of the third adder 24. At the output of the first subtractor, a voltage corresponding to the intersection of the input signals U ₈ (α) and U ₁₃ (α)

The output of the third adder is the voltage corresponding to the combination of these signals:

Signals of intersection and association are rigidly interconnected by a common point at which U ₈ (α) and U ₁₃ (α) intersect on the RSN. The position of this point in amplitude is ensured at a high energy level, which is determined by the choice of β values in phase shifters 5 and 6 and is highly independent of the angular displacement of the direction finding object with a weakly directed antenna in the vertical plane. The position of the RSN in the angle is determined by the bearing on the IRI.

Next, the intersection signal U ₂₁ (α) is amplified in the first amplifier 25 with a threshold coefficient C ₁ ≥1 and supplied to the first input of the first comparator 26, where it is compared with the combining signal U ₂₄ (α) supplied to the second input of the comparator. As a result, when the inequality С ₁ ⋅U ₂₁ ≥U ₂₄ is fulfilled, a narrow bearing reference region is formed that is symmetrical with respect to the RSN, which is the resulting HRP in the elevation plane.

The intersection signal at the output of the amplifier U ₂₅ combining U ₂₄ and the resulting PX in the elevation plane at the output of the comparator U ₂₆ are shown in FIG. 14 in coordinates, the relative amplitude is the elevation angle α in degrees.

The position of the RSN at each given point in time is determined by the voltage of the control voltage generator 28 u ₂₈ (t) supplied to the second input of the controlled phase shifter 7 and rearranging the phase.

At the same time, the voltage of the generator 28 is supplied to the second input of the key circuit 27, the first input of which receives the resolving signal of the direction-finding characteristic from the comparator 26. As a result, the signal corresponding to the elevation bearing at the time of formation of the HRF passes from the output of the key circuit to the first output of the direction finder.

The angular position of the IRI can be found at the time of reference from the relation

Based on the analysis of the behavior of the stresses U ₂₄ , U _25, and U ₂₆ in the area of RSN, it can be shown that the width of the direction-finding characteristic in the elevation plane will be determined by the relation

and the range of acceptable values of β lies within 10 ° ≤β≤45 °.

The scanning sector in the elevation plane ΔФ is determined from the condition of unambiguity of the bearing reading, since the voltages U ₈ (α) and U ₁₃ (α) are periodic in nature, as well as from the condition of an acceptable decrease in signal amplitude due to the directional properties of the antenna in the elevation plane. The condition of uniqueness is crucial in this case. Modeling and analysis show that the value of the scanning sector ΔФ for d / λ = 0.8 lies within -36 ° ≤ΔF≤36 ° of FIG. 15. Here in the polar coordinate system are presented: the normalized DND in the elevation plane for the adopted approximation F (α), the actual bearing at the IRI position θ = 36 ° (in fact, this is the resulting HRP in the elevation plane at C ₁ = 1.1) and the false bearing .

In addition, the value of the scanning sector ΔΦ depends on the relative antenna spacing d / λ. This relationship is shown in FIG. 16. The analysis shows that within 1≤d / λ≤0.6 it is inversely proportional and linear.

For d / λ≤0.5, a false bearing does not occur in the front hemisphere, however, at ΔФ> 60 °, the shape of the intersection signal is distorted and the PX expands. Therefore, for d / λ≤0.5, the factor limiting the scanning sector is the allowable extension of the direction-finding characteristic.

It should be noted that FIG. 15. Also illustrates the effect of the directional properties of the antenna on the magnitude of the scanning sector by elevation, which manifests itself in a decrease in the amplitude of the received signal. In this case, this influence is insignificant, since at the edge of the sector F (α) = 0.9F _max .

, где R,R₁ - коэффициенты пропорциональности. Значение R определяется типом управляемого фазовращателя. Коэффициент R₁ может быть определен как скорость сканирования, то есть

, где ΔT - период сканирования.To implement scanning along the elevation angle, the law of change in the control voltage of the generator 28 can be selected corresponding to the law of changing the difference in the wave path at the outputs of the irradiators, i.e.

where R, R ₁ - proportionality coefficients. The value of R is determined by the type of controlled phase shifter. The coefficient R ₁ can be defined as the scanning speed, i.e.

where ΔT is the scanning period.

The choice of the scanning period ΔT is related to the nature of the radio monitoring problem to be solved, the type of phase shifter, the type of signal, etc.

The simulation results, thus, confirm the operability of the proposed phase direction finder.

, вероятность ложной тревоги

, среднее значение ширины пеленга

, среднеквадратическая ошибка (СКО) измерения пеленга σ_ε(σ_α), определяемые путем имитационного моделирования.To assess the effectiveness of the proposed scheme, statistical quality indicators of the azimuthal and elevation channels are determined. For quality indicators taken: the probability of correct detection of the bearing

probability of false alarm

, average bearing width

, standard error (RMS) of bearing measurement, _{ε ε} (σ _α ), determined by simulation.

понимается вероятность принятия решения об обнаружении пеленга на фиксированном направлении и при заданном значении порогового коэффициента С₁ и С₂ в зависимости от отношения сигнал/шум на входе приемных каналов. Оценка указанной вероятности находилась как отношение числа положительных исходов n, определяемого счетчиком на выходе компараторов каждого канала, к общему числу опытов N. Среднее значение ширины пеленга определялось путем усреднения результатов М измерений при фиксированном значении

и значении порогового коэффициента. Оценка среднеквадратической погрешности проводилась аналогично при фиксированном значении порогового коэффициента в зависимости от отношения сигнал/шум на входе.Moreover, under

the probability of making a decision to detect a bearing in a fixed direction and for a given value of the threshold coefficient C ₁ and C ₂ depending on the signal-to-noise ratio at the input of the receiving channels is understood. An estimate of this probability was found as the ratio of the number of positive outcomes n, determined by the counter at the output of the comparators of each channel, to the total number of experiments N. The average value of the bearing width was determined by averaging the results of M measurements at a fixed value

and the value of the threshold coefficient. The root-mean-square error was estimated in a similar way with a fixed value of the threshold coefficient depending on the signal-to-noise ratio at the input.

при фиксированных значениях порогового коэффициента С₂. Здесь U_c - амплитуда сигнала, σ_ш - среднеквадратическое значение напряжения шума. Цифры, обозначающие кривые, являются значениями порогового коэффициента, для которого эти зависимости получены. Чем ниже отношение сигнал/шум, тем больше значение порогового коэффициента усиления требуется для реализации решающего правила обнаружения пеленга. Обращает на себя внимание высокая крутизна всех без исключения кривых, что свидетельствует о лавинообразном характере выполнения решающего правила обнаружения азимутального пеленга и подтверждает исключение ложного срабатывания в условиях действия только шума и вне пеленгационной характеристики.The evaluation results for the azimuth channel are shown in FIG. 17 ... 20. In FIG. 17 shows the dependences of the probability of correct detection of the bearing on the signal-to-noise ratio at the input in the form

for fixed values of threshold factor C _2. Here, U _c is the signal amplitude, σ _w is the rms value of the noise voltage. The numbers denoting the curves are the values of the threshold coefficient for which these dependencies are obtained. The lower the signal-to-noise ratio, the higher the threshold gain value is required to implement the decisive bearing detection rule. The steepness of all the curves, without exception, is noteworthy, which indicates the avalanche-like nature of the fulfillment of the decisive rule for the detection of the azimuth bearing and confirms the elimination of false triggering under conditions of only noise and outside the direction-finding characteristic.

следует заметить, что особенностью и достоинством предлагаемой схемы является исключение ложного срабатывания азимутального канала при действии только шума. Это следует из самого алгоритма реализации процедур пересечения и объединения над принятыми сигналами. Действительно, выбор большего из сопоставляемых сигналов при реализации процедуры объединения на выходе блока 14 и меньшего при реализации пересечения на выходе блока 13 приводит к тому, что в случае действия только шума на входах при любом его значении, как и значении порогового коэффициента усиления сигнала пересечения из области его рабочих значений

(блок 23), срабатывание компаратора 22 исключается.Relatively

It should be noted that the peculiarity and advantage of the proposed scheme is the elimination of false operation of the azimuth channel under the influence of noise only. This follows from the algorithm for implementing the intersection and union procedures over the received signals. Indeed, the choice of the larger of the compared signals when implementing the combining procedure at the output of block 14 and smaller when realizing the intersection at the output of block 13 leads to the fact that in the case of only noise at the inputs at any value, as well as the value of the threshold gain of the crossing signal from areas of its working values

(block 23), the operation of the comparator 22 is excluded.

In the presence of a useful signal at the inputs of the receivers, at the outputs of blocks 13 and 14, the signal component begins to appear, and when the signal-to-noise ratio is exceeded, it becomes possible to form a direction-finding characteristic in accordance with the decision rule C ₂ ⋅U _∩ (ε) ≥U _∪ (ε) , which depends on the ratio of signal and noise components and the value of the threshold coefficient C ₂ . This is illustrated by the diagrams in FIG. 18. Here are the output signals of blocks 13 - the signal of intersection U _∩ (ε)) and 14 - the signal of combining U _∪ (ε) for the input signal-to-noise ratio of ten.

сигнала пересечения U_∩(ε) в блоке 23, чтобы обеспечить формирование ПХ. Вне пеленгационной характеристики этого усиления будет явно недостаточно (требуется С₂≥10), чем исключаются ложные пеленги.As follows from FIG. 18, extremes U _∩ (ε) and U _∪ (ε) appear well in the direction of the bearing differ significantly _{(U ∩ (ε) ≅0,47,} U ∪ (ε) ≅0,55). Therefore, a small gain is enough

the intersection signal U _∩ (ε) in block 23 to ensure the formation of the HRP. Outside the direction-finding characteristic of this gain, it will be clearly not enough (C ₂ ≥10 is required), which eliminates false bearings.

The choice of operating values of C ₂ depends on the input signal-to-noise ratio and is associated with the analysis of the probability of the correct detection of the bearing, the required width of the HR and the standard error of the determination of the bearing.

в градусах от значения порогового коэффициента С₂ в виде

при фиксированном отношении сигнал/шум, обеспечивающем вероятность правильного обнаружения пеленга

, приведена на фиг. 19. Как видно из фиг. 19, эта зависимость почти линейная. Анализ показывает, что при заданном виде аппроксимации ДНА ширина ПХ практически не зависит от ширины ДНА, а полностью определяется значением порогового коэффициента С₂ при заданной вероятности правильного обнаружения. Это объясняется тем, что в формировании ПХ участвуют экстремумы функций U_∩(ε) и U_∪(ε), сближающиеся своей обостренной частью на РСН, фиг. 18.The dependence of the average value of the direction finding characteristic

in degrees from the value of the threshold coefficient C ₂ in the form

at a fixed signal-to-noise ratio, providing the probability of correct detection of the bearing

shown in FIG. 19. As can be seen from FIG. 19, this dependence is almost linear. The analysis shows that for a given type of approximation of the DND, the width of the HRP practically does not depend on the width of the DND, but is completely determined by the value of the threshold coefficient C ₂ at a given probability of correct detection. This is explained by the fact that extrema of the functions U _∩ (ε) and U _∪ (ε), which approach each other with their sharpened part on the RSH, participate in the formation of HRP, Fig. eighteen.

при фиксированном значении порогового коэффициента С₂ представлены на фиг. 20. Цифры, обозначающие кривые представляют собой значения порогового коэффициента С₂, для которых эти зависимости получены. Поведение этих зависимостей обратно поведению вероятностей правильного обнаружения. Увеличение отношения сигнал/шум и выбор порогового коэффициента позволяет обеспечить весьма малые значения СКО измерения азимутального пеленга, что и подтверждает соотношение (23).Dependences of the standard error of the determination of the azimuth bearing σ _ε in degrees on the signal-to-noise ratio in the form

at a fixed value of the threshold coefficient C _{2, are} presented in FIG. 20. The numbers indicating the curves are the values of the threshold coefficient C ₂ for which these dependencies are obtained. The behavior of these dependencies is inversely the behavior of the probabilities of correct detection. An increase in the signal-to-noise ratio and the choice of the threshold coefficient make it possible to ensure very small RMS values for measuring the azimuth bearing, which confirms relation (23).

The evaluation results for the elevation channel are shown in FIG. 21-24. The carbon channel, in contrast to the azimuth channel, uses phase information about the angular position of the radio source. A feature of the proposed scheme is the lack of a threshold device, as such. The role of the threshold device is played by the threshold gain factor C _{1 of} block 25. The crossing signal amplified by this block is compared in the first comparator 26 with the combining signal, thereby realizing the decision rule C ₁ ⋅U _∩ (α) ≥U _∪ (α), therefore, the probability of false alarm can be investigated only depending on the standard deviation of the input noise, fixing the value of the threshold coefficient C ₁ .

.In FIG. Figure 21 shows the averaged dependence of the probability of false alarm on the gain in the form

.

The given dependence makes it possible to make a reasonable choice of the threshold coefficient in various conditions of an interference environment.

при фиксированных значениях С₁ представлены на фиг. 22. Цифры, обозначающие кривые, представляют собой значения пороговых коэффициентов усиления сигнала пересечения, для которых получены указанные зависимости. Чем ниже отношение сигнал/шум, тем большее значение порогового коэффициента усиления требуется для реализации решающего правила обнаружения пеленга.Dependences of the probability of correct detection of the elevation bearing on the signal to noise ratio at the input in the form

at fixed values of C _{1 are} shown in FIG. 22. The numbers denoting the curves represent the values of the threshold gains of the intersection signal, for which these dependencies are obtained. The lower the signal-to-noise ratio, the higher the threshold gain value is required to implement the decisive bearing detection rule.

при фиксированном отношении сигнал/шум, обеспечивающем вероятность правильного обнаружения пеленга

, приведена на фиг. 23. Анализ показывает, что ширина ПХ в угломестной плоскости практически не зависит от вида аппроксимации ДНА и ширины ДНА, а полностью определяется значением порогового коэффициента С₁ при заданной вероятности правильного обнаружения.The dependence of the average value of the width of the direction-finding characteristic of the threshold coefficient C ₁ in the form

at a fixed signal-to-noise ratio, providing the probability of correct detection of the bearing

shown in FIG. 23. Analysis indicates that HRP width in the elevation plane is practically independent of the type of approximation beam width and beam, and is fully determined by the threshold value coefficient C ₁ for a given probability of correct detection.

This is explained by the fact that the phase characteristics are involved not in amplitude but in phase characteristics, as well as in the way the intersection and union functions are formed by counter extrema.

при фиксированном значении порогового коэффициента С₁ представлены на фиг. 24. Цифры, обозначающие кривые, являются значениями порогового коэффициента, для которого эти зависимости получены. Поведение кривых свидетельствует о том, что СКО определения пеленга практически мало зависит от порогового коэффициента, а, значит и ширины ПХ и полностью определяется отношением сигнал/шум.Dependences of the standard error of determining the elevation bearing on the signal-to-noise ratio in the form

at a fixed value of the threshold coefficient C _{1, are} presented in FIG. 24. The numbers denoting the curves are the values of the threshold coefficient for which these dependencies are obtained. The behavior of the curves indicates that the standard deviation of the determination of the bearing practically does not depend on the threshold coefficient, and, hence, the width of the HRP and is completely determined by the signal-to-noise ratio.

.The obtained dependences allow making a reasonable choice of the required value of the threshold coefficients C ₁ and C ₂ for the first 25 and second 23 amplifiers and determine the range of possible values of these coefficients in the interval

.

A comparison of the quality indicators obtained based on modeling of elevation and azimuth channels of the proposed direction finder with the known ones allows us to conclude that the proposed direction finder is effective.

The simulation results confirm the operability, feasibility and effectiveness of the proposed direction finder and, thus, the achievement of the technical result of the invention due to the introduced elements and relationships that implement the principle of detection-measurement, based on the joint use of functional procedures of intersection and association.

The possibility of practical implementation also follows from the fact that the circuit can be built on typical, well-known and technologically advanced elements. For example:

antennas 1, 2 - can be selected of various types, depending on the frequency range and the tactical and technical requirements for the direction finder. In particular, antennas of the vibrator-reflector type, or mirror antennas, as in the considered example, or horn antennas of the type described in [G.Z. Eisenberg, V.G. Yampolsky, O.N. Tereshin. Antennas VHF. CH 1. - M. "Communication", 1977. - 384 p.] P. 254, fig. 16.2;

receiving paths 3, 4 - can be built according to the standard scheme of radio communication or radar receivers with an output at an intermediate frequency as described in [M.K. Belkin, V.T. Belinsky, Yu.L. Mazor et al. A guide to the educational design of receiving amplifying devices. K., "Higher School", 1988. - 472 p.] P. 405, fig. 14.4;

phase shifters 5, 6 with a fixed phase shifter 7 with a controlled phase shift - can be performed depending on the value of the operating frequency on the delay lines, RC circuits, serial or parallel oscillatory circuits. At frequencies up to 30–40 MHz, phase shifters based on RC circuits as described in [A.P. Golubkov, A.D. Dalmatov, A.P. Lukoshkin et al. Design of radar receiving devices. Ed. M.A. Sokolova. - M., Higher. school., 1984, - 335 p.] p. 122-126. At lower frequencies - as a broadband phase shifter with stepless phase shift adjustment given in [P.M. Tereshchuk, K.M. Tereshchuk, S.A. Sedov. Semiconductor receiving and amplifying devices. Handbook of amateur radio. Kiev: Science, Dumka, 1987. - 800 p.] P. 701, fig. XI.II;

phase detector 8 - can be implemented as a balanced phase detector described in [M.K. Belkin, V.T. Belinsky, Yu.L. Mazor et al. A guide to the educational design of receiving amplifying devices. K., "Higher School", 1988. - 472 p.] P. 252, fig. 9.31, c;

the lower limiters at the zero level 10, 11 can be performed according to a simple diode detector circuit shown in [A.P. Golubkov, A.D. Dalmatov, A.P. Lukoshkin et al. Design of radar receiving devices. Ed. M.A. Sokolova. - M., Higher. school., 1984, - 335 p.] p. 140, fig. 5.12;

adders 9, 10, 14, 17, 18, 24 and subtractors 13 and 21 can be performed according to the usual scheme of amplifiers for two inputs or with direct and inverse inputs as described in [A.G. Alekseenko. The use of precision analog integrated circuits. - M., Radio and Communications, 1981, 354 pp.] P. 77, Figure 3.2;

device computing module 11, 12, 19, 20 can be implemented according to the scheme of a half-wave rectifier on operational amplifiers as described in [V.P. Bobrovsky, V.I. Kostenko, V.M. Mikhailenko et al. Handbook of circuitry for a radio amateur. Ed. V.P. Borovsky. - K: Tekhnika, 1989. - 480 p.] P. 241, fig. 12.4;

amplifiers 23 and 25 can be a conventional resistive-capacitive video amplifier with gain control in the collector circuit, given in [G.Z. Eisenberg, V.G. Yampolsky, O.N. Tereshin. Antennas VHF. CH 1. - M. "Communication", 1977. - 384 p.] P. 270, fig. 10.1;

comparators 22 and 26 can be implemented on an operational amplifier as an integral comparator described in [V.P. Bobrovsky, V.I. Kostenko, V.M. Mikhailenko et al. Handbook of circuitry for a radio amateur. Ed. V.P. Borovsky. - K: Tekhnika, 1989. - 480 p.] P. 251, fig. 12.22;

the key circuit 27 can be made in the form of a bipolar analog key according to the type given in [M.V. Halperin. Practical circuitry in industrial automation. - M.: Energoatomizdat, 1987. - 320 p.] P. 240, fig. 6.22;

a control voltage generator 28 for generating a control voltage of arbitrary shape the most universal is the synthesizer-based circuit given in [V.P. Bobrovsky, V.I. Kostenko, V.M. Mikhailenko et al. Handbook of circuitry for a radio amateur. Ed. V.P. Borovsky. - K: Tekhnika, 1989. - 480 p.] P. 296, fig. 10/14, g.

Analysis of the known technical solutions in the field of information transmission systems shows that the claimed invention, due to the essential features of the introduced elements and connections, which determined the way to achieve the technical result, does not follow for the specialist explicitly from the prior art in this subject area and meets the requirement of “inventive level. "

The applicant has not found an analogue characterized by features identical to all the essential features of the claimed invention. The definition of the prototype, as the closest in terms of the totality of the features of the analogue, allowed us to identify distinctive features that are significant in relation to the technical result in the claimed object, which allows us to consider the claimed invention to meet the criterion of "inventive novelty".

Claims (1)

A phase direction finder containing the first and second antennas spaced a certain distance, the first and second receiving paths connected by inputs to the first and second antennas, respectively, the first and second phase shifters having fixed and mutually opposite phase shifts and connected by inputs to the output of the first receiving path, a phase detector connected by one input to the output of the first phase shifter, and the second input through a third controlled phase shifter with the output of the second receiving path, the first and second limiters from the bottom at zero level, the first adder connected by the first direct input through the first limiter from the bottom at zero level with the output of the phase detector, and the second direct input with the output of the second limiter from the bottom at zero level, the second adder connected by the first direct input with the output of the first limiter from the bottom to zero level, and the second inverse input with the output of the second limiter from the bottom at zero level, the first and second module calculating devices, connected by the first input to the output of the first adder, the second input by the output the second adder, the first subtractor connected to the input of the module calculating unit being reduced with the output of the first device, and the module subtracting the input of the module calculating unit to the output of the second device, the third adder connected to the first and second direct inputs with the outputs of the first and second module calculation devices, respectively, the first amplifier connected by the input to the output of the first subtractor, the first comparator connected by the first input to the output of the first amplifier, and the second input to the output of the third adder, cl the hand circuit connected by the first input to the output of the first comparator, a control voltage generator connected by the output to the second inputs of the third controlled phase shifter and the key circuit at the same time, and the output of the key circuit is the first output of the direction finder, characterized in that a fourth adder is connected to it, connected by the first direct the input with the output of the second phase shifter, and the second direct input with the output of the third controlled phase shifter, the fifth adder connected by the first direct input with the output of the second phase shifter tel, and by the second inverse input with the output of the third controlled phase shifter, the second subtractor connected by the input of the module to be reduced through the third device to the output of the fourth adder, and the input of the module subtracted through the fourth device to the output of the fifth adder, the sixth adder connected by the first and second direct the inputs with the outputs of the third and fourth devices for calculating the module, respectively, the second amplifier connected to the input with the output of the second subtracting device and the input of the second of the lower limiter at the zero level at the same time, the second comparator connected by the first input to the output of the sixth adder, and the second input with the output of the second amplifier, the output of the second comparator is the second output of the phase direction finder.

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