RU2319170C1 - Digital multi-channel correlation-filtering receiving device with selection of moving targets - Google Patents

Abstract

FIELD: radiolocation systems which use probing signals with comber-type spectrum.

SUBSTANCE: claimed device contains n-bit analog-digital converter, digital selector of moving targets of compensation type, digital multiplexer and digital multi-channel device of correlation-filtration processing of signals received by Doppler-impulse radiolocation station, digital generator of weight function, which are all interconnected in appropriate manner, while digital selector of moving targets contains n-bit digital adder in direct channel, and in compensating channel - n-bit digital filter of low frequencies, which are also connected to each other and to appropriate instruments of claimed receiving device. As n-bit digital low frequency filter, a canonic recursive digital low frequency filter of first or second order is used.

EFFECT: increased stability of operation and expanded functional capabilities of device with insured interference protection of device.

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Description

The invention relates to a radio reception technique for processing quasi-continuous pulsed-Doppler signals and can be used in radar systems using sounding signals with a comb spectrum.

A multi-channel correlation-filter receiving device is known that contains a sequentially included compensator of spectral interference lines and a multi-channel correlation-filter processing device for received signals of a pulse-Doppler radar station, as well as a multi-channel synthesizer of a tunable equidistant heterodyne frequency grid and a weight function generator. The spectral interference line compensator contains a direct-connected algebraic adder, a broadband filter, a modulator and a first amplifier in the forward channel, and a heterodyne-tunable narrow-band comb filter, a second amplifier and a phase shifter, and each channel of a multi-channel heterodyne-tunable narrow-band combiner is sequentially connected in a direct channel the filter contains a series-connected first analog multiplier, a narrow-band filter and a second analog ne a multiplier, the first inputs of the first analog multipliers of all channels being combined and connected to the first input of the algebraic adder, which is the input of the device and connected to the output of the signal source of the pulse-Doppler radar, the outputs of the second analog multipliers of all channels of the heterodyne tunable narrow-band comb filter are combined and connected with the input of the second amplifier, and the output of the phase shifter with the second input of the algebraic adder, the second inputs of the first and second analog each channel's multipliers are combined and connected to the corresponding output of the multichannel synthesizer tunable equidistant frequency grid; the output of the generator of the weight function is connected to the second input of the modulator; A multichannel device for correlation and filter processing of received pulsed-Doppler radar signals contains in each channel a series-connected gating cascade in range and a set of band-pass filters with adjacent bands in the Doppler frequency range [1].

In the described device (prototype of the invention), the passive interference spectrum is suppressed, the input signal is time-gated and the Doppler frequency is filtered in each gate. In it, the output multi-channel device for correlation-filter processing of received signals (SLS) can be implemented according to well-known rules in both analog [2] and digital [3] forms with preliminary digitization of the received signal using analog-to-digital converters. In the analog implementation of a set of correlation filter channels (CFC) of the SLR, the main suppression of the spectral lines of passive interference is performed using an analog compensator for the spectral interference lines and, additionally, using a single-band analog filter that is turned on at the output of each gating stage before a set of band-pass Doppler filters. In the digital implementation of the KFK UOS kit, the main suppression of the spectral lines of passive interference is also performed using an analog compensator of the spectral lines of interference, and gating, additional suppression of the spectral lines of passive interference and Doppler filtering of the received signals are digitally performed.

The reason that impedes the obtaining of the technical result indicated below when using the well-known multichannel correlation filter receiver is the cumbersomeness of the hardware implementation of the analog compensator of the spectral interference lines, since it requires multichannel structure and the corresponding multichannel synthesizer tunable equidistant heterodyne frequency grid, as well as insufficient stability due to the influence of parametric and climatic factors in analog hardware. In addition, the prototype does not have universality, since the number of suppressed spectral lines of interference is always finite, and the notch band of each channel of the compensator for spectral lines of interference cannot be rebuilt during operation, which limits the functionality of the device when it is used.

The invention consists in the following.

The objective of the invention is to simplify the compensator for spectral interference lines. The technical result is to increase the stability of the compensator for spectral interference lines and expand the functionality of the device with ensuring its noise immunity by simple and reliable means.

The specified technical result is achieved by the fact that in the well-known multi-channel correlation-filter receiving device containing a sequentially connected modulator and multi-channel device for correlation-filter processing of received signals of a pulse-Doppler radar, as well as a weight function generator, the output of which is connected to the second input of the modulator, according to the invention introduced an n-bit analog-to-digital converter (ADC) and a digital moving target selector (SDC) of the compensation type, containing in in the pit channel, an n-bit digital adder, and in the compensating channel, an n-bit digital low-pass filter (LPF), while the modulator is made in the form of an n-bit digital multiplier, the weight function generator is in the form of a digital generator, a multi-channel correlation filter device processing the received signals of the pulse-Doppler radar is performed by n-bit digital; the input of the n-bit ADC is the input of the device and is connected to the output of the source of the received signals of the pulse-Doppler radar, and its output buses are connected to the first bit inputs of the n-bit digital adder and the input buses of the n-bit digital low-pass filter, the output buses of which are connected to the second bit the inputs of the n-bit digital adder, the output of which is connected to the first bit inputs of the n-bit digital multiplier; the control input of the n-bit ADC and the control input of the n-bit digital low-pass filter are combined and connected to the source of sampling pulses.

As an n-bit digital low-pass filter, a canonical recursive digital low-pass filter of the first or second order is used.

Causal relationships of the features of the invention with the technical result are as follows. In the claimed device, instead of an analogue compensator for spectral interference lines, a digital compensation type SDS is included, which has an infinite notch comb characteristic due to an n-bit digital low-pass filter. As will be shown below, in the steady-state mode of operation of the device in the digital SDS, the amplitude of the interference pulses is suppressed, which are allocated by the n-bit digital low-pass filter, and the spectral lines and the amplitude of the pulses of the Doppler signals remain unchanged, because they are not allocated by an n-bit digital low-pass filter and are not compensated. The digital SDC suppresses all spectral lines of passive interference, therefore, at the output of the digital adder, no bandpass filtering is required in comparison with the prototype. In addition, the introduction of a digital compensation SDS eliminates the influence of parametric and climatic factors on the operation of the device, since they are determined by the weight coefficients of the n-bit digital low-pass filter. This stabilizes all the characteristics of the device and makes it easy to change its parameters, which expands its functionality. Such a digital SDC allows you to work directly with radio signals at a higher intermediate frequency than the prototype, since due to frequency conversion in an n-bit ADC, sampling at the output of a digital SDC results in a significant reduction in the intermediate frequency without additional equipment, which simplifies further signal processing. At the same time, improving the quality of compensation for interference pulses is achieved both due to the identity of the shape of digital pulses in the forward and compensating channels, and due to the accuracy and stability of the n-bit digital adder. Since the level of the side lobes of the interference pulse spectrum directly depends on the amplitude of the pulses, due to their preliminary and substantial suppression using the n-bit digital SDS, the weighted processing of signals is simplified, the noise immunity of the device is increased, and its hardware implementation can be carried out by simple and reliable means.

The invention is illustrated by drawings, in which: FIG. 1 is a functional diagram of a digital multi-channel correlation-filter receiver with SDS; figure 2, figure 3 - respectively, the temporal and spectral characteristics that explain the operation of the device.

A digital multi-channel correlation filter receiver with SDC (FIG. 1) contains a n-bit ADC 1, a digital SDC 2 of compensation type, an n-bit digital multiplier 3 and an n-bit digital multi-channel device for correlation filter processing of received radar signals 4 as well as a digital generator of the weight function 5, the output bus of which is connected to the second inputs of the n-bit digital multiplier 3. The digital SDC 2 contains in the forward channel an n-bit digital adder 6, and in compensation uyuschem channel - n-bit digital LPF 7. (Since all the circuit elements other than digital weighting function generator 5 are n-bit, in the following, the term «n-bit" is not used to reduce the text). The ADC input 1 is the input of the device and is connected to the output of the signal source of the pulse-Doppler radar, its output buses are connected to the first bit inputs of the digital adder 6 and to the bit inputs of the digital low-pass filter 7, the outputs of which are connected to the second bit inputs of the digital adder 6. Output buses digital adder 6 are the output of the digital SDC and are connected to the first bit inputs of the digital multiplier 3, the output buses of which are connected to the bit inputs of the digital multi-channel device correlation-filter processing of received signals of a pulse-Doppler radar 4. The control input of the ADC 1 and the control input of the digital low-pass filter 7 are combined and connected to a source of sampling pulses following with a repetition frequency f _DC (not shown in the diagram).

A digital multichannel device for correlation-filter processing of received radar 4 signals contains a set of N identical digital CPKs, each of which consists of a series-connected digital gating range in range and a set of M digital narrow-band filters with adjacent bands, implemented, for example, using the fast algorithm Fourier transform (FFT).

As a digital low-pass filter 7, a well-known canonical recursive digital low-pass filter of the first or second order can be used [4]. In particular, the second-order low-pass filter can be made in the form of series-connected first, second adders and a multiplier by a normalizing coefficient; serially connected first and second delay devices; the first, second, third and fourth multipliers by the weight coefficient; the output of the first adder is also connected to the input of the first delay device, the output of which is also connected to the inputs of the first and third multipliers by a weight factor, the outputs of which are connected to the second inputs of the second and first adders, the output of the second delay device is connected to the inputs of the second and fourth weight factor multipliers, the outputs of which are connected to the third inputs of the second and first adders, respectively; the first input of the first adder is the first input of the low-pass filter, its control input is connected to the synchronizer, and the output of the multiplier by the normalizing coefficient is the output of the second-order low-pass filter. Delay devices in the digital low-pass filter can be made in the form of multi-link structures of digital shift registers, while the digital low-pass filter 7 can generally be made, for example, in the form of a well-known second-order block diagram [5] with the number of shift register links m in each digital category, equal to the product of the sampling frequency f _DK by the pulse repetition period of a quasi-continuous signal T = 1 / F, where F is the pulse repetition rate, i.e. m = f _DK T.

The digital adder 6 can be performed according to the known rules for constructing digital combination devices of the adder-subtractor of numbers (6). If the signals of the direct and compensating channels are out of phase at the first and second bit inputs of the digital adder 6, then it performs the function of summing the signals, otherwise, the function of subtracting signals. Similarly, the digital multiplier 3 can also be performed according to the known rules for constructing the number multiplier (6).

Digital multi-channel correlation and filter receiving device with SDC (figure 1) works as follows. At its input, respectively, at the input of ADC 1, an additive mixture (sum) of a packet of coherent passive jamming radio pulses and Doppler signals, generally coinciding in time, with a duration of τ and a repetition period T = 1 / F with the frequency of filling of the jamming radio pulses f is _received , and the radio pulses of the Doppler signal f _IF + F _D , where f _IF is the intermediate frequency at the input of the path, F _D is the Doppler frequency shift of the signal from a moving object. Moreover, the amplitude of the interference pulses significantly exceeds the amplitude of the signal pulses and they are indistinguishable in time (figa, oscillogram 1). However, their comb spectra differ due to the Doppler shift of the comb spectrum of the radio pulses of the Doppler signal with the center frequency of the spectrum f _IF + F _D relative to the stationary comb spectrum of the radio pulses of interference with the center frequency of the spectrum f _IF . From the output of the ADC 1, the digital signal converted to digital form is fed to the input of the digital SDC 2, i.e., to the input of the digital adder 6 and at the same time to the input of the digital low-pass filter 7, which has a narrow-band ridge amplitude-frequency characteristic (AFC) with a period of ridges F = 1 / T, which distinguishes the comb spectrum of the interference and does not pass the comb spectrum of the Doppler signal. Therefore, the output of the digital low-pass filter 7, i.e. at the output of the compensating channel of the digital SDC 2, only digital interference radio pulses are allocated, which in the digital adder 6 are subtracted from the mixture of the digital interference radio pulses and the Doppler signal. As a result, the interference pulses are suppressed a predetermined number of times, and the Doppler signal pulses pass further without suppression. It should be noted that in order to isolate interference pulses in the digital low-pass filter 7, it is necessary that the filling frequency f of the _{IF of} these pulses be a multiple of the frequency of the crests of the digital low-pass filter 7 equal to F, i.e. f _IF = n _f F, where n _f is an integer. In pulsed Doppler systems is usually performed, as they and intermediate frequencies and the pulse repetition frequency forming a multiple of the same reference frequency f _0.

As is known, when using a burst of pulses of a quasi-continuous signal to reduce the level of the side lobes of the spectrum of a burst of interference pulses, significantly exceeding the level of the main lobe of the spectrum of the Doppler signal, various types of weight processing of the burst of pulses of the mixture are used [7]. In particular, with a large dynamic range of signals, complex weight functions are used, for example, Dolph-Chebyshev or Taylor functions, which allow reducing the level of near side lobes to -90 dB or more. However, at the same time, the width of the main lobe of the interference spectrum at the level of the side lobes expands significantly (compared to the case where there is no weight processing), reducing the possibility of detecting weak low-speed Doppler targets. And the use of simpler weighting functions, for example, functions like cosine squared without a pedestal, does not allow to obtain a low level of near side lobes. The use of digital SDS 2 before weight processing of signals allows eliminating this contradiction and increasing the efficiency of weight processing. The fact is that the level of the side lobes of the spectrum of pulses of a burst of interference directly depends on the amplitude of these pulses for any weight processing of the burst. Therefore, in the absence of digital SDC 2 and with an interference exceeding the minimum Doppler signal, for example, by 90 dB, it is necessary to apply packet weighting using the Dolph-Chebyshev or Taylor functions with a level of near side lobes of -90 dB, which leads to the expansion of the main lobe of the interference spectrum at this level 7 times [8]. The presence of a digital SDS 2, which suppresses the amplitude of the interference pulses, allows one to use the same weight functions (or apply simpler weight functions), but with a higher level of side lobes and significantly less expansion of the main lobe of the interference spectrum. So, for example, when suppressing the amplitude of interference pulses in a digital SDC 2 by 40 dB, it is enough to apply the same weight functions, but with a level of side lobes of -50 dB and expanding the main lobe by 4 times, and when suppressing interference pulses in a digital SDC 2 by 60 dB, it is enough to apply the same weight functions with a level of side lobes of -30 dB and an extension of the main lobe of only 2.6 times. Thus, the use of digital SDC 2 in a digital correlation-filter receiving device allows you to expand the detection zone of weak low-speed Doppler targets at the indicated weight processing by 1.75-2.7 times.

A quasicontinuous signal is characteristic of a rectangular envelope of a pulse train of a mixture (Fig. 2a, waveform 1). Therefore, when the digital SDC 2 is operating, transients occur at the leading and trailing edges of the packet due to the inertia of the transmission of the narrow-band digital low-pass filter 7 (Fig. 2a, waveform 2). However, the beginning of the burst of pulses of the Doppler signal, as a rule, is significantly delayed relative to the beginning of the burst of pulses of interference, which makes it possible to efficiently use the result of suppressing interference pulses in digital SDC 2 by excluding transients from further signal processing. This provides a digital multiplier 3, controlled by the weight function of the digital generator 5, the beginning of which is delayed relative to the beginning of the interference packet for a time T ₃ , usually longer than the transient process T _PP , and the end coincides with the end of the received packet. Figure 2a shows the signal at the output of the digital multiplier 3 (waveform 3) for the case of a weight function of the cosine type squared without a pedestal (waveform 4), where the delay time T ₃ = 50 T, the burst duration T ₀ = 250 T, and the weight duration function T _B = 200 T, and also shows a significant, about 40 dB, suppression of interference pulses in the steady state.

The frequency conversion of the input radio signal to the ADC 1 is as follows. When the filling frequency of the pulses of the input radio signal f _IF and the sampling frequency of this signal f _DC at the output of the ADC 1, a set of direct and mirror spectra of the received signals is formed:

- direct spectra at frequencies f _IF -f _DC , f _IF , f _IF + f _DC , etc .;

- mirror spectra at frequencies 2f _DK -f _IF , 3f _DK -f _IF , etc.

Since the entire frequency grid in pulse-Doppler radars is a multiple of the reference frequency f ₀ , then, assuming f _IF = nf ₀ and f _DC = (n-1) f ₀ , we obtain direct spectra at frequencies f ₀ , nf ₀ , (2n -1) f ₀ , etc. and mirror spectra at frequencies (n-2) f ₀ , (2n-3) f ₀ , etc. Figure 3a shows spectrograms of the analogue equivalents of processes in ADC 1 and digital SDC 2 at n = 6. The spectrum of the input signal is located at a frequency of 6f ₀ (spectrogram 1), and the sampling spectrum at the output of the ADC 1 is located at frequencies f ₀ (spectrogram 2a), 6f ₀ (spectrogram 2c), etc. (direct spectra); at frequencies 4f ₀ (spectrogram 2b), 9f ₀ (spectrogram 2d), etc. (mirror spectra) - at a sampling frequency of 5f ₀ (spectral line 4). As can be seen from the spectrograms of FIG. 3a, the most intense is the direct sampling spectrum at a frequency f ₀ , the intensity of the remaining spectra rapidly decreases with frequency. This is due to the meander shape of the sampling pulses commonly used in standard ADCs. Spectrogram 3 shows the result of the compensation of the interference pulses in the digital SDC 2. It can be seen that the shape of the spectrogram 3 is similar to the shape of the spectrogram 2, and the intensity of the direct and mirror spectra is significantly lower due to the compensation of the interference pulses in the digital adder 6. To obtain the specified suppression depth P of the interference pulses, it is necessary so that the transmission coefficient of the compensating channel of the digital SDC 2 on all ridges of the frequency response is K ₀ = (P-1) / P, which, for example, at P = 100 (40 dB) is K ₀ = 99/100 = 0.99. This is achieved by fixing the desired value of the normalizing coefficient of the transfer function of the digital low-pass filter 7 of the compensating channel of the digital SDS 2.

On figb shows the waveforms explaining the process of compensating for impulse noise in the digital SDC 2 in steady state in the scale of the pulse duration:

waveform 1 - interference radio pulse at the input of the ADC 1 of duration τ = T / 10 with a filling frequency f _IF = 6f ₀ ;

waveform 2 - a digital pulse in analogue expression at the output of the ADC 1 in the form of a stepwise oscillation with a repetition rate f ₀ representing the sampling result of the input radio pulse 1 with a sampling frequency f _DK = 5f ₀ ; the length of the steps is τ _DC = 1 / f _DC ;

waveform 3 - a digital pulse at the output of the digital adder 6 of the digital SDC 2 also in the form of a stepwise oscillation with a repetition frequency f ₀ representing the difference of the digital oscillations 2 and the digital oscillations at the output of the digital low-pass filter 7, but with an amplitude 40 dB less than the oscillogram 2.

FIG. 3b shows spectrograms illustrating the process of clearing the spectrum of interference pulses from side lobes in the Doppler frequency range (from f ₀ to f ₀ + F) during weight processing by a cosine function squared without a pedestal:

spectrogram 1 represents the spectrum of the stepwise oscillations of the impulse burst pulses at the ADC 1 output, converted to a frequency f ₀ , in the absence of a digital SDC 2 and after weighing the signals in the form of main lobes following with a pulse repetition rate F = 1 / T, and part of the side lobes between them to a level of -100 dB relative to the level of the main lobes. It is seen that in this case, the zone of the side lobes is 13% of the interval between the main lobes;

spectrogram 2 represents a spectrum of step noise burst pulse oscillations at the output of digital MTI 2 at a frequency f ₀ after weighting signal processing, which shows that the main lobes of the spectrum are suppressed by 40 dB with respect to the spectrogram 1, whereby the half-width of the main lobe to sidelobe area on -100 dB decreased to 3% of the interval between the main lobes;

Spectrogram 3 represents a similar case when the suppression of interference pulses in digital SDC 2 is increased to 60 dB, as a result of which the half-width of the main lobe with the side lobe area at the level of -100 dB decreased to 1% of the interval between the main lobes.

Thus, the use of a digital SDS in a digital correlation filter receiver before weight processing of signals allows one to practically free the spectrum of interference pulses from the unwanted expansion of the main lobes and the presence of side lobes in the dynamic range of signals up to 100 dB even with a simpler weight function, which makes it possible use of pulse-Doppler radar to detect weak low-speed Doppler targets.

Information sources

1. RU No. 2205422, G01S 13/52, 13/5 26, H04B 1/10, 2003.

2. Reference radar. Ed. M. Skolnik, New York, 1970: Per. from English (in four volumes) / Under the general ed. K.N. Trofimova; Volume 3. Radar devices and systems / Ed. A.S. Vinitsky. - M .: Owls. Radio 1978, p. 369, Fig. 6.

3. P.A. Bakulev, V.M. Stepin. Methods and devices for moving targets selection. M: Radio and communications, 1986, pp. 140-141, Fig. 5.20.

4.S.I. Baskakov. Radio circuits and signals. M.: Higher School, 1983, pp. 489-491.

5. A. Anthony. Digital filters: analysis and design. M .: Radio and communications, 1983, pp. 292-293.

6. A. Anthony. Digital filters: analysis and design. M .: Radio and communications, 1983, pp. 273-289.

7. A.A. Trukhachev. Radar signals and their applications. M .: Military Publishing House, 2005, pp. 112-113.

8. A.A. Trukhachev. Radar signals and their applications. M .: Military Publishing House, 2005, pp. 125-138.

Claims (3)

1. A digital multichannel correlation filter receiver with moving targets, comprising a modulator and a multichannel correlation filter processor for receiving signals of a pulse-Doppler radar, as well as a weight function generator, the output of which is connected to the second input of the modulator, characterized in that introduced an n-bit analog-to-digital converter (ADC) and a digital moving target selector (SDC) of the compensation type, containing in the direct channel an n-bit chi background adder, and in the compensating channel - an n-bit digital low-pass filter (LPF), the modulator is made in the form of an n-bit digital multiplier, a weight function generator in the form of a digital generator, a multichannel device for correlation-filter processing of received pulse-Doppler signals The radar is made n-bit digital, the input of the n-bit ADC is the input of the device and is connected to the output of the signal source of the pulse-Doppler radar, and its output buses are connected to the first bit and the inputs of the n-bit digital adder and the input buses of the n-bit digital low-pass filter, the output buses of which are connected to the second bit inputs of the n-bit digital adder, the output of which is connected to the first bit inputs of the n-bit digital multiplier, the control input of the n-bit ADC and the control input of the n-bit digital low-pass filter is combined and connected to a source of sampling pulses. 2. The digital multi-channel correlation filter receiver according to claim 1, characterized in that the canonical recursive digital low-pass filter of the first order is used as a digital low-pass filter. 3. The digital multi-channel correlation filter receiver according to claim 1, characterized in that the canonical recursive digital low-pass filter of the second order is used as a digital low-pass filter.

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