RU2255349C1 - Device for formation of angular error at reception of noise signal - Google Patents

Abstract

FIELD: radiolocation, applicable in monopulse in pulse-Doppler radars that have one quadrature for provision of hard requirements in mass and overall dimensions in the receivers of summary and differential channels.

SUBSTANCE: in this case determination of the angular error of direction to the source of noise radiation should be made with due account for the particularities of the obtained signal spectrum. A correlation is based in the invention for estimation of the angular error of direction to the source of noise radiation, and a diagramma, is given of the device realizing the estimate of the angular error in case of availability of one quadrature in the receiving paths of each channel. The problem is solved due to decision of the task of determination of the angular error of direction to the source of noise radiation in pulse-Doppler radars with a single quadrature.

EFFECT: provided hard requirements in mass and overall dimensions in the receivers of summary and differential channels having a single quadrature.

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Description

The invention relates to radar and can be used to generate an angular error in radar coherent monopulse stations with a phase sum-difference direction finder when measuring the angular coordinates of a noise signal source or with automatic tracking of a noise signal source.

A device is known for generating angular error in a monopulse radar with a phase sum-difference direction finding method [A.I. Leonov, K.I. Fomichev. Monopulse radar. - M .: Radio and communications, 1984, p. 71], a block diagram of which is shown in figure 1. An angular error or error signal is a value functionally related to the angle between the axis of the antenna and the direction of the radiation source received by the antenna. The blocks included in the circuit of Fig. 1 used for standard purposes: antenna, mixer, reference signal generator, phase shifter, intermediate frequency amplifier, automatic gain control system, phase detector, are described there [A.I. Leonov, K.I. Fomichev. Monopulse radar - M: Radio and communications, 1984, p. 14-65].

As a result of the operation of the device of Fig. 1, an angular error α [see ibid., p. 72, expression /4.17/]:

where α is the angular error;

и

- комплексные сигналы суммарного и разностного каналов;

and

- complex signals of the total and differential channels;

- комплексно сопряженные величины сигналов суммарного и разностного каналов;

- complex conjugate values of the signals of the total and differential channels;

Re (...) is the real component of the value enclosed in brackets.

Also known is a device for generating an angular error in coherent radars with digital signal processing, a block diagram of which is shown in FIG. 2 [A.I. Kanaschenkov, V.I. Merkulov. Protection of radar systems from interference. - M .: Radio engineering, 2003, p. 38].

) и массивы значений действительной и мнимой составляющие разностного сигнала (

). Каждая из этих составляющих формируется своим приемным трактом.This device in the total and difference channels has two receiving paths, called quadrature. The angular error α of the target signal is calculated in the block for estimating the angular error after the fast Fourier transform (FFT), performed separately with the data of the total and difference channels. Arrays of values of the real and imaginary components of the total signal (

) and arrays of values of the real and imaginary components of the difference signal (

) Each of these components is formed by its receiving path.

The angular error α of the target signal is calculated after the FFT for the frequency ω _i (the ith value of the array of signals at the output of the FFT, called the outputs or samples of the FFT), on which there is a signal in the total channel. Performing the transformations of relation (1), we obtain

Denoting the signal values at the FFT output for the total and difference channels, respectively, through C _Σ , C _Δ , we rewrite the expression for the angular error α:

Where

ReC _Σ , ImС _∑ - real and imaginary components of the output signal of the FFT of the total channel;

ReС _Δ , ImС _Δ - real and imaginary components of the output signal of the FFT of the difference channel.

Relation (2) is used in the diagram of FIG. 2 to estimate the angular error. The angular error is calculated from the components of the ith value of the output FFT signal array corresponding to the frequency ω _i. on which the target is detected (observed).

In the presence of a noise signal source, the angular error α w is calculated by the relation (2 ^a ):

- действительная и мнимая составляющие шумового сигнала в суммарном и разностном каналах после БПФ на частоте ω i. В качестве частоты ω i обычно выбирается частота, на которой помеха в суммарном канале имеет максимальное значение. С целью уменьшения влияния флюктуаций для оценки угловой ошибки источника шума используется соотношение (3):Where

- the real and imaginary components of the noise signal in the total and difference channels after the FFT at a frequency ω i. As the frequency ω i, the frequency at which the interference in the total channel has a maximum value is usually selected. In order to reduce the influence of fluctuations, the relation (3) is used to estimate the angular error of the noise source:

where i = k + 1, k + 2, ..., k + n ₀ is the group of n ₀ frequencies taken from the occupied frequency band. By asking n ₀ ≥ 60, an angular error estimate is obtained with an acceptable confidence interval.

In practically used devices for generating the angular error when using relations (2), (3), the phase non-identity of the channel characteristics is taken into account. If the phase shift of the difference channel relative to the total channel is θ, then the values of the components of the spectrum of the difference channel are adjusted by recalculation as follows:

- составляющие спектра разностного канала, полученные на выходе блока БПФ разностного канала при фазовом рассогласовании разностного и суммарного каналов, равном θ .Where

- the components of the spectrum of the difference channel obtained at the output of the FFT block of the difference channel during phase mismatch of the difference and total channels equal to θ.

The device for generating an angular error according to the scheme of figure 2 with the calculation of the angular error in relation (3) is a prototype of the invention.

The disadvantage of constructing a device for generating an angular error according to the scheme of Fig. 2 is the need for two receiving paths (including an RF receiver, a phase detector, a low-pass filter, an ADC, an adder) in both the total and difference channels, which causes a deterioration in the overall dimensions. This may be unacceptable in the case of severe restrictions on weight and dimensions, which is the case in airborne aircraft systems.

The objective of the invention is to reduce the mass and dimensions of the device for generating the angular error of the interference signal.

The problem is solved in the present invention, the block diagram of which is shown in figure 3, through the use of only one receiving path in the total and difference channels, one FFT block for the simultaneous conversion of the signals of the total and difference channels, each of which is represented only by the real part, and introducing a block for estimating the angular error, in which this error is found in a ratio that takes into account the presence of only valid signal components at the inputs of the FFT.

Figure 1 presents a block diagram of a device for generating an error signal in a monopulse radar.

Figure 2 presents a block diagram of a device for generating an angular error in a monopulse radar with digital processing of information (prototype).

Figure 3 presents a block diagram of the invention.

Figure 4 presents a block diagram of the algorithm of the block recovery components.

Figure 5 presents the block diagram of the algorithm for estimating the angular error.

The device for generating an angular error when receiving a noise signal contains a monopulse antenna 1, a reference signal generator 2, a high-frequency (RF) receiver of the total channel 3, a phase detector of the total channel 4, a low-pass filter of the total channel 5, an analog-to-digital converter (ADC) of the total channel 6 , total channel adder 7, RF receiver of the differential channel 8, phase detector of the differential channel 9, low-pass filter of the differential channel 10, ADC of the differential channel 11, adder of the differential channel 12, fast conversion unit Fourier transform (FFT) 13, a unit for recovering the component signals of the sum and difference channels 14, a bipolar switch 15, a test signal generator 16, a phase mismatch estimation unit 17, a phase mismatch memory unit 18, a control unit 19, a first switch 20, a second switch 21, an angular error estimation unit 22, the output of which is the output of the device.

Included in the invention are the standard elements of bl. 1-12 are used in accordance with their purpose [A.I. Kanaschenkov, V.I. Merkulov. Protection of radar systems from interference. - M .: Radio engineering, 2003, p. 37, 39].

Bl.13 performs FFT conversion [Introduction to Digital Filtering. Ed. R. Bogner and A. Kostantinidis. - M .: MIR, 1976, pp. 116-121]. The FFT block has two inputs. An array of real components of the signal sample should be supplied to one of them (input of real components), and an array of imaginary components of the same sample to the second (input of imaginary components). In the proposed device, each of the two inputs of the FFT block is supplied with an array of M-values each, one of which is the real component of the sample signal of the total channel, and the second is the real component of the sample signal of the differential channel. If, when performing an FFT with the real and imaginary components of the same signal supplied to the inputs of the FFT block, the samples obtained at the output of the FFT block represent the spectrum of this signal, then when two valid components of two different signals are output to the inputs of the FFT block, the samples are a combination components of the spectrum of signals fed to the input by its real parts. According to [Introduction to Digital Filtering. Ed. R. Bogner and A. Kostantinidis. - M .: MIR, 1976., pp.128-131] the real and imaginary components of the FFT readings (real and imaginary components of the spectrum) obtained in this case at the output of the FFT block allow you to find the real and imaginary components of the spectrum of each of the input signals that were supplied to the input of the FFT block with its real parts.

The restoration of the spectrum components of the sum and difference channels is performed by block 14. Denoting X (n) the output values of the FFT transform, in accordance with [ibid., P. 131, relation (7.23)], the real and imaginary components of the spectrum of the total and difference channels are found by the ratios:

where n = 0, 1, 2, ..., M / 2

The implementation algorithm of bl.14 is shown in Fig.4. After the recovery operation, we obtain the arrays ReC _Σ , ImC _Σ , ReC _Δ , ImС _Δ . by M / 2 values each. The spectrum components obtained in bl.14 through a bipolar switch are received in bl.17 or bl.22.

The proposed device for generating an angular error when receiving a noise signal required solving problems associated with the rejection of the second receiving paths. It is known that when applying the FFT input only the real component of the converted quantity, the output spectrum has the property of complex conjugate symmetry [Introduction to digital filtering. Ed. R. Bogner and A. Kostantinidis. - M .: MIR, 1976, p. 128]:

where X _i is the complex signal of the i-th FFT output;

- комплексно-сопряженная величина (M-i)-му выходу БПФ;

- the complex conjugate value of the (Mi) -th FFT output;

M is the number of FFT points;

i = 0, 1, 2, ..., M-1.

In this case, the FFT output signals acquire the following property:

where ReX _i , ImX _i are the real and imaginary components of the i-th output of the FFT block (the components of the frequency signal ω i obtained at the output of the FFT);

- принятые сигналы, соответствующие частотам ω i и ω M-i.

- received signals corresponding to the frequencies ω i and ω Mi.

(при этом приходится работать с М/2 выходами БПФ). В этом случае в соответствии с соотношениями (4) составляющие ReX_i, ImX_i будут являться составляющими сигнала цели, что позволит по соотношению (2) найти угловую ошибку сигнала цели.This property does not complicate the formation of an angular error for one target, because, knowing the Doppler frequency range of the target, it is possible to select the FFT parameters so that the signals of all real targets lie in the range ω ₀ ≤ω _i <ω _{M / 2} , and then

(you have to work with the M / 2 FFT outputs). In this case, in accordance with relations (4), the components ReX _i , ImX _i will be components of the target signal, which will allow us to find the angular error of the target signal from relation (2).

In the direction finding of a noise source whose frequency band covers the entire reception frequency band, the noise interference signal is present at all frequencies, therefore, in accordance with relation (4), the real and imaginary components of each FFT output signal are the sum of the components of two input signals. As a result of this, the calculation of the signal of the angular error of the noise source from relations (2 ^a ) and (3) will lead to an incorrect result.

The ratio for calculating the angular error in the present invention is obtained as follows.

Let a continuous noise interference with an average value m = 0 and a standard deviation σ = σ _in be received at the input of a single-pulse antenna. The frequency band covered by noise interference is greater than Fп (Fп - radar repetition frequency).

For the case of a receiving path with one quadrature, the interference signal at the ADC input of the total channel can be represented by the expression (5)

where M is the number of points in the input data array (the number of FFT points).

Accordingly, at the input of the ADC of the difference channel, the interference can be represented by the expression (6),

where V _i is a random variable distributed according to Rayleigh;

φ _i is a random phase uniformly distributed over the interval 0-2π;

α is the angular error (a value functionally related to the angle between the axis of the antenna and the direction to the source of interference);

θ is the phase shift that occurs when the signal passes through the total and difference channels due to the hardware non-identity of the channels (phase mismatch of the channels);

k, i are integers

Signals from the sum and difference channels are sent to coherent processing. In each channel, 0.5 M filters are implemented that overlap the 0-0.5 Fp band.

Consider how a signal X with a frequency ω _i and a signal Y with a frequency ω _Mi fall into a filter tuned to a frequency ω _i .

According to (5), we can write:

We introduce the following notation:

Then:

and

Given (7), (8) and (9) and the equality e ^{j · 2 · π · k} = 1 (k is an integer), we can write: ω _м-i = -ω _i , and equation (10) can be written as:

We introduce the following notation:

Then we get:

From equalities (10) and (11) it can be seen that in the passband of the filter tuned to the frequency ω _i , there are two source signals with frequencies ω _i and ω _Mi.

After the Fourier transform, the signal at the output of the i-th filter of the total channel (signal of the i-th frequency) is written in the form:

In the difference channel at the output of the i-th filter after the Fourier transform, we similarly get:

considering that

we get the real and imaginary parts of the signals at the output of the i-th filter of the total and difference channels:

The quantities A _i · cosφ _i , B _i · cosφ _Mi , A _i · sinφ _i , B _i · sinφ _Mi are independent random numbers distributed according to the normal law with m = 0 and σ = _σout .

In order to determine α, we calculate the value of G - the average value of the sums of the products of the real and imaginary parts of the signals of the total and difference channels:

а выражение в круглых скобках соответствует числителю соотношения (2).Where

and the expression in parentheses corresponds to the numerator of relation (2).

We also find the average value of the square of the signal modulus of the total channel:

Given that the quantities

as the sum of the products of independent random variables, and the average values of the products

equal to the variance of these quantities and equal to each other, we finally get:

Where from

Where

In the proposed device, the ratio (24) is implemented by block 22 in accordance with the algorithm, a block diagram of which is given in Fig.5.

The phase mismatch of the total and difference channels θ included in relation (24) is constant over the time interval for estimating the angular error and is determined previously by the command generated by the control unit. Phase mismatch is when applying to the total and difference channels of the same test signal. There should be no noise signal.

To find the phase mismatch θ in accordance with the block diagram of Fig. 3 by the control signal generated by the control unit (bl.19), the first and second RF signal switches (bl.20 and bl.21, respectively) pass the test signal from the generator the test signal (bl.16) to the inputs of the RF receivers of the total and difference channels (blocks 3 and 8, respectively), and the bipolar switch (bl.15) supplies the components of the signal spectrum to the phase mismatch estimation block (bl.17). As a result, in the presence of a control signal, the operation of the proposed device in the measurement mode θ is ensured.

The control unit, by means of which the phase mismatch measurement mode is activated, is a device that, when an external control signal is received at its input, generates a control signal at the output. The control signal is an electric signal with predetermined parameters, for example, a constant voltage N volt, which occurs when a circuit existing in block 19 is closed. The external control signal represents the external action necessary to close the circuit for generating the output control signal. An external signal may be the inclusion of voltage supplied to an electronic relay that closes the above electrical circuit in Bl.19, or the operator presses a toggle switch that closes this circuit.

The switch is a device having two signal inputs and a control input, which, when applied to the control input of the control signal, passes the first input signal to the output. If there is no control signal at the control input, the second input signal is output. Possible implementations of the RF switch are given in [Design and calculation of strip devices. Ed. I.S. Kovaleva. M .: Sov. Radio, 1974, p.223-239].

The test signal generator is a constant frequency monochromatic signal generator in the radar reception range [I. Gonorovsky. Radio circuits and signals. - M .: Radio and communication. 1986, pp. 270-304], supplying the same test signal to both receiving channels.

To find θ in the proposed device, the test signal is processed by the sum and difference channels, FFT is performed on the real components of the signals of the sum and difference channels, the components of the spectra of the signals of the sum and difference channels are restored, which are fed to the phase mismatch estimation unit via a two-pole switch (bl.15) (bl.17).

A bipolar switch is a device that has a signal and control inputs and two outputs, which, by a control command, switches the input signal from one output to another [A. Flores. Organization of computers. Per. from English. - M., 1972, p.20-26].

In bl.17 phase mismatch (found by the ratio

where i is the frequency number corresponding to the test signal.

The above relation, implemented by Section 17, takes into account the phases of the total and difference channels, as well as the shift by π / 2 introduced by the waveguide bridge during the formation of the difference signal. Block 17 can be implemented in the form of an algorithm corresponding to the given relation in a computer or in the form of a special calculator that performs the calculation of θ in accordance with this relation.

The obtained θ value is stored in the phase mismatch memory block (Bl.18), which is a memory cell whose circuit design is known from the literature [A. Flores. Organization of computers. Per. from English. - M., 1972, p.20-26].

The proposed device of figure 3 works as follows.

A noise signal, the direction to which determines the angular error, is received by a monopulse antenna (Bl.1). Block 1 generates two components of the signal at the output: total ∑ and differential Δ, which, through the switches (total component through the first switch, differential component through the second switch), the control signal at which at this moment is "0", are each supplied to their RF -receiver (bl.3 and bl.8), where the signal is amplified. The signal from the output of each RF receiver (total and difference channels) is fed to the input of its phase detector PD (block 4, block 9). The signal of the reference signal generator (block 2) is supplied to the second input of each PD. As a result of the PD operation, a signal is generated at its output, having components with frequencies equal to the sum and difference of the frequencies supplied to the input. To isolate the difference frequency, the output signal of the phase detector of each channel is fed to its own low-pass filter (block 5 in the total channel, block 10 in the difference channel), which passes only the difference frequency signal, which is the Doppler frequency, to its output. From the output of the low-pass filter, the Doppler frequency signal is fed to the ADC (Bl.6 and Bl.11), where it is converted to digital form. After each ADC, its own adder is switched on (Bl.7 and Bl.12), where the digitally converted signal values are summed over an interval equal to the pulse duration, thereby ensuring consistent filtering of the pulse signal. The signals from the output of the adder of the total channel with a period of probing pulses are fed to one of the inputs of the FFT block (Bl.13), and the signals from the output of the adder of the difference channel are fed to the second input of the FFT block. Block 13 thus processes two sequences: the real component of the signals of the total channel and the real component of the signals of the differential channel. The output signals of bl.13, which are a combination of the spectra of the signals of the total and differential channels, are fed to the input of the recovery unit of the component signals of the total and differential channels (bl.14). In block 14, in accordance with the algorithm of Fig. 4, the real and imaginary components of the spectrum of the total and difference channels ReС _Σ , ImС _Σ , ReС _Δ , ImС _{Δ are} calculated.

The indicated components in the form of an array through a two-pole switch (bl.15), at the control input of which the signal is “0”, are fed to the input of the angular error estimation block (bl.22), in which the angular error is calculated in accordance with the algorithm of Fig. 5. To the other input of bl.22, the phase mismatch θ from bl.18 is received, where this value is stored after its determination in accordance with the device operation mode described below when the phase mismatch is found (the determination of the phase mismatch precedes the measurement of the angular error).

The operation of the device in the mode of finding the phase mismatch θ is as follows. Upon receipt of an external control signal at the input of the control unit (bl.19), a control signal is generated at the output of this unit.

The control signal is supplied to:

- control inputs of the first and second switches, which, in the presence of a control signal, pass the output of the test generator to the output;

- the control input of the bipolar switch, which, in the presence of a control signal, supplies to the input of bl.17 the components of the signals of the total and difference channels restored to bl.14.

The signal of a given constant frequency from the test signal generator through the first and second switches is fed to the inputs of the RF receivers of the total and difference channels. Processing of this signal is performed by blocks 2-14, as described above. From the output of unit 14, the values of the components of the spectrum of the test signal in the sum and difference channel are supplied through a two-pole switch (unit 15) to the phase mismatch estimation unit (unit 17), where the value of θ is calculated. The found value (supplied for storage to the phase mismatch memory block (bl. 18). The value stored in bl. 18 (used in bl. 22 when measuring angular error).

The value of the angular error calculated in Section 22 is fed to the output of the device.

A sample of the proposed device for generating an angular error when receiving a noise signal has successfully passed the test and is ready for industrial use.

Claims (1)

A device for generating an angular error when receiving a noise signal, including a monopulse antenna, an RF receiver of the total channel, the output of which is connected to the first input of the phase detector of the total channel, the output of which is connected to a low-pass filter, an analog-to-digital converter, and an accumulator of the total channel, RF -a receiver of the difference channel, the output of which is connected to the first input of the phase detector of the difference channel, the output of which is connected to the lower filter in series frequency, analog-to-digital Converter and the difference channel adder, a reference signal generator, the output of which is connected to the second inputs of the phase detectors of the total and difference channels, characterized in that a series-connected fast Fourier transform unit (FFT block) is introduced, the input of the real components of which is connected to the output of the adder of the total channel, and the input of the imaginary components of which is connected to the output of the adder of the differential channel, the recovery unit of the components of the total and differential about channels, a bipolar switch, a phase mismatch estimation unit and a phase mismatch storage unit, an angular error estimation unit, an output of the recovery unit of the sum and difference channel components are connected to the signal input of the bipolar switch, the first output of the bipolar switch is connected to the first input of the angular error estimation unit , the second output of this switch is connected to the phase mismatch estimation unit, the output of the phase mismatch storage unit It is connected to the second input of the angular error estimation unit, the test signal generator, the first switch, the second switch and the control unit are also introduced, the total output of the monopulse antenna is connected to the first input of the first switch, the differential output of the monopulse antenna is connected to the first input of the second switch, the output of the test generator connected to the second inputs of the first and second switches, an external control signal is supplied to the input of the control unit, the output of the control unit is connected to the control moves the first switch, second switch and a two-pole switch, the first switch output connected to the input sum signal RF receiver, the second switch output is connected to the input of the difference-channel RF receiver, the output evaluation unit angular error is the output device.

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