RU2143709C1 - Method of selection of moving targets - Google Patents

Abstract

FIELD: radiolocation. SUBSTANCE: proposed method is intended for use in radars selecting signals of moving targets against background of reflections from ground surface. Technical aim of invention lies in possibility of employment of system of complex phase-keyed signals with intradiscrete frequency modulation for selection of signals of moving targets received in additive mixture with signals reflected from underlying surface with simultaneous suppression of side lobes. This is achieved by formation, radiation, reception, compression of signals in matched filters with corresponding frequency modulation to form complex phase-keyed signal which is then compressed in proper matched filter. Even signal is subtracted from odd one which have maximum range according to Hamming. Subtraction results of system of signals are stored in two groups and obtained differences are multiplied in step. EFFECT: improved efficiency of method. 13 dwg

Description

 The proposed method relates to the field of radar and can be used in a radar station designed to distinguish signals of moving targets against reflections from the earth's surface.

 The known method, which is an analogue, including the formation (available in the claimed object), radiation (available in the claimed object), reception (available in the claimed object) of a continuous monochromatic signal and which is described in the book P. Bakulev. Radar of moving targets. M .: - Owls. radio, 1964, p. 107-110.

 However, this method has the disadvantage that when using such a signal, it is impossible to measure the distance to the target.

 The method, which is analogous, is free from this drawback, including the generation (available in the claimed object), radiation (available in the claimed object), reception (available in the claimed object) of a continuous signal with frequency modulation (FM), comparison with the emitted oscillation and filtering the required frequency . This method allows you to select the Doppler frequency signal against the background of reflections from the earth's surface and which is described in the book Bakulev P. A. Radar tracking of moving targets. M .: - Owls. radio, 1964, p. 117-125.

 However, this method has the disadvantage that although it allows to reduce the leakage of the probe pulse to the input of the receiving device, it does not allow to determine the distance to the target.

 The known method, which is an analogue, including the formation of a periodic sequence of signals at two frequencies, radiation (available in the claimed object), reception, including filtering, frequency offset, amplification (available in the claimed object), multiplication of signals to each other (available in the claimed object ) and which is described in the book of Whistlers V.M. Radar signals and their processing. M .: - Owls. Radio, 1978, p. 372 - 374.

 However, this method, eliminating the blind speeds of a moving target, has the disadvantage that it requires two transmitting and two receiving paths, the radiation frequency and the frequency of the local oscillators are interconnected.

 The method, which is analogous, is free from this drawback, which includes the formation of a periodic sequence of coherent pulses (available in the claimed object), radiation (available in the claimed object), reception (available in the claimed object), detection, filtering, amplification, indication, and which is described in the book Bakulev P.A. Radar of moving targets. M .: - Owls. radio, 1964, p. 162-164.

 However, this method has the disadvantage that in order to increase the resolution in range it is necessary to shorten the duration of the probe pulse, and this leads to a deterioration in the resolution in speed.

 The known method, which is an analogue, including the formation (available in the claimed object), radiation (available in the claimed object), reception (available in the claimed object), the accumulation of quadrature components of a packet of signals (available in the claimed object), followed by their modulation by LFM signals with opposite laws of modulation, compression of the obtained processing results (available in the claimed object), allocation of envelopes (available in the claimed object), subtraction of them among themselves (available in the claimed object) in order to isolate the signal in moving targets (RF Patent N 738450, cl. G 01 R 23/16. An apparatus for processing radar signals /V.I.Lityuk, 1993).

 However, this method has the disadvantage that simple signals are used, and this limits the possibility of obtaining high resolution in range while fulfilling the requirements of ensuring a given maximum detection range.

 The closest in technical essence and functional purpose is the method of sensing by a system of complex phase-shift keyed (PSK) signals with in-disc modulation, which is a prototype that allows suppressing the long-range side lobes (BL) of a system of compressed complex signals, that they form (have in the claimed object ) a sequence of phase-shifted signals, emit (available in the claimed object), receive (available in the claimed object), and each discrete of the emitted the manipulated signal with the dimension of code N of each of the sequence of phase-manipulated signals consisting of 2N signals is subjected to in-disc frequency modulation (available in the claimed object), the law of change of which is determined by the sign of the phase-manipulated signal code (available in the claimed object), the laws of variation of the frequency of the in-discrete frequency modulation are orthogonal or quasi-orthogonal (available in the claimed object), the range of variation of these frequencies for all discretes is the same (available in the claimed object) and all signals of the sequence have the same carrier frequency (available in the claimed object), and upon receipt, each discrete with its own law of in-discrete frequency modulation of the phase-shift keyed signal with the dimension of code N is processed in a system of matched filters (available in the claimed object), each of which is matched with the law of changing the frequency of the corresponding discrete phase-shift signal discrete (available in the claimed object), at the output of the matched filters, discrete envelopes are allocated (available in is the object), which corresponds to the sign of the discrete code code of the corresponding phase-manipulated signal (present in the claimed object), and the sequence of envelopes, whose signs correspond to the code of the corresponding phase-manipulated signal, are filtered in the corresponding matched filter of the phase-manipulated signal (available in the claimed object), the signals received on the outputs of all filters of the phase-shifted signals are appropriately delayed and synchronously summed over time interactions the vale, the duration of which is determined by the repetition period of the processed sequence, consisting of 2N phase-shifted signals with in-disc frequency modulation, consisting of two groups of N phase-shifted signals in each group, and each phase-shifted signal from the group of N phase-shifted signals corresponds to an inverse phase-shifted signal of the other group, consisting of from N phase-shifted signals, and phase-shifted signals in each group consisting of N phase anipulirovannyh signals orthogonal to each other (RF Patent N 2107926. Method impulse radar system PSK signal / R. V.Lityuk. Publ. in bull. fig. - 1998. N 9. - p. 377).

 However, this method has the disadvantage that it is impossible to isolate signals of moving targets received in an additive mixture with signals reflected from the underlying surface, which are simultaneously in the same range element with moving targets.

The technical result consists in the possibility of using a system of complex PSK signals with an intra-discrete FM to isolate signals of moving targets received in an additive mixture with signals reflected from the underlying surface, while completely suppressing BL in the entire plane (τ, F) with the exception of the ± τ _d relative to the point τ = 0, where τ _d is the duration of the discrete complex PSK signals.

 The technical result is achieved by the fact that in the method of selection of moving targets, a system of phase-manipulated signals with in-discrete frequency modulation is generated, emitted, and each discrete of the emitted phase-manipulated signal with a code dimension N of each of the sequence of phase-manipulated signals consisting of 2N signals is subjected to in-disc frequency modulation, the law the change of which is determined by the sign of the phase-manipulated signal code frequency modulations are orthogonal or quasi-orthogonal, the range of variation of these frequencies is the same for all discrete and all signals of the sequence have the same carrier frequency, and upon reception, each discrete with its own law of in-disc frequency modulation of the phase-shifted signal with code dimension N is processed in a system of matched filters, each of which is matched with the law of changing the frequency of the corresponding phase-shifted signal, and the sequence of envelopes whose signs correspond to to the corresponding phase-manipulated signal, they are filtered in the corresponding matched filter of the phase-manipulated signal, and the phase-manipulated signals with in-discrete frequency modulation are emitted sequentially in time so that the odd numbers of the signals in this sequence have a maximum Hamming code distance relative to the even signals following them, their results processing in the corresponding matched filters is subtracted from each other, the results of the subtraction accumulate in two groups, and the accumulation is carried out in such a way that in one group of accumulated differences all the first elements are positive (negative), and in the other group of accumulated differences all the last elements are positive (negative), the obtained groups of accumulated differences are appropriately delayed and synchronously summarize, and the obtained summation results in two groups are synchronously multiplied among themselves on a time interval, the duration of which is determined by the repetition period I have processed sequence.

Пусть на вход блока согласованных фильтров (СФ) одиночных ФМн сигналов (соответствующим образом настроенных) поступает периодическая последовательность импульсов системы сигналов с выходов блока согласованных фильтров дискретов (СФД) (Патент РФ N 2107926. Способ импульсной радиолокации системой фазоманипулированных сигналов /Л.В.Литюк. Опубл. в Бюл. изобр. - 1998. N 9. - с. 377) и которая может быть представлена в виде

где Δ_I и Δ_II - главные пики сжатых дискретов, имеющие противоположные направления частотно-временной зависимости на плоскости (τ,F) и знаки которых соответствуют знакам используемого ФМн кода;

- левые БЛ сжатых дискретов;

- правые БЛ сжатых дискретов. Знаки БЛ совпадают со знаками главных пиков Δ_I и Δ_II соответственно.Consider the proposed method for distinguishing signals of moving targets against the background of reflections from the earth's surface using the example of the H-matrix, which we denote by H _4/8

Let a periodic sequence of pulses of a system of signals from the outputs of a block of matched filter filters (SFD) be received at the input of a block of matched filters (SF) of single PSK signals (appropriately tuned) (RF Patent N 2107926. Pulse radar method of a phase-shifted signal system / L.V. Lityuk Published in Bul. Inventory - 1998. N 9. - p. 377) and which can be presented in the form

where Δ _I and Δ _II are the main peaks of the compressed samples having opposite directions of the time-frequency dependence on the (τ, F) plane and whose signs correspond to the signs of the used PSK code;

- left BL compressed discrete;

- right BC compressed discrete. The signs of BL coincide with the signs of the main peaks Δ _I and Δ _II, respectively.

Следовательно, матрица откликов будет иметь вид

В дальнейшем опустим в матрице (3) нулевые крайние столбцы, поскольку это не оказывает влияния на рассматриваемый алгоритм обработки. Произведем поэлементное вычитание из нечетных строк матрицы (3) тех ее четных строк, которые имеют максимальное кодовое расстояние по Хэммингу друг относительно друга. В результате получается разностная матрица ΔK/ , которая будет иметь вид

Выделяем разностные строки матрицы ΔK , у которой знаки обоих крайних разностных элементов в строках положительны. Данные, описываемые этими строками, в дальнейшей обработке используются без изменений.The impulse response matrix of the matched filters of the PSK signals will have the form

Consequently, the response matrix will have the form

In the future, we omit the zero extreme columns in the matrix (3), since this does not affect the processing algorithm under consideration. Let us perform element-by-element subtraction from odd rows of matrix (3) of those even rows that have a maximum Hamming code distance relative to each other. The result is a difference matrix ΔK /, which will have the form

We select the difference rows of the matrix ΔK, in which the signs of both extreme difference elements in the rows are positive. The data described by these lines is used unchanged in further processing.

После этого производится суммирование элементов столбцов матрицы ΔK_Л, что соответствует накоплению результатов обработки всей пачки импульсов. В результате получаем матрицу-строку ΔK_ЛC, у которой левые элементы относительно центрального, центральный элемент и элемент, описывающий БЛ, прилегающий к центральному элементу справа, не равны нулю, а все элементы, расположенные далее справа, равны нулю, и которая имеет вид

Далее умножаем на величину -1 те строки матрицы (4), у которых правые крайние элементы имеют противоположные знаки относительно знаков элементов, которые расположены в тех ее строках, где они имеют одинаковые знаки с обеих сторон строк. В результате получается матрица ΔK_{П =}, которая имеет одинаковые величины и знаки у правых крайних элементов и которая имеет вид

Аналогично, проводя суммирование элементов столбцов матрицы ΔK_П, что также соответствует накоплению результатов обработки, получаем матрицу-строку ΔK_ПC, у которой правые элементы относительно центрального, центральный элемент и элемент, описывающий БЛ и прилегающий к центральному элементу слева, не равны нулю, а все элементы, расположенные далее слева, равны нулю, и которая в результате принимает вид

Таким образом, сформированы две группы откликов, расположение которых на одноименных позициях на временной оси описывается выражениями (5) и (6). Переписывая (5) и (6) в виде диагональных матриц, главные диагонали которых являются элементами матриц-строк ΔK_ЛС и ΔK_ПС соответственно, и перемножая эти диагональные матрицы друг на друга, получим диагональную матрицу, имеющую вид

Из последнего выражения видно, что не равны нулю элементы, расположенные на главной диагонали, описывающие местоположение центрального пика, и БЛ, расположенных слева и справа относительно центрального.At the next stage, by multiplying all the elements of the corresponding rows by -1, positive signs are set for those elements of the leftmost column of the matrix (4) whose signs were negative. As a result, we obtain the matrix ΔK _L , which has the same values and signs of the left extreme elements and takes the form

After this, the summation of the elements of the matrix columns ΔK _L , which corresponds to the accumulation of the processing results of the entire packet of pulses. As a result, we obtain the row matrix ΔK _LC , in which the left elements are relative to the central, the central element and the element describing the BL adjacent to the central element on the right are non-zero, and all elements located further to the right are zero, and which has the form

Next, we multiply by the value -1 those rows of the matrix (4) for which the rightmost elements have opposite signs relative to the signs of the elements, which are located in those rows where they have the same signs on both sides of the rows. The result is a matrix ΔK _{P =} , which has the same values and signs at the rightmost elements and which has the form

Similarly, by summing the elements of the columns of the matrix ΔK _P , which also corresponds to the accumulation of the processing results, we obtain the row matrix ΔK _PC , in which the right elements are relatively central, the central element and the element describing the BL and adjacent to the central element on the left are not zero, but all elements located further to the left are zero, and which, as a result, takes the form

Thus, two response groups are formed, the location of which at the same positions on the time axis is described by expressions (5) and (6). Rewriting (5) and (6) in the form of diagonal matrices whose main diagonals are elements of the row matrices ΔK _LS and ΔK _PS, respectively, and multiplying these diagonal matrices by each other, we obtain a diagonal matrix of the form

It can be seen from the last expression that the elements located on the main diagonal describing the location of the central peak and BL located to the left and right relative to the central are not non-zero.

 Obviously, in the absence of a Doppler shift in the correlated noise, the correlation coefficient of which tends to unity, the main peaks and the corresponding BLs are completely subtracted.

 In the case of a Doppler shift, which corresponds to a signal of a moving target, due to the opposite nature of the frequency-time dependences of the uncertainty bodies (VTs) of the corresponding discrete, these signals are not subtracted from each other and are extracted.

Аналогичным образом на вход блока СФ одиночного ФМн сигнала (соответствующим образом настроенного) поступает последовательность, состоящая из отсчетов принадлежащих БЛ сжатых дискретов ФМн сигнала и главным пикам этих дискретов, знаки которых соответствуют кодам обрабатываемого сигнала и соответствующим образом совпадают. В этом случае, матрицу X можно представить в виде X = X_н + X_ч, где X_н описывает БЛ сжатых дискретов, X_ч описывает главные пики сжатых дискретов, причем в первой матрице нулевые элементы соответствуют главным пикам сжатых дискретов, а во второй матрице нулевые элементы соответствуют БЛ сжатых дискретов. Поскольку знаки не нулевых элементов крайних не нулевых столбцов обеих матриц совпадают, то без потери общности можно рассматривать только вторую матрицу X_ч.Next, we consider the proposed method as an example of an H-matrix, which we denote by H _8/16

Similarly, a sequence consisting of samples of the compressed BLs of the compressed QPSK signals and the main peaks of these samples whose signs correspond to the codes of the processed signal and correspondingly correspond to the input of the SF block of a single QPSK signal (appropriately tuned) is received. In this case, the matrix X can be represented as X = X _n + X _h , where X _n describes the BL of compressed samples, X _h describes the main peaks of compressed samples, and in the first matrix, the zero elements correspond to the main peaks of compressed samples, and in the second matrix zero elements correspond to BL compressed discrete. Since the signs of the non-zero elements of the extreme non-zero columns of both matrices coincide, without loss of generality we can consider only the second matrix X _h .

Нулевые столбцы в данной матрице можно опустить, т.к. они не оказывают влияния на дальнейший ход рассуждения, поскольку как было показано выше, знаки у элементов крайних столбцов матрицы X_н совпадают со знаками элементов крайних столбцов матрицы X_ч.

Zero columns in this matrix can be omitted since they do not affect the further course of the argument, since, as was shown above, the signs of the elements of the extreme columns of the matrix X _n coincide with the signs of the elements of the extreme columns of the matrix X _h .

Следовательно матрица откликов на сигналы X'_ч, описываемая матрицей автокорреляционных функций, будет иметь вид

Аналогичным образом, производя поэлементное вычитание из нечетных строк матрицы (8) ее четных строк, получим матрицу

, которая имеет вид

Аналогично, как было показано выше, выделяем разностные строки матрицы ΔK′, у которой знаки обоих крайних разностных элементов в строках положительны, и в дальнейшей обработке используются без изменений.Then the matrix X _h can be rewritten as a matrix X ' _h

Therefore, the matrix of responses to signals X ' _h , described by the matrix of autocorrelation functions, will have the form

Similarly, performing elementwise subtraction from the odd rows of the matrix (8) of its even rows, we obtain the matrix

which has the form

Similarly, as shown above, we distinguish the difference rows of the matrix ΔK ′, in which the signs of both extreme difference elements in the rows are positive, and are used without further changes in further processing.

, которая имеет вид

Аналогично, как описано выше, производится суммирование элементов столбцов матрицы

и получается матрица-строка

, имеющая вид

Далее умножаем на величину -1 те строки матрицы (9), у которых правые крайние элементы имеют противоположные знаки относительно знаков элементов, которые расположены в тех ее строках, где они имеют одинаковые знаки с обеих сторон строк. В результате получается матрица

, которая имеет одинаковые величины и знаки у правых крайних элементов и которая имеет вид

Аналогичным образом, описанным выше, проводим суммирование элементов столбцов матрицы

и получаем матрицу-строку

, имеющую вид

Таким образом, сформированы две группы откликов, расположение которых на одноименных позициях на временной оси описывается выражениями (10) и (11).Then, by multiplying all the elements of the corresponding rows by -1, positive signs are set for those elements of the leftmost column of the matrix (9) whose signs were negative. As a result, we obtain the matrix

which has the form

Similarly, as described above, the summation of the elements of the matrix columns

and we get a row matrix

having the form

Next, we multiply by the value -1 those rows of the matrix (9), for which the rightmost elements have opposite signs relative to the signs of the elements, which are located in those rows where they have the same signs on both sides of the rows. The result is a matrix

which has the same values and signs at the right extreme elements and which has the form

Similarly, as described above, we summarize the elements of the matrix columns

and get the row matrix

having the form

Thus, two response groups are formed, the location of which at the same positions on the time axis is described by expressions (10) and (11).

Из последнего выражения видно, что не равен нулю элемент, расположенный на главной диагонали, описывающий местоположение центрального пика.Rewriting (10) and (11) in the form of diagonal matrices, as shown above, and multiplying, we obtain a diagonal matrix of the form

The last expression shows that the element located on the main diagonal describing the location of the central peak is not non-zero.

 It is also obvious that the positions located next to the main peak and describing the BL of compressed discs will not vanish in the same way as shown in (7).

 Obviously, in the absence of a Doppler shift in the correlated noise, the correlation coefficient of which tends to unity, the main peaks and BL are subtracted completely.

 In the case of the presence of a Doppler shift, which corresponds to the presence of a moving target signal, due to the opposite nature of the time-frequency dependences of the VTs of the corresponding discrete, these signals are not completely subtracted from each other and are extracted.

 Similarly, results can be obtained for signal systems described by H-matrices of other sizes.

 In contrast to the method of sensing by the FMN system the signals with intra-discrete modulation described in RF Patent N 2107926. The method of pulsed radar system of phase-shifted signals / L.V. Lityuk. Publ. in bull. fig. - 1998. N 9. - p. 377, the proposed method allows you to highlight the signals of moving targets on the background of reflections from the earth's surface.

 The analysis of the proposed method for sensing space and comparing it with analogues and prototype allows us to conclude that the proposed method meets the criteria of "novelty", "inventive step", "industrial applicability".

 In FIG. 1 shows a structural diagram of a device that implements the proposed method; in FIG. 2 - time diagrams of codes generated by the block of digital formation of the PSK signal; in FIG. 3 - time diagrams in the form of quadrature components of the PSK signal at the output of the modulation block; in FIG. 4 - time diagrams in the form of quadrature components of the PSK at the output of the block for the formation of digital quadrature components 4 with Doppler shift; in FIG. 5 is a timing diagram in the form of quadrature components of the PSK at the output of the block for generating digital quadrature components 4 without a Doppler shift; in FIG. 6. - timing diagrams at the output of the adder (subtractor) 9 with Doppler shift; in FIG. 7. - timing diagrams at the output of the adder (subtractor) 9 without Doppler shift; in FIG. 8 is a timing chart at the output of a matched filter QPSK signal 10 with a Doppler shift; in FIG. 9 is a timing chart at the output of a matched filter of QPSK signal 10 without Doppler shift; in FIG. 10 - the resulting response of the processing of a packet consisting of eight signals at the output of the device in the presence of a Doppler shift; in FIG. 11 - the resulting response of the processing of the packet, consisting of eight signals at the output of the device in the absence of Doppler shift; in FIG. 12 - system responses to signals of moving targets having a positive Doppler shift; in FIG. 13 - system responses to signals of moving targets having a negative Doppler shift.

A device that implements the proposed method (Fig. 1) contains: an antenna (A) 1, the output of which through the antenna switch (AP) 2 is connected to the input of the radio receiving device (RPpU) 3, the output of which is connected to the input of the digital quadrature component generation unit (BOCKS) ) 4, the output of which is connected to the corresponding inputs of two parallel-connected channels, each of which consists of sequentially connected matched discrete filters (SFD) 5 and 6, the outputs of which are connected through the envelope allocation blocks (BVO) 7 and 8 to the inputs and an adder (subtractor) 9 (Σ9), the output of which is connected to a matched filter of a single phase-manipulated signal (SFOFMnS) 10, consisting of a digital delay line (DLC) 11 with N taps, each of which through a corresponding tunable weight coefficient block (BPVK) 12 connected to the corresponding input of the adder 13 (Σ13), the output of which is the output of the SFOFMnS 10, and which is connected to the block of selection of signals of moving targets (BVDC) 14, consisting of a switching unit (BC) 15, the input of which is connected to the output of the SFOFMnS 10, one of in moves BC 15 suitably connected to TSLZ 16 with a delay time equal to T _n, a second output is connected to one input of adder 17 (Σ17), and TSLZ output 16 connected to another input Σ17 and whose output is connected to the respective inputs of two parallel-connected channels, each of which consists of series-connected multipliers 18 (Umn 18) and 19 (Umn 19), the outputs of which are connected to the DLC 20 and DLC 21 with a delay time equal to NT _n each, and with three-input adders Σ22 and Σ23, respectively, and TsLZ outputs 22 and 23 are respectively connected to orymi input adder Σ22, and Σ23, and whose outputs are connected to the inputs TSLZ 24 and TSLZ 25 with a delay time equal to 2T _n each, and the inputs TC 26 and TC 27, whose outputs are appropriately connected to a multiplier 28 (Math 28), the output which is the output of the device, the outputs of the TsLZ 24 and TsLZ 25 are connected respectively to the third inputs Σ22 and Σ23, and the control inputs of all BPVK 12 are connected to the corresponding outputs of the digital phase-shift signal generating unit (BTsFMnS) 29 connected to the control input with the corresponding the output output of the synchronization unit (BS) 30, the second output of which is connected to the control input of the AP 2, the second input of which is connected to the output of the radio transmitting device (RPU) 32, connected through the input through the modulation unit (BM) 31 to the output of the BCFFMnS 29, and BS 30 connected to the corresponding control inputs BK 15, Umn 18 and Umn 19, BK 26 and BK 27, and BS 30 is connected to the corresponding outputs of BOTSKS 4, with blocks SFD5 and SFD 6, BVO 7 and BVO 8, TsLZ 11, TsLZ 16, TsLZ 20 and TsLZ 21, TsLZ 24 and TsLZ 25 and which in FIG. 1 are not shown.

A device that implements the proposed method and shown in FIG. 1, works as follows. At time t = t ₀ , AP 2, controlled by a signal from the second input of BS 30, connects antenna A 1 to the output of the RPU 32. At the same time, the control signal from the first input of BS 30 is fed to BTsFMnS 29, which generates the corresponding code PSK signal (for example, let the first signal be a signal having a code: 1, 1, -1, 1). The signals from the outputs of the BCFFMnS 29 arrive at BPVK 12 and set the corresponding weight coefficients in them. Simultaneously upon the output of the BCFFMnS 29, the generated code, bit by bit, is sequentially delivered to BM 31 in time. The time duration of each bit in the code is τ _{d =} . For the duration of this time interval equal to τ _d , a signal with a continuous phase function is generated in the BM 31, the law of change of which is determined by the parameters of the QPSK code of the signal coming from the output of the BCFFMnS 29. The complex QPSK signal generated in BM 31 in this way is fed to RPU 32 and modulates accordingly, the carrier frequency and through AP 2 is emitted A 1 into space. In FIG. 2a shows a signal in the form of a four-digit code coming from the output of the BTCPFMnS 29, and in FIG. 3a shows the quadrature components of the complex envelope of this emitted signal. At the time t = t ₁ = 4τ _d , corresponding to the time of the end of the radiation of the generated probing signal, BS 30 gives a control signal, by which the AP 2 connects the antenna A 1 to the input of the receiver 3. The device starts to receive.

где М - определяет количество отсчетов квадратурных составляющих на одном дискрете, N - число, определяющее величину бина, n - номер текущего отсчета. Обычно полагают М << N.Let at the time t = t ₂ ≥ t _{1 the} reflected signal comes to the input of the device. This signal, passing through A 1 and AP 2, will go to the RPrU 3, where it will be amplified accordingly and transferred to the intermediate frequency (IF). From the output of the inverter amplifier (IF), the received signal will go to the input of BOTSKS 4, at the output of which the quadrature components of the complex envelope of the received signal with Doppler shift (Fig. 4a) will appear, without Doppler shift (Fig. 5a). These quadrature components are multi-bit readings, the repetition rate of which is determined by the conditions of the Kotelnikov theorem. Samples of the quadrature components of the complex signal are fed to the SFD 5 and SFD 6, where there is a coordinated filtering of individual discrete received signals. In that case, if the used laws of intra-discrete modulation by a continuous FM signal are orthogonal, then by the end of each discrete at the outputs of the SFD 5 and SFD 6 there will be a complete separation of these signals. If the quasi-orthogonal laws of intra-discrete modulation by a continuous FM signal are used, the outputs of that SFD that is currently not consistent for the received signal will have an output response that is not equal to zero and proportional to the value of the convolution integral of the received discrete with the impulse response of this filter . Appropriately processed discrete samples in SFD 5 and SFD 6 are supplied to BWO 7 and BVO 8, at the outputs of which the envelopes of the transmitted SFD 5 and SFD 6 signals are highlighted. Let the laws of change in the frequency of individual discretes be described by the expressions:

where M - determines the number of samples of quadrature components in one discrete, N - the number that determines the bin, n - the number of the current sample. Usually put M << N.

 These two laws describe signals whose full phase functions are cubic parabolas with opposite laws of frequency change. Similar types of FM can be considered as quasi-orthogonal.

From the outputs of the BWO 7 and the BWO 8, the compressed signals are fed to an adder (subtractor) 9, at the output of which a signal is generated, the location and peak signs of which correspond to the PSK code generated by BTCPFMnS 29. In FIG. 6a shows the signal obtained at the output Σ9 in the presence of a Doppler shift, and in FIG. 7a - in the absence of a Doppler shift. The signal generated in this way enters the SFOFMnS 10 and is recorded in the CLZ 11, the input of which is the input of this filter. Appropriately processed in this filter, which is consistent for a single PSK signal, at the output Σ13, which is the output of this filter, a compressed PSK signal appears, which has the form shown in FIG. 8a (with Doppler shift) and in FIG. 9a (without Doppler shift). This signal is fed to the input of the BK 15, which is the input of the BVDC 14 and is controlled by the BS 30, and from the output of which the signal is fed to the CPL 16 with a delay time equal to T _p .

At time t = t ₀ + T _{p, in the} same manner as described above, the code of the second PSK signal is generated, having a maximum Hamming distance relative to the first. This code will be: -1, -1, 1, -1. In FIG. 2b shows a signal in the form of a four-digit code coming from the output of the BTCPFMnS 29, and in FIG. 3b shows the quadrature components of the complex envelope of this emitted signal. At time t = t ₁ + T _{p the} device begins to work on reception.

Let at the time t = t ₂ + T _p ≥ t ₁ + T _{p the} reflected signal comes to the input of the device. This signal is processed in a similar manner as described above, as illustrated in FIG. 4b, 5b, 6b, 7b, 8b, 9b. This signal from the output of Σ13 goes to the input of the BC 15, which is controlled by the BS 30. From the output of the BC 15, the signal goes to Σ17, where it is subtracted with the first signal delayed by T _p . Then, the resulting difference is fed to two controlled BS 30 multipliers Smart 18 and Smart 19, where they are multiplied with +1. From the outputs of these multipliers, the result of the multiplication enters the first channel on the DLC 20 with a delay time of 8T _p and Σ22, and in the second channel it arrives on the DLC 21 with a delay time of 8T _p and Σ23. These difference signals, passing through Σ22 and Σ23, arrive at the CLZ 24 and CLZ 25 with time
delays 2T _p and at the same time come to BK 26 and BK 27 in the first and second channels, respectively. The signals of the first and second channels do not pass to the output, because this is due to the fact that BK 26 and BK 27 controlled by BS 30 are turned off, because the processing of the entire generated and received burst of pulses has not been completed and can be called transitional.

At the time t = t ₀ + 2T _{p, in the} same manner as described above, the QPSK code of the signal is generated (for example, let the third signal be a signal having the code: 1, -1, 1, 1). In FIG. 2c shows a signal in the form of a four-digit code coming from the output of the BTCPFMnS 29, and in FIG. 3c shows the quadrature components of the complex envelope of this emitted signal. At time t = t ₁ + 2T _p , the device begins to work on reception.

Let at the time t = t ₂ + 2T _p ≥ t ₁ + 2T _{p the} reflected signal comes to the input of the device. This signal is processed in a similar manner as described above, as illustrated in FIG. 4c, 5c, 6c, 7c, 8c, 9c. This signal is fed to the input of the BK 15, which is the input of the BVDC 14 and is controlled by the BS 30, and from the output of which the signal is fed to the CPL 16 with a delay time equal to T _p .

At time t = t ₀ + 3T _{p, in the} same manner as described above, the code of the fourth PSK signal is generated, having a maximum Hamming distance relative to the third. This code will be: -1, 1, -1, -1. In FIG. 2g shows a signal in the form of a four-digit code coming from the output of the BTCPFMnS 29, and in FIG. 3d shows the quadrature components of the complex envelope of this emitted signal. At time t = t ₁ + 3T _{p, the} device starts to receive.

Let at the time t = t ₂ + 3T _p ≥ t ₁ + 3T _{p the} reflected signal comes to the input of the device. This signal is processed in a similar manner as described above, as illustrated in FIG. 4g, 5g, 6g, 7g, 8g, 9g. This signal from the output of Σ13 goes to the input of the BC 15, which is controlled by the BS 30. From the output of the BC 15, the signal goes to Σ17, where it is subtracted with a third signal delayed by T _p . Then, the resulting difference is fed to two controlled BS 30 multipliers Smart 18 and Smart 19, where they are multiplied with +1. From the outputs of these multipliers, the result of the multiplication enters the first channel on the DLC 20 with a delay time of 8T _p and Σ22, and in the second channel it arrives on the DLC 21 with a delay time of 8T _p and Σ23. These difference signals, passing through Σ22 and Σ23, arrive at the DLC 24 and DLC 25 with a delay time of 2T _p and at the same time arrive at BK 26 and BK 27 in the first and second channels, respectively. The signals of the first and second channels do not pass to the output, because this is due to the fact that BK 26 and BK 27 controlled by BS 30 are turned off, because the processing of the entire generated and received burst of pulses has not been completed and can be called transitional.

At the time t = t ₀ + 4T _{p, in the} same way as described above, the code of the fifth QPSK signal is generated (for example, let the fifth signal be a signal having the code: -1, 1, 1, 1). In FIG. 2d shows a signal in the form of a four-digit code coming from the output of the BTCPFMnS 29, and in FIG. 3d shows the quadrature components of the complex envelope of this emitted signal. At time t = t ₁ + 4T _{p, the} device begins to work on reception.

Let at the time t = t ₂ + 4T _p ≥ t ₁ + 4T _{p the} reflected signal comes to the input of the device. This signal is processed in a similar manner as described above, as illustrated in FIG. 4d, 5d, 6d, 7d, 8d, 9d. This signal is fed to the input of the BK 15, which is the input of the BVDC 14 and is controlled by the BS 30 and from the output of which the signal is fed to the CLZ 16 with a delay time equal to T _p .

At the time t = t ₀ + 5T _{p, in the} same way as described above, the sixth PSK signal code is generated having the maximum Hamming distance relative to the fifth. Such a code would be: 1, -1, -1, -1. In FIG. 2e shows a signal in the form of a four-digit code coming from the output of the BTCPFMnS 29, and in FIG. 3e shows the quadrature components of the complex envelope of this emitted signal. At time t = t ₁ + 5T _{p, the} device begins to work on reception.

Let at the time t = t ₂ + 5T _p ≥ t ₁ + 5T _{p the} reflected signal comes to the input of the device. This signal is processed in a similar manner as described above, as illustrated in FIG. 4th, 5th, 6th, 7th, 8th, 9th. This signal from the output of Σ13 goes to the input of the BC 15, which is controlled by the BS 30. From the output of the BC 15, the signal goes to Σ17, where it is subtracted with the fifth signal delayed by T _p . Then, the resulting difference is fed to two controlled BS 30 multipliers Smart 18 and Smart 19, where they are multiplied with +1 and -1, respectively. From the outputs of these multipliers, the result of the multiplication enters the first channel on the DLC 20 with a delay time of 8T _p and Σ22, and in the second channel it arrives on the DLC 21 with a delay time of 8T _p and Σ23. These difference signals, passing through Σ22 and Σ23, arrive at the DLC 24 and DLC 25 with a delay time of 2T _p and at the same time arrive at BK 26 and BK 27 in the first and second channels, respectively. The signals of the first and second channels do not pass to the output, because this is due to the fact that BK 26 and BK 27 controlled by BS 30 are turned off, because the processing of the entire generated and received burst of pulses has not been completed and can be called transitional.

At the time t = t ₀ + 6T _{p, in the} same way as described above, the code of the seventh PSK signal is generated (for example, let the seventh signal be a signal having the code: 1, 1, 1, -1). In FIG. 2g shows a signal in the form of a four-digit code coming from the output of the BTCPFMnS 29, and in FIG. Figure 3g shows the quadrature components of the complex envelope of this emitted signal. At time t = t ₁ + 6T _{p the} device begins to work on reception.

Let at the time t = t ₂ + 6T _p ≥ t ₁ + 6T _{p the} reflected signal comes to the input of the device. This signal is processed in a similar manner as described above, as illustrated in FIG. 4zh, 5zh, 6zh, 7zh, 8zh, 9zh. This signal is fed to the input of the BK 15, which is the input of the BVDC 14 and is controlled by the BS 30 and from the output of which the signal is fed to the CLZ 16 with a delay time equal to T _p .

At the time t = t ₀ + 7T _{p, in the} same way as described above, the code of the eighth QPSK signal is generated, having a maximum Hamming distance relative to the seventh. Such a code would be: -1, -1, -1, 1. In FIG. 2h shows a signal in the form of a four-digit code coming from the output of the BTCPFMnS 29, and in FIG. Figure 3c shows the quadrature components of the complex envelope of this emitted signal. At time t = t ₁ + 7T _{p, the} device begins to work on reception.

Let at the time t = t ₂ + 7T _p ≥ t ₁ + 7T _{p the} reflected signal comes to the input of the device. This signal is processed in a similar manner as described above, as illustrated in FIG. 4z, 5z, 6z, 7z, 8z, 9z. This signal from the output of Σ13 is fed to the input of the BC 15, which is controlled by the BS 30. From the output of the BC 15, the signal goes to Σ17, where it is subtracted with the seventh signal delayed by T _p . Then, the resulting difference is fed to two controlled BS 30 multipliers Smart 18 and Smart 19, where they are multiplied with -1 and +1, respectively. From the outputs of these multipliers, the result of the multiplication enters the first channel on the DLC 20 with a delay time of 8T _p and Σ22, and in the second channel it arrives on the DLC 21 with a delay time of 8T _p and Σ23. These difference signals, passing through Σ22 and Σ23, are supplied to the DLC 24 and DLC 25 with a delay time of 2T _p and at the same time, the accumulated difference signals arrive through the appropriately enabled BS 30 BK 26 and BK 27 to Umn 28. At the output of Umn 28, the response shown in FIG. 10 (there is a Doppler shift in the received signals), and in FIG. 11 (no Doppler shift). Subsequently, BC 26 and BC 27 open BS 30 in even periods of sounding for the duration of the period and close for odd ones.

At time t = t ₀ + 8T _{p, in the} same way as described above, the code of the ninth PSK signal is generated, which has the same code as the first: 1, 1, -1, 1. At time t = t ₂ + 8T _{n the} device starts to receive.

Let at the time t = t ₁ + 8T _p ≥ t ₁ + 8T _{p the} reflected signal comes to the input of the device. This signal is processed similarly to the first signal, as described above.

At the time t = t ₀ + 9T _{p, in the} same way as described above, the code of the tenth QPSK signal is generated, which has the same code as the second one having a maximum Hamming distance relative to the ninth. This code will be: -1, -1, 1, -1. At time t = t ₁ + 9T _{p, the} device begins to work on reception.

Let at the time t = t ₂ + 9T _p ≥ t ₁ + 9T _{p the} reflected signal comes to the input of the device. This signal is processed similarly to the second signal, as described above. This signal from the output of Σ13 goes to the input of the BC 15, which is controlled by the BS 30. From the output of the BC 15, the signal goes to Σ17, where it is subtracted with the ninth signal delayed by T _p . Then, the resulting difference is fed to two controlled BS 30 multipliers Smart 18 and Smart 19, where they are multiplied with +1. The difference signals that were recorded in DLC 20 and DLC 21 8T _p are fed back to the corresponding inputs of the adders Σ22 and Σ23 and thereby are subtracted from the resulting signals located in DLC 24 and DPC 25. From the outputs of Smart 18 and Smart 19 results multiplications are received in the first channel on the DLC 20 with a delay time of 8T _p and Σ22, and in the second channel they arrive on the DLC 21 with a delay time of 8T _p and Σ23, which will ensure the operation mode of the device with a sliding window. These difference signals, passing through Σ22 and Σ23, arrive at the CLZ 24 and CLZ 25 with a delay time of 2T _p and at the same moment of time they arrive through the appropriately switched on BS 30 BK 26 and BK 27 on Umn 28. Next, the operation of the device is repeated.

In the case of working with other codes (for example, eight-bit), the values of the corresponding codes will be repeated after 16T _p , and SFOFMnM 29 should be consistent with the eight-bit code. Similarly, you need to change the characteristics of the device when changing types of codes.

Regardless of the sign of the Doppler frequency offset of the signal reflected from the moving target, the responses at the output of the device will be the same at the same target speeds, as illustrated in FIG. 12 and 13. It can be seen from these figures that the BL of compressed complex signals, regardless of the magnitude and sign of the Doppler shift, are compensated on the plane (τ, F) except for the region ± τ _d relative to the point τ = 0. All the figures presented were obtained by modeling on a digital computer described processing algorithm with corresponding signals.

 A technical and economic comparison of the proposed method with the prototype shows that the proposed method allows to distinguish signals of moving targets when using systems of complex phase-shift keyed signals with intra-discrete modulation with simultaneous compensation of BL in the region of BL occupied by individual complex signals while maintaining high range resolution. It is also seen that, compared with the prototype, the proposed method allows to determine the speed of the target by measuring the distance between the peaks of the signal.

Claims (1)

 A method for selecting moving targets, which consists in generating, emitting, and accepting a system of phase-manipulated signals with in-disc frequency modulation, each discrete of a phase-manipulated signal emitted by the code dimension N of each of a sequence of phase-manipulated signals consisting of 2N signals, subjected to in-disc frequency modulation, the law of change which is determined by the sign of the phase-manipulated signal code, the laws of change in the frequency of the in-discrete frequency modulation are tohogonal or quasi-orthogonal, the range of variation of these frequencies is the same for all samples and all signals of the sequence have the same carrier frequency, and upon reception, each sample with its own law of in-discrete frequency modulation of a phase-shifted signal with code dimension N is processed in a system of matched filters, each of which is consistent with the law changes in the frequency of the corresponding phase-shift signal, the sequence of envelopes whose signs correspond to the code of the corresponding phase-manipulated signal is filtered in an appropriate matched filter of the phase-manipulated signal, characterized in that the phase-manipulated signals with in-discrete frequency modulation are emitted sequentially in time so that the odd numbers of the signals in this sequence have a maximum Hamming code distance relative to the even signals following them, their results the processing in the corresponding matched filters is subtracted from each other, the results of the subtraction accumulate two groups, and accumulation is carried out so that in one group of accumulated differences all the first elements are positive, and in the other group of accumulated differences all the last elements are positive, or in one group of accumulated differences all the first elements are negative, and in the other group of accumulated differences all the last elements were negative, the obtained groups of accumulated differences are appropriately delayed and synchronized, and the obtained summation results in two groups are synchronized of each other are multiplied in the time interval whose duration is determined by the repetition period of the processed sequence.

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