RU2109384C1 - Method of formation of coherent optical signal by summation of radiation beams of n lasers in vertex of conical surface and transmitter of coherent optical radiation realizing this method - Google Patents

Abstract

FIELD: radio engineering, formation of beams of coherent radiation with high density of power. SUBSTANCE: for obtainment of coherent summation of radiation beams of lasers frequencies of summed beams are leveled by heterodyning of parts of oscillations of each laser and of oscillations of reference laser, by isolation of difference frequencies from obtained spectrum of oscillations, by determination of values of these frequencies, by synthesizing N signals of acoustooptical modulation with obtained codes of difference frequencies, by modulation of elastic optically transparent media with these signals, by Bragg diffraction of laser oscillations illuminating these media and by Doppler shift of frequencies of oscillations of lasers to frequency of oscillations of reference laser. Phases of summed beams modulated acoustooptically are leveled by phase modulation of oscillations of each of N lasers with signal which number of steps amounts to $$$, by phase modulation of each of synthesized oscillations with signal which number of steps amounts to $$$, by heterodyning of parts of acoustooptically modulated oscillations of each of summed beams and parts of oscillations of same reference laser, by extraction of oscillations of signals of frequency $$$ from obtained spectrum, by phase detection of each of extracted signals of frequency $$$ with reference signal of same frequency and by addition of value $$$ to phase of each of synthesized oscillations. EFFECT: enhanced reliability of coherent summation of radiation beams. 10 cl, 27 dwg

Description

 The invention relates to the field of radio engineering and can be used to create beams of coherent radiation with a high power density.

 To create high power density interference to devices using laser radiation, a method of summing the radiation of N lasers at one point can be used, the lasers being located on a conical surface so that the optical axes of the laser radiation beams intersect at the top of the conical surface, and from the summing point, the total power is channeled. This method is so well known that it is practically not possible to refer to the source of information where it is mentioned. Nevertheless, this method, being the closest in its technical essence to the proposed invention, is taken as a prototype.

 The disadvantage of the prototype is that the summation taking place in it is incoherent, because each i-beam has both different frequencies and phases compared to other beams. For this reason, the energy potential at the summation point is proportional to the first power of the number of sources to be summed. The energy potential E (φ, θ) is defined as the product of all radiated power by the gain of the radiating system. With a coherent summation of N sources, the energy potential is proportional to the square of the number of summed sources. [1, p. 85].

E (φ, θ) = P _o g _or (φ, θ) • N ² ,
Where
P ₀ is the power of one source;
g _or (φ, θ) is the realized gain of the ith source in the direction φ, θ as part of the emitting system [2, p. 376].

 The purpose of the invention is to generate a coherent optical signal by summing the laser radiation beams, i.e. get a beam with a high power density in it.

пропорциональной величине соответствующего продетектированного напряжения, определяемой разностью фаз между выделенным колебанием частоты F_n и опорным колебанием той же частоты, где T_p - период функции с числом ступеней 2^p; ΔT_pij -временная задержка между i-й и j-й функциями с числом ступеней 2^p.This goal is achieved in that the frequencies of the summed radiation beams of N lasers are aligned by heterodyning the parts of the oscillations of each of the N lasers and the oscillations of the reference laser, isolating the signals of the "difference" frequencies (products of heterodyning) between the oscillations of each of the N lasers and the reference oscillations, determining and refining the digital the code of each of N difference frequencies, synthesizing N acousto-optic modulation signals with digital codes, and excite each i-th signal synthesized by the i-th ultrasonic wave waves in the ith elastic optically transparent medium, Bragg diffraction of the ith laser oscillations on the inhomogeneities of the ith elastic optically transparent medium due to the passage of the ith ultrasonic wave, and the shift of the ith laser oscillation frequency to the oscillation frequency of the reference laser, phase the summed acousto-optically modulated beams are aligned by modulating the oscillation phase of each of the N function lasers, the number of steps is 2 ⁿ , where n is the number of the smallest discrete state of the function φ _{n min} = π / 2 ⁿ , the period of the function is T _n = 1 / F _n module tion phase of each of the N synthesized oscillation function, the number of steps which is 2 ^p, where p - number of the smallest increment state function φ _{p min} = π / 2 ^p, the period function equal to T _p = 1 / F _p, heterodyning parts acoustooptic modulated oscillations of each of the summed beams and vibrations of the reference laser, isolation of frequency F _n signals from the heterodyning products, phase detection of each of the extracted signals of frequency F _n with a reference signal of the same frequency, introducing into the phase of each of the synthesized oscillations (signal acousto-optical modulation)

proportional to the value of the detected voltage detected, determined by the phase difference between the selected frequency oscillation F _n and the reference oscillation of the same frequency, where T _p is the period of the function with the number of steps 2 ^p ; ΔT _{pij is the} time delay between the ith and jth functions with the number of steps 2 ^p .

 The method for determining the code of the i-th difference frequency based on the excitation of the i-th ultrasonic wave by the difference frequency of the i-th ultrasonic wave in an elastic optically transparent medium, illumination of this medium with a coherent light wave, diffraction of this wave by inhomogeneities of the physical medium due to the passage of the i-th ultrasonic waves, focusing the first-order diffraction maximum of the diffracted light wave onto the image plane, converting optical signals focused on the image plane into electrical and determining the frequency code of the signal of the i-th difference frequency according to the location of the amplitude of the excited electric signal in the image plane, while in order to improve the accuracy of determining the frequency value of the signal of the differential frequency, the amplitudes obtained as a result of photoelectric conversion at K equal levels (K = 1, 2 ...), starting from the first, lying in the range from zero to minus 3 dB, with 0 dB coinciding with the maximum value of the amplitude of the i-th photoelectrically converted signal, and the value of the signal frequency code the difference frequency domain is determined by averaging the amplitude of the i-th photoelectrically converted differential frequency signal measured at K levels, the averaging results are presented in the form of a code containing integer and fractional parts, the integer part of the code corresponding to either the nearest maximum of the i-th photoelectrically converted amplitude of the difference frequency signal or to the nearest center of the half-distance between the two nearest maxima of the i-th and i + 1-st (i-1, i-th) photoelectrically converted equal-amplitude difference signals frequency, and the fractional part of the code corresponds to the average value of the frequency of the differential frequency signal, which is located between these two points.

The refinement of the frequency code of the i-th difference frequency signal is achieved by converting the fractional part of the frequency code of this signal into a frequency mismatch code, the direction of the maximum of the optical signal in the image plane is moved until this maximum coincides or with the direction corresponding to the closest maximum of the i-th photoelectrically converted signal in the image plane, or with a direction corresponding to the nearest half-center between the ith and (i + 1) -m (i-1, i-m) equipotential photoelectric converters ovannymi signals in the image plane by the frequency transfer difference frequency signal "up", "down" frequency by modulating the phase i-th difference frequency signal is a function of the number of steps which is 2 ^q, where q - number smallest increment state function with the number of steps 2 ^q φ _{q min} = π / 2 ^q , the period of the function with the number of steps 2 ^{q is} synthesized by the frequency mismatch code.

 After aligning the frequencies and oscillations of N laser radiation beams (N diffraction maxima of the first order), these beams are summed and channelized.

где λ - длина волны света в материале звукопровода;
Λ - длина акустической волны.The coherent summation of the radiation of N lasers is based on the phenomenon of acoustic Bragg diffraction [3, p. 324-336], the essence of which is illustrated in Fig. 1, where light beams 1 of frequency ν are shown, directed to a sound duct 2, in which a high-frequency (frequency ω) ultrasonic wave 3 propagates. At certain critical angles of incidence ± α _, an additional light beam excites an additional beam whose propagation direction differs from the incident beam by 2α _in . The angle α _in , called the Bragg angle, is determined by the relation

where λ is the wavelength of light in the material of the sound duct;
Λ is the acoustic wavelength.

 If the light beam falls at an obtuse angle to the direction of propagation of the sound wave fronts, an obtuse angle is also formed between the direction of propagation of the sound wave and the diffracted light beam (upper part of Fig. 1). The diffracted beam, called in this case minus the first order, turns out to be shifted downward in frequency by the frequency ω (negative Doppler shift). The lower part of figure 1 illustrates the Bragg diffraction with a frequency shift up, in this case, the diffracted beam is called plus the 1st order. It should be noted that the incident radiation can be completely deflected by the Bragg angle [3, p.331], i.e. almost all incident radiation diffracts into one diffraction maximum.

если
λ = 1 мкм, то ν = 3•10⁸ мГц.The laser frequency ν in Hz is related to the wavelength λ in μm by the following relation:

if
λ = 1 μm, then ν = 3 • 10 ⁸ MHz.

As experience shows, the spread of laser frequencies Δν _rel (relative instability of the radiation frequency) is [4, p. 156-227] Δν _rel = 10 ^-6 ... 10 ^-15 .

For values Δν _rel = 10 ^-5 ... 10 ^{-6 the} spread Δf will be within
Δf = 30 ... 300 MHz.

To equalize the oscillation frequencies of the summed beams with respect to the frequency of the signal of the reference beam, it is necessary to determine the difference frequencies between the frequencies of each of the summed beams and the frequency of the reference beam. Then, with the signal of the ith differential frequency, excite the ith acousto-optic modulator (AOM), which transfers the frequency of the ith beam to the frequency of the reference beam. In this case, the frequency differences of the signals of each of the lasers relative to the reference beam should not exceed the frequency band of the AOM. If the frequency of oscillations of the reference laser is ν ₀ , then (Fig. 2, a, b)
ν ₁ -ν ₀ = Δf ₁ ; ν _i -ν ₀ = Δf _i ; ν _N -ν ₀ = Δf _N.
In order for all the frequencies of the summed beams to be equal to the frequency ν ₀ , these frequencies must in this case be moved "up" to the values Δf ₁ ... Δf _i ... Δf _N (Fig. 2, c, d, e), which achieved by excitation of ultrasonic waves of the corresponding frequencies in the corresponding AOM, diffraction of laser beams on these waves and the alignment of the frequencies of their oscillations relative to the frequency of the reference beam.

Thus, to equalize the radiation frequencies of N lasers, it is necessary:
1. By heterodyning, select the signals of the difference frequencies f _pi of the radiation beams between the ith and the reference oscillation.

 2. Find the codes of difference frequencies and, if necessary, refine them, for example, using an acousto-optical deflector (AOD).

3. Synthesize signals U _{ci of} frequencies f _ci with difference frequency codes f _pi , with f _pi = f _ci ... f _pN = f _cN .

4. By modulating the AOM _i , which is excited in the Bragg diffraction mode with the signal f _ci, shift the frequency ν _i of the 1st order diffraction maximum (DM1) to the frequency ν _{0 of the} reference laser.

Frequency-aligned DM1 _i must be phase-aligned, which requires the following sequence of actions:
1. The oscillations of each laser are phase-modulated by a function with the number of steps 2 ⁿ [5], where n is the number of the smallest discrete phase state φ _{n min} = π / 2 ⁿ , the period of the function is T _n = 1 / F _n , ie the oscillation of each of the lasers is transferred in frequency by F _n , and the phase of the frequency signal F _n is introduced into the oscillation phase of each of the lasers.

2. Modulate the oscillations of the synthesized frequencies U _ci with a function with the number of steps 2 ^p , where p is the number of the smallest discrete phase state φ _{p min} = π / 2 ^p , the period of the function T _p = 1 / F _p . Thus, the frequencies of the synthesized oscillations are transferred to the same value F _p , and the phase of the oscillation F _p is introduced into the phase, which in turn allows you to introduce the phase of the oscillations of the frequency F _p into the phase of each acousto-optically modulated DM1.

3. Since the phase information contained in DM _i is completely stored in the function of the heterodyne field [6, p. 181], then this information is also stored in the phase of the extracted signals (heterodyning products) of frequency F _n , and when comparing the phase of the extracted signals of frequency F _n with the phase of the reference signal of the same frequency using the phase detection operation, a voltage proportional to the phase difference of the compared oscillations appears.

пропорциональная величине соответствующего напряжения, определяемого разностью фаз между колебаниями соответствующих ДМ_i и колебаниями опорного пучка (лазера), где T_p - период функции с числом ступеней 2^p, ΔT_pij - временная задержка между i-й и j-й функциями с числом ступеней 2^p.4. A quantity is introduced into the phase of each of the synthesized vibrations U _ci

proportional to the corresponding voltage, determined by the phase difference between the oscillations of the corresponding DM _i and the oscillations of the reference beam (laser), where T _p is the period of the function with the number of steps 2 ^p , ΔT _pij is the time delay between the i-th and j-th functions with the number of steps 2 ^p .

 The applicant and the author are not aware of technical solutions containing features equivalent to the distinguishing features of the proposed method, therefore, the proposed method meets the criterion of "novelty" and "significant differences".

 Introduction to the method of new sequences of actions on a material object-signal allows one to produce coherent summation of radiation beams of N lasers and to obtain a coherent beam with a high power density.

 It is a well-known device with lasers located on a conical surface so that the optical axes of the laser beams intersect at the top of the conical surface. The disadvantage of the prototype device is that this device does not allow coherent summation of laser beams, because the frequencies and phases of the summed beams are different; therefore, the summation is incoherent.

The purpose of the invention is to carry out coherent summation of laser radiation beams. This goal is achieved in that for the implementation of the proposed method, a first unit is introduced into the coherent optical radiation transmitter - a device for dividing, summing, aligning frequencies and phases of summed beams with N inputs of modulation frequency signals F _n , N inputs of acousto-optic modulation signals, N outputs of difference frequency signals , N outputs of the signals of the phase-locked loop signals of the summed beams and the output of the total coherent beam, which is the output of the entire device for summing the coherent beams blocks - devices for averaging and updating codes of difference frequencies with N inputs of signals of difference frequencies and N outputs of codes of difference frequencies, N third blocks are introduced - N devices of frequency and phase self-tuning of oscillations of summed beams with N inputs of phase-locked signals of oscillations of summed beams, with N inputs of codes difference-frequency signals and outputs n frequency F _n, n outputs acoustooptic modulation signals, wherein n input signals of the first frequency unit connected to the respective n outputs of frequency F _n signals tre unit, N inputs of acousto-optic modulation signals of the first unit are connected to the corresponding N outputs of acousto-optic modulation signals of the third unit, N outputs of difference frequency signals of the first unit are connected to corresponding N inputs of difference frequency signals of the second unit, N outputs of phase-locked signals of oscillations of the summed beams of the first unit are connected with corresponding N inputs of phase-locked loop signals of oscillations of summed beams of the third block, N outputs of difference frequency codes of the second block connected to the corresponding inputs of the codes of the differential frequencies of the third block.

The introduced device for dividing, summing, aligning the frequencies and phases of the summed beams - the first block - contains N optical channels (N optical inputs) of the reference laser, each of which contains the optically connected i-th translucent face of the N-sided pyramid, i-th first translucent mirror with the i-th mirror of the first i-th mixer with the i-th output of the difference frequency signal, the second i-th translucent mirror of the second i-th mixer with the i-th output of the phase-locked loop signal of the summed beams, contains N optical channels x beams, each of which contains the optically coupled i-th laser, the first i-th light coupler of the first i-th differential frequency mixer, i-th acousto-optic modulator, i-second light coupler of the second i-phase phase locked loop and beam combiner - first-order diffraction peaks - a circular conical horn excited at a wavelength of ₀₁ H, smoothly passing into the circular waveguide with the wave ₀₁ H, which is the output of division device for summing the frequency alignment phase and summed n chkov.

The introduced device for averaging and refinement of the difference frequency code — the second block — contains a filter in series, a phase modulator of the difference frequency signal with a function with the number of steps 2 ^q , where q is the number of the least discrete state of the function φ _{q min} = π / 2 ^q , the amplifier and the deflector contain optically coupled laser with forming optics, a line of photodetectors with integrating optics, contains a device for averaging the difference frequency code with the output of the strobe tracking the frequency code, contains a device for updating the difference code frequency with the output of the frequency correction code and the input of the strobe tracking the frequency code, and the outputs of the photodetector line are connected to the inputs of the averaging device of the differential frequency code, the first output of which is via the refinement device of the difference frequency code, the output of which is through the refinement of the difference frequency code device, the output of the frequency correction code and a control unit modulator phase of the difference frequency signal with the number of function stages 2 ^q modulator connected to the control inputs, the second output code refinement apparatus times ostnoy frequency is the output of the averaging unit and clarify the difference frequency code output strobe frequency tracking code connected with the second input device clarify the difference frequency code.

) соединен соответственно с j-м (j=

) входом мультиплексора, выход первого канала соединен с первым входом сумматора, второй вход которого соединен с выходом мультиплексора, выход сумматора соединен со входом усреднителя, управляющие входы мультиплексора, сумматора и усреднителя соответственно соединены с третьим, четвертым и пятым выходами блока управления и синхронизации, выход усреднителя является выходом блока усреднения кода разностной частоты.The introduced device for averaging the difference frequency code contains L channels, a switch, a multiplexer, an adder and an averager, contains a control and synchronization unit for the device for averaging the difference frequency code with the output of the strobe for tracking the frequency code, and the input of the device for averaging the difference frequency code through a switch, the control input of which is connected to the sixth output of the control and synchronization unit, connected to N outputs of the photodetector line, the output of the switch is connected to L parallel channels, each of which consists of a series connection of an adder and a comparator with averaging, the control inputs of the comparator and adder averaging each channel respectively coupled to first and second outputs and a synchronization control unit, an output of each j-channel (j =

) is connected respectively to the jth (j =

) the input of the multiplexer, the output of the first channel is connected to the first input of the adder, the second input of which is connected to the output of the multiplexer, the output of the adder is connected to the input of the averager, the control inputs of the multiplexer, adder and averager are respectively connected to the third, fourth and fifth outputs of the control and synchronization unit, output the averager is the output of the averaging block of the difference frequency code.

The introduced difference frequency code refinement device comprises a synchronization unit, the input of which is the second input of the frequency code refinement device, a zero encoder, a comparison element, a counter for increasing the integer part of the code by one, an arithmetic-logical unit for calculating the frequency mismatch code, a frequency mismatch code storage counter, synthesizer frequency grids loaded on the counter, read-only memory device for converting a relative frequency code to an absolute code, read-only memory device formations of the frequency mismatch code, an arithmetic-logical unit for calculating the true frequency code, gating the result storage register, the first inputs of the zero encoder, the comparison element, the counter increasing the integer part of the code by one, the arithmetic-logical unit for calculating the frequency mismatch code are connected to the output of the code averaging device differential frequency, the second input of the counter increases the integer part of the code by one connected to the first output of the synchronization unit, and the third with its third output, the second input q the zero encoder is connected to the second output of the synchronization unit, the output of the counter increasing the integer part of the code by one is connected to the second input of the comparison element and the input of the permanent storage device converting the relative frequency code into an absolute code, the output of which is connected to the first input of the arithmetic-logical unit for calculating the true code frequency, the output of the comparison element is connected to the first input of the counter storage frequency mismatch frequency, the second input of which is connected to the output of the arithmetic-logical frequency mismatch code calculation unit, and the third with the second output of the synchronization unit, the output of the frequency mismatch code storage counter is the output of the frequency mismatch code and connected to the input of the frequency grid synthesizer loaded on the counter, the output of which is connected to the second input of the difference frequency signal phase modulator with number of stages 2 ^q, connected to the input read-only memory error code conversion unit frequency, the output of which is connected to the second input arifmetiko- ogicheskogo unit calculating the true code frequency, the output of which is connected to a first input of a gating result storage register, a second input coupled to an output of the encoder zero, and the output of register is the output of the averaging unit and clarify the difference frequency code.

The introduced frequency and phase locked loop of summed beams contains a multi-channel frequency synthesizer connected in series, the input of which is the input of a frequency and phase locked loop of summed beams for a difference frequency code, a synthesized frequency phase modulator with a function with 2 ^p steps, an amplifier, a directional coupler, the output of which is acousto-optic modulation signal output, contains a series-connected mixer, filter, amplifier, phase detector, bl ok, the phase modulator control function with a number of steps 2 ^p , contains a frequency signal generator F _n and a laser radiation modulator, the first input of the mixer being the input of the frequency and phase self-tuning device of the summed beams of the entire device, the second input of the mixer connected to the second output of the directional coupler, the second input a phase detector coupled to the first output of the oscillator frequency signal F _n, a second generator output coupled to an input of the modulator, whose output is the output signal frequency F _n all have troystva.

The control unit for the synthesized frequency signal phase phase modulator introduced by the function with the number of steps 2 ^p and the time delay between functions with the number of steps 2 ^p contains N channels, each of which contains the element NOT connected in series, the AND element, and the counter, the input of the i-th element NOT connected to the i-th output of the phase detector, the output of the element is NOT connected to the first input of the element And, the second input of which is connected to the output of the frequency generator F _p , the output of the element And is connected to the counting input of the i-th counter, setting whose inputs are connected to the output of the reset generator, the counter output is connected to the control inputs of the i-th phase modulator of the synthesized frequency signal with a function with the number of steps 2 ^p .

In FIG. 1 shows a scheme of diffraction of light by ultrasound, where it is indicated: 1 - light beams of frequency ν; 2 - sound duct; 3 - ultrasonic wave of frequency ω; Λ is the acoustic wavelength; α _in - Bragg angle; +1 - diffraction maximum plus first order (DM1); -1 - the same minus the 1st order (DM-1); 0 - transmitted beam of frequency ν.

In FIG. 2 shows the frequency distribution of the lasers, where it is indicated: Δf is the bandwidth of the frequency distribution; ν ₀ is the frequency of the reference laser; ν ₁ ... ν _i ... ν _N are the frequencies of the summed lasers; f ₁ ... f _i ... f _N - difference frequencies; Δf ₁ ... Δf _i ... Δf _N is the frequency increment; ν + Δf ₁ ... ν _i + Δf _i ... ν _N + Δf _N are the frequencies of DM1.

In FIG. 3 shows a block diagram of a coherent optical radiation transmitter, where it is indicated: I — first block — a device for dividing, summing, equalizing frequencies and phases of summed beams; 1 - laser; 3 - acousto-optical modulator; 4 - the first differential frequency photo-mixer; 5 - the second phase mixer signal phase locked loop summed beams; 7 - DM _i ; 21 - reference laser; II - the second block is a device for averaging and refinement of the difference frequency; 6 - filter; 8 - phase modulator of the differential frequency function with the number of steps 2 ^q ; 9 - amplifier; 10 - deflector with forming optics; 11 - line photodetectors (FP) with integrating optics; 12 - averaging device of the differential frequency; 13 is a device for updating the differential frequency; 14 - control unit for the phase modulator of the differential frequency phase with a function with the number of steps 2 ^q (frequency grid synthesizer with counter); 22 - laser device for averaging the differential frequency; III - the third block is a device of frequency and phase self-tuning of summed beams; 2 - laser radiation modulator by a function with the number of steps 2 ⁿ ; 26 - generator; 25 - mixer; 17 - filter; 18 - amplifier; 19 - phase detector; 20 - control device for the phase modulator of the signal of the synthesized frequency by a function with the number of steps 2 ^p and the amount of time delay between functions with the number of steps 2 ^p ; 15 - phase modulator of the signal of the synthesized frequency function with the number of steps 2 ^p ; 16 - amplifier; 23 - N-channel frequency synthesizer; 24 - directional coupler.

In FIG. 4 shows the optical diagram of the summation device, where it is indicated: 1 - laser; 2 - AOM; 4 - 1st differential frequency signal mixer; 5 - 2nd photo-mixer of the phase-locked loop signal of the summed beams; 1.1 - light coupler of the 1st mixer; 21.1 - translucent mirror of the 1st photo mixer; 21.2 - translucent mirror of the N-faced pyramid; 21.3 - mirror of the 1st mixer; 3.1 - light coupler of the 2nd photo mixer; 21.2 - translucent mirror of the 2nd photo mixer; 7 - summing circular conical horn; Λ ₁ ... Λ _N are the radiation beams of the reference laser.

In FIG. 5 shows a multipath structure; in FIG. 6 - various cases of excitation of the AF line; in FIG. 7 - a histogram of errors; in FIG. 8 is a structural diagram of a difference frequency averaging device, where it is indicated: 1 — laser, 2 — collimator; 3 - deflector; 4 - amplifier; 5 - focusing device DM1; 6 - line FP; 7 - synchronization unit; 8 - control unit; 9 - switch on keys; 10 ₁ -10 _m - level comparators; 11 ₁ -11 _m - blocks of summing and averaging the frequency code; 12 - multiplexer; 13 - adder frequency code levels; 14 - block averaging the frequency code.

- модулятор фазы сигнала разностной частоты функцией с числом ступеней 2^q;

- устройство уточнения частоты.In FIG. 9 shows various cases of excitation of the AF line; in FIG. 10 is a functional diagram of an averaging device; in FIG. 11 is a functional diagram of an averaging device; in FIG. 12 is a timing diagram of an averaging device; in FIG. 13 is a structural diagram of a refinement device, where the designations 1 to 14 are the same as in FIG. 8, but additionally indicated:

- phase modulator of the signal of the difference frequency with a function with the number of steps 2 ^q ;

- frequency refinement device.

In FIG. 14 shows a functional diagram of a refinement device, where it is indicated: 1 - zero encoder; 2 is a comparison diagram; CT1 - counter for increasing the integer part of the frequency code by one; ALU1 - arithmetic-logic device for calculating the mismatch of the frequency code Δf; CT2 - counter counter frequency mismatch storage; PROM1 - read-only memory device for converting a relative frequency code to an absolute code; PROM2 - read-only memory device for frequency mismatch code conversion; ALU2 - arithmetic-logic device for calculating the true value of the frequency code; RG1 - result storage register; NU - initial installation; SSCH - strobe tracking frequency code; 14 - control unit for the phase modulator of the differential frequency signal with a function with the number of steps 2 ^q , consisting of: NNS - frequency grid synthesizer; CT3 - counter for controlling the modulators of the phase signal of the differential frequency signal 8 function with the number of steps 2 ^p .

 In FIG. 15 shows a timing diagram of the operation of the refinement device.

 In FIG. 16 is a structural diagram of a frequency grid synthesizer, where it is indicated: 1 — input code (frequency correction code Δf); 2 - decoder; 3 - divider with a variable division ratio (DPKD); 4 - reference generator; 5 - a divider with a fixed division ratio (DPCD); 6 - pulse frequency-phase detector (ICHFD); 7 - low pass filter; 8 - tunable generator.

 In FIG. 17 is a timing diagram of a digital frequency carrier; in FIG. 18 is a structural diagram of a digital frequency carrier; in FIG. 19 shows a multi-stage function; in FIG. 20 shows a structural diagram of a multi-channel frequency synthesizer, where it is indicated: 1 - reference generator; 2 - a system of pulsed phase-locked loop (IFAPCH); 3 - control device multi-channel frequency synthesizer; 4 - counter; 5 is a diagram of the formation of a reset pulse; 6 - trigger; 7 - switches.

 In FIG. 21 shows the structural diagram of IFAPCH, where it is indicated: 1 - control device; 2 - frequency divider with a variable division ratio (DPKD); 3 - reference generator; 4 - pulse frequency-phase detector (ICHFD); 5 - low-pass filter (low-pass filter); 6 - tunable generator.

 In FIG. 22 is a structural diagram of a control device for a multi-channel frequency synthesizer, where it is indicated: 1 - microcomputer (microprocessor); 2 - communication channel; 3 - radial parallel interface; 3.1 - bidirectional linear formers; 3.2 - channel receiver-shapers; 3.3 - channel transmitters-shapers; 3.4 - output data register; 3.5 - output status register; 4 - decoder.

 In FIG. 23 is a structural diagram of a modulator, where it is indicated: 1 — a clock generator; 2 - zero pulse generator; 3 - counter; 4 - p-bit switch; 5 - voltage stabilizer; 6 - resistive matrix; 7 - operational amplifier.

In FIG. 24 is a timing diagram of a modulator; in FIG. 25 is a structural diagram of controlling phase shifters, where it is indicated: 18 ₁ ... 18 _N - amplifiers; 19 ₁ . ..19 _N - phase detectors; 2 ₁ ... 2 _N - elements NOT; 3 ₁ ... 3 _N - AND elements; 4 ₁ ... 4 _N - counters; 5 - clock generator; 6 - reset pulse generator; 15 ₁ ... 15 _N - phase modulators of the signals of the synthesized frequency by a function with the number of steps 2 ^p , controlled phase shifters; 26 - frequency generator F _n .

In FIG. 26 is a structural diagram of modulation DM1, where indicated: 3 - AOM; 24 - directional coupler; 16 - amplifier; 15 - phase modulators, controlled phase shifters; 23 - multi-channel frequency synthesizer; 13 ₁ ... 13 _N - difference frequency refinement devices; 20 ₁ ... 20 _N - control devices for phase shifters, phase modulators 15 ₁ ... 15 _N.

на выходах АОМ_i и распределение электромагнитного поля H₀₁ в круглом волноводе, где обозначено: 3 - АОМ.In FIG. 27 shows the spatial arrangement of vectors

at the outputs of AOM _i and the distribution of the electromagnetic field H ₀₁ in a circular waveguide, where it is indicated: 3 - AOM.

 The applicant and the author are not aware of technical solutions containing features equivalent to the distinguishing features of the proposed device that implements the proposed method, therefore, the proposed device meets the criterion of "novelty" and "significant differences". Introduction to the device of new elements and relationships allows us to implement the proposed method of summing the radiation beams of N lasers and to obtain a coherent beam with a high power density.

 Consider the essence of the proposed method and device that implements it.

 Let the bandwidth of the spread of the laser radiation frequency be several hundred MHz (100 ... 300). This value corresponds to the values of the dispersion of the laser reproduction frequencies encountered in practice, and laser samples with an “almost” coinciding frequency (“almost” = ± 150 MHz) can be easily selected from a series using existing methods for measuring the frequency of laser radiation [4, p. 156-222], because

Таким образом, рабочая полоса частот АОМ должна быть равна нескольким сотням МГц, что всегда имеет место [4, с. 86-88, 95]. С другой стороны после гетеродинирования излучения i-го лазера и излучения опорного лазера необходимо определить значение (код) разностной частоты. Эти значения также лежат в пределах 100-300 МГц, т.е. полоса рабочих частот дефлектора, используемого в устройстве усреднения разностной частоты, должна также составлять 100-300 МГц, что на практике имеет место.

Thus, the operating frequency band of the AOM should be equal to several hundred MHz, which always takes place [4, p. 86-88, 95]. On the other hand, after heterodyning the radiation of the ith laser and the radiation of the reference laser, it is necessary to determine the value (code) of the difference frequency. These values also lie in the range of 100-300 MHz, i.e. the operating frequency band of the deflector used in the difference frequency averaging device should also be 100-300 MHz, which is the case in practice.

, а точность определения частоты не превышает ξ_f= ±0,5δ. Ясно, что значение частоты определяется по расположению амплитуды фотоэлектрически преобразованного сигнала в плоскости изображения.To determine the value of the difference frequency, the phenomenon of the interaction of light and sound is also used. On this basis, a fairly wide class of instruments has been developed for spectral and correlation signal analysis [4]. In such devices, DM1 focuses on one of the phase transition phase transitions, i.e. if the number of phase transitions in the line is N, the studied frequency band is Δf, the "frequency" distance between the phase transitions in the line is ΔF = δ, then

, and the accuracy of determining the frequency does not exceed ξ _f = ± 0.5δ. It is clear that the frequency value is determined by the location of the amplitude of the photoelectrically converted signal in the image plane.

где f_j - отсчитанное значение частоты;
f_i - номинальное значение частоты i-го ФП;
n_i - количество ФП, сигнал в которых достиг данного уровня;
k_j - порядковый номер первого ФП, сигнал в котором достиг данного уровня;
n_j - количество ФП, сигнал в котором достиг j-го уровня.If DM1 of the deflector excites several phase transitions, then when the signals are fixed in the image plane at an arbitrarily chosen level Z _{j, the} frequency can be determined from the expression

where f _j is the measured frequency value;
f _i is the nominal value of the frequency of the i-th phase transition;
n _i is the number of phase transitions in which the signal has reached a given level;
k _j - serial number of the first AF, the signal in which has reached this level;
n _j is the number of phase transitions in which the signal has reached the jth level.

,
где f - усредненная частота;
M - используемая зона в дБ, в пределах которой ведется сравнение сигналов в каналах;
Δ - интервал в дБ между уровнями зоны, в пределах которой ведется сравнение сигналов в каналах.To reduce the error in determining the frequency, the AF amplitude readings are made for a number of signal levels in channels, and the results are averaged:

,
where f is the averaged frequency;
M - usable area in dB, within which the comparison of signals in the channels;
Δ is the interval in dB between the levels of the zone within which the signals in the channels are compared.

 As is known, in deflectors the deflection of the laser beam is carried out by changing the acoustic frequency, i.e. frequency of the analyzed signal. AOD works with light beams, the divergence of which is much less than the divergence of the sound field [7]. Therefore, it can be assumed that the distribution of the field of diffracted light is identical to the distribution of incident light. When analyzing the characteristics of the deflector, this feature allows one to operate only with the field of the incident wave, and also formally assume that the laser light field deflected by the deflector is nothing more than the field of incident light, diffracted at the final aperture of the deflector.

причем каждой из частот входного сигнала соответствует свое направление, т.е. своя точка в Фурье-плоскости.The most important characteristic of a deflector is the resolution, which is understood as the number of light elements (spots) or light positions allowed in space, which the deflector provides when the frequency of the acoustic field (signal frequency) changes. The number of allowed spots N is determined by the ratio of the maximum angular interval Δθ, within which the deviation of DM1 is possible when the signal frequency (acoustic frequency) changes to the angular size of the diffracted field in the far zone

moreover, each of the frequencies of the input signal corresponds to its direction, i.e. its point in the Fourier plane.

 The described device for determining the frequency can be attributed to linear devices. The main characteristic of any linear device, as is known, is its response to the corresponding delta effect in the frequency domain.

при этом в дефлекторе возбуждается бегущая упругая волна, распространяющаяся в направлении оси Z₁ (в направлении от точки возбуждения дефлектора). Падающая на дефлектор плоская световая волна модулируется гармонической упругой волной и на выходе дефлектора имеем

где K_зв= ω_зв/v - волновое число упругой волны;
I_n(A) - функция Бесселя.To do this, consider the passage through a device of harmonic oscillation
S (t) = cosω _sv t,
whose spectral density is the sum of two delta effects in the frequency domain

in this case, a traveling elastic wave propagating in the direction of the Z ₁ axis (in the direction from the point of excitation of the deflector) is excited in the deflector. The plane light wave incident on the deflector is modulated by a harmonic elastic wave and at the deflector output we have

where K _sv = ω _sv / v is the wavenumber of the elastic wave;
I _n (A) is the Bessel function.

где B₁ - размерный коэффициент пропорциональности;

- пространственные частоты;
F - фокусное расстояние линзы, осуществляющей преобразование Фурье;
L - апертура дефлектора.Since the light field distributions in the front and rear focal planes of the collective lens are connected by the Fourier transform, we have in the output plane of the deflector

where B ₁ - dimensional coefficient of proportionality;

- spatial frequencies;
F is the focal length of the lens implementing the Fourier transform;
L is the aperture of the deflector.

Учитывая, что vω_z= ω - текущая частота спектра и вводя обозначение L/v=T - длительность анализируемой выборки, получим

Амплитуда e₊₁ линейно связана с амплитудой входного сигнала при малых индексах фазовой модуляции света, при этом

а спектральное распределение в выходной плоскости дефлектора может быть записано в виде

.Information about the signal spectrum is contained in DM1, the expression for which has the form

Given that vω _z = ω is the current frequency of the spectrum and introducing the notation L / v = T is the duration of the analyzed sample, we obtain

The amplitude e _{+1 is} linearly related to the amplitude of the input signal for small phase modulation indices of light, while

and the spectral distribution in the output plane of the deflector can be written as

.

 This is the hardware function of the deflector and it describes the spectral distribution, which we can attribute to the analytical signal corresponding to the input harmonic oscillation.

We determine the spectral distribution in the region of DM1 when an arbitrary signal S _in (t) is applied to the input of the device in question.

а спектральное распределение в выходной плоскости дефлектора

где ω_o - средняя частота полосы пропускания дефлектора, совпадающая со средней частотой спектра анализируемого сигнала;
2Δω_o - ширина полосы пропускания дефлектора (ширина полосы анализа).Such a physical signal corresponds to an analytical signal Z (t), the spectrum of which [8, p. 122-123]

and the spectral distribution in the output plane of the deflector

where ω _o - the average frequency of the passband of the deflector, which coincides with the average frequency of the spectrum of the analyzed signal;
2Δω _o - bandwidth of the deflector (analysis bandwidth).

Следовательно, в области формирования ДМ1 спектральное распределение определяется мгновенным спектром входного сигнала.If ω _o -Δω _o <ω _sv <ω _o + Δω _o , and 2Δf _o T≫1, then the last expression can be written in the following form

Therefore, in the region of formation of DM1, the spectral distribution is determined by the instantaneous spectrum of the input signal.

а сам дефлектор работает при малых уровнях входных сигналов (sin0,5A ≅ 0,5A), он может быть отнесен к классу линейных приборов и в области DM1 он формирует световое распределение, соответствующее мгновенному спектру анализируемого сигнала.Since the hardware function of the deflector operating in the Bragg diffraction mode has the form

and the deflector itself operates at low levels of input signals (sin0.5A ≅ 0.5A), it can be assigned to the class of linear devices and in the DM1 region it forms a light distribution corresponding to the instantaneous spectrum of the analyzed signal.

 Thus, the deflector, on the basis of the interaction of light and sound, “parallelizes” the input signal, and each spectral component of the spectrum of this signal corresponds to its own direction DM1 and, therefore, its spot in the Fourier plane, while all these spots in the Fourier plane exist simultaneously .

где f - текущее значение частоты;
f_m - направление максимального значения F(f);
K - постоянное число для данного DM1;
Δf_0,5 - ширина DM1 на уровне минус 3 дБ.It is known that the distribution of light intensity in DM1 is Gaussian [3, p. 9, 23 - 28] and can be approximated by an expression of the form

where f is the current frequency value;
f _m is the direction of the maximum value of F (f);
K is a constant number for a given DM1;
Δf _0.5 is the width of DM1 at the level of minus 3 dB.

In the coordinates g (f) = logF (f), f, DM1 can be represented as a parabolic curve, the generalized equation of which has the form [9, p. 113]
g (f) = af ² + bf + c, (2)
where a, b, c are coefficients.

то уравнения парциальных лучей структуры могут быть записаны в виде
g_N(f) = (f∓Nδ)²,
где знак плюс берется для f<0, минус - для f>0, N = 0, ±1, ±2, ... (фиг. 5). Для характеристики с N = 0
g₀(f) = f²
в точках

представляет собой прямую параллельную оси f, которая пересекает характеристики g_N(f) в точках
f = ±δ(N±1/2)
Характеристики g₀(f) = f² и g₁(f) = (f-δ)²; g₀(f) = f² и g_-1(f) = (f+δ) соответственно пересекаются в точках f_0;1= δ/2 и f_0;-1= -δ/2, а разность f_{0; 1} - f_{0; -1} = δ равна ширине парциального луча на уровне (δ/2)².If signals of several frequencies are equally excited in the duct of the deflector by such a value that a multilobal structure is formed in the plane of the FP line, and the adjacent structure diagrams intersect at the level of minus 3 dB, denote by δ the width of the partial beam of the structure at the level of minus 3 dB, take into account that the coordinates of the vertex of the parabola (2) can be found from the relations [9, p. 113]

then the partial ray equations of the structure can be written as
g _N (f) = (f∓Nδ) ² ,
where the plus sign is taken for f <0, minus for f> 0, N = 0, ± 1, ± 2, ... (Fig. 5). For characteristics with N = 0
g ₀ (f) = f ²
at points

represents a straight line parallel to the f axis, which intersects the characteristics g _N (f) at points
f = ± δ (N ± 1/2)
Characteristics g ₀ (f) = f ² and g ₁ (f) = (f-δ) ² ; g ₀ (f) = f ² and g _-1 (f) = (f + δ) respectively intersect at points f _{0; 1} = δ / 2 and f _{0; -1} = -δ / 2, and the difference f _{0; 1} - f _{0; -1} = δ is equal to the width of the partial beam at the level (δ / 2) ² .

.Characteristics g ₀ (f) = f ² and g ₂ (f) = (f-2δ) ² ; g ₀ (f) = f ² and g _-2 (f) = (f + 2δ) ² intersect at the points f _{0; 2} = -δ; f _{0; -2} = -δ, through which the line passes

.

m = 0, 1, 2, . .. (при m = 0 это ось f), располагаются через равные промежутки и разграничивают уровни, в пределах которых могут сравниваться значения g_N(f).So direct

m = 0, 1, 2,. .. (at m = 0 this is the f axis), are spaced at equal intervals and delimit the levels within which g _N (f) can be compared.

где N - номер характеристики:
m - номер линии, с которой пересекается данная характеристика (фиг. 5).The points of intersection of the lines g _m = m (δ / 2) ² with the characteristics g _N (f) = (f∓Nδ) ² are the reference frequency values and are found from the relation

where N is the characteristic number:
m is the number of the line with which this characteristic intersects (Fig. 5).

 Then we proceed as follows. In the frequency range from 2δ to -2δ, we find the values of the reference frequencies for the multipath structure.

В нашем случае Δf = δ. В секторе f_k= 4δ (фиг. 5) характеристика g₀(f) = f² пересекается с соседними характеристиками (3) g(f) = (f∓Nδ)² (N = 1, 2) в точках f = ±δ; ±δ/2 и в соответствии с (5) число отсчетов m = 5, ФП линейки необходимо размещать в точках f = 0; ±δ/2; ±δ, через частотное расстояние, равное δ/2 (фиг. 5).The FP line produces spatial sampling of the optical signal DM1 used to determine each of them using expression (1). The degree of correspondence between “continuous” and “discrete” is established by the sampling theorem of V. A. Kotelnikov-Shannon, according to which smooth and discrete movement of DM1 of the ith deflector will be equivalent if in the sector f _{k the} DM1 beam with a width Δf of minus 3 dB provides obtaining m readings [10, p. 333 - 338]

In our case, Δf = δ. In the sector f _k = 4δ (Fig. 5), the characteristic g ₀ (f) = f ² intersects with the neighboring characteristics (3) g (f) = (f∓Nδ) ² (N = 1, 2) at the points f = ± δ; ± δ / 2 and in accordance with (5) the number of samples m = 5, the linear phase transition must be placed at points f = 0; ± δ / 2; ± δ, through a frequency distance equal to δ / 2 (Fig. 5).

Let us evaluate the accuracy characteristics of a deflector with a line of phase transitions, the distance between which is half the width of DM1 at the level of minus 3 dB, when expression (1) is used to determine the frequency. Compatible DM1 maximum with f reference values $\genfrac{}{}{0ex}{}{\text{uh}}{\text{i}}$ (Fig. 6) and find for each of these positions DM1 the average value f _iy in accordance with (1). In this case, it is necessary to determine at what reference level the signal amplitude at the FP output will be located using the expression
g (f) = (f - f _i ) ² ,
in which the values of f _i correspond to the values of the reference frequencies, and the current value of f to the values of reference points on the frequency axis established on the basis of (5).

f_i = 0,1180339 δ ; уравнение характеристики g(f) = (f - 0,1180339)²;
точки:

Используя фиг. 5 и информацию о распределении амплитуд по уровням, определим усредненные значения частот для каждого из f_i.Thus, we successively obtain:
f _i = 0; equation of characteristic g (f) = f ² ;
points:

f _i = 0.1180339 δ; characteristic equation g (f) = (f - 0.1180339) ² ;
points:

Using FIG. 5 and information on the distribution of amplitudes by levels, we determine the averaged frequency values for each of f _i .

Если обозначить f_i,δ - эталонное значение частоты в δ,f_iy,δ - усредненное значение частоты в δ.For example,

If we denote f _i , δ is the reference value of the frequency in δ, f _iy , δ is the average value of the frequency in δ.

Δ, δ = f _i -f _iy ,
n is the number of the frequency reference value, and the obtained values are tabulated, then according to the data in this table it is possible to construct a histogram of errors (Fig. 7) and evaluate the probability of determining the frequency with a given error.

 To do this, we calculate the area limited by the histogram in the range, say, from 2δ to 1,5δ.

Rectangle Base Size:
2δ - 1,882δ = 0,118δ;
1.882δ - 1.866δ = 0.016δ;
1.866δ - 1.707δ = 0.159δ.

The heights of these rectangles are respectively:
0.018δ; 0.066δ; 0.007δ.

The area of the rectangles included in the histogram (Fig. 7):
S ₁ = 0.118δ • 0.018δ = 2.124 • 10 ^-3 δ ² ;
S ₂ = 0.016δ • 0.066δ = 1,056 • 10 ^-3 δ ² ;
S ₃ = 0.159δ • 0.007δ = 1.113 • 10 ^-3 δ ² .

Total area:
S _Σ = 4.293 • 10 ^-3 δ ² .

.The probabilities of determining the frequency with a given error will be:

.

We introduce the quantity
E = 1 / K
where K = Δ _f is the accuracy of determining the frequency and define it as the efficiency of determining the difference frequency. For a known method for determining the frequency, this value is equal to
E = 2 / δ.

Таким образом, точность определения частоты на один, несколько порядков выше, чем при отсутствии усреднения.For the proposed method
E ₁ = 55.6 / δ with a probability of 0.5;
E ₂ = 15.2 / δ with a probability of 0.24;
E ₃ = 142.2 / δ with a probability of 0.26;
and their relationship

Thus, the accuracy of determining the frequency is one, several orders of magnitude higher than in the absence of averaging.

To determine the difference frequency code, the block diagram shown in FIG. 8. Functional diagram implementing the block diagram shown in FIG. 8 is shown in FIG. 10 and 11 for the case of the 31st AF in the ruler and 15 measurement levels, and in FIG. 12 is a timing diagram of the operation of this functional diagram. We will keep in mind that block 9 in FIG. 8 corresponds to the key block in FIG. ten; block 10 in FIG. 8 correspond to the comparators K ₁ ... K ₁₅ in FIG. ten; block 11 in FIG. 8 correspond to blocks ALU ₁ ... ALU ₁₅ ; RG9 ... RG29; RG24 ... RG38 in FIG. 10, 11; block 12 in FIG. 8 corresponds to the MS block in FIG. eleven; block 13 in FIG. 8 corresponds to ALU ₁₆ in FIG. eleven; block 14 in FIG. 8 correspond to blocks RG30, PROM in FIG. eleven.

 All blocks of FIG. 10, 11 are standard microcircuits of the series 100, 1500, 590.

Block 8 in FIG. 8 correspond to blocks T ₁ , CT1 ... CT16, T ₂ , T ₃ , CT17, CT18 and the remaining non-numbered elements in FIG. 10, 11. Block 7 in FIG. 10, 11 is not depicted due to its triviality, it is a 50 MHz crystal oscillator with a divider on the counters.

The operation of the averaging device. The signals from the outputs of the FP line arrive at the switch on the keys. After turning on the device, the system signal NU (initial setting) comes, which sets the triggers T ₁ ... T ₃ to its original state. The key management registers RG1 ... RG8 are supplied with the clock frequency CLK1, with which the left logical unit is shifted to the left, which through the inverters levels PU1 ... PU31 is fed to the control inputs of the keys, opening them sequentially. From the outputs of the keys, the voltage is supplied to the comparators K ₁ ... K ₁₅ . Level comparators are used with gating frequency CLK1 and memorization. Thus, the sampling time of the signal from the channels and the processing time of its voltage values are not summed, but the largest of them is selected. Signal processing from the output of each channel is carried out simultaneously at all measurement levels. If there is an excess of the signal level over a predetermined threshold, the CT2 ... CT16 level counters fix this, and the arithmetic-logic devices ALU ₁ ... ALU ₁₅ summarize the channel frequency code expressed in terms of the relative number in the system with the sum of codes obtained when viewing previous channels (FP). The channel frequency code comes from the outputs of the CT1 counter, which counts along the leading edges of the sequence CLK1, and only those CT2 ... CT16 counters whose comparators give a signal that the threshold level is exceeded are counted. After viewing all the channels, a cycle of normalization of frequency codes for each level begins at the same time. The thirty-first pulse CLK1 trailing flip flops the trigger T ₁ and blocks the flow of the sequence CLK1, allowing the flow of the sequence CLK2. At one of the inputs ALU ₁ ... ALU ₁₅ , the number of excesses of the threshold of the corresponding level from the counters CT2 ... CT16 is applied, and the sum of frequency codes from the registers RG9 ... RG23 is supplied as the dividend to the second inputs. The division is carried out with a clock frequency of CLK2, and if you select the division accuracy up to seven decimal places, then after each clock cycle CLK2, you need one left shift of the result stored in the registers RG9 ... RG23. The shift occurs in the registers when a logical unit occurs from the output of the counting trigger T ₂ (Fig. 12, diagram 6). On the leading edge of the 24th pulse CLK2, the decoder collected on the CT17 counter generates a pulse (Fig. 12, diagram 4), which sets the trigger T ₁ to its initial state, thereby triggering two cycles simultaneously. On the one hand, a cycle of a new viewing of the 31st channel begins, described above, on the other hand, a cycle of summing frequency codes (in this case, 15 levels) located in the registers RG24 ... RG38. 16 pulses arrive at the counting input of the CT18 counter (Fig. 12, diagram 8), by which the counter, changing its state, controls the multiplexer MS, which alternately supplies the frequency codes from the registers RG24 ... RG38 to the input of ALU ₁₆ for summing. According to the 16th pulse f _{T1, the} counter CT18 generates a pulse (Fig. 12, diagram 9), which ends the cycle of summing the frequency codes by 15 levels and writes the result of the summation in the register RG39. The contents of the register is the address for the PROM ROM, in which the corresponding frequency code is recorded at each address. RG39 and PROM blocks are frequency code averaging blocks and the average value of the difference frequency code is obtained at the output of the PROM block.

 Despite the fact that the averaging operation increases the accuracy of determining the difference frequency by one to several orders of magnitude, this accuracy may not be enough.

В случаях б и в соответственно получаем
f_б = -0,25δ; f_в = 0,25δ.Consider the possibility of increasing the accuracy of determining the frequency. In FIG. Figure 9 shows several cases of excitation of the FP1 DM1 line (cases a, b, c, d). Using expression (1) for case a gives

In cases b and c, respectively, we obtain
f _b = -0.25δ; f _in = 0.25δ.

Thus, if the value of the difference frequency coincides with the frequency coordinates f = 0; ± 0.25δ, then the value of its code is determined exactly; in other cases, the code of the averaged frequency does not correspond to its true value. So, if f _p = -0.08 δ, then f _g = -0.1δ (Fig. 9, 10).

Points f = 0; ± 0.25δ corresponds to the integer part of the frequency code (ch.ch.k.), at these points the fractional part of the code (ch.ch.) is equal to zero, at other points the frequency code contains both parts. If the frequency coordinate of the difference frequency code containing the c.ch. and d.ch. combine with the nearest of the points f = 0; ± 0.25δ (transfer the difference frequency signal "up", "down") by modulating the phase of the signal of this frequency with a function with the number of steps 2 ^q [5] using a digital frequency carrier (CPC), then the frequency value will be determined by one of the frequency coordinates f = 0; ± 0.25δ and the value of the modulation frequency, i.e. it is possible to refine the difference frequency code, which is carried out by the device, the functional diagram of which is shown in FIG. 14, and structural in FIG. 13.

модулятором фазы сигнала разностной частоты функцией с числом ступеней 2^q (блок переноса частоты). Если после первого переноса частоты дробная часть кода частоты не равна нулю, а целая часть не равна целой части, записанной в счетчик СТ1, то ALU1 произведет суммирование дробной части кода частоты с содержимым счетчика СТ2. Если после дальнейших переносов дробная часть кода частоты не равна нулю, а целая часть равна содержимому счетчика СТ1, то с выхода схемы сравнения подается импульс, который заблокирует прием информации счетчиком СТ2 со входа Д и вычтет единицу из содержимого счетчика. Если в результате дальнейшего переноса частоты дробная часть кода частоты равна нулю, то шифратор нуля выработает импульс записи кода частоты в регистр RG1. Код частоты получается в результате вычитания в ALU2 кода поправки частоты, хранящейся в ПЗУ PROM2, из кода частоты, хранящегося в ПЗУ PROM1. В ПЗУ PROM1 и PROM2 осуществляется переход от относительного кода частоты в системе к абсолютному значению кода частоты.The operation of the device refinement code differential frequency. From the output of the frequency averaging device ch.k. frequency is fed to the parallel load input CT1 and CLK1 is increased by one by the pulse; at the same time, the synchronization block from the output of the frequency averaging device receives the frequency code tracking strobe (CKSC, output 9 of CT18 block, Fig. 11, diagram 9 of Fig. 12). The controlled synchronization unit begins to generate signals CLK1 ... CLK3 (Fig. 15) with a delay relative to the trailing edge of the SSCH equal to the time of establishing the information at the output of the PROM ROM. The integer part of the frequency code also goes to the input of the comparison circuit 2. The fractional part of the frequency code goes to the input of encoder 1 and to the input of the arithmetic-logic device ALU1, where the fractional part is subtracted from unity, and the difference is written to the counter CT2, from the output of which the difference code arrives to the input of the control unit

the phase modulator of the differential frequency signal with a function with the number of steps 2 ^q (frequency transfer unit). If after the first frequency transfer the fractional part of the frequency code is not equal to zero, and the integer part is not equal to the integer part recorded in the CT1 counter, then ALU1 will sum the fractional part of the frequency code with the contents of the CT2 counter. If after further transfers the fractional part of the frequency code is not equal to zero, and the integer part is equal to the contents of the counter CT1, then an impulse is supplied from the output of the comparison circuit, which will block the reception of information by the counter CT2 from input D and subtract one from the contents of the counter. If, as a result of further frequency transfer, the fractional part of the frequency code is equal to zero, then the zero encoder will generate an impulse to write the frequency code to register RG1. The frequency code is obtained by subtracting the frequency correction code stored in ROM PROM2 from the frequency code stored in ROM PROM1 in ALU2. In the PROM1 and PROM2 ROMs, a transition is made from the relative frequency code in the system to the absolute value of the frequency code.

 The operation of the frequency grid synthesizer, the block diagram of which is shown in FIG. 16 is as follows.

The signal of the reference generator 4 of frequency f _{or is} fed to the input of the DPCD, at the output of which pulses are generated with a frequency
f = f _or / R
where R is the division coefficient of DFCD.

Сигналы этой частоты поступают на первый вход ИЧФД6, а на другой - импульсы от перестраиваемого генератора 8, деленные делителем с переменным коэффициентом деления ДПКД3. В результате сравнения временного положения входных импульсов ИЧФД6 на его выходе образуется напряжение ошибки, которое управляет генератором 8. Под воздействием напряжения ошибки ПГ8 устанавливается в такое состояние, при котором временное рассогласование входных импульсов по входным импульсам ИЧФД6 равно нулю. В этом случае частота сигнала на выходе ПГ8 определяется выражением

,
где N - коэффициент деления ДПКД. Работой ДПКД управляет дешифратор 2, который преобразует входной код 1 (код поправки частоты Δf). При необходимости изменения шага перестройки частоты выходного сигнала следует изменить коэффициент деления R ДФКД и заменить ФНЧ7.The IFAPCH system determines the step of tuning the frequency of the output signal of the MSS. So, for example, if you set the frequency f _or = 5 MHz, and R = 5000, then the tuning step will be

Signals of this frequency are fed to the first input of ICHFD6, and to the other, pulses from the tunable generator 8, divided by a divider with a variable division coefficient DPKD3. As a result of comparing the temporary position of the input pulses of ICHFD6, an error voltage is generated at its output, which controls the generator 8. Under the influence of the voltage of the error, PG8 is set to a state in which the temporary mismatch of the input pulses with respect to the input pulses of ICHFD6 is zero. In this case, the signal frequency at the output of PG8 is determined by the expression

,
where N is the division coefficient of the DPKD. The operation of the DPKD is controlled by a decoder 2, which converts the input code 1 (frequency correction code Δf). If you need to change the pitch of the output signal frequency tuning, you must change the division coefficient R DPCD and replace the low-pass filter.

 The output signals of the MSC control the central frequency converter through the CT3 counter (Fig. 18). The digital frequency converter represents q successively connected discrete phase shifters into two states [5]. For example, for q = 4 (Fig. 17) it is 0 ... π / 8; 0 ... π / 4; 0 ... π / 2; 0 ... π.

 A timing diagram of the operation of the DSC is shown in FIG. 17, and in FIG. 19 is a view of a multi-stage function.

,
где Δf_ссч - шаг перестройки ССЧ;
q - число дискретных фазовращателей в ЦПЧ.At points f _c = ± 0.25Nδ, where N is the characteristic number of the multipath structure, the difference frequency is determined exactly. The error in determining the frequency by the averager is on average no more
ξ _f ≤ 0,032δ
and this error occurs between neighboring points at which the frequency is determined accurately. On the other hand, the step of tuning the synthesized frequency with the qualifier is

,
where Δf _ssc - the tuning step of the MSS;
q is the number of discrete phase shifters in the CPC.

,
где f_c - значение синтезированной частоты. Так, если Δf_ссч = 1 кГц, q = 4, то f_ист = f_c ± 62,5 Гц и эта ошибка намного меньше естественной ширины спектральной линии каждого из лазеров [3, с. 21-28].Thus, the true value of the frequency f _East at the output of the qualifier can be written in the form

,
where f _c is the value of the synthesized frequency. So, if Δf _scc = 1 kHz, q = 4, then f _source = f _c ± 62.5 Hz and this error is much less than the natural width of the spectral line of each laser [3, p. 21-28].

The equality of the frequencies of all DM1 _{i to the} frequency of the reference oscillation is maintained automatically due to the presence of feedback from the output of each laser to the modulating input of the corresponding AOM _i laser. Any change in the frequency ν _{i of the} i-th laser leads to a change in the modulation frequency of the AOM _{i of} this laser and to "pull" the frequency DM1 _{i of} this laser to the frequency ν _{o of the} reference oscillation. Indeed, a change in the frequency ν _{i of the} ith laser causes a change in f _pi , which in turn will cause a change in f _ci , i.e. will lead to a change in the modulation frequency of the AOM _{i of} this laser.

 Consider the work of a multi-channel frequency synthesizer (MSC), i.e. modulating frequency synthesizer, the block diagram of which is shown in FIG. 20, a block diagram of the IFAP is shown in FIG. 21 and in FIG. 22 is a block diagram of an MSC control device.

 The difference frequency code is supplied to a multichannel synthesizer (Fig. 20), which includes a reference oscillator 1, the frequency of which is determined by the required discrete (step) frequency modulation of the MSS, IFAPCH2, MSCH3 control device, counter 4, pulse generating circuit 5, trigger 6 and N switches 7.

In stationary mode (difference frequency codes are defined and known), the output signals IFAPCH1 ... IFAPCH are fed to the signal inputs of the corresponding switches K ₁ . ..K _N. From the output of trigger 6 to the control inputs of the switches receives a logical unit, under the action of which the switches K ₁ ... K _N are open for the passage of signals. The frequency of the signals of each of the IFAH systems is determined by the control code, which is supplied to the control inputs of each IFAH system from the corresponding difference frequency refinement units.

When you change one or more difference frequencies, the codes of these frequencies change at the corresponding outputs of the refinement blocks. In this case, the reset pulse generation circuit generates a pulse that resets the counter and sets a logical zero at the trigger output, which closes the corresponding switches K ₁ ... K _N. At the same time, the control unit code changes the division factors of the dividers with a variable division coefficient, which leads to a mismatch in the corresponding IFAPCH1 systems. . . IFAPCHN. As a result, an error signal is generated at the outputs of the low-pass filters 5, which sets the frequency of the tunable generator (output frequency of the IFAPC system) in accordance with the input code. The restructuring of the corresponding IFAH systems is completed in 25-40 periods of oscillation of the reference generator.

After resetting the counter, it counts the pulses of the reference generator and after 40 pulses, when the transient processes in the corresponding IFAPCH1 ... IFAPCHN systems are completed, a pulse is generated that goes to the trigger, setting it to the state of the logical unit, the corresponding switches K ₁ . . . K _N open and at the outputs of the MSC at the same time, the frequencies corresponding to the codes of difference frequencies are set at the corresponding outputs.

 The MSC control device can be implemented on the basis of a microcomputer or a special processor with a set of hardware and is interfaced with the MSC through a radial parallel interface (IRPR).

 An embodiment of the control device is shown in FIG. 22.

 The operation of the device. When executing the program for writing frequency codes to the corresponding buffer registers of the control units of IFAPCH2 devices, frequency synthesizers in the “output” cycle, the central computer writes the IFAPCH number code (equal to the channel number) to the IPRF 3.5 status register, into which the code will be written to the buffer register (IFAPCH) frequency. This code is converted by the decoder 4 (Fig. 22) into the positional code of the IFAPCH number, according to which an impulse for selecting the microcircuit of the corresponding DPCD is generated. Then, also in the “output” cycle, the central processor writes a frequency code to data register 3.4, which must be written to the selected IFAPC and, using the Ready signal from the selected IFAPC system, which signals the system is ready and records the frequency code, generates a “Strobe” signal , according to which the frequency code is written from the data register 3.4 in the buffer register of the DPKD selected IFAPCH. Before this, a “Reset” signal is preliminarily generated for installation in the initial state of counter 4 and trigger 6 of MSC. To write the frequency code to other PLL, the indicated operations are repeated.

Thus, the device for averaging and refinement of the code of the different frequency together with the MSS (Fig. 3, 20) and the corresponding AOM _i automatically align the frequencies DM1 _i relative to the frequency of the reference beam with an error substantially less than the natural width of the spectral line of each laser.

The alignment of the oscillation phases of the summed beams is as follows. First, the oscillation phase of some auxiliary frequency is introduced into the oscillation phase of each laser by modulating the oscillation phase of each of the lasers with a function with the number of steps 2 ⁿ , which can be done by applying a step voltage to the piezoelectric holder of one of the mirrors of the laser cavity [4, 156-222]. The step voltage can be obtained using a modulator, the functional diagram of which is shown in FIG. 23, and in FIG. 24 shows an example of step voltage generation for n = 4.

 In this case, the laser radiation frequency is transferred to the modulation frequency [5].

In FIG. 25 shows a phase shifter control circuit, and FIG. 26 for explanation, part of the block diagram of FIG. 3, the radiation DM1 _{i being} directed perpendicular to the plane of the figure.

In the absence of signals from phase detectors 19 ₁ ... 19 _N at the inputs of elements 2 ₁ . . .2 _N NOT (Fig. 25) is the logical zero level. After elements are inverted by NOT, a signal with a logic unit level is fed to the first inputs of AND 3 ₁ ... 3 _N elements, the second inputs of which are supplied with clock pulses from generator S, and from the outputs of these elements, signals are fed to the counting inputs of counters 4 ₁ ... 4 _N. If, when the entire device is turned on, the initial installation pulse from the generator 6 is supplied to the counter installation inputs, then all the counters work synchronously and at the outputs of the phase shifters 15 ₁ ... 15 _{N the} phase shift is zero, since at the same time moment between all counters ΔT = 0, i.e., there is no delay, therefore, the phase advance is the same. When a signal appears at the output of any of the phase detectors 19 ₁ ... 19 _N , the inputs of the corresponding elements And 3 ₁ ... 3 _N receive logic zero signals and clock pulses to the input of the counter connected to the corresponding element And 3 ₁ ... 3 _N , do not arrive.

Эта задержка, а значит и фазовый сдвиг Δφ, увеличивается до тех пор, пока разность фаз между упомянутыми сигналами не станет равна нулю, в результате чего исчезает сигнал на выходе соответствующего фазового детектора 19₁. . .19_N, импульсы от генератора 5 снова начнут поступать на соответствующие входы счетчиков, фиксируя полученную временную задержку, т.е. разность фаз. Таким образом, разность фаз между DM1_i обратится в ноль и будет поддерживаться равной этой величине, обеспечивая синфазное суммирование пучков в суммирующем устройстве 7 (фиг. 4).This will lead to a corresponding time delay ΔT and, consequently, a phase shift between the signal supplied from the generator 26 (Fig. 3) and the signals supplied from the outputs of the amplifiers 18 ₁ ... 18 _N (Fig. 25) equal to

This delay, and hence the phase shift Δφ, increases until the phase difference between the mentioned signals becomes zero, as a result of which the signal at the output of the corresponding phase detector 19 ₁ disappears. . .19 _N , the pulses from the generator 5 will again begin to arrive at the corresponding inputs of the counters, fixing the received time delay, i.e. phase difference. Thus, the phase difference between DM1 _i will turn to zero and will be maintained equal to this value, providing in-phase summation of the beams in the adder 7 (Fig. 4).

Since the phase information contained in DM1 _i is completely stored in the function of the heterodyne field [6, p.181], i.e. in the phase of the signal supplied to one of the inputs of the phase detectors 19 ₁ ... 19 _N , the voltage at their output automatically controls the value ΔT _ij , compensating for the emerging phase difference between DM1 _i .

перпендикулярен направлению распространения ультразвуковых волн [7, с. 651], а пространственная ориентация векторов

во всех DM1_i показана на фиг. 27,а и подобна распределению волны H₀₁ в круглом волноводе (фиг. 27, б). Диаграмма направленности (DH) конического рупора с волной H₀₁ имеет 2-х лепестковую структуру с нулем по оси рупора, а в направлении максимума DH должны быть ориентированы продольные оптические оси DM1_i.In all DM1 _{i the} vector

perpendicular to the direction of propagation of ultrasonic waves [7, p. 651], and the spatial orientation of the vectors

in all DM1 _i shown in FIG. 27, a similar distribution and H ₀₁ wave in circular waveguide (FIG. 27b). The radiation pattern (DH) of a conical horn with wave H ₀₁ has a 2-lobe structure with zero along the axis of the horn, and the longitudinal optical axes DM1 _i should be oriented in the direction of the maximum DH.

Each of AOM _i is an emitter emitting power P _i with a gain g _ri (φ, θ) characterizing the directivity of each of DM1 _i . The set of AOM _i is an annular array of N emitters, the radiation of which is focused at the location of the summing device 7 (Fig. 4), i.e. focused on infinity. It is known that the gain of such a grating G _r (φ, θ) can be written in the form [2, p.376].

G _r (φ, θ) = Ng _ri (φ, θ),
where the index r means that both G _r (φ, θ) and g _ri (φ, θ), the quantities realized in the direction φ, θ. If we assume that both P _i and g _ri (φ, θ) are the same for all AOM _i and, accordingly, are equal to P ₀ and g _or (φ, θ), then the energy potential E (φ, θ) of the lattice can be written in the form [ 1, p. 85].

E (φ, θ) = P _o g _or (φ, θ) N ² ,
which is a generalization of the coherent addition of N oscillations.

 The main characteristic of any antenna (the summing device is the receiving antenna) is the function Ф (φ, θ), which describes the dependence of the field strength of the wave radiated (received) by the antenna on the angles φ, θ.

где (φ,θ) - азимут и угол места соответственно;

- амплитудная DH;
ψ(φ,θ) - фазовая DH.In our case, this dependence has the form

where (φ, θ) is the azimuth and elevation angle, respectively;

- amplitude DH;
ψ (φ, θ) is the phase DH.

Within the main lobe, the summing device has a phase characteristic that does not differ from the constant function [10, p.142]. Thus, the summing device, having a constant phase characteristic, does not introduce phase errors into the phase of each of DM1 _i and does not violate the obtained coherence of the summed beams.

The wave H ₀₁ in the circular waveguide is the longest, its critical length is 1.64 • a, where a is the radius of the waveguide, the field structure is shown in FIG. 27, b [12, p. 61-72]. Electric lines of force are circles, and magnetic lines of force in the plane of the cross section are radii, i.e., there are no longitudinal currents. As the frequency increases, the currents on the walls of the waveguide decrease, which leads to a decrease in the attenuation coefficient, and the propagation conditions of the wave in the waveguide approach the propagation conditions of a plane wave in space.

 A conical horn is used to match the circular waveguide to the space. As studies show [13, p. 207-210], if a round waveguide passes into a conical horn, then the waveguide wave that arrives at the mouth of the horn is almost completely converted into the corresponding horn wave, which is essentially a spherical wave, the latter degenerates into sufficiently large distances compared to the wavelength flat. In accordance with the reciprocity theorem [14, p. 123], the converse is also true, that is, when a plane wave falls on the opening of a horn, it is converted into a horn, and then into a waveguide.

 It should be noted that the laser coherence length is much larger than several tens of meters [15, p. 187] (interference is observed when the path difference exceeds 1000 m), therefore, the alignment of the phases of the beams can be performed both in the plane of their summation and in a plane remote from the plane of summation of the beams.

 Like any optical device, the claimed device must be aligned [16, p. 425].

 Adjustment - a set of operations to bring a measure or measuring device into working condition, ensuring its proper accuracy, correctness and reliability of the action. Adjustment consists in establishing the correct interaction, relative position and relative movement of parts, assemblies and systems of aligned objects. The main operations during adjustment: determination of defects, regulation of the arrangement of parts and assemblies with screws, gaskets, etc., correction of worn-out areas by grinding, lapping, finishing, changing individual parts and assemblies.

 Alignment of optical systems (devices) is a special adjustment or overlay of optical systems, which consists mainly in installing optical parts (lenses, prisms, mirrors, reading scales, etc.) in such mutual positions that they best perform their intended functions. The need for such special adjustment in the process of assembling optical devices is due to the strong dependence of some properties of these devices on the relative positions of the optical parts and their parameters, which cannot be accurately performed during their manufacture. The alignment of optical systems is usually done by sequentially installing individual optical parts along the light beam in the device and monitoring their action.

The adjustment methods of various optical devices are very diverse, but usually they come down to the following operations:
a) to a change in the mutual distance between the individual optical parts (telescopes, cameras, etc.);
b) to install various prisms and mirrors in certain positions at which the prism (mirror) must deflect the incident light beam at a certain angle (prismatic binoculars, prism spectral devices, etc.);
c) to the exact installation of certain surfaces strictly parallel to each other (interferometers, etc.);
d) adjustment of the readings of various scales or reading devices (measuring optical instruments, etc.).

 Moreover, the design of the device usually provides for the possibility of performing these operations and fixing the parts in the positions established during the adjustment process. A general rule for alignment of most devices is the alignment of the device, i.e., the installation of all optical parts on the same optical axis.

Currently known methods and devices that allow you to change the value of local surface roughness with an accuracy of 0.015-0.03 microns [17, p. 196], there are indicators that allow you to record the linear displacement of the plane of the part (the surface of the AOM _{i of the} mixer) by a value of 0.007-0.014 μm [17, p. 185-187]. We also note that at present the coherence of some lasers reaches 10 ^-2 s, i.e. about 10 ⁶ times more than any classical source of radiation [3, p. 36].

 Compared with the prototype, the proposed method for generating a coherent optical signal by summing the laser radiation beams at the apex of the conical surface and the coherent optical radiation transmitter that implements the inventive method make it possible to obtain a coherent beam with a high power density.

 The inventive method and device implementing it can find the widest application in various fields of science and technology - metallurgy, medicine, physics, etc.

Literature
1. Privalov EM The input characteristics of the waveguide emitter in the composition of the final lattice. In the book: Scattering of electromagnetic waves. Vol. 5. - Taganrog: TRTI, 1985, p. 84-90.

 2. Scanning microwave antenna systems. T. 2. / Ed. G.T. Markova and A.F. Chaplin. - M .: Owls. radio, 1969.

 3. Pakhomov II, Rozhkov OV, Rozhdestven VN Optoelectronic quantum devices. - M .: Radio and communications, 1992.

 4. Measurement of spectral-frequency and correlation parameters and characteristics of laser radiation / Ed. A.F. Kotyuk and B.M. Stepanova. M .: Radio and communication, 1982.

 5. G. Klein, Z. Dubrowsky. Digilotor - a new broadband microwave translator. 9EEE Trans. on Microwave Theory ahd Techn. March, 1967, v - mTT - 15, N 3.

 6. Galliardi R., Karp S. Optical communication. - M.: Communication, 1978.

 7. Born M., Wolf E. Fundamentals of optics. - M.: Science, 1970.

 8. Gonorovsky I.S. Radio circuits and signals. - M .: Owls. Radio, 1971.

 9. Bronstein I.N., Semendyaev K.A. Math reference. - M.: Science, 1986.

 10 Vendik O.M. Antennas with non-mechanical beam movement. - M .: Owls. radio, 1965.

 11. Markov G.T., Sazonov D.M. Antennas - M .: Energy, 1975.

 12. Efimov I.E., Shermina G.A. Waveguide transmission lines. - M.: Communication, 1979.

 13. Weinstein L.A. Electromagnetic waves. - M.: Radio and Communications, 1988.

 14. Markov G.T., Petrov B.M., Grudinskaya G.P. Electrodynamics and radio wave propagation. - M .: Owls. radio, 1979.

 15. Kalitievsky N.I. Wave optics. - M.: Science, 1971.

 16. The Great Soviet Encyclopedia. T. 49. - M.: Second edition, ed. TSB.

 17. Kolomiytsev Yu.V. Interferometers - L .: Engineering, 1976.

Claims (10)

1. A method of generating coherent optical radiation by summing the phased laser radiation beams, characterized in that the frequencies of the summed beams are aligned by heterodyning the parts of the vibrations of each of the lasers with the vibrations of the reference laser, extracting from the spectra of the optically converted signals of each of N difference frequencies between the oscillations of each of N lasers and oscillations of the reference laser, determining and refining the digital code of each of N difference frequencies in a digital way, synthesizing digital codes of these frequencies N signals of acousto-optical modulation, excitation by the i-th synthesized signal of the i-th ultrasonic wave in the i-th elastic, optically transparent medium, Bragg diffraction of the i-th radiation beam of the i-th laser on inhomogeneities of the i-th elastic, optically transparent medium caused by the passage of the i-th ultrasonic wave excited by the i-th synthesized acousto-optic modulation signal, the Doppler shift of the oscillation frequency of the i-th laser accompanying Bragg diffraction to the frequency of the reference laser, the phases of the summed diff first order promotional maxima of the same frequency aligned by modulating the phase fluctuations of each of the N lasers in the same multi-stage signal, the number of stages of which is 2 ^n, where n - number of the smallest increment the state of phase of the signal φ _{n min} = π / 2 ⁿ , the signal period is T _n = 1 / F _n , by modulating the phase of each of the synthesized acousto-optic modulation signals with a multi-stage signal, the number of steps of which is 2 ^p , where p is the number of the least discrete phase state of this signal φ _{p min} = π / 2 ^p , p The signal period is T _p = 1 / F _p , by heterodyning each of the i-parts of the first-order diffraction maxima and i-oscillations of the same reference laser, extracting i-signals of frequency F _n from the converted signal spectra, detecting each phase from the selected i-th signals of frequency F _n with a reference signal of the same frequency, introducing into the phase of each of the i-th signals of acousto-optic modulation of the phase shift determined by the phase difference between the i-th frequency oscillation F _n extracted from the spectrum and the reference oscillation of same frequency equal

where Δ _ij is the phase shift between the ith and jth oscillations of acousto-optical modulation;
T _p - the period of the modulating signal with the number of steps 2 ^r ;
ΔT _pij is the time delay between the ith and jth modulating signals with a number of steps of 2 ^r .  2. The method according to claim 1, characterized in that the code value of each of the i-th difference frequencies is determined by exciting the i-th difference frequency signal of the i-th ultrasonic wave in an elastic, optically transparent medium, illuminating this medium with a coherent light wave, diffraction of this wave by inhomogeneities of the physical medium caused by the passage of the ith ultrasonic wave, focuses the first-order diffraction maximum of the diffracted light wave onto the plane of its image, transforms focused onto the plane into optical signals into electric ones, determine the frequencies of i-difference signals by the location of the amplitudes of the excited electric signals in the image plane, while measuring the amplitudes obtained as a result of photoelectric conversion at K equal levels (K = 1,2, ...), starting from the first, lying within 0 ... - 3 dB, with 0 dB coinciding with the maximum value of the amplitude of the i-th photoelectrically converted signal, the frequency value of the difference frequency signal is determined by averaging the levels measured at K the amplitudes of the i-th photoelectrically converted differential frequency signal, the averaging results are presented in the form of a frequency code containing integer and fractional parts, the integer part of the frequency code corresponding to either the nearest maximum of the i-th photoelectrically converted amplitude signal of the difference frequency or the nearest half-center between the two nearest the maxima of the i-th and (i + 1) -th or (i-1) -th and i-th photoelectrically converted equal-amplitude signals of difference frequency, and the fractional part of the frequency code corresponds to t the average value of the frequency of the differential frequency signal, which is located between these two points. 3. The method according to claim 2, characterized in that the frequency code of the i-th difference frequency is refined so that the fractional part of the frequency code of this signal is converted into a frequency mismatch code, this code synthesizes a multi-stage voltage with the number of steps 2 ^q , where q - the number of the smallest discrete state of the phase voltage with the number of steps 2 ^q , φ _{q min} = π / 2 ^q , multi-stage voltage with the number of steps 2 _q modulate the phase signal of the difference frequency, transferring it either "up" or "down" in frequency, changing the period T _q eg multi-stage according to the mismatch code, as a result of which the direction of the maximum of the optical signal in the image plane is moved until this maximum coincides either with the direction corresponding to the closest maximum of the i-th photoelectrically converted signal in the image plane, or with the direction corresponding to the nearest half-center between the ith and (i + 1) -th, (i-1) -th and i-th equal-amplitude photoelectrically converted signals in the image plane. 4. A coherent optical radiation device containing N lasers, characterized in that it is made in the form of a device for dividing the reference radiation beam into N parts, a device for summing N phased beams, devices for aligning frequencies and phases of summed beams with N inputs of modulation frequency signals F _n , N inputs of acousto-optic modulation signals U _{c i} , N outputs of difference frequency signals f _{p i} , N outputs of phase-locked loop signals of summed beams and output of the total coherent beam, made in the form of N devices Comparison and refinement of the values of difference frequencies with N inputs of signals of difference frequencies, with N outputs of codes of difference frequencies, made in the form of N devices of frequency and phase self-tuning of oscillations of summed beams, with N inputs of signals of phase self-tuning of oscillations of summed beams, with N inputs of codes of difference frequencies, n output signals with frequency F _n, with n outputs acoustooptic modulation signals U _{c i,} wherein n input signals of frequency F _n dividing device summation, the frequency alignment phase and summed beams are connected with a tvetstvuyuschimi outputs signals of frequency F _n devices the frequency and phase locked summed beams, N acoustooptic modulation signal device inputs division, summation and smoothing of frequencies and phases summed beams are connected to respective N-output signals akustooptitcheskoy modulation device frequency and phase avtopostroyki oscillations summed beams, N signal outputs the difference frequencies of the devices for dividing, summing, aligning the frequencies and phases of the summed beams are connected to the corresponding inputs and signals of the phase auto-tuning oscillations of the summed beams of the device for frequency and phase auto-tuning oscillations of the summed beams, N outputs of the difference frequency codes are connected to the corresponding inputs of the codes of difference frequencies of the devices of the frequency and phase auto-tuning oscillations of the summed beams, and the output of the total coherent beam is the output of the coherent optical radiation device. 5. The device according to claim 4, characterized in that the device for dividing, summing, equalizing frequencies and oscillation phases of the summed beams contains N optical channels of the reference laser, which are N optical inputs of the coherent optical radiation device, each of the N optical channels of the reference laser contains optically coupled i-th translucent face of the N-faced pyramid, i-th first translucent mirror, i-th mirror of the first i-th mixer with the output of the i-th difference frequency signal through the filter, second i-th translucent mirror of the ith i-th mixer with the output of the i-th phase-locked loop signal of the summed beams through the filter, contains N optical channels of the summed beams, each of which contains optically coupled i-th laser, the first light coupler of the first i-th differential frequency mixer with the same first i-th translucent mirror, i-th acousto-optic modulator, i-second light coupler of the second i-th phase mixer of the auto-phase signal with the same second i-th translucent mirror and adder of diffraction maxima of the first order , N acousto-optical modulators in the form of series-connected circular conical horn and circular waveguide, the inner diameter of the circular waveguide and the diameter of the lower base of the said horn being equal to each other, and the radius of the circular waveguide has a value corresponding to the propagation condition of the wave H _{0 1 in it} , the open end of the round the waveguide is the optical output of the device for dividing, summing, aligning the frequencies and phases of the summed beams, that is, the optical output of the coherent optical device and radiation, and the longitudinal axis of the summed beams are oriented along the maxima of the two-petal radiation pattern of a circular conical horn. 6. The device according to claim 4, characterized in that the device for averaging and updating the difference frequency code contains a phase modulator of the difference frequency signal in series with a multi-stage signal with a number of steps 2 ⁰ , where q is the least discrete state of the phase of the step signal φ _{q min} = π / 2 ^q , the input of which is the input of the device, an amplifier and a diffractor, contains an optically coupled laser with forming optics, an integrating optics and a line of photodetectors with S outputs, located in the plane of the diffraction image of a maximum of the first order of the diffuser, contains a differential frequency averaging device with an output of a frequency code tracking gate, contains a difference frequency refinement device with a frequency correction code output, an input of a frequency code tracking strobe, a second output, and the outputs of the photodetector line are connected to the input of a differential frequency averaging device, whose first output is through the difference frequency refinement device, through the output of the frequency correction code and the control unit of the phase difference signal modulator pilots at a stepped signal with the number of steps of 2 ^q, connected to the control inputs of the phase modulator of the difference frequency signal stepwise signal with the number of steps of 2 ^q, the second output device specification of the difference frequency is the output of the averaging unit and clarify the difference frequency, the output strobe tracking code frequency difference frequency averaging device connection to the second input of the differential frequency refinement device. 7. The coherent optical radiation device according to claim 4, characterized in that the difference frequency code averaging device comprises P channels, a switch, a multiplexer, an adder and an averager, and comprises a control and synchronization unit of the difference frequency code averaging device with the output of the frequency code tracking gate, wherein the input of the difference frequency code averaging device through a switch, the control input of which is connected to the sixth output of the control and synchronization unit, connected to the S outputs of the photodetector line, One switch is connected to P parallel channels, each of which consists of a serial connection of the comparator and the adder with averaging, the control inputs of the comparator and adder with averaging of each channel are respectively connected to the first and second outputs of the control and synchronization unit, the output of each j-th channel

connected respectively

the multiplexer input, the output of the first channel is connected to the first input of the adder, the second input of which is connected to the output of the multiplexer, the output of the adder is connected to the input of the averager, the control inputs of the multiplexer, adder and averager are respectively connected to the third, fourth and fifth outputs of the control and synchronization unit, the output of the averager is the output of the difference frequency code averaging device. 8. The coherent optical radiation device according to claim 4, characterized in that the difference frequency code refinement device comprises a synchronization unit, the input of which is the second input of the difference frequency code refinement device, a zero encoder, a comparison element, a counter for increasing the integer part of the code by one, arithmetic -logic block for calculating the frequency mismatch code, a counter for storing the frequency mismatch code, a frequency grid synthesizer whose outputs are connected to the counter inputs, read-only memory converting a relative frequency code to an absolute code, a read-only memory device for converting a frequency mismatch code, an arithmetic-logical unit for calculating the true frequency code, gating the result storage register, the first inputs of the zero encoder, the comparison element, the counter increasing the integer part of the code by one, arithmetic-logical the frequency mismatch code calculation unit is connected to the output of the difference frequency code averaging unit, the second input of the counter increases the integer part of the code per unit the source is connected to the first output of the synchronization unit, and the third to its third output, the second input of the zero encoder is connected to the second output of the synchronization unit, the output of the counter increasing the integer part of the code by one is connected to the second input of the comparison element and the input of the read-only memory of the relative frequency code conversion in the absolute code, the output of which is connected to the first input of the arithmetic-logic unit for calculating the true frequency code, the output of the comparison element is connected to the first input of the storage counter to a frequency mismatch ode, the second input of which is connected to the output of the arithmetic-logical block of the frequency mismatch code calculation, and the third to the second output of the synchronization block, the output of the frequency mismatch code storage counter is the output of the frequency mismatch code and connected to the input of the frequency grid synthesizer, the outputs of which are connected with the inputs of the counter, the output of which is connected to the second input of the phase modulator of the differential frequency signal with a step signal with the number of steps 2 ^q , connected to the constant frequency conversion device of the frequency mismatch code, the output of which is connected to the second input of the arithmetic-logic unit for calculating the true frequency code, the output of which is connected to the first input of the gating register for storing the result, the second input of which is connected to the output of the zero encoder, and the output of the register is the output of the averaging device and refinement of the difference frequency code. 9. The coherent optical radiation device according to claim 4, characterized in that the frequency and phase locked loop of the summed beams contains an N-channel synthesizer of vibrations of acousto-optic modulation with N inputs of codes of difference frequencies and N outputs of synthesized vibrations of acousto-optic modulation, contains N first channels, each of which includes a phase modulator of acousto-optic modulation signals connected in series with step signals with a number of steps of 2 ^r with a control input, an amplifier directed a coupler with direct and branch signal output, the i-th output of the direct signal of which is the i-th output of the acousto-optic modulation signal of the device for frequency and phase self-tuning of summed beams, and the i-th output of the N-channel acousto-optic modulation signal synthesizer is connected to the corresponding i-th input the first channels, contains N second channels, each of which includes a mixer with first and second inputs, the output of which is connected to the first input of the phase detector, through a filter and amplifier, the second i-th input to is connected to the ith output of the reference signal generator of the frequency F _n , the ith output of the phase detector is connected to the ith input of the control unit of the phase modulator of acousto-optic modulation signals by step signals with the number of steps 2 ^p and controlling the amount of time delay between step signals, i the i-th output of which is connected to the i-th control input of the said modulator, and the first i-th input of the mixer is connected to the i-th output of the phase-locked signal of the device for dividing, summing, equalizing frequencies and oscillation phases summable beams, a second i-th input of the mixer is connected to the i-th output of the branched signal of the directional coupler comprises a generator with a first N outputs of the reference signal frequency F _n phase detectors and second N outputs radiation modulators signals N lasers, where i-th output radiation modulator i-th laser step voltage with the number of steps 2 ⁿ is the i-th output of the modulation signal of the i-th laser. 10. The coherent optical radiation device according to claim 4, characterized in that the device for controlling the synthesized frequency signal phase modulator with a step signal with a number of steps 2 ^p and controlling the amount of time delay between modulating signals with a number of steps 2 ^p contains N channels, each of which represents the element NOT connected in series, the AND element, and a counter, the input of the i-th element NOT connected to the output of the i-th phase detector, the output of the element NOT connected to the first input of the AND element, the second input of which about coupled to i-th output of the oscillator frequency F _n, output of i-th AND gate is connected to the counting input of the i-th counter, adjusting input of which is connected with i-th output of the reset generator, counter outputs are connected to control inputs of i-th phase modulator synthesized frequency signal with a step signal with the number of steps 2 ^r .

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