RU2087072C1 - Data transmission method - Google Patents

Abstract

FIELD: computer engineering; digital data transmission systems, such as local computer networks. SUBSTANCE: method involves use of double-frequency coding in which signal transmitted is shaped by alternating W sections including signal periods with carrier frequency Fo and V sections incorporating half-periods of signals with frequency Fo/2; W sections in source signal are shifted in time (delayed) with respect to V sections through θ ranging within 0 < θ ≅ To/4; length of first half-period of V section at boundary of transition from W to V is reduced from To to To - θ and at boundary of transition from V to W length of first half-period of section W is increased from To/2 to To/2+θ, where To = 1/Fo is carrier frequency period. EFFECT: improved signal stability relative to zero line and provision for maintaining validity of decoding results in long transmission lines thereby ensuring twice as long transmission distance and more compared to phase-keyed code. 4 cl, 7 dwg

Description

 The invention relates to computer technology and can be used in digital information transfer systems, for example, in local area networks.

 When transmitting data over communication lines, dual-frequency coding is widely used. The most famous implementation of dual-frequency coding is a phase-shifted code of the Manchester II type (Shevkoplyas B.V. Microprocessor structures. Engineering solutions. M. "Radio and Communication", 1986, pp. 93-97, Fig. 6. 2c).

 The main advantages of this code include: ease of formation, self-synchronization of data in each code interval, the absence of a constant component in the spectrum and the presence of only two voltage levels in the output signal. These advantages are due precisely to the two-frequency structure of the signal (Fig. 1a). At the same time, dual-frequency coding requires a doubling of the carrier frequency compared to the usual representation of binary information in the form of a BVN code (without returning to zero), which leads to faster attenuation of the signal in the transmission line. However, the transmission range of information is limited mainly not by the amount of signal attenuation, but by the degree of distortion of the signal shape in the transmission line, which leads to an increase in complexity and a decrease in the reliability of information decoding.

 The distortion of the waveform is associated with the frequency dispersion of the propagation velocity and attenuation coefficient for various components in the signal spectrum. The greatest influence on the shape is exerted by the relative increase in the low-frequency components of the signal spectrum due to their lower attenuation in the transmission line, which leads to a loss of signal stability relative to the zero line (Fig. 2, a-d). The latter circumstance, in turn, leads to the fact that in real data transmission systems using a phase-shifted code, the transmission range in most cases is limited by the length of the transmission line corresponding to the degree of attenuation of the carrier frequency in it of the order of 10-15 dB in voltage. For example, the standard for an ETHERNET local area network using a Manchester II code with a carrier frequency of 10 MHz limits the amount of signal attenuation in the transmission line to 8.5 dB.

 Known are modifications of the phase-manipulated and other codes (US patent N: 4219890, NKI: 395-98, 1980 and A.S. USSR N: 1259493, MKI (4) H 03 M 5/12, 1984), which make it possible to increase the signal stability with respect to zero line by eliminating the so-called "local constant component" of the signal by introducing additional voltage levels in the form of a signal.

 The presence of more than two voltage levels in the output signal is a drawback, since in this case the requirements for the accuracy of forming the shape of the output signal of the transmitter increase. In addition, multilevel signals are more critical to the presence of reflected signals and interference signals in the transmission line.

The objective of the invention is the embodiment of a modification of a two-frequency code while maintaining a two-level signal by introducing into the signal an additional relative phase shift of the two signal components of the frequency sequences F ₀ and F _0/2 , where F ₀ is the carrier frequency.

 The technical result of the invention is to increase the stability of the signal with respect to the zero line and maintain decoding reliability at transmission line lengths corresponding to attenuation of the signal carrier frequency in it of more than 30 dB in voltage, which corresponds to an increase in the transmission distance by more than two times compared to the phase-shifted code.

The specified technical result in the proposed method of transmitting information using two-frequency coding, in which the transmitted signal is formed by alternating sections "W" containing periods of the signal with a carrier frequency F ₀ , with sections "V" containing half-periods of a signal with a frequency F _0/2 , is achieved in that the portions of "W" in the original signal are shifted in time (delayed) with respect to the portions "V" on the value of θ being in the range of 0 <q ≅ T _0/4, while at the interface of the "W" to " V "the duration of the first half-life stka "V" is reduced from a value T ₀ to a value T ₀ q, and at the interface of the "V" to "W" duration of the first half-cycle portion "W" increases from a value T _0/2 to a value of T _0/2 + q where T ₀ 1 / F ₀ period of the carrier frequency.

To achieve an additional technical result, which consists in obtaining the maximum resolution when decoding the signal, the value of the delay q is chosen equal to the relative phase delay of single-frequency signals with frequencies F ₀ and F _0/2 having the same pulse shape as the "V" sections and "W", when passing the last between the most remote in the sense of maximum signal attenuation - the receiver and the transmitters in the line.

On the other hand, the value of the phase shift q may be chosen equal to the maximum value of q _max T _0/4. This value is numerically equal to the limiting value of the relative phase delay Φ _max during the passage of rectangular signals with frequencies F _0/2 and F _o cable through the perfect infinite length and close in value for the same Φ maximum delay _before a real transmission lines. Using the maximum value of θ _max for the value of the additional phase shift, the design of the encoder can be simplified by eliminating the delay line from it.

 In addition, to achieve the simplicity of encoding and decoding information, the logical unit levels of the transmitted information are encoded in bit intervals containing two half-periods in the "W" sections, and logical zero levels are encoded in bit intervals containing one half-period in the "V" sections.

In FIG. 1 is an example explaining for some arbitrary sequence of bits the essence of signal formation with a code modified according to the invention and comparing it with a Manchester II code, where ac is the original encoded signal for three phase shift values θ _a 0; θ _b T _0/8 ; θ _c T _0/4 and its corresponding arrangement of bit intervals; d waveform in the endless cable with the ideal original signal attenuation parameter to shift θ _max T _0/4; e allocation signal durations in infinite attenuation at the perfect cable original signal with shift parameter θ _max T _0/4 bit intervals corresponding to the location; in Fig.2, the waveform of the signal with the Manchester II code when the latter passes through a transmission line with distributed parameters (real coaxial cable, carrier frequency F ₀ 30 MHz) for different line lengths corresponding to the values of the attenuation value of the voltage of the carrier frequency signal, where a 0 dB , b 10 dB, c 20 dB, d 35 dB; figure 3 oscillograms of a signal with a modified, according to the invention, two-frequency code, when the latter passes through a transmission line with distributed parameters (real coaxial cable, carrier frequency F ₀ 30 MHz) for different line lengths corresponding to the attenuation values for the voltage of the carrier frequency signal where a 0 dB, b 10 dB, c 20 dB, d 35 dB; figure 4, ae the change in the shape of the signals during the attenuation of rectangular signals with frequencies F ₀ and F _0/2 in an ideal coaxial cable of various lengths; figure 5 the dependence of the relative phase delay Φ of signals with frequencies F ₀ and F _0/2 on the line length for the ideal (1) and real (2) cable; in FIG. 6 areas (I and II) of admissible q values for various transmission line lengths, bounded by curves: 1 q values lying on the code stability boundary of the relative zero line; 2 values of q, providing maximum resolution when decoding the signal and corresponding to the values of the relative phase delay v of the signals F ₀ and F _0/2 ; 7 is an example implementation of an encoding device.

To clarify the essence of the method previously consider the passage of two monofrequency signal (F ₀ and F _0/2) of rectangular shape along the transmission line representing an ideal coaxial cable.

In FIG. 4 (ae) shows a sequential change in the waveform and relative phase delay v for these two signals. As the attenuation in the cable increases the relative phase delay of the signal F _0/2 with respect to F ₀ . The nature of the change in the relative phase delay v is shown in FIG. 5 (curve 1). The value of v is monotonically approaches the value of v _max T _0/4. This value is numerically equal to the limit of the relative phase delay in an ideal cable of infinite length.

In a real cable, due to the presence of a frequency dispersion of the propagation velocity and nonlinearity of the attenuation processes, the limiting value Φ _{before the} relative phase delay (Fig. 5 curve 2) is slightly less than Φ _max This value is an experimental characteristic of the entire transmission system and depends on the transmission medium , the form of the output signal of the transmitter and the value of the carrier frequency F ₀ .

The invention consists in the following. Since the attenuation of the dual-frequency signal occurs relative phase shift Φ monofrequency constituting portions (portions with a frequency F _0/2 delayed in time relative portions of frequency F _0/2 delayed in time relative portions of frequency F ₀₎ (Figure 4, be), which leads to a loss of signal stability relative to the zero line (Fig.2d), pre-emphasis is introduced into the original signal to compensate for this shift in the form of an oppositely directed relative phase shift q = Φ of tk signal. The method of generating a signal with a code modified due to an additional phase shift, as well as the principle of dividing the signal into mono-frequency sections "W" and "V" and their bit intervals "H", "L" are illustrated in Fig. 1.

In FIG. 1a shows an initial signal with q _a 0. In FIG. 1b, c shows a modified signal to two values, respectively θ _b T _0/8 and θ _c T _0/4.

Bit intervals belonging to sections "W" are indicated as "H", and bit intervals belonging to sections "V" are indicated as "L". For θ> 0, the bit intervals have an unequal length, however, as the signal attenuates, the bit intervals will gradually become equal. The waveform in an infinite ideal cable with the attenuation of the original signal with the parameter q T _{0/4 is} shown in FIG. 4d. In it, bit intervals and durations of periods (Fig. 4e) correspond to the original signal with q 0.

If we compare the bit intervals for the modified code at q 0 (Fig. 1a) and the bit intervals of the phase-shifted code of the Manchester II type shown at the bottom of FIG. 1a, it can be seen that they are shifted relative to each other by T _0/2 . It is possible to present information in a modified code in such a way that, for the case q 0, the output waveform coincides with the waveform with a phase-shifted code for the same bit sequence. However, from the point of view of achieving simplicity of encoding and decoding information, encoding with a logical “1” with an interval “H” and a logical “0” with an interval “L” is more convenient. The following representation is used below.

The range of allowable values of the additional phase shift introduced into the original signal corresponds to the range of possible values of the relative phase delays of mono-frequency signals during attenuation in lines of different lengths, i.e. 0 <q ≅ T _0/4. In this case, the maximum length depends on the selected quantity q at which the signal remains stable with respect to the zero line and, accordingly, the possibility of decoding information is preserved.

 In FIG. 6 shows a region (I + II) of acceptable phase shift values q for various transmission line lengths. This area is bounded on the right by curve 1 — the code stability boundary with respect to the zero line. However, the points on this curve correspond to zero margin of noise immunity, therefore, the real limit values for the length of the transmission line, taking into account the margin of noise immunity, are inside region II, bounded to the left of curve 2. This curve contains points that correspond to the maximum resolution for decoding .

 The maximum resolution is possible if the lengths of the bit intervals “H” and “L” are equal, which occurs at the moment when the phase shift q in the original signal is compensated by the relative relative phase delay v arising from the attenuation of the signal. Curve 2 is characteristic of a particular transmission system and its shape depends on the characteristics of the transmission line, the value of the carrier frequency, and the shape of the output signal of the transmitter. This curve can be used to select the best signal coding option.

When the transmission line lengths do not exceed the length that provides a guaranteed margin of noise immunity and resolution of decoding at q 0, conditionally shown on the graph l _mii , it is advisable to use the Manchester code. In transmitting systems, where l _max > l _mii , it is optimal to use a modified code with the q value lying on the lower branch of curve 2. In this case, the maximum resolution for decoding is achieved and, accordingly, the maximum for a given noise immunity length. It should be noted that when using the modified code at lengths l _max <l _mii , an additional margin in noise immunity can be obtained.

Limiting the distance l _before achieved when using the modified code to this transmission system corresponds to the inflection point of the curve 2, and q respectively _{before the} shift is a limit value for which it is possible to obtain the maximum transmission distance.

On the other hand, if we limit the maximum transmission distance to L _0.25 , which corresponds to the intersection of curve 2 with the horizontal line θ = θ _max, then for the magnitude of the phase shift, it is possible to use the value θ = θ _max . θ _{max /} unlike θ _pre does not depend on the specific characteristics of the transmitting system. This delay is easier to generate in the encoder.

 The possibility of flexible selection of the phase shift θ allows you to build adaptive transceiver systems that are tuned to a specific distance between the receiving and transmitting devices. On the other hand, if one transmitting medium, which is usual in local area networks, uses several transceivers at the same time, or if it is required for all transceivers to provide a single parameter setting, in this case, the specific value of q for this system is selected based on the tasks set during the design: achieving the maximum range, or achieving maximum noise immunity, or simplifying the encoding device, etc.

Directly from the essence of the invention and the above presentation of information, the bit intervals "H" and "L" follow a simple algorithm for constructing an encoding device: to obtain an output signal, two signals delayed relative to each other with frequencies F ₀ and F _0/2 , or with multiple them frequencies.

Figure 7 shows an example implementation of an encoding device that contains a counting trigger 1, a D-trigger 2, a counting trigger 3, a logic element 4, a logic element (multiplexer) 5 and a delay unit 6. Here, signals with a double frequency are used to form the final signal F _2w 2F ₀ and F _2v F ₀ . The signal F _{0 is} received from the output of trigger 1. The signal F _{2v is} received at the output of element 4 as a function of signals 2F ₀ and F ₀ . The signal F _{2w is} obtained from the signal 2F ₀ delayed in the delay unit 6. The signal delay value in block 6 is chosen equal to the sum of the relative phase shift q and the signal delay values in trigger 1 and logic element 4. Trigger 2 synchronizes the phase of the incoming data along the falling edge of the signal F ₀ and generates control signals for the multiplexer 5. Depending on the incoming data, the multiplexer 5 connects to the input of the trigger 3 either a signal with a frequency of F _2w or a signal with a frequency of F _2v . At the output of trigger 3, the frequency of the input signal is divided by 2 and a signal is formed with the required phase shift. Paraphase signals from the output of trigger 3 are used to control the end stage of the transmitter.

Due to the periodic nature of the signals F ₀ and 2F _{0, the} relative delay can also be obtained due to the additional delay of the signal with a lower frequency.

In the particular case of q T _0/4 of the encoder can be generally excluded delay line and the corresponding phase shift can be generated using a signal with carrier frequency quadrupled - 4F _0.

Described modified code uniquely decoded for any value 0 <q ≅ T _0/4. This is a consequence of the redundancy of the two-frequency code, in which one logical state of the transmitted information, indicated by "H", is transmitted by two signal intervals with shorter durations, and the other, indicated by "L", is transmitted by one interval with a longer duration. For the modified code under consideration, the assignment of intervals to the “H” or “L” state depends on the identification of previous states:
if the previous interval corresponds to a bit interval of "L", then the interval of duration t> (3T _0/4 q / 2) refers to the bit interval of "L", and the interval of duration t <(3T _0/4 θ / 2) refers to the first half-period of the bit interval "H";
if the previous interval was the first half-period of the bit interval "H", then the current interval is considered, regardless of the duration, as the second half-period of the bit interval "H";
if the previous slot was the second half-bit interval "H", then the interval of duration t> (3T _0/4 + θ / 2) refers to the bit interval of "L", and the interval of duration t <(3T _0/4 + θ / 2) refers to the first half-period of the bit interval "H".

Presentation of information in the form of two-frequency code, formed by alternating sections of "W", containing periods of the signal with a carrier frequency F _0, and plots "V", comprising half cycles of the signal with frequency F _0/2 are shifted relative to each other in phase by an amount θ located in the range of 0 <q ≅ T _0/4, where T ₀ 1 / F ₀ between the carrier frequency, so that at the interface of the "W" to "V" duration of the first half-cycle portion "V" decreases from a value T ₀ to q value T ₀ and at the interface of the "V" to "W" duration of the first half cycle lan and "W" increases from a value T ₀ to a value of T _0/2 + q in which the transmitted information levels of logic-one coded bit intervals of "H", comprising two half-period composed plots "W", and the levels of logic-zero encoded bit intervals of "L "containing one half-cycle in the structure of sections" V ", it is proposed to call BASK-code (BASK-code) or two-frequency code with a delayed phase.

Claims (4)

1. A method of transmitting information using two-frequency coding, in which the transmitted signal is formed by alternating sections of W containing periods of the signal with a carrier frequency of F ₀ , with sections of V containing half-periods of the signal with a frequency of F _0/2 , characterized in that the sections of W in the initial signal is shifted in time with respect to sections V by a value of θ located in the interval 0 <θ ≅ T _o / 4, while at the boundary of the transition from W to V, the duration of the first half-period of section V is reduced from T ₀ to T _o - θ, and at the boundary n Navigate from V to W duration of the first half-cycle portion is increased with the value W ₀ T / 2 to the value T _o / 2 + θ, where T ₀ 1 / F ₀ between the carrier frequency. 2. A method according to claim 1, characterized in that the value of the shift amount θ is selected at a value of the relative phase delay of signals at frequencies F ₀ / F ₀ 2 and having the same pulse shape as the corresponding portions of V and W, the passage these signals between the most distant in the sense of maximum signal attenuation by the receiver and transmitter. 3. A method according to claim 1, characterized in that the shift amount q is chosen equal to the maximum value of q _max = T _o / 4, which is numerically equal to the limit value of the relative phase delay of signals at frequencies F ₀ and F _0/2, having pulses with an envelope rectangular in shape, when passing through the perfect coaxial cable of infinite length.  4. The method according to claim 1, characterized in that the logical unit levels of the transmitted information are encoded in bit intervals containing two half-periods in the sections of W, and the logical zero levels are encoded in bit intervals containing one half-period in the sections of V.

Images