RU2006044C1 - Receiver - Google Patents

Abstract

FIELD: measurement technology. SUBSTANCE: panoramic receiver has aerial 1, two sweep generators 2, 22, local oscillator 3, mixer 4, two intermediate frequency amplifiers 5, 7, detector 6, two meters 8, 9 of width of spectrum, comparator 10, threshold unit 11, key 12, delay line 13, multiplier 14, n processing channels each composed of band-pass filter 15, amplitude detector 16, video amplifier 17, vertical electrode 19, (n+1)-th amplitude detector 20, differentiation unit 21, second sweep generator, (n+1)-th cathode-ray tube 23, n+2 band-pass filters 24, 26, frequency divider 27, phase inverter 27, (n+2)-th cathode-ray tube 28. EFFECT: improved operational characteristics. 3 dwg

Description

 The invention relates to radio engineering and can be used to search and detect phase-shift (PSK) signals, as well as a visual assessment of their main parameters.

Of the known devices, the closest to the proposed one is a panoramic receiver selected as a prototype, which provides the search and detection of PSK signals in a given frequency range Df, as well as a visual assessment of their carrier frequency, but does not allow to visually evaluate such basic parameters as the law of phase manipulation , the magnitude of the phase jumps Δφ, the duration τ _n and the number N of chips.

 The aim of the invention is to expand the functionality by visually assessing the law of phase manipulation, the magnitude of the phase jumps, the multiplicity of phase manipulation, the duration and number of chips of the detected phase-manipulated signal.

This goal is achieved by the fact that an (n + 1) -th amplitude detector, a frequency detector, a differentiating unit, a second sweep generator, an (n + 1) -th and (n + 2) -th band-pass filters, a frequency divider eight, a phase shifter of 90 ^° , an (n + 1) -th and (n + 2) -th cathode ray tube, and an (n + 1) -th amplitude detector, a differentiating unit, a second sweep generator and the horizontal electrode of the (n + 1) -th cathode ray tube, the vertical electrode of which through the frequency d Tector connected to the output switch to the output of the frequency multiplier are connected sequentially into eight (n + 1) -th band-pass filter, a frequency divider by eight, (n + 2) -th band-pass filter, phase shifter ⁹⁰ and a vertical electrode (n + 2 ) th cathode ray tube, the horizontal electrode of which is connected to the output of the (n + 2) th bandpass filter, and the control electrode is connected to the output of the frequency detector.

 The block diagram of the proposed receiver is presented in FIG. 1. A view of the possible waveforms is shown in FIG. 2. Timing diagrams illustrating the operation of the receiver are shown in FIG. 3.

The panoramic receiver contains an antenna 1, a first sweep generator 2, a local oscillator 3, a mixer 4, an intermediate frequency amplifier 5, a detector 6, an eight frequency amplifier 7, a first spectral width meter 8, a second spectral width meter 9, a comparison unit 10, a threshold block 11 , key 12, delay line 13, multiplier 14, n processing channels, each of which consists of a series-pass bandpass filter 15i, amplitude detector 16i, video amplifier 17i and vertical CRT electrode 18i, the horizontal electrode of which is connected to Odometer generator 2 scan (i = 1, 2,..., n), frequency detector 9, (n + 1) -th amplitude detector 20, differentiating unit 21, second scan generator 22, (n + 1) th CRT 23, (n + 2) -th band-pass filter 24, frequency divider 25 by eight, (n + 2) -th band-pass filter 26, phase shifter 27 by 90 ^о and (n + 1) -th CRT 28. Moreover, to the output of the generator 2 sweeps, a local oscillator 3, a mixer 4, the second input of which is connected to the output of the antenna 1, an intermediate frequency amplifier 5, an eight frequency multiplier 7, a spectral width meter 9, a comparison unit 10, and a second input are connected in series through a spectral width meter 8 connected to the output of the intermediate frequency amplifier 5, a threshold block 11, a key 12, the second stroke of which is connected to the output of the intermediate frequency amplifier 5, a delay line 13, a multiplier 14, the second input of which is connected to the output of the key 12, and n processing channels. The output of the intermediate frequency amplifier 5 is connected in series with the (n + 1) th amplitude detector 20, the differentiating unit 1, the sweep generator 22, and the horizontal electrode of the (n + 1) th CRT 23, the vertical electrode of which is connected through the frequency detector 19 to the key output 12. The output of the frequency multiplier 7 by eight is connected in series with the (n + 1) -th band-pass filter 24, the divider 25 of the frequency by eight, the (n + 2) -th band-pass filter 26, the phase shifter 27 by 90 ^°, and the vertical electrode (n + 1) th CRT 28, the horizontal electrode of which is connected to the output of the strip fil tra 26, and the control electrode - output from the frequency detector 19.

. . 18_n, которая используется как ось частоты, причем ее длина соответствует полосе обзора частотного диапазона Дf. Ключ 12 в исходном состоянии всегда закрыт. Принимаемый ФМн-сигнал
U_c(t) = V_c cos[ω_ct + φ_c(t) + φ_c] , 0 ≅ t ≅ Tc, где V_c, ω_c, φ_c, T_c - амплитуда, несущая частота, начальная фаза и длительность сигнала;
φ_K(t) - манипулируемая составляющая фазы, отображающая закон фазовой манипуляции в соответствии с модулирующим кодомM(t) (см. фиг. 3а), причем φ_K(t) = const при K τ_n < t < (K + 1) τ_n и может изменяться скачком на Δ φ при t = K τ_n , т. е. на границах между элементарными посылками (K + 1, 2, . . , . N-1);
τ_n, N - длительность и количество элементарных посылок, из которых составлен сигнал длительностью Т_с(Т_с = N τ_n); с выхода антенны 1 поступает на первый вход смесителя 4, на второй вход которого подается напряжение гетеродина 3 линейно изменяющейся частоты
U_г(t) = V_г cos(ω_гt + π γt² + φ_г), 0 ≅ t ≅ Т_п, где V_г, ω_г, φ_г, Т_п - амплитуда, начальная частота, начальная фаза и период повторения напряжения гетеродина,
γ=

- скорость изменения частоты гетеродина (скорость изменения частоты первой гармоники гетеродина).Panoramic receiver operates as follows. View a given frequency range Df is carried out using a sweep generator 2, which periodically with a period T _p according to the sawtooth law tunes the frequency of the local oscillator 3. At the same time, sweep generator 2 forms a horizontal scan of the CRT 18

. . 18 _n , which is used as the frequency axis, and its length corresponds to the frequency span Df. The key 12 in the initial state is always closed. Received QPSK signal
U _c (t) = V _c cos [ω _c t + φ _c (t) + φ _c ], 0 ≅ t ≅ Tc, where V _c , ω _c , φ _c , T _c - amplitude, carrier frequency, initial phase and signal duration;
φ _K (t) is the manipulated component of the phase that displays the law of phase manipulation in accordance with the modulating code M (t) (see Fig. 3a), and φ _K (t) = const for K τ _n <t <(K + 1) τ _n and can change abruptly by Δφ for t = K τ _n , i.e., at the boundaries between elementary premises (K + 1, 2,.... N-1);
τ _n , N is the duration and number of chips that make up a signal of duration T _s (T _s = N τ _n ); from the output of the antenna 1 is fed to the first input of the mixer 4, the second input of which is the voltage of the local oscillator 3 of a ramp frequency
U _g (t) = V _g cos (ω _g t + π γt ² + φ _g ), 0 ≅ t ≅ T _p , where V _g , ω _g , φ _g , T _p - amplitude, initial frequency, initial phase and heterodyne voltage repetition period,
γ =

- the rate of change of the local oscillator frequency (the rate of change of the frequency of the first harmonic of the local oscillator).

K₁U_с·U_г ;
K₁ - коэффициент передачи смесителя;
ω_пр = ω_с - ω_г - промежуточная частота;
φ_пр = φ_с - φ_г - промежуточная начальная фаза; которое представляет собой сигнал с комбинированной линейной частотной модуляцией и фазовой манипуляцией (ЛЧМ-ФМн). Это напряжение поступает на вход обнаружителя 6, состоящего из умножителя 7 частоты на восемь, измерителей 8 и 9 ширины спектра, блока 10 сравнения, порогового блока 11 ключа 12.At the output of the mixer 4, voltages of combination frequencies are generated. Amplifier 5 is allocated voltage intermediate (differential) frequency
U _pr1 (t) = V _pr cos [ω _pr t + φ _K (t) -
-π γ ₁ t ² + φ _CR ], 0 ≅ t ≅ T _s , where U _CR =

K ₁ U _s · U _g ;
K ₁ - gear ratio of the mixer;
ω _CR = ω _s - ω _g - intermediate frequency;
φ _CR = φ _s - φ _g - intermediate initial phase; which is a signal with a combined linear frequency modulation and phase shift keying (LFM-PSK). This voltage is fed to the input of the detector 6, consisting of a frequency multiplier 7 by eight, meters 8 and 9 of the spectrum width, block 10 comparison, threshold block 11 of the key 12.

At the output of the multiplier 7, a frequency of eight forms a voltage
U ₁ (t) = U _pr cos [8 ω _pr t - 8 π γ ₁ t ² + 8 φ _pr ],
0 ≅ t ≅ T _s ,
Since 8 φ _K (t) = 0.8 when receiving a signal with a single phase shift keying [FMN-2, φ _K (t) = 0, π], 8 φ _K (t) = 0.4π, 8π, 12π for receiving a signal with double phase shift keying [FMN-4, φ _K (t) = 0, π / 2, π, 3 / 2π], 8 φ _K (t) = 0.2 π, 4π, 6π, 8π, 10π, 12π, 14π when receiving a signal with three-phase phase shift keying [FMn-8, φ _K (t) = 0, π / 4, π / 2, 3 / 4π, π, 5 / 4π m 3 / 2π, 7 / 4π], then in the indicated oscillation phase manipulation is already absent.

The width of the spectrum Δf _{1 of the} eighth harmonic is determined by the duration T _{s of the} signal (Δf ₁ = 1 / T _s ), while the width of the spectrum Δf _{s of the} QPSK signal is determined by the duration of its elementary premises (Δf _c = 1 / τ _n ), i.e., the width of the spectrum Δf _{1 of the} eighth harmonic of the signal is N times smaller than the width of the spectrum Δf _{from the} output signal: Δf _c / Δf ₁ = N. Therefore, when the frequency of the QPSK signal is multiplied by eight, its spectrum "folds" N times. This allows the PSK signal even when its power at the receiver input is less than the noise power.

The spectral width Δf _{from the} input QPSK signal is measured using meter 8, and the spectral width Δf _{1 of the} eighth harmonic of the signal is measured using meter 9. Voltages V and V ₁ proportional to Δf _c and Δf _1, respectively, from the outputs of meters 8 and 9 of the spectrum width arrive at the two inputs of block 10 comparison. Since V >> V ₁ , a positive voltage is generated at the output of the comparison unit 10, which exceeds the threshold level of V _pores in the threshold unit 11.

The threshold level V _pore is chosen so that it does not exceed random interference. When the threshold level V _pores is exceeded, a constant voltage is generated in the threshold block 11, which is supplied to the control input of the key 12, opening it. In this case, the voltage U _pr1 (t) from the output of the intermediate frequency amplifier 5 through the public switch 12 is simultaneously supplied to the first input of the multiplier 14 and to the input of the delay line 13, at the output of which a voltage is generated
_Np2 U (t) = U _pr1 (t -τ _s) = V ^_ave. cos [ω _ol (t -
-τ _h ) + φ _K (t - τ _h ) - π γ ₁ (t-τ _h ) ² + φ _pr ],
0 ≅ t ≅ T _c , where τ _s is the delay time of the delay line 13.

=

K₂U $\genfrac{}{}{0ex}{}{\text{2}}{\text{п}}$ _р;
K₂ - коэффициент передачи перемножителя 14;
ω_б1 = 2π γ₁ τ_з - частота биений;
φ_K1(t) = φ_K(t -τ_з) - φ_K(t);
φ_б1 = ω_пр τ_з + π γ₁ τ_з ² .The output of the multiplier 14 produces a voltage
U _b1 (t) = Vb. cos [ω _b1 t + φ _K1 (t) -
- φ _b1 ], 0 ≅ t ≅ T _c , where U

=

K ₂ U $\genfrac{}{}{0ex}{}{\text{2}}{\text{P}}$ _p ;
K ₂ - transfer coefficient of the multiplier 14;
ω _b1 = 2π γ ₁ τ _s - beat frequency;
_{φ K1 (t) = φ K} (t -τ _s) - φ _K (t);
φ _b1 = ω _pr τ _z + π γ ₁ τ _z ² .

The carrier frequency of the beat voltage is ω _b1 = 2π γ ₁ τ _s = const. Consequently, with a fixed delay time τ _s , a mono-frequency beat signal is generated at the output of the multipliers 14, the carrier frequency ω _{b1 of} which depends on the rate of change of the frequency γ _i (i = 1, 2, ..., n local oscillator 3. The rate of change of the signal conversion frequency, coming to the input of the autocorrelator, depends on the harmonic number of the local oscillator frequency 3, interacting with the carrier frequency of the received PSK signal.

The tuning frequency of the bandpass filter 15 ₁ is chosen equal to
ω _n1 = ω _b1 = 2π γ ₁ τ _s , the tuning frequency of the bandpass filter 15 ₂ is chosen equal
ω _n2 = ω _b2 = 2π γ ₂ τ _s , and the tuning frequency of the bandpass filter 15n is chosen equal to
ω _nn = ω _bn = 2π γ _n τ _s , where γ _n = nγ ₁ is the rate of change of the nth harmonic of the local oscillator frequency.

The voltage U _bi (t) from the output of the bandpass filter 15i is supplied to the input of the amplitude detector 16i, where it is detected and, after amplification in the video amplifier 17i, is applied to a vertical CRT electrode 18i, on the screen of which a pulse is generated (frequency mark).

The position of the frequency mark on the horizontal scan of the CRT 18i uniquely determines the carrier frequency ω _{from the} received PSK signal (i = 1, 2,..., N).

 Therefore, the harmonic number of the local oscillator frequency 3, with which the carrier frequency of the received PSK signal interacts, is determined by the number of the band-pass filter of that channel on the CRT screen of which a frequency mark is observed (see Fig. 2a).

The voltage U _pr1 (t) (see Fig. 3b) from the output of the intermediate frequency amplifier 5 is simultaneously supplied to the input of the amplitude detector 20 and through which the key 12 to the input of the frequency detector 19. At the output of the latter, a sequence of short bipolar pulses is generated (see Fig. 3c) the temporary position of which corresponds to the moments of an abrupt change in the phase of the input QPSK signal (see Fig. 3b). The amplitude detector 20 emits an envelope of the spectrum (see Fig. 3g), which is differentiated by the unit 21 is converted into two short bipolar pulses (see Fig. 3h). Moreover, a short positive pulse starts, and a short negative pulse closes the oscillator 22 of the sweep, which generates a sawtooth voltage (see Fig. 3i). This voltage is used as the scan voltage and is supplied to the horizontally-deflecting plates of the CRT 23, on the screen of which a horizontal scan is formed. The length of the specified sweep is proportional to the duration T _{s of the} detected PSK signal. A sequence of short bipolar pulses (see Fig. 3c) from the output of the frequency detector 19, which can be visually observed on its screen (see Fig. 2b), is fed to the vertically-deflecting plates of the CRT 23.

Moreover, the minimum "distance" between short bipolar pulses corresponds to the duration τ _{n of the} elementary premises (see Fig. 3a). By counting the number of phase jumps ν (the number of short bipolar pulses), we can estimate the number N of the elementary premises. The following relationship exists between the number of phase jumps ν and the number N of chips
ν = 0.5 (N-1).

 By the number and sequence of short bipolar pulses, the law of phase manipulation is estimated.

For a visual assessment of the magnitude of the phase jumps Δ φ and the multiplicity of phase manipulation of the detected PSK signal, a circular CRT 28 is used. Moreover, a circular scan is formed using voltage extracted directly from the detected PSK signal. To this end, the voltage U ₁ (t) (see Fig. 3d) from the output of the frequency multiplier 7 by eight is allocated by a band-pass filter 24 and applied to the input of the frequency divider 25 by eight, the output of which produces a voltage
U ₂ (t) = V _pr cos (ω _pr t - π γ ₁ t ² + φ _pr ),
0 ≅ t ≅ Tс which is allocated by a band-pass filter 26 and used to form a circular scan of a CRT 28. For this, the voltage U ₂ (t) (see Fig. 3d) is applied directly to the horizontally deflecting plates, and through the phase shifter 27 to 90 ^about vertical deflecting plates of a CRT 90. A sequence of short bipolar pulses (see Fig. 3c) that modulates the electron beam in brightness arrives at the control (modulating) electrode of a CRT 28. On the screen of the CRT 28, an image is formed in the form of bright points (see Fig. 2, c, d, e). Moreover, the number of bright points is equal to the multiplicity m of phase manipulation, and the angular distance between them determines the magnitude of the phase jumps Δ φ. When the next PSK signal falls into the passband of the amplifier 5 of the intermediate frequency, the receiver operates in a similar way.

Thus, the proposed receiver in comparison with the prototype provides a visual assessment of not only the carrier frequency of the detected phase-manipulated signal, but also its other parameters such as the law of phase manipulation, the magnitude of the phase jumps Δ φ, the multiplicity m of phase manipulation, the signal duration T _s , duration τ _n and the number N of chips. Thus, the functionality of the panoramic receiver is expanded. (56) Copyright certificate of the USSR N 1661661, cl. G 01 R 23/00, 1989.

Claims (1)

A RECEIVER containing a receiving antenna in series, a mixer, the second input of which is connected through the local oscillator to the first output of the first sweep generator, an intermediate frequency amplifier, an eight frequency multiplier, a second spectral width meter, a comparison unit, the second input of which is connected to the first spectral width meter with the output of the intermediate frequency amplifier, a threshold block, a key, the second input of which is connected to the output of the intermediate frequency amplifier, a delay line, a multiplier, the second input of which connected to the key output, and n channels, each of which consists of a series-pass bandpass filter, an amplitude detector, a video amplifier and a vertical electrode of a cathode ray tube, the horizontal electrode of which is connected to the second output of the first scan generator, characterized in that (n + 1) th amplitude detector, frequency detector, differentiating block, second sweep generator, (n + 1) th and (n + 2) th band filters, frequency divider by eight, phase shifter by 90 ^o , (n + 1) - and (n + 2) -th electron of the urethral tube, moreover, the (n + 1) -th amplitude detector, the differentiating unit, the second scan generator and the horizontal electrode of the (n + 1) -th cathode ray tube, the vertical electrode of which is connected to the by the key output, the (n + 1) -th band-pass filter, the frequency divider by eight, the (n + 2) -th band-pass filter, the phase shifter by 90 ^o and the vertical electrode (n + 2) -th are connected in series to the output of the frequency multiplier by eight cathode ray tube, horizon ny electrode connected to the output (n + 2) -th band-pass filter, and a control electrode - output from the frequency detector.

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