RU116293U1 - RADIO RECEIVER OF HIGH SPEED INFORMATION OF SPACE RADIO LINE - Google Patents

Abstract

The device for receiving high-speed information of a space radio link contains:! bandpass filter (BPF), matched radio signal reception filter with one side (SFOB), balanced modulator (BM), first demodulator (DM), first block decoder (DC), information receiver (PI), receiver signal sample memory unit (PAMP) , a device for compensating a parasitic phase shift (UC SPS), a device for compensating for a parasitic shift of spectral components (UC SSS), a matched filter for receiving a radio signal from one side (SFDB), a second balanced modulator (BM), a second demodulator (DM), a second block decoder (DC), PC of the control unit (PC), PC bus interface (interface), software (software), and the input of the device is the PF input, the output of which is connected to the series-connected SFOB, BM, DM, DC, PI, the second output of the DM connected to the PAMP, the output of which is connected to the input of the phase signal readings of the CC SFS and CC CC, the outputs of which are connected to the second input of the PI, the output of the BM is connected to the first input of the exemplary signal of the CC CCS, the SFOB input is connected to ohm SFDB, the output of which is connected to the input of the second BM, the second DM, the second DC, the output of which is connected to the third input of the PI, the software is connected to the PC, which is connected by the interface to the first DC, the second DC, PI, UK SFS, UK SSS, PAMP.

Description

The utility model relates to radio engineering and can be used in radio systems for promptly delivering high-speed information via a space radio link to a consumer.

Information delivery involves the passage of a signal in communication and data transmission (DSPD) from the spacecraft to the consumer. The addressable, multi-channel system of data storage systems operates in a network of information lines and terminals, contains satellite, ground relay of a communication system, a fiber optic communication line, which form the basis of the MIL-STD-1553A standard for multiplexed information lines of airborne information collection and distribution systems. The weak point of information delivery occurs in the radio link from the spacecraft to the receiving point of the SDKD, since the probability of reception depends not only on the signal-to-noise ratio of the radio link, but also on the Doppler effect arising from the motion of the spacecraft.

It is known that the Doppler effect complicates the reception of information, fading and distortion of the signal are formed in certain parts of the flight path, the illustrations in figures 1-7.

In the prior art, real-time digital information radio reception schemes are known in which the carrier frequency is entered into the communication and the carrier frequency is monitored, the reception of blocks and information bits is synchronized, the received signals are filtered, the information blocks are decoded by error-correcting decoding. Information transmission via a radio link is provided by radio systems that carry a message s [λ, φ (t), t] with a continuous parameter vector λ and a discrete parameter φ (f) in the radio signal. The signal is transmitted in total with the synchronization signal s ₁ [λ, t], which in turn is used to extract the discrete parameter φ (t) from the radio signal. In this case, the clock receiver is synchronized based on the equations of nonlinear filtering, and the receiver that selects the discrete component is constructed in accordance with the theory of optimal solutions [1, p. 100]. To combat interference, complex signal receivers with low-noise high-sensitivity elements, new types of modulation and interference filtering methods are used. The transmission probability is increased by error-correcting coding of the radio signal of the message [2, 3]. The receiving block diagram contains a bandpass filter (PF), a matched filter (SF), a balanced modulator (BM), a demodulator (DM), a block decoder (DK), and an information receiver (PI).

In addition to the signal reception schemes s [λ, φ (t), t] with real-time noise-correcting decoding, there are schemes that record the signal before digitalizing, find and eliminate distortions in it (until the received symbols are extracted from the signal and information is decoded ), allowing to increase the probability of receiving information. For example, the circuit of Fig. 8 [6], containing the reception of the signal s [λ, φ (t), t] by the PF, SF, BM, DK, PI devices, records the phase signal in the memory unit (PAM _P ) 7, if it is determined status information DC. With the signal block breaker (ППБС) 8, when the status information of the DC is turned on, the operation of the device for compensating the stray phase signal offset (CC SPS) 9, which determines and compensates for the stray offset of the DC component of the signal from the Doppler effect and noise interference. CC SFS, the circuit of which is depicted in Fig. 8, contains a stray signal bias calculator (SPS) 10, a digital signal offset compensation device (CCU) 11, a symbol solving unit (RBS) 12, and a second recreation center 13. The CC SFC determines and compensates, determines the initial bias φ _g and the parameter Ω _{g of the} linear law of the change in the bias in time, an estimate of the stray bias of the phase signal is obtained using the phase frequency measurement method [4, p. 428] from the phase signal samples e _c (t _i ). TSUK compensates for spurious signal bias. According to the block symbols generated by the EinSS, the second recreation center performs noise-resistant decoding and transmitting PI information.

The circuit of Fig. 8 is selected by the prototype.

The Doppler effect creates distortion of the signal, FIGS. 1-7, in separate sections of the spacecraft flight path when transmitting information on a channel with a given band W and signal-to-noise ratio.

The technical result of the claimed utility model is to reduce distortion and signal loss in circuits with noise-resistant coding in a communication session, depending on the Doppler effect.

The technical result is achieved in that the device for receiving high-speed information of a space radio link contains:

band-pass filter (PF) 1, matched filter for receiving a radio signal from one side (SOPF) 2, balanced modulator (BM) 3, first demodulator (DM) 4, first block decoder (DK) 5, information receiver (PI) 6, memory block samples of the signal of the recipient (PAM _P ) 7, the compensation device for the stray phase shift signal (CC SFS) 9, the device for compensation for stray shift of spectral components (CC SSS) 14, the matched filter for receiving a radio signal from one side (SPSB) 22, the second balanced modulator (BM ) 23, the second demodulator (DM) 24, the second decoder blo (DK) 25, PC of the control unit (PC) 26, PC bus interface (interface) 27, software (PO) 28, and the input of the device is the PF input, the output of which is connected to SFO 2, BM 3, and DM 4 connected in series , DC 5, 6 PI, a second output connected to the SID 2 PAM _II, whose output is connected to an input of phase samples CC SPS signal 9- CCC CC ₃ and 14 _-3, the outputs of which are connected to the second input UI BM output connected to a first input exemplary CC CAS signal 14 _-1 SFOB input coupled to the input SFDB whose output is connected to the input connected placentas tion, BM second, 23 second DM 24, a second DC 25, whose output is connected to a third input of the PI 6 _-3, PO is connected to a PC, which is connected to a first DC interface, the second DC, PI, CC SPS, MC CCC PAM _P .

The features and essence of the utility model are explained in the detailed description illustrated by the drawings, which shows the following:

in figure 1. Fading of the signal from the Doppler effect;

figure 2. Distortions of the meander;

figure 3. Carrier frequency reception

figure 4. Signal modulation meander;

figure 5. Square wave distortion from phase shift;

in Fig.6. The signal of the component vectors;

in Fig.7. The phase shift of the components;

on Fig. The prototype scheme, where indicated:

1 - band-pass filter (PF);

2 - matched filter (SF);

3 - balanced modulator (BM);

4 - demodulator (DM);

5, 13, - the first and second block decoders (DC);

6 - recipient of information (PI);

7 - memory block (PAM) of the samples of the recipient signal.

8 - chopper receiving signal block (PPBS);

9 - a device for compensating for stray phase signal displacement (CC SPS);

10 - calculator stray signal bias (IPN);

11 - digital signal offset compensation device (CCU);

12 - decisive block character (RBS);

Fig.9. Scheme of the claimed device for receiving high-speed information of a space radio link, where it is indicated:

1 - band-pass filter (PF);

2 - a consistent filter for receiving a radio signal from one side (SFOB);

3, 23 - the first and second respectively balanced modulator (BM);

4, 24 - the first and second, respectively, demodulator (DM);

5, 25 - the first and second, respectively, block decoder (DC);

6 - recipient of information (PI);

7 - memory block phase samples of the recipient signal (PAM _P );

9 - a device for compensating for stray phase signal displacement (CC SPS);

14 - a device for compensating for stray shift of spectral components (CC SSS);

22 - matched filter for receiving a radio signal with two side (SFDB);

26 - PC;

27 - interface;

28 - software unit control the device and display its status (software);

figure 10. Scheme of the Criminal Code SPS 9, where it is indicated:

10 - calculator stray signal bias (IPN);

11 - digital signal offset compensation device (CCU);

12 - decisive block character (RBS);

13 - the second block decoder (DK);

figure 11. The scheme of the CC SSS 14, where it is indicated:

15 - sample signal distortion calculator (VIS)

16 - block distortion compensation (BKI),

17 is a block of samples of the corrected phase signal (BFS),

18 - the first decision block character (RBS),

19 - the third block decoder (DK);

20 - memory block samples of the reference signal (PAMos)

21 - third demodulator (DM);

on Fig. Spectral representation of a modulating sinusoid;

on Fig. Vector representation of the components of an FFT;

on Fig. Decoding scheme formed by a set of codecs;

on Fig. Vector image of a reference signal;

in Fig.16. Example configuration search;

on Fig. The formation of a series of errors.

The following notation is used in the text:

SSPD - means of communication and data transfer, containing communication systems for space systems: on-board signal transmitters and ground communication stations, exchanging relevant types of information (digital, telephone, etc.) [5, p.16];

"Message" - a high-frequency or low-frequency signal of the receiving device in a temporary form or spectral form, carrying information transmitted to the consumer (software);

“Reference signal” - reference signal s _arr [λ, t], a high-frequency or low-frequency signal of the receiving device in a temporary form or spectral form, used to determine the discrepancy of the spectrum components from the Doppler effect, transmitted in the radio link in total with the radio signal of the message s [λ, φ (t), t] and the synchronization signal s ₁ [λ, t];

s _fs [λ, t] - characteristics of the transmitted message — signal in the block (initial displacement φ _g and parameter Ω _{g of the} linear law of the change in time displacement);

f _s (t _i ) - samples of the phase signal in the transmitted data block.

In the structural diagram of the radio reception device for high-speed information of the space radio link, Fig. 9, the devices making up the circuit, enter into communication by the carrier frequency, synchronize the clock frequency of the reception of bits, synchronize the reception and reception of messages in real time. Blocks PF 1, SFDB 22, BM 23, DM 24, DC 25 receive the signal with two side, information from the DC 25 receives PI 6. Blocks PF 1, SFOB 2, the second blocks BM 3, DM 4, DC 5 receive the signal from one side, information from the second recreation center 5 receives PI 6. The matched filters for receiving the radio signal SFOB 2 and SFDB 22 respectively filter the radio signal with one and two side frequencies. The first and second DM perform analog-to-digital conversion of the phase signal. Reception BM, DM, DC is built in the traditional way using optimal reception schemes and high-performance noise-correcting coding schemes. The status information of the data block of the second recreation center, if errors are detected by the second decoder of the recreation center 5 and are not corrected, is transmitted via the interface to the PC, by the command of which the samples of the phase signals of the message and the reference signal are recorded when the information is received, respectively, in the PAM _P and PAM _OS . The samples of the PAM _P signals are transmitted to the CC SFS and CC SSS.

Examples of the execution of the schemes of the Criminal Code of SFS and the Criminal Code of SSS.

The circuit of the CC SFS 9, Fig. 10, contains a calculator of parasitic signal bias (SPS) 10, a digital device for compensating signal bias (CCU) 11, a decisive symbol block (RBS) 12, a second block decoder (DC) 13; the input bus of the interface of the CC SFS 9 _{-1 is} connected to the control inputs of the DK \, VPS and the CSC, the output of the CSC is connected to the first decision block symbolic (RBS), the output of which is connected to the input of the second DC, the signal output of which is the output of the CC SFS 9 _-2 , the IPN output is connected to the second input of the CCM, the input of the phase signal samples of the CC SFS 9 _-3 , is connected to the second input of the IPN and the third signal input of the CCM.

The CC CC 14 scheme, FIG. 11, contains a distortion calculator of an exemplary signal (VIS) 15, a distortion compensation unit (BCR) 16, a corrected phase signal (BFS) sampler 17, a first symbol decider (RBS) 18, and a third block decoder ( DC) 19, the memory block of samples of the sample signal (PAM _OS ) 20, the third demodulator (DM) 21, the input of the model signal of the CC SSS 14 _{-1 is} connected to the input of the DM 21, the output of which is connected to the PAM _OS whose input-output is connected to the VIS, the output of which is connected to the second input of the BKI, the output of which is connected to the signal input of the BFS 17, the output which is connected to the RBS 18, the output of which is connected to a DC 19, the output is an output keotorogo CCC CC 14 _-2, the signal input - CC CCC _-3 message 14 is connected to the third input BCH 16, CHB output connected to a second input of the BFS, the output is connected BFS with the input of the second RBS 18, CC SSS 14 _-3 , the bus interface input of the CC SSS 14 _{-4 is} connected to the first input of the BKI by the first input of the BFS

CC SPS team PC, calculates the parameters of the parasitic displacement of the phase of the signal level according to PAM _P s [λ, φ (t) , t] and compensates a phase signal samples e _c (t _i), which RBS determines the bits of the data block (“0” or “1”). An error-free information block received by error-correcting decoding is transmitted by PI 6. The status information of the DC of the CC SFS block is transmitted to the PC, from which the “On” command comes to the CC CCC block, which proceeds to signal processing. An example of the work of the Criminal Code of the SPS, containing the IPN, CCA, RBS, DC is described in [5].

CCC CC 14 according to the PC 26 command, calculates the stray shift of the spectral components in the fast Fourier transform (FFT) of the received signal, compensates them in the spectrum and, by converting it to the time domain, receives the phase signal samples of the data block. To determine the distortion, a reference signal is used. Parasitic distortion calculations use PAM _OS and PAM _P data. The spurious distortion of the phases of the spectral components is determined by the difference in the spectral components obtained by the FFT, the received s _sample [λ, t] and the emitted (undistorted by the Doppler effect) reference signal. In the signal spectrum, s [λ, φ (t), t] compensate for spurious phase distortions of the spectral components, and the inverse FFT transform produces a vector of time-domain signal samples. The samples of the corrected signal arrive at the decisive device, which forms a sequence of bits of the data block (characters "0" and "1"). The decoder performs noise-correcting decoding, the received block of information is transmitted by PI 6.

The VIS 15, BKI 16 and BFS 17 devices use the fast Fourier transform (FFT) to compensate for distortion. The FFT in the frequency domain has a frequency grid. A grid of N frequencies is formed by frequencies ω ₀ + ω (i), where ω ₀ = 2π · f ₀ , ω (i) = i · Ω, i is the frequency number in the grid, i = 0, 1, 2, ... N-1 . The grid frequency step Ω is unchanged. The difference in neighboring frequencies is the same, the Doppler shift of the difference in neighboring frequencies is the same, we denote it by Ω _D. With the Doppler effect, a grid of difference frequencies i · Ω _D and a grid of discrepancies i · (Ω _D -Ω) are formed on the frequency axis for i = 0, 1, 2 ... N-1.

The Doppler effect ascertains the linearity of the Doppler frequency shift from the velocity component of a satellite or spacecraft, directed at the point of its location along the tangent to the wave path, curved in the case of inhomogeneous propagation medium.

The linearity property is manifested in a change in the distance between the grid frequencies and a change in each grid frequency with a coefficient a _D from the Doppler effect. We take two grid frequencies ƒ ₁ = k ₁ · Ω, ƒ ₂ = k ₂ · Ω, k ₁ , k ₂ are integers, the difference frequency F ₁ = ƒ ₂ -ƒ ₁ , when there is no Doppler effect. Frequencies with the Doppler effect ƒ _1D , ƒ _2D , the difference frequency F _1D = ƒ _2D -ƒ _1D lies on the frequency axis, where the coefficient of linear frequency change from the Doppler effect a _D , the Doppler effect for points of the frequency axis allows you to write 2πƒ _1D = a _D 2πƒ ₁ , 2πƒ _2D = a _D 2πƒ ₂ , the discrepancy between the rotational frequencies Ω _rО = 2π (F _2D -F _1D ) = 2π (ƒ _2D -ƒ _1D -ƒ ₂ + ƒ ₁ ) forms the discrepancy of the vibration vectors.

The device uses a reference signal (OS) with frequencies from the frequency grid. OS harmonic frequencies will be called “adjustable” f ₁ and “control” f ₂ . The OS is transmitted via a radio link, is extracted from the input signal from the BM 3 output, and the second DM 13 is recorded in the PAM memory block of the samples of the reference signal 14. The OS frequencies do not go beyond the spectrum of the information signal. The difference frequency F ₁ = ƒ ₂ -ƒ ₁ when there is no Doppler effect. The difference frequency F _1D = ƒ _2D -ƒ _1D with the Doppler effect. The discrepancy Ω _KP = 2π (ƒ _2D -ƒ _1D -ƒ ₂ + ƒ ₁ ). The discrepancy gives a parasitic phase shift of frequencies, causing signal distortion, depending on the time t _S.

An exemplary signal from the output of BM 3 is transferred by the amplitude phase second DM 13 to the low-frequency region. An example of a transformation scheme is the basic scheme [7, p. 241], which belongs to the category of classical schemes. In the device, the local oscillator frequency is taken to the sum of the frequencies ω ₀ + 2πf ₁ . In the region of zero frequencies, a combination of the frequency and phase of the “tunable” frequency with the local oscillator is achieved by the self-tuning system, and an OS frequency difference signal is generated.

The first transformation by the second DM 13 is taken for free space in the absence of Doppler shift, when there is no reception delay, t _S ≈ 0, a frequency oscillation F _{1 is formed} . The second conversion is performed in a communication session, where the frequency fluctuation F _{1D is} formed from the Doppler effect. Digital samples of the phases of the oscillation of the frequencies F ₁ and F _1D of the OS block are stored in the PAM reports of the reference signal 14.

The operator for calculating the distortion of the reference signal in the VIS 15 determines the stray phase shifts of the spectral components of the OS for the time from radiation to reception t _S.

VIS 15 translates the fast Fourier transform (FFT) into the frequency domain of the frequency fluctuations F ₁ and F _1D , which determines the discrepancy of the control frequency.

FFT transformation operators are known, for example [6]: cfft (Y), icfft (F), transformation vectors Y, F. F: = cƒƒt (Y) with respect to the arguments xi: = i · Δ, i: = 0 ... N-1 , . The vector Y is formed by: modules M _i : = | Y _i | and phase Φ _i : = arg (Y _i ). Modules are the values of the samples of the amplitudes in the data block of the phase detector, the phases are the samples of the phases of the phase detector in the data block. Sample vectors with the number of elements N = 2 ^{n are} accepted for processing, missing elements are supplemented with zeros, samples at regular intervals. As a result of the direct transformation, the Fourier vector of the spectrum F is obtained from the vector Y. The components of the vector F: phases ΦF _i : = arg (F _i ) and the modules MF _i : = | F _i |. The inverse FFT transform is performed by the operator Y: = icƒƒt (F), F is the Fourier vector of the spectrum. In the frequency domain, frequency decomposition where i: = 0 ... N-1. Complex numbers:

- Y: = 19.785j + 0.1;

- Im (Y) = 19.785;

- Re (Y) = 0.1;

- | Z | = 23;

- arg (Z) = 0.1;

- J is a complex unit.

The phase difference of the control frequency φ _П = {Ω _iD · t _S ) for time t _S , in the spectral decomposition of the radian measure we find by the expression φ _П = n _C · 2π + φ _K , where n _C is the number of integer vibrations (2π), φ _K is the phase of frequency discrepancy during the session. The discrepancy n _C = N _MPA -N _MP , N _MP is the maximum number of the module in the spectrum of the emitted signal (recorded in the SAM of the sample signal reports 14), N _MPA is the maximum number of the module in the session. The modules and phases of the spectral decomposition of N _MP and N _MPA in the three versions “a”, “b”, “c” in Fig. 12 [8, p. 182] determine the total phase difference for time t _{S of the} control frequency.

Option "A" of Fig.13 shows the module of the resulting vector and its phase components of Fig.12. Option B, FFT decomposition spectrum, vectors of spectral components are shown. The number of components of the spectral decomposition by the fast Fourier transform i: = 0 ... N-1. The resulting vector is represented by the sum of N-1 frequency decomposition vectors, the figures of Fig. 12 show the decomposition vector modules. To determine the phase of the resulting vector, we will summarize a part of the component vectors that sufficiently fully reflect the length of the resulting vector, for example, take the sum of the numbers of the “complex form” that form the extremum.

Option "a" - the oscillation period contains an integer number of oscillation periods of the spectral decomposition, ie the oscillation frequency coincides with the frequency of the grid i of the frequency decomposition of the FFT, the module MF _i : = | F _i | maximum, N _MPA = i is taken, phase φ _K = ФF _i : = arg (F _i ), the modules of the remaining components are equal to zero.

Option “b” - the oscillation frequency differs from the grid frequencies by less than 0.5 steps, for example, by 0.25 steps, the module MF _i : = | F _i | maximum, N _MPA = i is taken. Modules of neighboring components are formed, decreasing in their range, phases of the components ФF _i : = arg (F _i ). The determination of the phase φ _K is made by the sum of the group of components of the "control" frequency, forming the extremum, by the numbers of the "complex form".

Option “c” - the oscillation frequency differs from the grid frequencies by 0.5 steps. Module MF _i : = 0. Equal modules of neighboring components are formed, the remaining modules decrease in terms of location, component phase, ФF _i : = arg (F _i ). The resulting vector is on the verge of jumping, either left or right by 0.5 steps, we believe that it did not jump, remained in the middle of N _MPA = i, where the maximum modulus is zero. The determination of the phase φ _K is made by the sum of the group of components of the "control" frequency, forming the extremum, by the numbers of the "complex form". On Fig shows that in the transformation you can take about five components.

The total phase difference of the control frequency during time t _S is equal to φ _PC = n _C · 2π + φ _K.

We write the frequency discrepancy causing the phase discrepancy Δφ over time Δt as a frequency . When Δφ = φ _PC , Δt = t _S we get . Substituting φ _PC , we determine the discrepancy by the grid step .

Grid of frequency discrepancies Ω _iD = i · Ω _r , i: = 0 ... N-1

The total phase difference φ _Пi (spurious displacements) during the time t _{S of the} spectral components Ω _{iD is} determined using the formula for the grid of frequency differences

, i: = 0 ... N-1

BKI 16 uses the vector φ _{Пi of the} parasitic phase displacement of the VIS unit 15, translates, with the PPBS command 7, the operator F: = cƒƒt (Y) the vector Y with the components: phases Ф _i : = arg (Y _i ), modules М _i : = | Y _i |, i: = 0 ... N-1. Phases - samples of the phase signal PAM reports of the signal of the recipient 12. We assume the units of samples to unity, M _i = 1. We get the vector F with the components: phase Ф _Fi : = arg (F _i ) and modules М _Fi : = | F _i |, i: = 0 ... N-1. Phase _Fi contains distortion.

From the Doppler effect at the time of reception, only the frequency changes, the signal modules from the relative speed of the receiver and transmitter do not change, therefore, the modules of the vectors M _{Fi of the} frequency decomposition under the Doppler effect are considered unchanged. Frequency changes over time t _{S are} taken into account by the complete phase difference φ _{Пi of the} components of the decomposition frequencies.

Distortion compensation is performed by changing the phases of the components of the frequency decomposition. Distortion Compensation Operator:

Ф _i : = mod _2π (arg (Y _i ) -φ _Пi ) for i: = 0 ... N-1

The output of the block BKI 16 is the phase vector Φ _Fi and the vector of modules M _Fi i: = 0 ... N-1.

The sampled block of the corrected phase signal (BFS) 17 uses the vectors Ф _i , M _Fi , i: = 0 ... N-1 obtained in the BKI 16 at the second input, converts the frequency domain signal into the time domain by the inverse transformation operator Y: = icƒƒt (F ), where:

vector F - vector of Fourier spectrum data with components;

F _i - phase vector;

M _Fi is the vector of modules.

The output of the BFS 17 is the phase reference vector Ф _i : = arg (Y _i ), i: = 0 ... N-1.

In the signal conversion operators of the receiving device, the number of samples N does not change, the normalization of the base system is not violated, which corresponds to the requirements of transforms in the FFT [9, p. 224].

Block 28 contains electrical circuits for FPGA, RAM, ROM, microprocessors, made in the form of loading modules for spacecraft signals with an a priori known structure.

The control unit of the device and displaying its status of the personal computer 26, performs the selection of the boot module from the software (software) 28 and includes the configuration of the reception in the communication session.

It is known to use configurations for receiving plug-in types of information flow, type of coding, data format in the subsystem SE985. Satellite Services B.V. Subsystem SE985 (Netherlands) - a single-board version that implements several functions, including transmission, coding (TM / TS) options that work in parallel in real time, with simultaneous processing of several request / response information streams that operate at different speeds. High-speed applications up to 150 Mbps, meet industry standards. Each section of the coding scheme can be selectively turned on or off, the control equipment ensures the transfer of commands to the spacecraft of the NRZ form transition to the Biφ form in the radio channel.

The multi-channel receiving and demodulating device of phase-shifted signals [10] uses data reception formats, a variable composition of the codecs of Fig. 14 in data reception configurations. If the installed configuration of the circuit does not receive, then the command changes the configuration. The formation of configurations is performed by the host PC with the interface. A multi-channel device uses the signals of one sideband in the channels. The input signal, filtered by a filter to protect against mirror and side channels of reception, is decoded in digital form.

An increase in the probability of message transmission is achieved by spreading signals (complex signals), coding, i.e. modulation by a code sequence of series chopping (CPIS) of a stream of transmitted bits.

Two types of coding are applied, intrasystemic and noise-immune to increase the probability of receiving information [3]. Intrasystem coding solves the problem of information compression, i.e. elimination of natural redundancy of information in order to replace uncontrolled redundancy with managed, increasing the reliability of information transfer. Intrasystem coding by alphabetical coding is used in the selection of NRZ data formats; start code; a synchronization code for an adaptive block synchronizer (frame); time markers, etc. Interference-free coding of signals is used for radio line signals as uniform coding with “detection” and “detection and correction” of errors from exposure to noise interference [3, p. 81, 92].

Spread spectrum coding yields good error correction results with error-correcting decoding. K. Shannon's formula establishes the relationship between the possibility of error-free transmission of information over a channel with a given band W on the signal-to-noise ratio. According to this formula, noise-correcting coding uses the use of spread-spectrum signals and the extension of the frequency band of signal transmission [1, p. 18]. Practically in such communication systems, the frequency band expands 2-6 times due to the use of noise-resistant codes with a code rate of R = 1/2 ÷ 1/6. The development of error-correcting coding schemes in control and communication channels [5, p. 233] in recent years has allowed the creation of codes, the implementation of which brought the communication channel throughput closer to the theoretical Shannon limit.

Widening the spectrum does not always give a good result. Analysis of the occurrence of errors from the Doppler effect, when fading and distortion of the signal occurs, shows that they are due to the expansion of the signal spectrum. Pictures of fading and distortion are built by a computer in the Delphi 3 Standart system in figure 1, figure 2. In the construction of vector diagrams of the signals of Fig.3 - Fig.7. complex signals used. Fading with the oscillation components of the upper side frequency band ω ₀ + Ω = f _B and the lower side frequency band ω ₀ -Ω = f _{H is} illustrated by the vector diagram of Fig.6, Fig.7. Fading results from a divergence of side frequencies and a phase shift as a result of the Doppler effect. Vector diagrams, FIG. 4, FIG. 5. show the discrepancy causing a change in waveform. Strong distortions arise from the divergence and stray phase shift of frequencies of the order of π, form a distortion zone. The distortion is deterministic in nature, the amount of distortion depends on the reception range S, or the propagation time of the signal before reception.

An example of a discrepancy between two coherent frequencies of harmonic oscillations with a frequency ratio f ₂ = kf ₁ [12, p. 185], form a discrepancy ƒ _r = Δf _and -Δf, difference frequency Δf and Δf _and , where Δf = f ₂ -f ₁ - difference the frequency in free space before the passage of the ionosphere, Δf _and is the difference frequency after the passage of the ionosphere. The discrepancy also arises from the Doppler effect. The difference frequencies Δf and Δf _d , where the difference Δf = f ₂ -f ₁ when there is no Doppler effect; Δf _d , there was a Doppler effect, form a discrepancy ƒ _r = Δf _d -Δf. The discrepancy ƒ _r forms a phase shift of the oscillations φ _r , during the time t of the signal propagation before reception, φ _r = mod _2π | 2π · ƒ _r · t |, The values of φ _r can be different, including zero and π, with a period of circular frequency discrepancies Ω _D = 2π · ƒ _r .

The discrepancy between the frequencies and the change in the waveform can be seen when the carrier frequency is modulated by the meander, the case of emission of oscillations in one side frequency band, distortions of Fig. 2 c, ω ₀ + Ω = 2π · ƒ _H , ω ₀ + kΩ = 2π · ƒ _B , k = 3. When receiving a signal in the absence of Doppler offsets, the shape of the received signal is shown in figure 2 "a". Dimension of frequency values Hz. The carrier frequency f ₀ = 11000 × 10 ⁶ , the components ƒ _H = 11000 × 10 ⁶ + 5 × 10 ⁶ , ƒ _B = 11000 × 10 ⁶ + 15 × 10 ⁶ , the difference frequency Δf = ƒ _B- = _H = 10 * 10 ⁶ . Doppler shift of the carrier frequency 50 × 10 ³ , ƒ _0Д = 11000 × 10 ⁶ + 50 × 10 ³

ƒ _ND = 11000 × 10 ⁶ + 50 × 10 ³ + 5 × 10 ⁶ +22.72

ƒ _VD = 11000 × 10 ⁶ + 50 × 10 ³ + 15 × 10 ⁶ +68.18

Difference frequency Δf _d = ƒ _VD -ƒ _ND = 10 × 10 ⁶ +45.46

The discrepancy f _r = Δf _d -Δf = 45.46.

The discrepancy in the coherently radiated frequencies leads to the formation of a phase shift φ _r = π, we obtain t _S = 11 ms. For 11 ms, the wave travels 3.3 thousand km. The oscillation phase of figure "b" of figure 5 has changed relative to the "b" of figure 4. The divergence of the component frequencies of the meander at a receiving range of 3.3 thousand km is illustrated in Fig.5 "b", the distortion in Fig.5 "c", Fig.2 "guards".

Mirror frequencies. Change in the signal value (fading) from the Doppler frequency shift when receiving a radio signal with vibrational components of the upper side frequency band ω ₀ + Ω = f _B and the lower side frequency band ω ₀ -Ω = f _H.

Carrier frequency ƒ ₀ = 11000 × 10 ⁶

Modulating frequency Ω = 15 × 10 ⁶

Lateral frequencies ƒ _H = 1l000 × 10 ⁶ -15 × 10 ⁶ , ƒ _B = 11000 × 10 ⁶ + 15 × 10 ⁶ .

The difference frequency of the component frequencies Δf is Δf = f _B -f _H = 30 × 10 ⁶ .

Doppler effect. Doppler shift of the carrier frequency 50 × 10 ³ .

Frequencies from the Doppler effect:

ƒ _0Д = 11000 × 10 ⁶ + 50 × 10 ³

ƒ _ND = 11000 × 10 ⁶ + 50 × 10 ³ -15 × 10 ⁶ -68.18

ƒ _VD = 11000 × 10 ⁶ + 50 × 10 ³ + 15 × 10 ⁶ +68.18

The difference frequency Δf _d = ƒ _VD -ƒ _ND = 30 × 10 ⁶ +136.36

The discrepancy between the frequencies f _r = Δf _d -Δf = 136.36.

The discrepancy in the frequencies f _r leads to the formation of a phase shift φ _r in the plane Pl (t = 0 + t _S ) versus time φ _r = mod _2π [2πƒ _r t _S ]. We determine t _S , with a phase shift π, we obtain t _S = 3.7 ms. For 3.7 ms, the wave travels the path of 1.1 thousand km. With a phase shift of π, the signal amplitude decreased and became equal to zero, Fig. 7, the signal fading. Figure 6 shows the signal of the demodulator when the phase shift of the frequency components of the meander signal was not. The signal fading - a parasitic decrease in amplitude to zero at the time of reception is illustrated in figure 1 "g", where the frequency offset during the simulation was taken into account by the coefficient Δω / ω = 0.015.

Set forth with the illustrations of figure 4 - figure 7 shows that the signal components receive a discrepancy from the Doppler effect, leading to fading and distortion of the signal. The Doppler effect in vector diagrams creates a circular frequency Ω _{D of} rotation of a vector with a period and the initial phase of the oscillation, equal to zero φ _r = 0 at the time of radiation t = 0. The rotation of the vector is illustrated by the vector diagram of FIG. 15, constructed for an example of demodulation of a reference signal.

It is known in the art to use a signal with one upper sideband or lower sideband of the spectrum in microwave equipment with frequency division multiplexing for transmitting a large number of channels in a given frequency band [13, p. 14]. The circuit contains a bandpass filter following the modulator, which passes one of the two side frequencies.

Positive results from the use of CPIS are achieved when receiving, when the modulating signal of CPIS matches the signal of CPIS used in demodulation. The disadvantage of applying CPIS is when CPIS interference is added to the distortion from noise interference, CPIS distortion from the Doppler effect is added to the radio link, and undistorted CPIS signals are removed during demodulation, the difference signal creates interference.

The use of randomization gives the approximation of the average received bits to zero for the application of signal processing with a known zero level [4, p.197]. Shredding long series increases the number of high-frequency components of the spectrum and increases the signal distortion from the Doppler effect. The effect of the Doppler effect can be reduced by eliminating randomization by transmitting to the receiving device an average level of the emitted signal. In this case, parasitic bias compensation can be used, an example of a circuit [5], where the deviation of the average value from the samples of the phase signal is used. By evaluating the mathematical expectation (parameter), the average value is used when there are only values of n observations [11, p. 53],

,

where x _i (i = 1, 2, ..., n) are the values of n observations.

In the radio line, with the Doppler effect, zones are formed at different ranges: signal fading, zones of signal spectrum distortion and zones where the influence of the Doppler effect is not manifested. Signal fading zones and signal spectrum distortion zones are detected by the decoder when status information is generated. If the receiver has the ability to receive a signal in several configurations, then the reception configuration is the one in which the decoder transmitted the received signal to the PI (no status information was found).

The device has the following configurations:

- PF, SFDB, BM, DM, DK, PI in real time in the receiving frequency band of two side;

- PF, SFOB, BM, DM, DK, PI in real time in the frequency band with one side;

- PF, SFOB, BM, PAMP, CC SFS, PI according to the phase signal received in the frequency band with one side signal and recorded in memory;

- PF, SFOB, BM, PAMP, CC SSS, PI according to the phase signal received in the frequency band from one side and recorded in the memory.

An example of a change in configurations in the claimed device when receiving, from the receiving range D, is shown in Fig.16.

The first configuration “a” is real-time reception, a radio signal with two side signals. Marked areas 0-D ₁ , D ₂ -D ₃ , D ₄ -D ₅ slight fading, where distortion of the signal from the CPIS is not observed, and zones D ₁ , -D ₂ , D ₃ , -D ₄ - zone fading signal, where the ratio P _C / P _W decreases and reaches a threshold value due to a decrease in the power of the received signal, detected by the decoder according to the status information signal. Reconfiguration of reception according to the status information of the decoder at a range of D ₁ -D ₂ , D ₃ -D ₄ .

The second configuration “b” - real-time reception, a radio signal from one side. Distortion from the Doppler effect is not observed at a range of up to D ₁ , from D ₂ to D ₃ . Distortions are detected by the status information of the decoder at a range of D ₁ -D ₂ . Reconfiguration of reception according to the status information of the decoder at a range from D ₁ to D ₂ .

The third configuration “c” is the compensation of the parasitic displacement of the phase signal φ _g , the processing of the phase signal from the PAM memory block, the radio signal from one side. Distortion from the Doppler effect is not observed at a range of up to D ₁ , from D ₂ to D ₃ . Distortions are detected by the status information of the decoder at a range of D ₁ -D ₂ . Reconfiguration of reception according to the status information of the decoder at a range from D ₁ to D ₂ .

The fourth configuration “g” is the compensation of spurious shifts of the spectral components from the Doppler effect in the phase signal of the PAM memory block, a radio signal from one side. Implementation of the reception is depicted when, at the reception range within the estimated range, when decoding the data block, no errors were found in the information or all errors were corrected.

The expansion of the spectrum, with the Doppler effect, can give a fading and distortion of the phase signal, from which single and series of errors arise. On Fig shows the formation of a series of errors in the signal from the stray shift of the frequency components of the signal containing series "1" and series "0" as a result of the Doppler effect. Line "a" is numbered 16 bits of the block of the received number. String "b" of the value of the digits, binary number system. The transmitted signal of the meander is “c”. The transmitted meander signal, consisting of two frequency components, the first and third harmonics is “g” and its transmitted bits are “d”. When receiving, a distorted signal is obtained, as a result of the harmonics diverging from the Doppler effect - curve “e”, from the parasitic shift of the third harmonic by π. The RBS determined the received digits - line "g". The content of the discharges is determined by the average value of the amplitude of the phase signal equal to π / 2. The decisive rule for determining the code for bit-wise reception of the signal (between the labels of the clock synchronization signal, the average value of the amplitude is -U; the decisive rule is ifU≥π / 2, then “1”; ifU <π / 2, then “0”). Line “z” shows two series of errors that occurred.

The device for compensating for the parasitic shift of the spectral components (CCCSS) containing DM, PAM _OS , VIS, BKI, BFS, SSR, and DC, in the case of the status information of the second DC 13, determines and compensates for the parasitic shifts of the spectral components from the Doppler effect. After the status information DK 13 appears, the VIS and BKI blocks transfer the model signals and the PAM _P signal to the frequency domain using the fast Fourier transform, calculate the spurious shift of the spectral components of the signal s [λ, φ (t), t]. The BFS converts the spectrum to the time domain and obtains samples of the phase signal of the data block. The second decisive block character RBS 17 forms a sequence of bits of the data block (characters "0" and "1"). The third DC 19 performs noiseless decoding, the received block of information is transmitted by the PI.

The status information of the DC 11 is transmitted to the second input of the BSS, from the second output of which the “On” command receives the distortion compensation block (BKI) 16 and the corrected phase signal (BFS) sample block 17, which determine the phase signal distortion from the distortions of the reference signal and compensate them. The distortion of the reference signal is determined by the VIS 15. The readout of the corrected signal is sent to the second RSB 18, which forms a sequence of bits of the data block (characters "0" and "1"). The third DC 19 performs noiseless decoding. A correctly received information block by the third DC 19 is transmitted by PI 6.

Known schemes in which reception is provided by changing the configuration of the reception, for example, the reception scheme SE985 and scheme [10], the configurations differ in data presentation formats and noise-resistant coding.

The introduced configurations of receiving information from the PF to the PI provide signal reception in case of distortions from noise interference and distortions from the Doppler effect. A similar solution is not described in the literature, therefore it meets the criteria of novelty and inventive step.

Signal structures and data transmission formats recommended by the National Space Agency by the International Advisory Committee on Space Transmission Systems (CCSDS) are known in the art. The following can be used in transceiver devices: six data presentation formats, the first three formats - NRZ variants, the rest Bi φ; Viterbi unsystematic convolutional code decoders, systematic convolutional codes, trellis code decoder, turbo code decoder, turbo decoder, Reed-Solomon code decoder, universal self-synchronizing descrambler, deinterleaver, etc.

It is advantageous to have the smallest possible spread of the signal spectrum, providing the maximum distance from the radiation source of the zones of fading and distortion arising from the Doppler effect. Codecs for minimum bandwidth expansion with a high error correction result are created, codecs are implemented and created [5, p. 233], including:

in 1991, an implementation of the convolution-block code was obtained (384, 288), which made it possible to reduce the probability of error per bit from 10 ^-3 to 10 ^-5 , which was used for the exchange between the control center (MCC) and the space station (ISS);

implementation of the non-linear Nordstrom-Robinson code with a byte code decoder (6, 4) for radar mapping of the surface of Venus, the Venera-15 and Venera-16 in 1983 spacecraft were developed and implemented at the Evpatoria Center by the Kvant-D complex with spacecraft. The information stream (100 kbit / s) was recorded on magnetic recorders (MZU), errors were corrected by the byte code decoder (6, 4), the gain in error probability when using encoding is more than two orders of magnitude (with a probability of 1.5 × 10 ^-4 without encoding up to 1.3 × 10 ^-6 with encoding).

The claimed technical solution allows to reduce the possibility of distortion and signal loss in circuits with noise-resistant coding in a communication session, depending on the Doppler effect and on the reception range, with bitwise transmission of numerical information. This effect of the circuit is obtained by reconfiguring the reception of a signal with two side signals, a signal from one side, with compensation for stray phase signal displacement and compensation for stray shifts of the spectral components from the Doppler effect.

The proposed scheme increases the probability of receiving information at Doppler shifts by eliminating the causes of errors by compensating for spurious distortions of the phase signal before decoding.

Literature.

1. Aces G.I. and others. Interference immunity of radio systems with complex signals. Radio and Communication, 1985.

2. I.M. Teplyakov, B.I. Roshchin, A.I. Fomin, V.A. Weitsel. Radio transmission systems. Moscow, Radio and Communication, 1982

3. Markov A.A. Introduction to coding theory. Moscow, "Science", 1982

4. Berezin L.V. and Weitzel V.A. Theory and design of radio systems, M., "Soviet Radio", 1977

5. E.P. Molotov. Terrestrial radio control systems for spacecraft. Moscow, FIZMATLIT, 2004. Proceedings of FSUE RNII KP.

6. Patent for invention No. 2371845. Digital Information Radio 2009

7. Kolchinsky V.E. and others. Autonomous Doppler devices and aircraft navigation systems. Moscow "Soviet Radio", 1975

8. Vollerner N.F. Hardware spectral analysis of signals. Moscow "Soviet Radio", 1977

9. Trakhtman A.M. Introduction to the generalized spectral theory of signals. Moscow "Soviet Radio", 1972

10. Patent for invention RU 2305375. Multichannel receiving and demodulating device of phase-shifted signals, 2007

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12. Grudinskaya G.P. Propagation of radio waves. Moscow Higher School, 1975

13. Handbook of radio relay communication. Radio and Communication, 1981

Claims (1)

A device for receiving high-speed information of a space radio link contains: a band-pass filter (PF), a matched filter for receiving a radio signal from one side (SOPF), a balanced modulator (BM), a first demodulator (DM), a first block decoder (DK), an information receiver (PI), a receiver signal sample memory block (PAM _P ), a device for compensating for stray phase signal displacement (CC SFS), a device for compensating for stray spectral components (CC SSS), a matched filter for receiving a radio signal from one side (SPSB), a second balanced modulator (BM), a second demodulator (DM), and a second decoder block (DK), personal computer b eye control (PC), a bus interface PC (interface), software (SW), wherein the input device is an input PD, whose output is connected to the joined series SFOB, BM, DM, DC, PI, a second output DM is coupled to PAM _II, the output of which is connected to the input of the samples of the phase signal of the CC SSS and CC SSS, the outputs of which are connected to the second input of the PI, the output of the BM is connected to the first input of the reference signal of the CC SSS, the SFR input is connected to the SPSB input, the output of which is connected to the input of the second BM connected in series second DM, second DC, the output of which is connected to the third input of the PI, the software is connected to the PC, which is connected by an interface to the first DC, the second DC, PI, CC SFS, CC SSS, PAM _P.