NZ329782A - Low-pass filter for integrated circuit and stepwise-adjustable notch frequency filter - Google Patents

Description

New Zealand No. 329782 International No. PCT/ TO BE ENTERED AFTER ACCEPTANCE AND PUBLICATION Priority dates: 14.09.1994; Complete Specification Filed: 14.09.1995 Classification:^) H03H1/02; H03H7/01; H03C1/64 Publication date: 26 August 1998 Journal No.: 1431 NEW ZEALAND PATENTS ACT 1953 COMPLETE SPECIFICATION Title of Invention: Filters Name, address and nationality of applicant(s) as in international application form: ERICSSON INC., a Delaware corporation of 7001 Development Drive, PO Box 13969, Research Triangle Park, North Carolina 27709, United States of America 329782 Under the provlelone of Regulation 23 (1) the Specification hae been ante-dated t0 |* 19 J&WZEALAND ./Wflfrfl EAIENTS ACT, 1953 Initials (Divided out of NZ Patent Application No. 293170 filed 14 September 1995) COMPLETE SPECIFICATION FILTERS We, ERICSSON INC., a corporation of the State of Delaware, of 7001 Development Drive, PO Box 13969, Research Triangle Park, North Carolina 27709, United States of America, do hereby declare the invention for which we pray that a patent may be granted to us, and the method by which it is to be performed, to be particularly described in and by the following statement: /r ii «* , x RECEIVED (followed by page - la -) inteiieuuni Propsrty orrice 1 7 FFB 19S8 of New Zealand 329 782 - la - B3<?Kzrwn4 1) Field of the Invention.

The invention relates to methods and apparatuses for the realization of frequency band-stop, band-pass and low pass filters as integrated circuit elements on a silicon chip for use in a quadrature modulator which can generate complex modulated radio signals. 2) Discussion of Related Art.

The prior art for the construction of frequency selective filters includes: (1) passive inductor-capacitor filters; (2) passive resistor-capacitor filters; (3) acnve RC filters; (4) distributed RC filters; (5) gyrator-capaciior filters; (6) transmission line or waveguide filters; (7) switched capacitor filters; and (8) digital filters, each of which is discussed below.

The construction of inductor-capacitor filters on a silicon chip is constrained by the very small inductance values that can be achieved with spiral metallization patterns within a frequency range above about 2 GHz.

Passive resistor-capacitor filters can only synthesize a limited subset of the possioie frequency responses that might be required, and at low frequencies are limited by the available RC product that can be made while also being limited at high frequencies by stray (parasitic) capacitance and resistance. 329782 Active RC filters can provide useful performance up to a few megahertz, but are limited by the performance and bandwidth of amplifiers as well as the parasitic effects mentioned above. Unfortunately, the amplifiers consume power and limit the dynamic range.

Distributed RC filters are, on the other hand, inherently based on the parasitic capacitance and resistance parameters, such as described in "Tidskonannreliga Ligpass Filteri CMOS", by Katarina Hansson and Mats Torkelsson, LUTEDX/(TETE-7029)/ pp. 1-26 (1987).

Gyrator-capaciior filters use an active impedance inverting circuit to 10 make a capacitor function as an inductor, so that LC equivalent filters may be built. These circuits are useable for bandpass filters up to a few megahertz. The Gyrator-Capacitor filler can be classed as a form of active RC filter.

Transmission line or waveguide filters require elements that are 15 typically a quarter wavelength long so their construction on a chip is limited to the micro-wavelengths above 2 GHz.

Switched capacitor filters operate according to a number of different principles, but all require transistor switches to operate at a very much higher frequency than the operating frequency range of the filter. This 20 restricts their use to a few hundred kilohenz. Moreover, the dynamic range of switched-capacitor filters is limited by their high noise levels.

Digital filters are very flexible in the frequency response repertoire they can realize, and have the advantage of no tolerances. On the other hand, the signal to be filtered must first exist in digital form and the required 25 analog-to-digital convenors restrict both the dynamic range and speed.

Digital logic power consumption is also a factor which restricts such filters to the 300 kHz region or below in practical applications.

The frequency range upon which the present invention focusses is the 0.3 MHz to 300 MHz region. This is above the range of most of the 30 techniques mentioned while being below the range for transmission line 329782 solutions. Hitherto there has been no practical silicon-iniegrable solution for these three decades of frequency, which encompass virtually the entire radio communications frequency spectrum. Accordingly, the present invention % was conceived to address this important range of frequencies. The present 5 invention makes use of concepts of the distributed RC techniques mentioned above.

Summary The present specification discloses methods and apparatuses for the realization of frequency band-stop, band-pass and low pass filters as integrated circuit elements on a silicon chip. Several inventions are 10 described.

The methods allow the manufacture of continuous-time, analog filters in frequency ranges not conveniently covered by other, known techniques. Such filters are typically required in the generation of complex-modulated radio signals with the aid of digital signal processors and 15 quadrature modulators. The inventive filters are aimed to be suitable for construction as part of integrated circuits for analog or mixed analog/digital radio communications signal processing applications.

In the present specification, new distributed RC filter structures and applications are disclosed, and in 20 particular, means to overcome the problems created by manufacturing tolerances in the resistive and dielectric layer properties. The distributed RC filter structure include means for selectively switching in and out of the circuit incremental RC lines and for selectively switching in and out of the circuit incremental nulling resistors. Several embodiments are disclosed.

The present specification claims a low pass filter and a notch filter, and has been divided from parent specification NZ 293170 which claims a quadrature modulator. 329782 Brief Description of the Drawings The inventions will now be described with reference to the accompanying drawings in which: Figure la is a partially schematic diagram of the structure of a 5 distributed RC line; Figure lb is the circuit symbol for a distributed RC line filter such as shown in Figure la; • Figure 2 is a schematic diagram of a prior an distributed RC^y, device; Figure 3 is a schematic diagram of a prior an quadrature modulator arrangement for synthesizing an arbitrary modulated signal; Figure 4 is a schematic diagram of a quadrature modulator arrangement- Figure 5 is a schematic diagram showing prior an Gilbert mixers as balanced modulators; Figure 6 is a schematic diagram of a section of a balanced filter; Figure 7 is a schematic diagram of a complete balanced filter; Figure 3 is a schematic diagram of a stepwise-adjustable RC line; Figure 9 is a schematic diagram of a switched milling resistor combinabie with the switched RC line of Figure S ; Figure 10 is a schematic diagram showing the use of an ad;useable notch filter in a feedback loop for obtaining an adjustable bandpass amplifier response; Figure 11 is a schematic diagram of a preferred arrangement of a switcfaed-ninable RC-^u. device; 32978? Figure 12 is one possible stepwise-adjustable milling resistor for use with the switched-tunable RC^u. device shown in Figure 11; Figure 13 is a schematic diagram of a preferred arrangement of a 5 stepwise-adjustable nulling resistor for use with the switched-tunable RC^u. device shown in Figure 11; and Figure 14 is a graph of the frequency response of the filter shown in Figure-7.

Detailed Description of the Preferred Embodiments 10 The inventive filter construction includes a distributed RC line as shown in Figure la which utilizes the sheet-resistivity properties of deposited conductive films such as a polysilicon film resistive filter 10, and the capacitance-per-unit-area properties between the resistive filter 10 and a conductive plate 14 (connected to a common by a connection point 14a) with 15 a thin dielectric layer 12 interposed between the layers 10 and 14. The resistive filter 10 includes an input connection point 10a and an output connection point 10b.

In order of the layers' appearance from substrate to top level, the filter is composed of a substrate 13 comprising, e.g., silicon, alumina, 20 gallium arsenide, sapphire or polyamide, an insulating film 11 comprising silicon dioxide, alumina, gallium arsenide, sapphire, poiyamide, etc., a conductive plate 14 of heavily doped polysilicon, aluminum, gold or the like, a thin film dielectric layer 12, and a resistive filter 10 composed of polysilicon or the like.

Resistors formed by the polysilicon film 10 are treated as distributed over and insulated from a capacitor plate 14 and, thus, as a distributed RC line that may be described by the resistance per unit length, capacitance per unit length, and length.

The circuit symbol for a distributed RC line is shown in Figure lb. 329782 Such RC lines have an inherent low-pass type of frequency response that attenuates higher frequencies, but the cut-off is rather gentle. Sharper cut-off low-pass filters generally achieve their characteristics with the aid of notches in the stop* band.

A notch in the frequency response may be formed using a distributed RC line by connecting its capacitor plate 14' terminal to ground 22 through a resistor 21 of specific value, such as shown in Figure 2. For uniform RC lines, the notch is complete when the resistor connected to ground has the approximate value 0.056 times the total through-resistance Riot of the resistive filter 10', and the notch frequency is approximately 11.2/RC radians per second where Riot is the total through-resistance and C is the total distributed capacitance.

Once a complete or partial notch can be formed, other frequency responses can be synthesized, such as bandstop, or bandpass, the latter by including the notch device in the feedback loop of an amplifier such as shown in Figure 10, discussed below.

According to a first aspect of the invention, matched, balanced, low-pass filters are provided in conjunction with a so-called quadrature modulator for the purposes of synthesizing an arbitrarily modulated radio frequency signal.

According to a second aspect of the invention, means to overcome the high production process spreads (i.e.; deviation from ideal values on resistive and dielectric layer properties) are provided. In some processes, typical spreads on the sheet resistivity and capacitance per unit area parameters can be up to 15% on capacitance and as much as 100% max/min ratio on sheet resistivity. Without the inventive means, the notch frequency given by the RC product could not be guaranteed' to within an octave. The inventive means can be used to bring the notch frequency within a desired tolerance when such processes are used. The present invention achieves this 329782 by effectively providing a stepwise-variable line length that can be programmed in-circuit to set the filter frequency to a desired value.

Figure 3 shows a prior art arrangement of a quadrature modulator for synthesizing an arbitrarily modulated signal. A digital signal processor 5 (DSP) 30 calculates time-spaced samples of the real and imaginary parts of a desired complex modulation. The real part is given by the desired amplitude times the cosine of the desired phase angle, while the imaginary part is given by the amplitude times the sine of the phase angle. In this way both Amplitude Modulated (AM) signals or Phase Modulated (PM) signals can be 10 generated, or signals comprising both, the result of which is generally known as complex modulated signals. The numerical samples calculated by the DSP 30 are transferred to a pair of Digital-to-Analog (D-to-A) convenors 31 that conven each numerical sample pair into a pair of analog voltages known as I (In-phase) and Q (Quadrature) signals. A sequence of such numerical 15 samples generates I and Q waveforms but in a stepwise fashion.

The steps in the wavefonns cause undesirable spectral components that would interfere with adjacent radio channels unless suppressed. Some techniques for D-to-A conversion provide interpolation between samples giving sloping waveforms between adjacent sample values, which reduces 20 but does not sufficiently eliminate the undesired components. Consequently, I and Q smoothing filters 32 are necessary. These are low-pass filters that pass all modulation spectral components of interest but suppress the higher frequency components of the spectrum associated with the stepwise or piecswise linear I, Q waveforms from the D-to-A convenors 31. 25 The smoothed I, Q waveforms are applied to a pair of balanced modulators 33 together with cosine and sine carrier frequency signals, this arrangement being known as a quadrature modulator. The arrangement described so far and illustrated in Figure 3 belongs to the well-known prior an. 329 It is important for accurate signal generation that (1) the two balanced mixers are accurately matched, (2) the levels of the I and Q signals are accurately controlled relative to each other, and (3) the balanced mixers have low earner leakage or offset, that is, the output signal of a balanced modulator should be zero when its respective I or Q modulating signal is zero.

Since the I and Q signals vary from positive to negative, if a circuit is required to operate only from a single positive supply, then the zero point of an I or Q waveform cannot be defined to he zero voltage, but must be defuied to be some positive reference voltage such as half the supply voltage. Then when an I or Q waveform swings below this reference voltage it will be interpreted as negative, and positive when it swings above.

Unfortunately, it is difficult to generate a reference voltage from the DSP 30 that is exactly equal to the voltage the D-to-A convenors supply with an input numerical value of zero. This problem is overcome in the invention by use of the balanced configuration shown in Figure 4, which uses special D-to-A conversion techniques to generate I and Q signals as well as their complements I and Q.

In accordance with the present invention as shown in Figure 4, the numerical I and Q signals from DSP 30' are transferred to a delta-sigma (A-E) convenor 41. This device is built according to known an to generate a high bitrate stream of binary 'l's and 'O's having a shon-term average value proportional to the numerical input value. With a maximum possible numerical input vaiue the bit stream produced would be 11111 ... (the voltage of a '1' condition being equal to the chosen supply voltage) while the minimum numerical input value will generate the bit pattern 00000 .... A half-scale numerical input will produce the bit stream 1010101010 ... having an average voltage equal to half the supply voltage. According to an aspect of the present invention, extra invenor gates 42 are provided at the output of each delta-sigma convenor 41 to additionally generate the complementary 329782 bicsrams. That means when delta-sigma convenors 41 produce a bit stream 100100100100 ... having a mean of 1/3 the supply voltage, the complementary bit stream will be 011011011011 ... having a mean of 2/3rds the supply voltage. The difference between these two is 1/3-2/3 = -1/3 of 5 the supply voltage. If the convenor produces 111011101110 ... having a mean of +3/4 of the supply voltage then the complementary signal 000100010001 ... will have the mean 1/4, so that the difference is 3/4-1/4 * + 1/2 supply. Consequently, by using the difference between the convenor output signal and its complement to represent an I or Q signal, the value 10 represented can be positive or negative even with a single positive voltage supply, and no reference voltage need be generated. The balanced mixers 43a and 43b are therefor provided with balanced, two-wire inputs rather than single-ended inputs, that are responsive to the difference in the signals on the two wires and unresponsive to the absolute or common-mode voltage (sum 15 of the voltages) on the two wires.

High bitrate delta-sigma modulation bitstreams are simply converted to the analog voltage they represent by forming the moving average voltage over a large number of bits. This may be done using a continuous-time, low-pass filter having a bandwidth which is a small fraction of the bitrate, 20 but still sufficient to pass all desired modulation components. For the balanced signal configuration developed in this invention, balanced filters 44 axe interposed between the delta-sigma convenor outputs and the I, Q balanced modulators 43.

The balanced modulators 43 may include so-called Gilben mixers 43a 25 and 43b such as shown in Figure 5. As shown in Figure 5, the balanced I or Q inputs 50a and 50b of the Gilben mixers is applied to the bases of two transistors 51a and 51b. The emitters of the two transistors 51a and 51b are commonly connected through respective resistors 52a and 52b to a common bias current source 53. Each of the collectors to the two transistors 51a and 30 5 lb are respectively connected to a pair of commonly connected emitters of 329782 two pairs of transistors 54a, 54b and 55a, 55b. The base of one transistor 54a, 55b from each of the transistor pairs 54 and 55 are commonly connected to one side of a cosine or sine signal generator 56, with the other base of one transistor 54b, 55a of each of the transistor pairs 54 and 55 5 being commonly connected to the other side of the cosine or sine generator 56. The collectors of one transistor of each of the two transistor pairs 54a and 55a are commonly connected to one output line 57a, with the other collectors of one FET of each of the two transistor pairs 54b and 55b being commonly connected to the other output line. These balanced modulators 10 can be formed in the same substrate as the balanced low pass filters.

The outputs of the balanced modulators 43a and 43b of Figure 4 are added together by an adder 43c, to result in a complex modulated radio signal.

The balanced I or Q input signals swing around a mean voltage of 15 half the supply (e.g., around 2.5 volts for a 5 volt supply) but the amplitude of the peak-to-peak voltage excursions should be somewhat less, for example, +/-250mV. The delta-sigma convenor output, however, can at its extremes swing between 0 and 5 volts on each output line or its complement, and therefore a 1720 attenuation of the balanced mode signal is called for 20 while no attenuation of the common mode voltage (2-5 volts) is required in this embodiment.

A basic filter section 44 according to the invention which accomplishes the appropriate attenuation of the balanced mode signal is shown in Figure 6. Two identical devices 61, 62 provide a low-pass filtering action to both balanced (push-pull) and common mode signals, with a notch in the frequency response.

The balanced filter includes two input terminals for I, I or Q, Q and two output terminals 50a and 50b, and a common terminal connected to ground. A series resistor Rs is connected between an input terminal I or Q 30 and an ourout terminal 50a, and an identical resistor Rs is connected between 329 a second input terminal Q or Q and a second output terminal 50b. T?ach of the resistors Rs can be formed by depositing a pattern of resistive material over respective conducting plates with an intervening dielectric layer, such as shown in Figure 1, to provide distributed capacitance between the resistive pattern and the conductive plate.

A resistor or resistors iRWUm are connected between each of the conducting plates to the common terminal or between the conducting plates, or both. A shunt resistor Rp is connected between the output terminals of at least one of the filter sections (when cascaded, as discussed below with reference to Figure 7).

This filter 44 has a common mode attenuation of unity at direct current and low frequencies, as there is no resistance to ground. In other words, the pair of balanced, lowpass filters arr^iiaro to a different degree, or not at all, the common mode signal defined as the sum of the voltages on the two input lines or two output lines, compared to the balanced mode defined as the difference of the voltages on the two input or output lines. In the balanced mode, the attenuation is Rp/(2Rs-i-Rp) due to the resistor Rp connected between the output terminals. This may be set to 1/20 or other desired value less than unity by choice of the shunt resistor Rp relative to resistive filter Rs. The desired value is defined as the difference in voltages between the two input lines or the two output lines.

Another effect of the shunt resistor Rp is to emphasize the high frequency response in the balanced mode relative to the low frequency response, as the high frequency attenuation tends to unity. This has the desirable effect of sharpening up the rate of cutoff. The rate of cutoff may be further sharpened by exponential tapering of the RC line.

A complete filter design consisting of a cascade of such balanced sections is shown in Figure 7. A series of balanced RCiWI_r. devices 70, 71, 72, 73, each characterized by a starting line width, an exponential tapering factor (MAX/MIN width ratio), a total resistance Rtot and a total capacitance 329782 Crot are cascade connected by depositing distributed RC line* having resistive patterns deposited over a conductive plate over an intervening dielectric film on a substrate, such as shown in Figure 1. Shunt resistors * Rpl, Rp2, Rp5, Rp4 are connected between the output terminals of each 5 section to provide graduated attenuation. The total attenuation in the balanced mode is set to the desired value by choice of these shunt resistors, but there is a continuum of ways to do the ranging from all attenuation in the first section Rpl to all attenuation in the last section Rp4. An optimum distribution of the attenuation between the sections can be found by trial and 10 error using computer simulation which gives the sharpest rate of cutoff. Likewise, an optimum set of line widths and tapers can be found within constraints on minimum line width and maTimum allowed filter area that gives the sharpest rate of cutoff. The values of a near optimum design for a cutoff frequency of 150 kHz are shown in Table 1 and its resulting 15 frequency response is shown in Figure 14. 329 - 13 -TABLE 1 70 ^MAX7\ LINE WIDTH AT THICK END - 20.00000 MICRONS TAPERING FACTOR- 20.00000 R •TOTAL RESISTANCE- IIS 83687 kfl c TOTAL CAPACITANCE- 47 53474 oF R*UlL7l NULLING RESISTOR- 3.16380 kfl ATTENUATION FACTOR- 1.41410 R~.

SHUNT ATTENUATION RESISTOR - 286.97623 kfl 71 Wmaxt: LINE WIDTH AT THICK END- 1.01000 MICRONS TAPERING FACTOR - 1.010000 R TOTAL RESISTANCE- 271.01572 kfl r TOTAL CAPACITANCE- 5.47452 nF R*uixr: NULLING RESISTOR- 15.22687 kfl ATTENUATION FACTOR- 2.82820 R*- SHUNT ATTENUATION RESISTOR » 194.20894 kfl 72 W.MAX73 LINE WIDTH AT THICK END = 1.01000 MICRONS TAPERING FACTOR = 1.01000 R TOTAL RESISTANCE- 171.40543 kfl C TOTAL CAPACITANCE - 3.46239 oF R*ULLT3 NULLING RESISTOR- 9.63032 kfl ATTENUATION FACTOR- 4.00000 R*- SHUNT ATTENUATION RESISTOR - 98.98187 kfl 1 / j Wy(AX74 LINE WIDTH AT THICK END = 1.01000 MICRONS TAPERING FACTOR = 1.01000 R TOTAL RESISTANCE- 98.96096 kfl C TOTAL CAPACITANCE- 1.99901 oF RwU.74 NULLING RESISTOR- 5.56007 kfl ATTENUATION FACTOR - 1 25000 Rf7« SHUNT ATTENUATION RESISTOR- 692.78949 kfl 329 782 A practical problem is how to control in mass production the resistivity of the deposited films to be equal to the target value 1 in the design. If the resistivity varies, the whole frequency response scales proportionally. For example, double the resisdvity would halve the cutoff 5 and null frequencies while half the resisdvity would double all frequencies. In the case where practical production tolerances are too wide to permit the frequency response to be held within desired limits, the second aspecTof the invention may be applied to adjust the frequency response to be within limit* after manufacture. This is done by means of an inventive means for 10 stepwise variation of the line length.

A first configuration according to this aspect the invention is shown in Figure 8. It is to be understood that the circuit of Figure 8 can replace any of resistive lines of the balanced null devices 70, 71, 72, 73 shown is Figure 7.

The step-wise-adjustable line or notch frequency filter includes at least one input 70a, at least one output 70b and a common terminal 70c. A number of resistive elements 80, 81, 82 and 83 may be formed as thin films deposited over a corresponding number of conducting plates with an intervening dielectric layer, such as shown in Figure 1. The resistive 20 elements are connected in series between the input 70a and output 70b terminals.

A number of switches 85, 87, 89 are arranged to be able to selectively bypass or shortout a respective resisdve element. A corresponding number of switches 84, 86, 88 are arranged to be able to 25 seiecrively connect together respective conducting plates with resistive elements which have not been bypassed and hence through a resistance connected to common terminal 70c. The value of the series connected resistances is changed according to which of the resistive elements is bypassed. 32 An RC line 80 of a normalized length of one unit is permanently in circuit, while other RC lines 81, 82, 83 of lengths, for example, 1/2, 1/4, 1/8, etc., units may be switched in or out of circuit by switching selected pans of switches 84, 85; 86, 87; and 88, 89. The effective line length may thus be switched between the values 1, 1.125, 1.25, 1.375, 1.5, 1.625, 1.75, and 1.875 in this example. Since increasing the length increases both the total through-resistance and capacitance, the RC product follows the square of these values, and thus is controlled over almost a 4:1 range.

If it is only desired to vary the RC product over a 2:1 range, the maximum line length need only be root(2) - 1.414 times the minimum line length and this is achievable with switched sections of length 0.207, 0.1035, 0.052 units, etc. With only three such switched sections, 5% Line-length steps corresponding to 10% frequency steps are achievable, and if the nearest frequency step to a desired value is selected, the error is only ±5%.

To create a nimble notch filter with the above arrangement, the resistor from the capacitor plate to ground is also varied to maintain a certain fraction (e.g., 0.056) of the through-resistance. Thus, a switched resistor to ground is also used, such as shown in Figure 9 for example. It is to be understood that the switched nulling resistor circuit of Figure 9 can replace one or more of the resistors R^u-i through Rmai.74 of Figure 7.

As shown in Figure 9, the switched nulling resistor usable in conjunction with the circuit of Figure 8 includes a first, non-switchable null resistor 90 with a relative value R*wu.>t0 which is connected one end of three series connected null resistances 91, 92, and 93. The three null resistances 91, 92, 93 having relative values of, e.g., 1/2 R^^, 1/4 R^u. and 1/8 R^, respectively. The three resistances 91, 92 93 are selectively switchable into and out of the circuit by parallel connected switches 94, 95 and 96.

While the circuits shown in Figure 8 and Figure 9 may be manuracairable using field effect transistor switches, there can be problems 329 with the capacitance and resistance of the switches, as well as limitations on the dynamic range of signal voltage swing through the filter imposed by the switch transistor characteristics.

% The preferred implementation of a tunable notch filter in accordance with the present invention, such as shown in Figures 11 and 13, largely eliminates these problems and gives a notch filter that can operate with a rail-to-rail signal swing.

. It will be appreciated that once a notch filter can be formed on a desired frequency, low-pass filters can be constructed by the cascade connection of such devices to position notches in the stop band so that all frequencies above a certain range are suppressed to a desired extent. Such filters may not have the same sharpness of cutoff as, for example, LC filters, but the present invention does allow practical and useful filters to be made in the frequency range 0.3 to 300 MHz, and such a filter has been successfully fabricated that passes frequencies up to about 3 MHz with little ancnuarion but has high attenuation at 12 MHz and above by the positioning of notches at 12.5 MHz, 35 MHz, 52 MHz and 300 MHz. Because tolerances on the higher frequency notches further away from the passband have little effect on the passband, it was determined that in this instance they did not need to be tunable, and that only the filter having its notch frequency nearest the passband had to be tunable to remove the effect of process spreads.

Figure 10 shows how an adjustable notch filter can be used to obtain a bandpass amplifier response suitable for frequency-selective, intermediate frequency amplification in radio systems of appropriate bandwidth. A tunable notch device 102 according to the invention is connected as the feedback path around an amplifier 101 such that the gain is suppressed outside the notch frequency when the filter 102 allows a strong, negative feedback signal through, while the gain is high around the notch frequency when the negative feedback effect is reduced. A cascade of such tunable selective amplifiers can be used to form an integrated circuit intermediate 329 frequency strip for small, portable radio receivers. The "tuning bits" shown in Figure 10 refer to control signals which operate switches 125-132 of Figure 11 and 140-143 of Figure 13.

A preferred arrangement for an adjustable notch filter that does not suffer loss of dynamic range due to the transistor switches is described below.

Adjustment of the notch frequency is provided by means of a* stepwise adjustable line length using an advantageous, inventive configuration. This is used with a matching stepwise adjustable resistor to form the adjustable notch device.

A preferred implementation of the adjustable RC line is shown in Figure 11. A main, permanently in-circuit line section 110 is cascade-connected with switchable sections 111, 112, 113, 114 on either side. Two switchable sections 111, 112 on the left hand side as shown in Figure 11 have linelengths that are a first fraction dL of the main line length L. The two switchable sections 113, 114 on the right hand side have fractional lengths 3dL. Thus, various effective line lengths can be achieved by switching by corresponding switches 115, 116, 117, 118, 119, 130, 131, 132 the switchable sections in or out of circuit in the following combinations: 111 112 113 114 Effective line length out out out out L out in out out L+dL in in out out L+2dL out out in out L+3dL out in in out L+4dL in in in out L+5dL out out in in L+6dL out in in in L+7dL in in in in L+8dL An important feature achieved by the above arrangement is that the line sections switched into circuit are always contiguous, i.e., no 329 combination of lines such as "in out in" is used. This enables simplification of the switching so chat the capacitor plates only of the ii««t need to be switched., In other words, the switching is thereby simplified as it is only % necessary to switch the capacitor plate terminals of the line sections, and not the series resistive part. To switch a line section dL or 3dL to add to the main line length, its capacitor plate is connected to the capacitor plate of the main line (e.g., by switch 115). To prevent the line section adding to the main line length, its capacitor plate is either left unconnected or connected to ground (e.g., by switch 119). The switched out sections therefore appear as separate, short RC lines or series resistors that are in cascade with the device and not addidve to the effective main line length L. Thus, when the main line terminal is connecad to ground via the nulling resistor of Figure 13, for instance, the frequency to the null in the frequency response so created is not affected by the switched-out sections.

To provide a marching, stepwise-adjustable nulling resistor, the arrangement of Figure 12 could in principle be used. This has a main resistor Rfwu. 123 of nominal value 0.056 of the resistance of the main RC line total resistance. Two switchable sections (126, 127) of fraction dIVL times the main nulling resistor and two switchable sections (124, 125) of value 3dL/L are provided, enabling the same control signals that select the line sections to be used to select corresponding switchable parts of the nulling resistor of Figure 12.

A disadvantage of the arrangement of Figure 12 is that the resistance of the switch transistors that can be fabricated on a silicon chip is appreciable compared to the switched resistance. Therefore, the improved arrangement of Figure 13 is disclosed.

In Figure 13, adjustment of the total effective resistance R is accomplished by switched shunt resistors of high value instead of switched series resistors of low value. The main resistor value R in Figure 12 is now shown in Figure 13 divided into a fraction aR and a fraction (a-l)R. In 329 parallel with the first fraction aR are connected two resistors R1 and R2 switchable into and out of the circuit by two transistors 135, 136. Switching in R1 will reduce the effective value aR to aR-dR where dR is equal to (aR)2/(aR+Rl), while switching in both R1 and R2 will reduce the effective value aR to aR-2dR. Likewise, the two addinonal resistors R3 and R4 connected in parallel to (l-a)R and switchable into and out if the circuit by two additional transistors 137, 138 allow the resistance (a-l)R to be reduced to (a-l)R-3dR or (a-l)R-6dR. Thus, all values of total resistance from R to R-SdR in steps of -dR can be achieved.

Since the adjustment of R is in the downward direction, the value of R must initially be set to 8dR ohms higher than in Figure 13, and the switching transistors 135-138 must be operated by inverse control signals to those of the switches 115 to 118 of Figure 11. The value of the fraction "a" may be chosen so that smallest of the four switchable resistors Rl, R2, R3 and R4 is as great as possible in order to minimize the influence of series switch resistance. If "a" is too small, then Rl and R2 will be unnecessarily small while R3 and R4 are large, and vice versa if "a" is too large.

Therefore, an optimum exists that can be found by calculation.

The construction of the notch filters and adjustable notch filters and their applications has been described here under the assumption that integration on a silicon integrated circuit is the aim, but one skilled in the arc can readily adapt the invention to other forms of fabrication or applications, such adaptations nevertheless being considered to be within the scope of the invention as set forth in the claims. The above discussion of the exemplary embodiments is for purposes of explanation and not limitation. The scope of the invention should be determined by reference to the appended claims. 329782

Claims (1)

1. WHAT WE CLAIM IS: 1 • A low-pass filter suitable for construction on an integrated circuit, comprising: a number of filter sections each having two input terminals and two output terminals and a common terminal, each filter section 5 comprising: a series resistor connected between a first input terminal and a first output terminal of said filter section and a substantially identical resistor connected between a second input terminal and a second output terminal of said filter section, each of said resistors being formed by depositing a pattern 10 of resistive material over respective conducting plates with an intervening dielectric layer to provide distributed capacitance between said resistive pattern and said conductive plate; a resistive means connected between each of said conducting plates to said common terminal, or connected between said conducting 15 plates, or both; a resistor connected between the output terminals of at least one of said filter sections, said filter sections being cascade-connected such that the output terminals of one section are connected to tbe input terminals of the 20 next. 2 • A low-pass filter according to claim l wherein at least one of said resistive patterns is tapered. 3. A device according to claim 2 wherein said taper is exponential. 25 4 • A device according to claim 1 wherein a product of a value of said series resistor with a total of said distributed capacitance of at least 329 - 21 - one of said sections can be adjusted to a desired value in order to determine a notch frequency after formation of said low pass filter. *;5 * A stepwise-adjustable notch frequency filter comprising: at least one input terminal and at least one output rgrminai and a ground terminal; -j. a number of resistive elemenis constructed as thin ffl"" deposited over a corresponding number of conducting plates with an intervening dielectric layer, said resistive elements being connected in series between said at least one input and said at least one output terminal of yfd filter; a number of switches arranged to be able to selectively bypass certain of said resistive elements; a corresponding number of switches arranged to be able to selectively connect together certain of said conducting plates associated with unbypassed resistive elemenis and to a common terminal which is connected through a resistance to said ground terminal; and means to change the value of said resistance depending on which of said resistive elements are switch-bypassed. 6. A stepwise-adjustable notch frequency filter comprising: at least one input terminal and at least one output terminal and a common terminal; a first number of resistive elements constructed as thin films deposited over a corresponding number of conducting plates with an intervening dielectric layer, said resistive elements being comiected in series between an input and an output terminal of said filter; a second number of switches arranged to selectively connect certain of said conducting plates together to a resistance connected to said common terminal; and 329 - 22 - means to change the value of said resistance depending on which of said conducting plates are switch-connected to said resistance. 7. A device according to claim 6 wherein said switches can be operated to connect selected conducting plates alternatively to said common terminal or through said resistance to said common terminal. 8 • A device according to claim 6 wherein said switches can be operated independently to select each conducting plate to be connected either to said common terminal through a separate resistance for each plate or to be connected to a common resistance connected to said common terminal. 9 • A device according to claim 6 wherein said switches are operated only to select conducting plates associated with sequentially contiguous resistive elements to be connected together. 10. A device according to claim 7 wherein said switches are operated only to select conducting plates associated with sequentially contiguous resistive elements to be connected together. 11. A device according to claim 8 wherein said switches are operated only to select conducting plates associated with sequentially contiguous resistive elements to be connecad together. 12. A device according to claim 5 wherein the lengths of said switch-selectable resistive elemenis are in the binary ratios 1:1/2:1/4 ...1/2", where n is equal to the number of resistive elements minus one. 32 9 7 82 - 23 - 13. A device according to claim 5 wherein one of said resistive elements is permanently in circuit and the remainder of said resistive elements are switch selectable and have length ratios in a binary progression. 14. A device according to claim 6 wherein one of said conductor plates is permanently connected to permanently connected to said common terminal and the remainder of said conducting plates can be selectively switch-connected to said common terminal. 15. A device according to claim 14 wherein said remainder of conducting plates being switchable are disposed on each side of said 10 permanently connected plate as dispersed on a semiconductor substrate. 16. A device according to claim 15 wherein said switchable plates disposed on one side of said permanently connected line have associated resistive elements of length one unit increment and the switchable plates disposed on the other side have associated resistive element lengths equal to N+l unit increments where N is the number of switchable plates disposed on said first side. 17. A low-pass filter suitable for construction on an integrated circuit substantially as hereinbefore described with reference to the accompanying drawings. 18. A stepwise-adjustable notch frequency filter substantially as hereinbefore described with reference to the accompanying drawings.