MXPA98003029A - Digital determination of speed in ultrasonic measurements of fl - Google Patents

Abstract

A method and a device (60) for measuring the speed of flowing fluid (F) is described, the measurement is made by measuring the difference of time it takes for an ultrasonic signal to travel first upstream and then downstream of the fluid. In each direction, the device (60) calculates the time it takes for an ultrasonic wave packet, emitted by an ultrasonic transducer (62), to be received by another transducer (63). The method used consists in the digitization (85) of the received waveform and the subsequent identification of the characteristics of the waveform, by comparison with a normal template of the waveform. The time position of these characteristics is then determined with respect to a high-speed clock (70). The results are then used in a weighted computation to determine the arrival time of the waveform in the transducer (6

Description

DIGITAL DETERMINATION OF SPEED IN ULTRASONIC O-MPO FLOW MEASUREMENTS OF THE INVENTION The present invention relates to a device for measuring the flow velocity of a fluid in motion and using ultrasound and digital speed determination.

DESCRIPTION OF THE RELATED ART The measurement of the difference in propagation time for the upstream and downstream travel of an ultrasonic signal in a moving fluid is an established method, used to measure the flow velocity of that fluid. The ultrasonic signal used for these purposes typically consists of a sinusoidal wave packet that reaches the maximum of different amplitudes. An example of this wave packet shown in Figure 3 shows an electronically received ultrasonic signal 40. The problem in accurately determining the arrival time of this signal is considered, in general, to comprise two specific criteria. The first is the unique identification of a particular part of the received signal 40 that can be used as a synchronization reference. The chosen part P1182 / 98MX is usually a point at which a cycle of the wave crosses the axis of signal 42, a so-called "zero-axis crossing". The identification of a particular zero-axis junction is conventionally made with reference to the magnitude of the largest maximum 41 in the received signal 40. This method has several difficulties. The first is that the size of the largest maximum 41 can vary considerably, and is dependent on the conditions under which the signal is being transmitted. For example, if ultrasonic transducers are being used based on piezoelectric materials such as PVDF, the size of the maximum 41 can change by a factor of 30 as the fluid temperature changes from +60 to -20 ° C. More importantly, for a wave packet in a tube, the particular cycle of the received signal 40 where the largest maximum 41 occurs changes frequently, and is dependent on conditions such as temperature and frequency. This is largely due to the fact that the maximum of this signal frequently occurs, but not always, where the secondary acoustic modes constitute a major part of the received signal 40. Represented in general in 43 in Figure 3 is an example of the influence of secondary acoustic and other higher order modes. These secondary modes are much more affected by the temperature and frequency of what is the plane wave (primary mode).

P1182 / 98MX The condition where the two maxima within the wave packet are identical in magnitude is one that the zero-axis crossing method finds particularly difficult to adapt. The second criterion is the identification of the arrival time of the intersection of the identified zero axis, which is in general the crossing of the zero axis 44 immediately after the largest maximum 41, in the signal detector with respect to the time scale that is used . The accuracy of the synchronization of the wave arrival is usually limited to a clock pulse, this time interval becomes, therefore, the uncertainty of the measurement. International Patent Publication No. WO 93/00569 entitled "An Electronic Fluid Flow Meter" describes a detection arrangement of the acoustic wave packet and the associated flow measurement apparatus that incorporates a solution to the above problems. This arrangement uses envelope detection and an armament method for the detection of the zero-axis junction. The publications of International Patents Nos. WO 93/00570 and WO 94/20821 each describe different methods for reducing the propagation of higher order acoustic modes which, as discussed above, can contribute significantly to the P1182 / 98MX wave packet received and, in this way, cause synchronization errors. U.S. Patent No. 5,206,836 describes a digital array for determining, based on a linear regression around an individual zero-crossing, the arrival of a wave packet.

SUMMARY OF THE INVENTION It is an object of the present invention to provide a measurement device and method for ultrasonic synchronization, which allows both the unique identification of a part of an ultrasonic signal and the timing of the arrival of that part of the signal in a detector. . According to a first aspect of the present invention, a method for detecting the arrival time in a transducer of a packet of acoustic waves transmitted to a first frequency is described, the method comprises the steps of: (a) converting, to a sampling frequency, an analog signal output that comes from the transducer in digital signal data; (b) determining a measurement portion of the digital signal data, the measurement portion responding to a response of the transducer to the acoustic wave packet; P1182 / 98MX (c) determining from the moving portion a plurality of slopes (Si) for a corresponding plurality (i) of measurement segments of the response, each of the segments is approximately a signal level value (V); and (d) determining from the at least one of the selected slopes (Si), the arrival time (t) of the acoustic wave packet. In general, the method comprises the additional step, between steps (c) and (d) of: (ca) determining, for one of the selected segments, a corresponding measurement time (Ai) indicative of a time in which the response intercepts the signal level value (V); wherein step (d) comprises determining the arrival time (t) from the corresponding measurement times (Ai.). Preferably, an additional step is provided, between steps (c) and (ca) of: (ca) matching the slopes (Si) with a corresponding reference set of slopes (P) to determine a measuring position of each one of the segments in the measurement position; wherein the step (ca) comprises determining the corresponding measurement times (Ai) from one of the corresponding measuring positions.

P1182 / 98MX In general, step (c) comprises assigning a set of () points of the digital signal data to each measurement segment (i), each measurement segment (i) being centered around two adjacent points whose values they are on either side of the signal level value (V), fitting a straight line to each set of points, and then determining the slope (Si) of each straight line. Advantageously, the straight line comprises a line connecting the two adjacent points. Preferably, the value of (k) is related to the ratio of the sampling frequency to the first frequency, and to the consequent separation of the digital signal data in the measurement portion. Typically, k is equal to one eighth of the number of points per cycle. Most preferably, the value of k is 10. In an advantageous array, each slope (P) of the slope reference set is associated with a specific position parameter (n) which marks the position of a polarity transmission of signal within the wave packet, the parameter (n) that corresponds to the number of half-wave periods between the beginning of the wave packet and the transmission of the particular signal polarity. In a preferred implementation, step (d) comprises the following sub-steps: (da) assign a weighting factor (W to each P1182 / 98MX one of the measuring times (A) based on the corresponding position parameter (n); (db) selecting (m) from the measurement times (Ai) those to be used in determining the arrival time (t); (de) determining a half average wave period (t) from time intervals between one of the selected (m) measuring times (At), adjacent; (dd) determining, for each of the selected (m) measurement times (Ai), the selected arrival time of the wave pack in the transducer, using the expression tt = A - n.t; and (de) determining a sum, over the (m) measurement times (Ai), of a product of the weighting factor (W) assigned to each measurement time and the estimated arrival time, corresponding (tJ; ) determine the arrival time (t) of the acoustic wave packet by dividing the sum of the step (de) by a sum of the (m) weighting factors (W,.) In general, each weighting factor (W) is proportional to the slope (Si) of one of the corresponding segments Typically, m can take a value from 2 to 20. A preferred value of m is 6. Advantageously, step (db) comprises selecting the measurement times (A dependent on to the P1182 / 98MX minus one of the precision of measurement time determination, the closeness of the connection to the reference set of slopes (Pj), and the freedom from the interference of the measurement portion from acoustic modes of order higher. In the preferred embodiment, the sampling frequency is used to generate the acoustic wave packet in a preferential manner the first sequence is a sub-multiple of the sampling frequency. In general, the value of the signal level is derived from the average of those digital signal data received in advance of the measurement portion. Also, step (a) typically comprises the continuous conversion of the signal. According to a second aspect of the present invention, a method for determining the propagation time of an acoustic wave packet transmitted between two ultrasonic transducers is described, the method comprises the steps of: (e) starting a clock operating at a default frequency, (f) simultaneously with a clock transition, the steps of: (fa) energizing one of the transducers to extract the packet of sound waves at a frequency P11B2 / 98MX lower than the determined frequency; and (fb) keeping the clock as a synchronization reference for the subsequent detection of a wave pack arrival in another of the transducers; (g) detecting the arrival time of the wave pack using the method of the first aspect, wherein the sampling frequency is the predetermined frequency and is provided by the clock, and the digital signal data including the measurement portion is stored in a memory whose address determination for consecutive samples of the digital signal data are altered by the array, such that each address in the memory corresponds to an identifiable time after generation of the acoustic wave packet. In general, a plurality of acoustic wave packets are transmitted and, except for the first transmission, in the transmission of all subsequent or subsequent acoustic wave packets are synchronized to the reception of the immediately preceding acoustic wave packet. Typically, in the transmission of a particular acoustic wave packet, a delay time is counted and the transmission of the next acoustic wave packet is instituted after the delay time has elapsed and when the measurement portion corresponding to the packet of particular acoustic waves P1182 / 98 X next crosses the value of the signal level. In an alternative configuration, a packet of acoustic waves is transmitted simultaneously from each of the transducers for the reception of the transducers by the other and the propagation time of each of the acoustic wave packets is determined from this way. Typically, in this arrangement step (a) further comprises compensating for phase differences between the digital signal data converted from analog signals extracted from one of the respective transducers. According to a third aspect of the present invention, a system for measuring the velocity of a fluid in motion is described, the system comprising: a measuring tube through which the fluid whose velocity is to be measured passes; two ultrasonic transducers placed in the tube and configured to transmit ultrasonic signals to each other; a transmission means for driving at least one of the transducers to generate a packet of acoustic waves propagating at a first frequency along the tube to the other of the transducers; a receiving means connected to at least the other transducer to detect the wave pack P1182 / 98MX acoustic and to convert an analog signal output, which comes from at least the other transducer, to digital signal data, the receiving means includes a "first" memory for storing the digital signal data; a processor means connected to the receiving means for determining the velocity of the fluid, the processing means comprises: a first means for determining a measurement portion of the digital signal data, the measurement portion corresponds to a response of another transducer to the package of acoustic waves; a second means for determining from the measurement portion a plurality of slopes (S_) for a corresponding plurality (i) of measurement segments of the response, each of the segments is approximately a signal level value (V); a third means for determining from the at least one of the slopes (Si) selected the arrival time (t) of the acoustic wave packet. Typically, the system further comprises four means for determining, for one of the selected segments, a corresponding measurement time (Ai) indicative of a time in which the response intersects the value of the signal level, wherein the third means P1182 / 98MX determines the arrival time (t) from the corresponding measurement times (Ai). Preferably, the system further comprises a (second) memory means connected to the processor means for storing a reference signal, the processor means includes a fifth-medium arranged to equalize the portions of the reference signal with the measurement portion to determine a measurement position of each of the segments in the measurement portion, wherein the measurement times (Ai) are determined using one of the measurement positions. Advantageously, the system further comprises a means for adjusting the magnitude of the analog signal from at least the other transducer, so that it falls between certain limits, these limits are also preferably displaced on either side of the level value of signal, the signal level value is preferably an average of one output of the other transducer before the measurement portion. Typically, in either aspect, the sampling frequency is between 500 kHz and 10 GHz. Preferably, the sampling frequency is 10 MHz. In general, in either aspect, the first frequency is between 20 kHz and 5 kHz. MHz. More preferably, the first frequency is 125 kHz.

P1182 / 9BMX In accordance with a broad aspect of the present invention, a method for detecting the arrival time in a transducer of a packet of acoustic waves transmitted at a first frequency is described, the method is characterized by compensating derivations from the first sequence of the acoustic wave packet as detected by the transducer.

BRIEF DESCRIPTION OF THE DRAWINGS A number of preferred embodiments of the present invention will now be described with reference to the drawings in which: Figure 1 depicts the arrangement of two ultrasonic transducers placed in the fluid flow field, the ultrasonic transmissions of which are used to measure the flow velocity of the fluid according to the described modality; Figure 2 is a schematic block diagram representation of the functional units of the preferred embodiment; Figure 3 depicts a typical ultrasonic waveform emitted by the ultrasonic transducers in Figure 1; Figure 4 is an information flow diagram for the preferred detection method; P1182 / 98MX Figure 5 illustrates the signal amplification and digitization process used in the described modes; Figure 6 illustrates the effect of the increased gain on the digitized waveform; Figure 7 is a schematic block diagram of the electronic components for a first embodiment; Figure 8 is a schematic representation of a multiplexer unit of Figure 7; Figure 9 is a schematic block diagram of a second embodiment using the so-called "false trigger" method; Figures 10A and 10B illustrate the ignition sequence for the transducers in the embodiment of Figure 9; Figures HA and 11B illustrate the timing of that ignition sequence of Figures 10A and 10B for the two drivers of the transducers in the embodiment of Figure 9; Figure 12 is a schematic block diagram representation of a third embodiment in which the ultrasonic transducers transmit simultaneously; and Figure 13 is a schematic representation of the part of the microprocessor used in the described embodiment.

P1182 / 98MX BEST MODE AND OTHER MODES FOR CARRYING OUT THE INVENTION Figure 1 shows two ultrasonic transducers, TI and T2, each of which can act as either a transmitter or a receiver, placed oriented towards each other as along the axis of a cylindrical measuring chamber C through which a fluid F, whose flow velocity v is to be measured, is flowing. The measuring chamber C is part of a total measurement circuit 30 shown schematically in Figure 2. The measurement system 30 includes a control microprocessor 33 linked to a main common bar 37 to a microelectronic unit 31 containing specialized electronic products that are interconnected to the TI and T2 transducers, and to a temperature probe 32, which can be placed inside the measuring chamber C to determine the temperature of the fluid F with the flows it is measuring. A battery unit 36 supplies electrical power to the microprocessor 32 and the microelectronic unit 31, the last of the two generation signals and the reception signals from the CT and T2 transducers, and the individual times of flight determination (propagation) of the acoustic wave packets, the microprocessor 32 provides the average times of flight and the calculation of the P1182 / 98MX flow rate for displaying on a liquid crystal display (LCD) 34 via a common display bar 38. A communication interface 35 is provided to transfer the flow rate of other data to an appropriate data acquisition system (not illustrated) but known in the art, for example from the application of International Publication No. WO 93/00569. The value for the fluid flow velocity from the measured flight time of an ultrasonic signal with and against the fluid flow can be derived using the expression: v. { = 0.5L (1 / Td-1 / Tμ) where: vf = Fluid velocity in the measuring tube, L = Measuring tube length Td - Flying time of the downstream acoustic pulse, and Tu = Flying time of the acoustic impulse upstream. Therefore, the exact measurement of flight time (or propagation) is the central task in determining the flow rate by the preferred embodiment. In the preferred embodiment, a row of short acoustic impulses is transmitted from an ultrasonic transducer to one end of the measuring tube, which is detected (received) by a similar transducer in the P1182 / 98MX other end. After passing through a variable gain amplifier chain, the received signal is digitized at a rate of 10 MHz by an A / D converter to give the reception data. The reception data is then entered into 4 Kbyte of random access memory (RAM). To achieve the required accuracy, the arrival time of the acoustic signal must be determined to be within approximately lns, that is, 1% of the time between the digitizing samples. Therefore, it is desirable to interpolate between the points in the digitized signal. This is done by performing linear regressions on the reception data of a band around the quiescent signal level, and using the intercepts obtained in this way, together with knowledge of the received signal frequency, to make a number of calculations separate from the arrival time. A weighted average of these results forms the final estimate of the acoustic transit time. To reduce the uncertainty of the measurement, a number of specific features are included in the preferred embodiments. First, the A / D converter used is an 8-bit device that only operates on a window that covers approximately one-third of the received signal, so that the effective accuracy of the conversion is increased from 8 bits to (8 + log2 (3)) P1182 / 98MX bits, that is, approximately 9.6 bits. This arrangement is preferred because data points outside this average range of a sinusoid carry more amplitude information than the synchronization information, and thus are of little use for synchronization measurement. Second, pulse generator for the transmission transducer that links directly to the 10 MHz clock. All crystal clocks suffer fluctuation to some degree, and if it were activated in a stable oscillator, separated from a single clock transition, the complete uncertainty of the fluctuation would result in the measurement. By joining each transition of the acoustic impulse general to an edge of the 10 MHz clock signal, it is substantially averaged at the center of the clock jitter. Third, because the transducers are resonant devices, although briefly resonant, the frequency of the received signal can not be the same as that of the driven signal. Under the conditions of the currently described application, if the received frequency differs from the transmitted frequency by saying 10, and no correction is made, an overestimated or underestimated 0.35% over the entire flow interval will occur. The resonant frequency of many ultrasonic transducers changes markedly with temperature, thus in the P1182 / 98MX calculation of the arrival time of the start of the received signal, the period of oscillation is not assumed to be equal to the frequency transmitted, but instead it is derived from the data of the received waveform. A specific modality that comprises the advantages of this method in the measurement of the velocity of a gas, integrating this to measure the volumetric flow and exhibiting that result, is represented in Figure 7. In Figure 7, a measurement system is shown 60 including a tube section 61 adapted for fluid flow measurement, two transducers 62 and 63 positioned within section 61 and separated by a predetermined distance for transmitting or receiving ultrasonic signals, a battery unit 64 (not illustrated for the purposes of clarity, but whose connection would be well understood in the art), a microprocessor 65, a liquid crystal display (LCD) 66, a serial communication interface 67, electronic measurement components 68 for driving and receiving from transducers and synchronizing the ultrasonic flight time, and a probe 101 for measuring the temperature inside the tube 61. The microprocessor 65 is preferably a di 16-bit device built using a "von-Neumann" architecture and include 4 Kbytes of read-only memory (ROM) and an internal RAM of P1182 / 98MX 256 bytes. Also contained in the microprocessor 65 is a receiver and a universal asynchronous transmitter (UART), an analog-to-digital converter (later in the present ADC) with a current source, an interruptible synchronizer usable as an event switch and an oscillator crystal of the sequence controller synchronizer which is a day time device that provides a signal of 32,768 kHz. The latter is used as a real-time clock and, when multiplied by frequency-synchronized circuit techniques, is used as a clock for the microprocessor 65. The LCD 66 incorporates a driver that interconnects the microprocessor 65 in a known manner. A unique, programmable, electronically erasable readout memory (EEPROM) 72 is connected to a common bus 79 of the system, which is controlled by the microprocessor 65 and provides the calibration information for the measurement system 60. The serial interface 67 allows the system 60 is coupled to the devices through either an optical link, or some common patented communications bar such as common M-bar. The battery unit preferably comprises a lithium cell of size D of 3.5 volts, single.

P1182 / 98 X A controllable crystal oscillator 70 and crystal 71 provide, via a common clock bar 99, a high frequency clock, at a preferred frequency of 10 MHz, to a high speed analog-to-digital converter 85 ( instant type) (later in the present instant ADC) and the logic circuit of the memory control 74 comprising an address counter 76 and a read / write control logic circuit 75. The oscillator 70 is started before a flow measurement and allows time to be established. When it is not required, it is closed to conserve energy. A transmission clock generator (transmission counter) 80 is connected to common busses 79 and 99 and divides the high frequency clock to provide a clock signal 100 for a transmission pulse sequence generator 81. The transmission frequency may be varied under the control of the microprocessor control 65 by changing the number of high frequency clock cycles comprising each transmission clock pulse. The transmission pulse sequence generator 81 is formed by a shift register which retains a bit pattern which, when synchronized, forms the transmission pulse sequence. The bit pattern is initialized by the microprocessor P1182 / 98MX 65 via the common bus 79 to which the sequence generator 81 is also connected. More complex waveforms can be generated by having more bits in the shift register and synchronizing at a high speed. The edges of the outgoing signal are synchronized by the high frequency clock. A quantization error is prevented at the start of transmission by having the transmission sequence synchronized by the same clock that controls the instant ADC 85. The transmission pulse sequence is supplied to four 94-97 tri-state drivers, arranged in two pairs 94 and 95, and 96 and 97, each pair that is connected to a transducer different from the transducers 62 and 63. The pair of impellers (e.g., 94, 95) connected to the transducer (62) that is beginning to transmit the ultrasonic signal is left while the other pair (96, 97) is placed in a state of high impedance, allowing the transducer (63) connected to it to act as a receiver. The two outputs of the pair of impellers (94, 95) are driven in anti-phase such that when one goes high the other goes low. This effectively applies twice the drive signal to the transducer (62). Each of the ultrasonic transducers 62, 63 when they receive an acoustic signal appear to the rest P1182 / 98MX circuit as a source of differential current. To allow reception of the acoustic signal, each of the transducers 62, 63 is connected to two analog multiplexers 92 and 93, which in turn are connected to the corresponding current integration preamplifiers 90 and 91, respectively. A schematic representation of the analog multiplexers 92 and 93 is given in Figure 8. The multiplexers 92, 93 switch to select one of the two different ultrasonic input sources and connect it to the input of the preamplifier 90 or 91. The switches in the "off" state receives the drive signal from the transducer. Each preamplifier 90 and 91 amplifies the difference between the two current signals coming from transducer 62 and 63. This minimizes the effect of any interference that occurs on both inputs, by canceling common mode signals. The preamplifiers 90, 91 also provide a DC bias for the analog multiplexers 91, 93, and in turn, the corresponding transducer 62, 63. The gain of the preamplifiers 90, 91 is sufficiently high to ensure that only the source of significant noise is in the amplifier of the front end of it. The signal is filtered in a PU82 / 98MX low pass filter while in the preamp stage to reduce the amplitude of the noise. It is advantageous to have a pole approximately 65 kHz. A differential amplifier 89 adds the outputs of the preamplifiers 90 and 91, and provides an input to a variable gain amplifier 83. The gain amplifier 83 is controlled by a gain control signal output 82 of the microprocessor 65 to give an interval of signal amplification from 0 dB to 60 dB. This interval is required because the amplitude of the received signal varies with the direction of the ultrasonic signal, with respect to the flow rate, the amount of flow, and the temperature. The received signal transferred from the amplifier 83 is then filtered by the bandpass filter 84 operating between 80 kHz and 170 kHz. The output of the filter 84 has a DC level centered on approximately a reference voltage, which is generally about half the supply voltage provided by the battery unit 64. The bandpass filter 84 is transferred to the instantaneous ADC 85. which preferably has a resolution of 8 bits. The instant ADC 85 converts the received signal to a digital signal at a sampling rate P1182 / 98MX equal to the high frequency clock generated by oscillator 70 (ie, 10 MHz). The operation of the instantaneous ADC 85 is depicted with respect to the waveform in Figure 5. If the received signal is located between the designated signal levels Superior Reference (i.e., Top Reference) and Background Reference (i.e., Reference of the Fund), then the digital value is a measure of this position. If the signal exceeds the Upper Reference then the output is FF (Hex) and if it is smaller than the Reference of the Fund the output is 00 (Hex). The Superior Reference at a CD level generated by the circuit 87 that provides the upper reference voltage for the instantaneous ADC 85. The Background Reference is a CD level generated by a circuit 88 that provides the lower reference voltage for the Ade instantaneous 85. The distance between the Upper Reference and the Background Reference is preferably set to 0.375 of the maximum height that the received, amplified signal can achieve without significant distortion. The position of these voltages must be such that when the signal is at the maximum height it is centered, so that the amplifiers 83, 87, 90 and 91 do not cut or distort the maxima of the waveform. It is important to avoid any clipping in the waveform by amplifiers 83, 87, 90 and 91 because it would introduce a CD misalignment in the P1182 / 98MX signal received, processed. A signal reference 86 is a CD level generated from the levels of the upper reference signal and the background reference and is in between these two voltage levels. The signal reference 86 is used as the DC bias for the output of the bandpass filter 84 as discussed above. The data samples taken by the instant ADC 85 are transferred into a common signal data bar 78 which connect a high-speed static memory (SRAM 77), which preferably has a capacity of 32 kbytes, and a latch buffer 73 , the above which provides the storage of the waveform of the part of the received signal. A memory read / write control unit 75 produces the read and write pulses required by the high-speed memory 77. An address counter 76 within the control unit 75 is connected to the high-speed memory 77 and includes a 12 bit counter with the associated logic circuit. The waveform data from the instant ADC 85 is stored in the high-speed memory 77 under the control of the address counter 76 and the memory read / write control logic circuit 75. The memory 77 has two modes of operation. P1182 / 98MX In a "measurement mode", the data is transferred from the instant ADC 85 to the static memory 77 in real time. One byte is transferred for each cycle of the high-frequency clock, which corresponds to the data transfer rate of 10 megabytes per second. The address counter 76 provides the address to the memory 77. In this way, the address counter 76 is automatically incremented to make transaction. This means that the address has a one-to-one correspondence at the time the waveform is displayed after the transmission of the ultrasonic signal. In a "processing mode", the recovery and processing of the data is performed by the microprocessor 65. The data can be transferred to or from the microprocessor 65 from a selected address simply by loading the address counter 76 with the address required The address counter 76 is incremented in each reading or writing of the high-speed memory 77 by the microprocessor 65. The data from a series of addresses can therefore be read, or written, to another address when the meters are loaded. address 76 with the first address, then reading or writing the bytes in the sequence. The transfer of data between memory 77 and the P11B2 / 98MX microprocessor 65 is via latch buffer 73. Microprocessor 65 reads from or writes to latch memory 73 instead of directly from or to memory 77. This allows memory 77 to remain in a low mode. energy, most of the time while the slower microprocessor 65 is reading or writing data. With reference to Figure 13, which illustrates only part of the architecture of the microprocessor 65, the temperature of the fluid is measured by means of a microprocessor current source 69 that drives a resistive temperature sensor 101. The magnitude of the current source 69 is programmed by the external resistor 102. The input from the resistive temperature sensor 101 can then be differentiated back to the supply voltage, provided by the battery unit 64, in the same way as the current source 69, allowing This way a metric relationship measurement takes place. As mentioned above, the microprocessor 65 includes a μADC 103 that is formed from a comparator 104, which receives the analog input, a successive approximation register 105, and a digital-to-analog converter (DAC) 106, a well-defined configuration. known in the art. A bidirectional buffer 107 allows the coupling of the P1182 / 98MX data converted into the internal common bars 108 of the microprocessor 65. For temperature measurement, the voltage is taken directly from the sensor 101 and applied to an analog input of μADC 103. The sensor 101 is a sensor of resistive temperature supplied by the current source 69. The magnitude of the current source 69 is a function of the supply voltage and is programmed by the external resistor 102, such that: dfuente = (0, 25, Vdd) / Rext. The input from the resistive temperature sensor 101 thus differentiates back to the supply voltage in the same way as the current source 69: V_n = (0.25, VJd). (R, ßna / R.xt), (1) where R.! (- n3 is sensor 101, a resistive element of temperature variation.The output of μADC 103 is a linear function of supply voltage: where a and K they are known constants and N is the multiplier of the DAC 106. The comparator 104 compares the output of the DAC 106 with the sensor voltage 101. This determines the value N for which Vdac = Vin.

P1182 / 98MX The measured value N can then be transferred via buffer 107 for calculations of temperature compensation. The measurement of R3en3 is a relationship metric. Combining equations (1) and (2) above, the supply voltage is removed from the calculation, to give: (0, 25, Vdd). (R3en3 / Rβ?) = (A + K. N). V.d Rsens - (a + K. N). Four . Rβ? T Also, the buffer 107 can be used for automatic gain control whereby the microprocessor 65 determines the appropriate value, which is then input to the DAC 106 via the buffer 107. The DAC 106 then performs a direct conversion to provide the gain control voltage 82. The switch 109 is provided to isolate the gain control functions of the temperature functions. In view of the temperature that is relatively stable over more than one sample, which requires the determination of the gain control, the temperature functions can be analyzed when the flow measurements are idle. In an alternative configuration, a parallel or serial (eg I2C) output of the microprocessor 65 can drive an external DAC (not shown) that can directly supply the control voltage 82 P11B2 / 98MX gain. The conservation of energy is important for the long life of the battery. The current consumption is reduced by keeping the electronic products in a low energy mode while it is possible. The address counter 76 can be used to help achieve this conservation if it doubles as a counter in the attention delay. If the address counter 76 is initialized to a non-zero value, then the transport can be used from the counter 76 as an attention interruption to the microprocessor 65. The microprocessor 65 serves and turns on the electronic products of the signal path consisting of the preamplifiers 90, 91, in differential amplifiers 89, the amplifier Variable gain 83, bandpass filter 84, enables memory 77 and instantaneous ADC 85. Microprocessor 65 then returns to a low power mode while the data is being collected. Having described the structure of the measurement system 60, the flight time measurement will now be described in detail with reference to Figure 4, which schematically represents the flow of information and a number of method steps resulting from the mode of Figure 7 P1182 / 98MX At point (1), the clock signal 10 MHz, generated by oscillator 70, provides the synchronization reference information. At point (2), to initiate a measurement, the 10 MHz clock is used, by the counter 80 and the generator 81 to generate a row of three square wave pulses of nominally equal mark / space duration, which are fed to the transmission transducer in the measuring tube 61 for the generation of the ultrasonic signal. In the present example, the ultrasonic signal preferably has a frequency of approximately 125 kHz, so that each cycle in the ultrasonic signal has a duration of 80 clock cycles of 10 MHz. The use of the 10 MHz clock to generate the transmitted signal ensures that the synchronization of the transmitted waveform is known exactly. At point (3), starting at the same clock transition as the transmitted signal, the address counter 76 is enabled. In this way, a synchronous counter is used to generate sequential addresses for the memory 77 at the speed of one direction for 100 ns. As indicated in point (7) in Figure 4 and as discussed above, the output of the reception transducer, which includes the desired analog waveform, is passed to the instantaneous ADC 85 in which the conversions to the digital form by P1182 98MX clock transitions of 10 MHz. In this way, the memory 77 receives its address control information from the address counter 76 and its data from the instant ADC 85, at the rate of one byte every 100 ns. The static memory 77 can therefore store 409.6 μs of digitized waveform. Some time later transmission begins, and while the acoustic wave packet is still propagating along the measuring tube 61, the instant ACD 85 and the static memory 77 are enabled and the incoming data, initially at the CD level , begin to be written in the static memory 77. To conserve energy, the microprocessor 65 does not activate the static memory 77 until close to the expected arrival time of the acoustic signal. If the conditions have changed and the signal is omitted, the measurement is repeated with more conservative settings. Because the required part of the received waveform is much shorter than 409.6 μs, the rotation of the address counter 76 and the subsequent overwriting of the static memory 77 do not constitute a disadvantage. The acoustic wave packet generated by the transmission of the transducer travels along the measuring tube 61, and after approximately 500 μs, (in a measuring tube of a length of 175 mm and a P1182 / 98MX sound velocity in natural gas crosslinking system of approximately 350 ms "1), is received by the receiving transducer, resulting in an analog electrical signal, such as that shown in Figure 3, which is transferred from The transducer of reception This signal is amplified in a number of stages, including a variable gain stage, the result is a signal, indicated in point (5), of a standardized maximum amplitude, known The derivation of variation of gain described below, the received acoustic wave packet is thus converted by the instantaneous ADC 85 and stored in the static memory 77. The microprocessor 65 for data collection after the anticipated arrival time of the acoustic wave packet is stopped. 10 MHz clock or counter clock and the 10 MHz clock is stopped to the instant ADC 85, which also works as a strobe mark to write to me static memory 77. if the wave packet is absent, then the measurement is repeated using a larger delay. The largest possible delay will correspond to the largest possible delay of the acoustic wave packet due to the maximum fluid flow in a fluid of density greater than the lowest temperature. The address counter 76 must be allowed to run for the entire period from the initiation of the acoustic signal, P1182 / 98MX at the moment in which the microprocessor 65 disables the static memory 77, otherwise the position of a data byte in the static memory 77, will not correspond to its arrival time. With a digitized representation of the signal received in the static memory 77, the microprocessor 65 and its associated computer program can then be extracted an estimate of the arrival time of the acoustic signal. This is done by readjusting the address counter 76, and switching its input from the 10 MHz clock to an output line of the microprocessor 65. A start address is loaded in the address counter 76 and, when the output line is tilted of the aforementioned microprocessor, the computer program can access the subsequent bytes of the static memory 77 in sequence. The start address is also loaded into a memory location in the microprocessor 65, and this "shadow" address value is incremented each time a data value is extracted from the static memory 77. To establish a quiescent signal level 45 (see Figure 3), which precedes the arrival of the acoustic wave packet and which acts as a reference against which all other measurements are compared, a number of bytes are read, typically an energy of 2 such as 128 or 256, from static memory 77, they add up and then they P11B2 / 98MX average. The starting point for the address counter 77 should be sufficiently low to allow this determination of the quiescent signal level 45 of the data received prior to the previous anticipated arrival of the acoustic signal. The quiescent signal level 45 in this way is representative of a non-received signal state (NRS) of the measurement system 60. Also, the previous anticipated arrival of the acoustic signal represents the beginning of a measurement portion of the measurement data. stored digital signal that are representative of an analog signal of the transducer indicating a response of the transducer to the acoustic wave packet.The measurement portion may extend for approximately 10 to 20 cycles of the received signal 40. Once the level has been determined of quiescent signal 45, at point (8), a circular buffer of the computer program is established by the microprocessor 65, which has a duration that preferably corresponds to approximately 10 to 15% of a cycle of the acoustic signal. present example, where approximately 80 samples of the signal level are taken per cycle, a circular buffer size of 9 or preferente All 10 bits is adequate. The circular buffer technique P1182 / 98MX allows data to be updated without the need to physically change each entry along a location each time a new data point is entered. A buffer indicator, initially zero, indicates the position in the buffer where the next entry is to be made. The indicator increases after each new entry is made, and when the value of the indicator exceeds the length of the circular buffer, the indicator is reset to zero. In this way, the indicator "circular" therefore around the circular buffer, and once the circular buffer is filled, the position of the indicator at any time denotes the previous entry in the circular buffer. When each byte of static memory is read 77, and before it is placed in the circular buffer, it is compared with the quiescent signal level 45 obtained previously, and the sign of the difference is compared with the value obtained from the previous byte. If the sign is the same as that calculated from the previous byte, the received signal has not crossed the quiescent signal level 45 and the current sign is stored for a future comparison. If the current sign differs from the previous sign, the received signal has crossed the quiescent signal level 45 and a P1182 / 98 X limit that is equivalent to the current value of the "shadow" value and direction mentioned above, plus half the size of the circular buffer. The bytes are then read from the static memory 77 and placed in the circular buffer until the value of the shadow address reaches the limit mentioned above, at which point the circular buffer contains a set of data points placed approximately equal around the quiescent signal level 45. This represents a individually manipulable measurement segment for the synchronization calculations. In this step, the contents of the circular buffer are used to perform a linear regression, indicated at point (9) in Figure 4, using the shadow direction values that correspond to the time axis as the independent variable, and the data points in the circular buffer as the dependent variable. Linear regression comprises the ato of the data pairs of voltage and time to a straight line using a least squares technique. The regression line technique allows an estimate of the slope or slope of the signal as it passes through the quiescent signal level 45, and also a time intercept value in that P1182 / 98MX point time, which are determined for each measurement segment. At point (10) both slope and intercept estimates are placed in a first-in, first-in (FIFO) buffer of the computer program in the internal memory of the microprocessor 65, for later use. The computationally less efficient FIFO buffer is used for this application because it is short, access is relatively simple, and is updated at only one-fortieth the speed of the circular buffer. Because, as seen in Figure 3, because the received waveform begins imperceptibly and accumulates in amplitude over the various cycles, it is not possible to directly determine its starting point. The starting point must be inferred by locating an earlier point in the waveform (this position used as a synchronization marker) and by subtracting a period of time corresponding to a known number of waveform cycles. (period that in turn corresponds to the time elapsed from the starting point of the wave packet) from that moment. Typically, the position in the received signal 40 used as a synchronization marker in synchronization PU82 / 98MX of the packet transit of both along the tube 61 is a particular position in the waveform when the waveform passes through the quiescent signal level 45. The intercepts determined from the linear regression in the point (9) thus serve as the synchronization markers if their absolute position can be determined within the waveform as a whole. To do this, the successive slope values within the FIFO buffer are compared with a stored set of slopes, referred to herein as a "spreadsheet" and are indicated at point (13). This stored spreadsheet is representative of the anterior portion of the waveform, the shape of which can vary depending on the fluid flow velocity, fluid properties, and temperature. A suitable size for this template is from two to four elements, either of successive magnitudes of slope of alternating sign, or of alternating magnitudes of slope of the same sign. Once the FIFO buffer of the slope has been filled, as each new slope value is entered (including the magnitude and sign), the elements selected in point 14 are compared (with the template for P1182 / 98MX obtain a sum of squares of differences. This measurement goes through a minimum when the template better matches the selected elements of the FIFO buffer of earrings, and the process stops the matching process when the newly calculated sum of squares exceeds the previous value. In this stage, the intercepts have been correctly located within "the waveform as a whole, based on the known template, with each intercept being nominally an integral number of the new periods from the beginning of the received signal. In this way, the equalization of the slopes determines a measurement position for each slope within the received waveform, which is related to the beginning of the acoustic wave packet.As previously indicated, the frequency of the received waveform can not equal to that of the impulse signal, meaning that the half-period of the impulse waveform can not be a good estimate for the new period of the received waveform.This is particularly so where the signal is in the form of a short increment instead of a long wave row Where transducers 62 and 63 are made of a material such as PVDF, the resonant frequency of the combination lada P1182 / 9TMX transducer / camera / fluid, dominates the shape of the received signal, and this resonant frequency can vary markedly (from 10 to 15%) with temperature and fluid properties. In addition, the effects of aging can alter the piezo and elastic and elastic properties of the transducer, producing changes in the sensitivity and resonant frequency of the system. To provide these changes, at point (15) an average of slow movement of the received waveform period is maintained to calculate the starting point of the received signal. This average can be conveniently updated from the differences in the interception values of the FIFO buffer. A fraction of the difference between the newly calculated half-period and the value of the old half-period is added to the old value. If the fraction is, for example, 0.1, then after 20 repetitions, the moving average will cover 88 l of the step change in the middle period. Using the smallest fractions of the difference that results in correspondingly slower responses in moving average. This moving average is then used, in the movement (16), as the half period of the received signal. Any memory interception value P1182 / 9TMX intermediate FIFO can be used to estimate the arrival time of the start of the waveform, by subtracting the appropriate number of half-periods from the waveform. To improve accuracy, more than one intercept can be used and the results can then be averaged. It is preferred to use a weighted average, where the weighting factors, indicated in point (17), are derived from the slope corresponding to each intercept. The above intercepts are from a low amplitude part of the received signal and therefore are relatively more affected by noise. Since the maximum slope of a sinusoid is proportional to its amplitude, the weighting factors are proportional to the absolute slope at each intercept, averaged over the temperature and the type of fluid. The weighted-average estimate of the start of the waveform, indicated at point (18), is calculated by the microprocessor 65 in terms of the sample number and the fractions thereof due to the last rotation of the address counter 76. In the present case, the address counter 76 would have "regirado" to 409.6 μs, so that the total time of flight of the acoustic signal, indicated in the point (19) is: Td, u = (409.6 + (weighted estimate -average P1182 / 98MX of the start of the waveform) / 10) μs. In point (6), a new gain estimate referred to above is calculated, starting from the maximum slope, derived from point (11), found in the FIFO buffer of the slopes. The value of the maximum slope allows the amplitude of the maximum 41 to be estimated and therefore, by adjusting the gain of the amplifier 83, for the amplitude of the maximum 41 that remains constant, or at least substantially constant within narrow limits . These limits are generally within ± 5%, and preferably about ± 2.5%. Allowing the profit to deviate beyond these limits may result in a mismatch of the template to the measured slopes, and therefore an inaccurate location of these interceptions. If the gain of the variable gain amplifier stage produces a signal output directly proportional to its input control voltage, it can obtain the re-estimation of the gain from the equation: Gn = (G0) (Maximum Slope Objective) / (Sn) where: Gn = Re-estimate of the required gain, G0 = previous gain value, and P1182 / 98MX Sn = most recent maximum slope. However, depending on its design, the response of the gain control stage can be best approximated by an exponential or polynomial curve. In the present case, a cubic curve gives a satisfactory adjustment and after the linearization, the equation of prediction of gain can be expressed as: Gn = G0 + Ci - C2. (3 VS «- C 3)) where Cl, C2 and C3 are constants to adjust the particular combination of DAC 106 and amplifier, with the present case being 59, 10.24 and 10 respectively. The result of this gain control is that the maximum slope, found in any received wave packet, remains substantially constant for consecutively transmitted wave packets. The pre-estimated gain value Gn, indicated at point (12) in Figure 4, is a number between 0 and 225, which is fed to a DAC indicated at the point (20), to produce the gain control voltage 82, indicated in point (6) for the step of the gain control amplifier. At start-up, there is no previous maximum slope value from which to derive the new P1182 / 98MX gain. Therefore, a typical value is stored in the memory, the initial estimate from which one works. There are times when the required gain must change abruptly, for example, when the temperature of the fluid changes rapidly. A particularly difficult condition occurs when the required gain falls very suddenly and the estimate derived from the previous measurement is too large. Figure 6 shows seven examples of the effect of the gain increase in the digitized waveform, where the gain is increased by a factor of 1.5 from one graph to the next. In graph 1, the gain is too low, while graph 7 the gain is too high. The objective condition is similar to that shown in graph 3 or graph 4, with 9 or preferably 10 data points that fall between the upper and lower saturation values. This condition maximizes the adjustment sensitivity for the linear return exercise that follows the gain adjustment process. One consequence of this, and an advantageous feature of the preferred embodiment, is that the procedure for adjusting the gain is P1182 / 98MX extremely strong, being able to respond well to all conditions that are likely to be under real conditions. The measured slope is an increasing function of the gain, so that the gain adjustment process is able to adjust the gain in any direction. In particular, "it is able to quickly correct for a situation with an excessively high gain." An additional situation that should be avoided is that of zero gain, since the arithmetic of the processor 65 is fixed point, it can be truncated to zero a small gain value Because the new gain estimate is derived from the previous value by multiplication, the gain adjustment process can not recover from a zero gain situation, therefore, the process includes a verification from zero, where a non-zero value is substituted for the next gain if the calculated gain value is zero.It will be apparent from the method described above that it is not necessary to measure the maximum or minimum values of the received signal and therefore It is therefore not necessary to have access to that part of the waveform, this allows the electronic gain to be adjusted so that the maximums of the wave are by P1182 / 98MX above and below the voltage range of the instantaneous ADC 85. When operating in this state, the middle part of the waveform can be digitized with an improved amplitude resolution. This allows the accuracy of the entire process to determine when the signal has passed through the NRS state. Also, while the absolute position of the synchronization markers in the waveform has been determined, it is possible to use any of the equations when the signal passes through the NRS state as the centralization markers. Then you can take an average of the estimated arrival time of the signal based on a number of those markers. The average is advantageously weighted according to the slope of the waveform at that time, since the attitude of the time estimate when the signal passes through the NRS state is better than when the slope of the signal is higher. Using the developed method, it is possible to perform all the necessary processing of the data in an individual scan. This has the advantage of requiring less time and less power for its execution than for methods that require multiple passes of the data through the processor. The method also makes it possible to take into account the aging of transducers, or other processes that P11B2 / 98MX can change the waveform, by slowly updating the template against which the waveform is compared. This is possible because there is a large degree of difference allowed while still obtaining a correct determination of the absolute position in the waveform. In addition, by using a synchronized shift register to provide the transmission pulses, the edge jitter in the quartz clock pulses can be reduced and therefore an increase in synchronization accuracy can be obtained. In a second embodiment of the invention, the method and arrangement described above are combined with the method called "false triggering" described in International Patent Publication WO 93/00569, mentioned above. In the configuration of the measuring tube and transducers described in WO 93/00569, it is required to synchronize an ultrasonic pulse to an accuracy of 1.56 ns to achieve the required flow accuracy. If an individual zero-crossing was used for this synchronization, a clock running at 640 MHz will be required. The same accuracy can be achieved at much lower power consumption by synchronizing at 10 MHz and sending the signal down the tube 64 times. In this implemented scheme that uses the P1182 / 98MX teachings of the present invention, the received pulse can be examined each time to identify the crossing of the quiescent signal level 45 in which to retransmit the next pulse. A sudden ultrasonic increase and a determined delay that acts to inhibit the retransmission until the arrival of a nominated intersection of the ultrasonic pulse is digitally sampled. The transmission time of the ultrasonic increments is then more exactly terminated by transmitting a sequence of ultrasonic increments, each increment after the first that is transmitted at the reception of the immediately preceding increment. The delay is initialized at each transmission and retransmission occurring at a crossing of the quiescent signal level 45 of the received ultrasonic pulse only after it has inspired this delay. This method has the advantage that the delay at crossing the quiescent signal level 45 is determined with an individual ultrasonic increase so that no higher order acoustic modes are present from the previous transmissions. In addition, the method requires less energy to run a slower counter and transmit the ultrasonic signal several times. The identification of a specific crossover is best done from a synchronization delay. The acoustic modes that remain in the tube from the previous transmissions will modulate the P1182 / 98MX received ultrasonic signal and will cause its envelope to vary in shape. The retransmission junction is little affected by the residual modes because the signal energy is relatively high at that position. A crossing of level 45 of quiescent signal by the received ultrasonic increment is selected as the "retransmission junction". This crossing must be one that arrives before any of the significant secondary modes, but also arrives far enough in the waveform to make it little affected by the noise. A suitable junction for the measuring tube described here has been found to occur two and half cycles in the waveform. A measurement system 120 that realizes the advantages of this measurement scheme is mentioned in Figure 9, where components having similar structure and function similar to that of Figure 7 are provided, with the corresponding reference numbers, some of the other common components that are omitted for the purpose of clarity, which will be understood by those skilled in the art. During a scan, a transmission oscillator 127 is enabled at a quiescent signal level crossing 45 by the received signal 40. The oscillator 127 replaces the pulse generator 81 of Figure 7 and is asynchronous to P1182 / 98MX a scan synchronization counter clock 125, and is transferred to the 94-97 thrusters of the tri-state. The clock 125 of the scan synchronization counter is provided in a transmission controller 121, which replaces the transmission controller of FIG. 7, and comprises a 24-bit resettable counter used to measure the cumulative time it takes to send pulses from a transducer to another. The counter 125 is readjusted at the start of the first pulse transmitted in a scan and counting is disabled after receipt of the last pulse received in a scan. The logic control circuit 124 is provided in the controller 121 to integrate the functions of the transmission controller 121. The controller 121 also includes a load register 122 that receives a delay value from the microprocessor 65 via the common bus 79. That The value is then placed in a delay counter 123 which is used to synchronize the delay before the retransmission is enabled. This counter 123 is charged each time there is an ultrasonic transmission. The flow measurements are made specifically in the following stages. 1. The transmission address is selected using the address lines 98a and 98b.

P1182 / 98MX 2. The received gain is adjusted, under the control of the buffer 65, until the slope of the transmissions of the received ultrasonic elements is within an established specification. 3. The crossover detection delay is initialized. In this first step, the waveform is digitally sampled, and the time is determined from the ultrasonic transmission to the arrival at the crossing detector that has been selected to initiate a retransmission. A delay value less than this for half of a period of the ultrasonic increment is loaded into the delay counter 123 by the microprocessor 65. 4. At this stage of the measurement, the transit time of the ultrasonic pulses is determined from more exact way when transmitting a sequence of ultrasonic increments, each increment after the first that is transmitted in the reception of the immediately preceding increment. The generation, emission and detection of an individual ultrasonic increase will hereinafter be referred to as a "false trigger", sometimes referred to as "warning in P11B2 / 98MX False. "An individual exploration of the fluid flow consists of two sets of a predetermined number of false shots, first in one direction and then in the other, a sequence can include any number of false shots, but will advantageously include 64. The delay counter 123 and the synchronization counter 125 are started when the first ultrasonic increment is sent.The leading edge of this first ultrasonic increment is synchronized to the synchronization clock 70. The start of the second and subsequent ultrasonic increments is activated asynchronously by the arrival at the detector end of the measurement tube 61 of a specific junction corresponding to the preceding ultrasonic increase.The specific junction that activates the retransmission order is the first crossing that the quiescent signal level 45 of the required polarity (positive gear) or negative) to derive after the delay counter 123 has been synchronized The polarity required for a crossing to generate a retransmission depends on the polarity of the preceding transmission. A comparator 126 is provided to detect these junctions and includes inputs connected to the signal reference 86 and the bandpass filter 84. An output of the comparator 126 is connected to the transmission controller 121 to activate a P1182 / 98MX retransmission order. The delay is re-loaded in the delay counter 123 at the start of the next ultrasonic increment and immediately begins counting. The synchronization counter 125 is stopped by the arrival in the detector of the specific junction corresponding to the last ultrasonic increment. This specific junction is also identified by the delay counter 123. Upon completion of a series of false trips, the value in the high-speed synchronization counter 125 is transferred to the microprocessor 65. The second transmission direction (opposite) the previous sequence is selected and repeated.

The stored high-speed counter values 125 for both directions are used by the computer program in the microprocessor 65 to calculate the flow rate during the scan period. The time taken by the sequence as a whole can be divided by the number of increments transmitted to determine the average travel time. A sequence of false shots usually consists of groups of four shots. This sequence of shots is depicted in Figures 10A and 10B. the sequence of four is composed of three pulses and a plurality while the third is inverted with respect to the first three pulses. P1182 / 98MX To keep the polarity reversal away from important cases, the polarity change is achieved by adding an extra edge at the end of the transmission pulse sequence before the transmission has a different polarity. In a group of four shots, the signal is reversed after the third and is restored in the fourth shot. A false trigger sequence consists of, typically, 64 acoustic transmissions in each direction, but the first one is activated by the reception of the subsequent transmission. To assist in canceling the coherent acoustic scream buildup in the measurement tube as the false trigger sequence proceeds, the polarity of the transmitted signal is reversed on a regular basis throughout the sequence. The sequence is constituted imaginatively of groups of four transmissions, with the plurality of each fourth transmission inverted with respect to the other three. In this way, if the two possible polarities are designated "A" and "B", a sequence of 64 transmissions could have a polarity pattern of "AAABAAABAAAB ...", the pattern "AAAB" that is repeated 16 times in total . The polarity of the cross comparator 126 in the received signal is reversed just after a polarity transmission "B" and changed back again just after the next P1182 / 98MX first transmission of polarity "A" to ensure that it matches the plurality of received signal. In addition, to cancel the errors that could be introduced due to misalignment in the comparator 126, the full polarity of the false trigger sequence is reversed for each full gas velocity measurement. In this way, the above polarity sequence becomes "BBBABBBABBBA ..", again with the appropriate inversions of the plurality of the cross comparator 126 in each fourth transmission. A typical timing diagram for this measurement sequence is shown in Figures HA and 11B. An additional mode of measurement comprises a measurement system 140 depicted schematically in Figure 12 and in which the transmission of the ultrasonic signal occurs simultaneously in both directions with respect to the direction of fluid flow. Again, similar reference numbers are used for similar components that have similar functions, some components that have been omitted for the sake of clarity. This mode is suitable for rapidly changing flows as occurs with maximum flow meters. The transmission occurs from both transducers 62 and 63 at the same time, then a time P1182 / 98MX suitable posterior (advantageously about 500 μs in natural gas) both transducers 62 and 63 are used as receivers for the signal transmitted by the other transducer. Two separate sets of electronic reception components are required, which correspond to those of Figures 7 and 9, but the components are identified respectively using the suffixes A and B. It is extremely difficult to make the two sets of electronic reception products sufficiently identical in order to avoid the introduction of significantly different phase delays. These phase delays vary with conditions such as temperature and gain. To cancel this introduced phase difference, a filtered and attenuated version of the transmission signal is connected to both receivers. This is achieved by passing the outputs of the transmission pulse generator 81 first through a filter 141 and then to an attenuator 142 controlled by an output 143 of the control signal of the microprocessor 65. The variable attenuator 142 is transferred to an attenuator fixed 144 that supplies two analog multiplexers 92A and 93B, which are similar in construction to the multiplexers 92 and 93 of the previous modalities, accepting that they are each of a double pole construction, double shot, instead of one P1182 / 98MX single pole construction, double shot, as previously used. This signal is processed by both sets of electronic reception products and the memory 77 is stored. Due to the need to store the received data sets, the memory 77 is modified to simultaneously store double bytes (16 bits) of data and this way has a preferred total capacity of 64 kbitios. The signal is filtered so that its transmission time is greater than eight cycles of the high frequency clock. This allows each instantaneous ADC 85A, 85B to obtain enough samples for interpolation. The difference in phase delay can be calculated from this stored signal and used to calculate the corrected time difference for the upstream and downstream directions. The actual measurement procedure for this mode can then follow the following sequence: 1. Tri- State 94-97 drivers or activators are enabled to the up and down transducers 62, 63. Both receivers A and B are connected to a signal transmission that has been filtered, attenuated and adjusted in amplitude. Both sets of electronic reception products are enabled and a first block of memory 77 is selected.

P1182 / 98MX 2. The address counter 76 is enabled and the transmitted signal is transmitted in both directions. Data, typically, corresponding to only a few cycles, from both sets of receiving electronic products are stored in the first block of memory 77. These data allow the calculation of phase lag discussed above. 3. The electronic reception products are disabled and a second much larger second block of the memory 77 is selected. The multiplexers 92A, 93B are changed so that the transmitters are connected to the electronic reception products. 4. To conserve energy, electronic reception products are disabled just before the ultrasonic pulses are at the top. 5. Both sets of electronic reception products are enabled and the data of both sets of electronic reception products are stored in the second block of the memory 77. In this scheme, during the transmission sequence, a very small version is coupled in the signal P1182 / 98MX on the two preamplifiers. The amplitude of this signal is variable to accommodate different gains in the reception amplifiers. This transmission signal passes through the chain of amplifiers and instant ADCs and is stored in memory. The multiplexers are then switched to receive the signal from the transducers and a higher order memory address line is changed, not counting. The lower bits of the address counter 76 continue to be counted. The electronic products of reception (preamplifier, variable gain amplifier, filter, instant ADC in memory) are all disabled until this moment to receive the ultrasonic impulse. The data in the first block of memory 77 is then used to determine the current phase delay in the manner described above. The same process is then applied to the data in the second block of the memory 77. The phase difference for each direction is then determined. Actual propagation times are then determined using that data from the second block of memory 77, and adjusted where appropriate by the calculated phase difference and then retained for the average. In each of the embodiments, once the microprocessor 65 determines the flow rate P1182 / 98MX average, it is evaluated and can then be used to calculate the flow volume that can be increasingly retained. The flow rate and the increasing volume over a period of time can then be displayed by the LCD 66. This information can also be communicated via the serial interface 67 to a data acquisition system. Typically, flow rate calculations can be enabled at periodic intervals, generally between 0.5 seconds and 60 seconds. Preferably, the interval is about 2 seconds. In this way, it will be apparent from the above embodiments that the general method depends on the conversion of the analog wave packet signal to the digital information, and the subsequent use of this digital information, by comparison with a normal digital template of the characteristics of the selected waveform, both to identify only a number of particular cycles of the waveform and to combine this information to give a measurement of the elapsed time of the significantly reduced uncertainty for the arrival of the received wave packet. In addition, a preferred implementation allows the responses of the electronic products of the system to be measured to the signal state not received (NRS). This is achieved by averaging the digitized data taken before P1182 / 98MX of the arrival of the signal. The preferred implementation also allows the slope of the voltage waveform as a function of time to be determined by a technique of fitting the data pairs of voltage and time to a straight line using the least squares technique. The value of the maximum slope allows the amplitude of the signal to be estimated and therefore, by adjusting the gain of the amplifiers, for the amplitude of the received signal to be kept constant within narrow limits. Using the slopes of this signal of essentially constant amplitude, it is possible to determine the absolute position in the waveform of any characteristic with reference to a stored template. The characteristic used as a synchronization marker in the synchronization of signal transit down the tube is a particular occasion of the various occasions when the waveform passes through the NRS state. The exact time that the waveform passes through the NRS state is determined from the function fitted to the data pairs mentioned above. This allows the synchronization accuracy to be improved by effectively interpolating between the clock pulses, thereby overcoming the quantification that would otherwise occur.

P1182 / 98MX would be present. The preferred implementation of the present invention confers certain advantages in its use as an ultrasonic signal synchronization technique. First, the preferred method is less affected by the temperature of other methods because it uses the guide part of the wave pack. This guide part of the wave pack consists mainly of the wave in the plane and this is much less affected by the temperature of the upper order modes that form the back of the wave packet. The preferred method also makes better use of a quantization of the slope of the waveform since it makes a transition from a negative maximum to a positive maximum or vice versa. An array of slopes is used to describe the waveform in a simple and concise way. This also gives the user the ability to compare waveforms for a minimum of data. The transition slope also confers the ability to identify the position of the transmission within the waveform and thus calculate the position of the beginning of the waveform. Each transition slope can also be used to infer the amplitude of the maximum companion and can be used as an alternative parameter to the maximum value, and in this way the transition slope provides a parameter P1182 / 98MX useful for adjusting the gain in the system. In addition, the selection of only a portion of the amplitude of the waveform can be judiciously adjusted to increase the accuracy obtainable from the method. In the preferred embodiment, an increase in accuracy corresponds to approximately 1.6 bits is obtained by operating the analog-to-digital converter on a window covering about half a third of the received signal. The data describing the transitions in this way is defined more precisely and therefore it is possible to obtain more resolution in the synchronization. This is possible because the preferred method does not use the maximum values of the envelope for any measurement purpose and therefore it is possible to ignore them and use the transition slope only. The preferred method also confers the ability to accommodate any of the changes in the ultrasonic signal that could accompany the aging of, or any damage to, a transducer assembly, by slowly updating the template against which the waveform is compared. This is possible because the method allows a large degree of difference in the size and shape of the waveform while still obtaining a complete determination of the absolute position of the synchronization marker within it.

P1182 / 98MX In addition, by using a number of transition slopes, the preferred method makes more use of a greater percentage of the information in the waveform, making it more versatile and increasing the accuracy with which the synchronization measurement is made. Using a specific implementation, it is possible to process all the data in an individual scan. This requires less time and energy, and in this way results in longer unattended battery life for any system that uses it, than methods that require data to pass through the processor more than once. The foregoing describes only a number of embodiments of the present invention and modifications, obvious to those skilled in the art, can be made to it without departing from the scope of the present invention.

P1182 / 98MX

Claims (42)

1. NOVELTY OF THE INVENTION Having described the present invention, it is considered as a novelty and, therefore, the content of the following is claimed as property.
2. CLAIMS: 1. A method for detecting the arrival time in a transducer of a packet of acoustic waves, transmitted to a first frequency, the method comprising the steps of: (a) converting, at a sampling frequency, an output of transducer analog signal in digital signal data; (b) determining a measurement portion of the digital signal data, the measurement portion corresponds to a response of the transducer to the acoustic wave packet; (c) determining from the measurement portion a plurality of slopes (SJ for a corresponding plurality (i) of response measurement segments, each of the segments is approximately a signal level value (V); and (d) determining from at least one of the selected slopes (Si), the arrival time (t) of the
3. P1182 / 9TMX package of acoustic waves. A method according to claim 1, comprising the additional step, between steps (c) and (d) of: (ca) determining, for one of the selected segments, a corresponding measurement time (AJ indicative of a time) in which the response intercepts the signal level value (V), wherein step (d) comprises determining the arrival time (t) from the corresponding measurement times (Ai) 3. A method according to claim 2, comprising the additional step, between steps (c) and (ca) of: (ca) matching the slopes (SJ with a corresponding reference set of slopes (PJ to determine a measurement position of each of the segments in the measurement portion, wherein the step (ca) comprises determining the corresponding measurement times (Ai) from one of the corresponding measuring positions 4. A method according to claim 2, wherein the step (c) ) comprises assigning a set of (k) point s of the digital signal data to each measurement segment
4. (i), each measurement segment (i) is centered around two adjacent points whose values are on either side of the signal level value (V), fit to a straight line
5. P1182 / 98MX each set of points, and then determine the slope (Si) of each straight line. A method according to claim 4, wherein the straight line is a line joining the two adjacent points
6. 6. A method according to claim 4 or 5, wherein the value of (k) is related to the frequency relationship sampling at the first frequency, and the consequent separation or spacing of the digital signal data in the measurement portion.
7. 7. A method according to claim 4, 5 or 6, wherein k is equal to one eighth of the number of points per cycle.
8. A method according to claim 7, wherein the value of k is 10.
9. 9. A method according to claim 3, wherein each slope (Pi) of the slope reference set is associated with a specific position parameter (n) ) which marks the position of a signal polarity transition within the wave packet, parameter (n) corresponds to the number of half wave periods between the beginning of the wave packet and the transition of the particular signal polarity.
10. 10. A method according to claim 9, wherein the set of slopes is updated during the time
11. P1182 / 98MX to compensate for changes in the analog signal. A method according to claim 10, wherein the slope reference set is updated to follow a slow moving average of the slopes (Si).
12. 12. A method according to claim 9, wherein step (d) comprises the following sub-steps: (da) assign a weighting factor (Wi) to each of the measurement times (Ai) based on the position parameter corresponding (n); (db) select (m) of the measurement times (Ai) as those to be used in the determination of the arrival time (t); (de) determining a half average wave period (t) from time intervals between one of the selected (m) measuring times (Ai), adjacent; (dd) determining, for each of the selected (m) measurement times (Ai), an estimated arrival time (ti) of the wave packet in the transducer, (of) determining a sum, during the (m) times of measurement (Ai), of a product of the weighting factor (Wi) assigned to each measurement time and the corresponding estimated arrival time (ti); and (df) determining the arrival time (t) of the acoustic wave packet by dividing the step sum (of)

    P1182 / 98MX between a sum of the (m) weighting factors (Wi).
13. 13. A method according to claim 12, wherein step (dd) comprises determining the estimated arrival time (ti) using the expression ti = Ai-n.t.
14. A method according to claim 12 or 13, wherein each weighting factor (Wi) is to provide the slope (Si) of one of the corresponding segments.
15. A method according to claim 12, 13 or 14, wherein m has a value from 2 to 20.
16. 16. A method according to claim 15, wherein the value of m is 6.
17. 17. A method according to any of claims 12. to 16, wherein the step (db) comprises selecting the measurement times (A) dependent on at least one of the precision of measuring time determination, the closeness of attachment to the reference set of slopes (PJ, and the freedom of the interference of the measurement portion from higher order acoustic modes
18. 18. A method according to claim 1, wherein the sampling frequency is used to generate the acoustic wave packet and preferably the first frequency is a sub-frequency. multiple of the sampling frequency
19. 19. A method according to claim 1, wherein the signal level value (V) is derived from the

    P1182 / 98MX average of the digital signal data received in advance of the measurement portion.
20. 20. A method according to claim 1, wherein step (a) comprises the continuous conversion of the signal.
21. 21. A method according to claim 1, wherein a maximum magnitude of one of the slopes (Si) provides an estimate of a maximum amplitude of the analog signal, the estimate is used to maintain the maximum amplitude substantially constant.
22. 22. A method according to claim 1, wherein step (a) comprises, before conversion, the step of: (aa) adjusting the amplitude of the analog signal such that a maximum amplitude in the analog signal remains substantially constant over a number of acoustic wave packets received.
23. A method according to claim 22, further comprising the use of the magnitude of one of the maximum slopes (Si) to adjust the amplitude to maintain the maximum amplitude substantially constant and, thus, the maximum slope within a range default of magnitudes.
24. 24. A method for determining the propagation time and a packet of acoustic waves transmitted between two ultrasonic transducers, the method comprising the steps of:

    P1182 / 98MX (e) start a clock that operates at a predetermined frequency; (f) simultaneously with a clock transition, the steps of: (fa) energizing one of the transducers to extract the acoustic wave packet at a first frequency lower than the predetermined frequency; and (fb) keeping the clock as a synchronization reference for the subsequent detection of a wave packet arrival in the other of the transducers; (g) detecting the arrival time of the wave packet using the method as claimed in any of claims 1 to 23, wherein the sampling frequency is the predetermined frequency and is provided by the clock, and the digital signal data which include the measurement portion are stored in a memory whose address determination for consecutive samples of the digital signal data are altered by the array, such that each address in the memory corresponds to an identifiable time after generation of the packet acoustic waves.
25. 25. A method according to claim 24, wherein a plurality of acoustic wave packets are

    P1182 / 98MX transmit and, with the exception of the first transmission, the transmission of all subsequent acoustic wave packets is synchronized to the reception of the immediately preceding acoustic wave packet.
26. 26. A method according to claim 25, wherein in the transmission of a particular acoustic wave packet, a delay time is counted and the transmission of the next acoustic wave packet is instituted after the delay time has elapsed and when the measurement portion corresponding to the particular acoustic wave packet then crosses the signal level value.
27. 27. A method according to claim 24, wherein the acoustic wave packet is transmitted simultaneously from each of the transducers for the reception of the transducers by the other and the propagation time of each of the acoustic wave packets is determined in this way.
28. 28. A method according to claim 27, wherein step (a) further comprises compensating for phase differences between the digital signal data converted from analog signal outputs of one of the respective transducers.
29. 29. A method to estimate the arrival time in a transducer of a transmitted acoustic wave packet

    P1182 / 98MX at a first frequency, the method comprising the steps of: (a) converting an analog signal output of the transducer into digital signal data; (b) determining from the digital signal data a waveform period for different segments of the analog signal; (c) averaging the periods of the waveform, and

    (d) use the average and number of segments to estimate the arrival time.
30. 30. A method according to claim 29, wherein step (b) comprises determining a plurality of slopes corresponding to the transitions of the analog signal around a predetermined signal level, and using the slopes to identify a period of time of wave between one of the adjacent slopes.
31. 31. A method according to claim 30, wherein the slope plurality is not compared with a set of reference slopes to identify a specific location in the analog signal, from which, using the average period, it can determine the time of arrival.
32. 32. A method according to claim 31, wherein the set of reference slopes are modified over time to follow the changes in the plurality

    P1182 / 98MX of slopes to compensate in this way the leads from the first frequency of the acoustic wave packet as detected by the transducer over time.
33. 33. A system for measuring the velocity of a fluid in motion, the system comprising: a measuring tube through which the fluid whose velocity is to be measured passes; two ultrasonic transducers placed in the tube and configured to transmit ultrasonic signals therebetween; a transmission means for driving at least one of the transducers to generate a packet of acoustic waves propagating at a first frequency along the tube to the other of the transducers; a receiving means connected to at least the other transducer for detecting the acoustic wave packet and for converting an analog signal output from at least the other transducer to digital signal data, the receiving means including a (first) memory for storing the digital signal data, a processor means connected to the receiving means for determining the speed of the fluid, the processor means comprises: a first means for determining a portion of

    P1182 / 98MX measurement of the digital signal data, the measurement portion corresponding to the response of the other transducer of the acoustic wave packet; a second means for determining from the measurement portion a plurality of slopes (SJ for a corresponding plurality (i) of response measuring segment, each of the segments being approximately a signal level value (V) a third means for determining, from at least one of the selected slopes (Si), the arrival time (t) of the acoustic wave packet
34. 34. A system according to claim 33, further comprising a fourth means for determining , for one of the selected segments, a corresponding measurement time (A) indicative of a time in which the response intercepts the signal level value, where the third means determines the arrival time (t) from the corresponding measuring times (AJ
35. 35. A system according to claim 34, further comprising a (second) memory medium connected to the processor means for storing a reference signal, the processed medium r including a fifth means arranged to equalize the parts of the reference signal with the measurement portion to determine a measurement position of each of the segments in the measurement position, in

    P1182 / 9BMX where the measurement times (Ai) are determined using one of the corresponding measurement positions.
36. 36. A system according to claim 33, comprising means for adjusting the magnitude of the analog signal from at least the other transducer, so that it falls within certain limits, these limits that also preferably displace any side of the signal level value, the signal level value that is preferably an average of one output of the other transducer before the measurement portion .
37. 37. A system according to claim 33, further comprising means for altering the amplitude of the analog signal such that a maximum amplitude of the analog signal is maintained substantially continuously for a number of received acoustic wave packets.
38. 38. A system according to claim 37, wherein the alteration means uses the magnitude of the maximum slope (Si) to adjust the amplitude to maintain the maximum amplitude substantially constant and thus the maximum slope within a predetermined range of magnitudes .
39. 39. The invention according to claim 1 or 33, wherein the sampling frequency is between 500 kHz and 10 GHz. The invention according to claim 39 in
40. P1182 / 98MX where the sampling frequency is 10 MHz.
41. 41. The invention according to claim 39 or 40, wherein the first frequency is between 20 kHz and 5 MHz.
42. 42. The invention according to claim 41 when it is dependent on the claim 40, where the first frequency is 125 kHz.

    P1182 / 98MX