MXPA01007520A - Photodiode light sensor - Google Patents

Abstract

Light sensors having a wide dynamic range are used in a variety of applications. A wide dynamic range light sensor (48, 52) includes an exposed photodiode light transducer (106) accumulating charge in proportion to light (104) incident over an integration period (158). Sensor logic (306) determines a light integration period (158) prior to the beginning of integration and the charge is reset. Charge accumulated by the exposed light transducer (106) over the light integration period (158) is measured and a pulse (122) having a width (170) based on the accumulated charge is determined.

Description

LIGHT SENSOR OF FOTODIODO TECHNICAL FIELD The present invention relates to light sensors that incorporate a charge integration photodiode as a light transducer.

TECHNICAL BACKGROUND A light sensor generates a light signal that indicates the intensity of the light that falls on the light sensor. The light sensor includes a light transducer for converting light into an electrical signal and may also include electronic circuits for conditioning signals, compensation for cross-sensitivities such as temperature, and signal formatting. Light sensors are used in a wide range of applications including remote sensing, communications and controls. An application for light sensors is in rear-view mirrors of automatically dimmable vehicles. Vehicle operators use interior and exterior rear view mirrors to see scenes behind the vehicle without having Ref: 132092 to flip in a backward direction and to see areas around the vehicle that would otherwise be blocked by vehicle structures. In this way, the rear view mirrors are an important source of information for the operator of the vehicle. Bright lights that appear in a scene behind the vehicle, such as those of another vehicle approaching from behind, can create reflections in a rearview mirror that can temporarily deteriorate or dazzle the operator's vision. This problem is generally worsened during low light conditions such as at night, when the operator's eyes have adjusted to the dark. Automatic dimming mirrors eliminate the need for the operator to manually adjust the mirror. The first designs used a single reflex sensor that looked back to detect the level of light that hits the mirror. This design proved inadequate since the threshold perceived by the operator to attenuate the mirror, known as the reflex threshold, varied as a function of the ambient light level. An improvement included a second light sensor to detect the level of ambient light. The threshold of reflection in these systems is based on the amount of ambient light detected.

Among the proposed double sensor designs include those described in the patents of E.U.A. Nos. 3,601,614 to Platzer; 3,746,430 to Brean et al.; 4,580,875 to Bechtel et al .; 4,793,690 to Gahan et al.; 4,886,960 to Molyneux et al.; 4,917,477 to Bechtel et al.; 5,204,778 from Bechtel; 5,451,822 to Bechtel et al. and 5,715,093 to Schierbeek et al, each of which is incorporated herein by reference. A key element in the design of an automatic dimming mirror is the type of light transducer used to implement detection of ambient light and reflections. A feature of primary interest in the selection of a type of light transducer is the dynamic range. The relationship between the intensity of bright sunlight and lunar light is approximately 1,000,000: 1, indicating the wide range that must be detected by the ambient light sensor. Both the ambient light and reflective light sensors must operate within temperature, humidity, shock and vibration scales experienced within a vehicle passenger compartment. If a sensor is to be mounted on an external mirror, even more aggressive operating conditions can be expected. Sensors and electronic support circuits must not be costly to allow the cost of an attenuated mirror to automatically be within the range considered acceptable by a car buyer. Transducers must have adequate noise immunity or be compatible with electronic noise compensation circuits inside the sensor for sensitivity to low light levels. The transducers must also have a spectral response similar to the frequency response of the human eye. As a final desirable feature, the sensor must be easily integrated into the types of digital control systems commonly found in automotive applications. The photodiode light sensors incorporate a silicon-based photodiode and electronic conditioning circuits on a single substrate. The photodiode generates charge at a speed proportional to the incident light load. This charge induced by light is collected during a period of integration. The resulting potential indicates the level of light to which the light sensor is exposed during the integration period. Light sensors with integral load collection have many advantages. By varying the integration time, the dynamic range of the sensor is widely extended. Also, the ability to incorporate additional electronic circuits on the same substrate as the photodiode increases the noise immunity and allows the sensor output to be formatted to be used by a digital circuit. The - integration of the components further reduces the cost of the system. The silicon light sensors are relatively invariant in temperature and can be packaged to provide necessary protection against moisture, shock and vibration. One disadvantage of silicon-based light transducers is that they have a different frequency response than the human eye. A variety of charge integration photodiode devices have been described, including those of the U.S.A. Nos. 4,916,307 to Nis ibe et al.; 5,214,274 to Yang; 5,243,215 to Enomoto et al.; 5,338,691 to Enomoto. et al. and 5,789,737 a Street, each of which is incorporated herein by reference. One difficulty with all types of light sensors is the occurrence of operational anomalies at high temperatures. Some devices become extremely non-linear at high temperatures. Some devices may suffer a permanent change in the operating characteristics. The devices can even provide completely false readings, such as indicating bright light in low light conditions due to excessive thermal noise. Traditionally, the only way to handle this problem has been to incorporate a temperature sensor and associated electronic circuits in systems that use light sensors. What is needed is a light sensor with a wide dynamic range that can be incorporated in cost sensitive digital systems such as automatically dimmable mirrors. The light sensor must compensate for cross-sensitivity at temperatures and, preferably, provide an indication of the temperature of the light sensor. A load integration light sensor having an externally determined integration period is also desirable.

DESCRIPTION OF THE INVENTION An object of the present invention is to provide a charge integration light sensor with a wide dynamic range. Another object of the present invention is to provide a packaged light sensor that is economical to produce.

A further object of the present invention is to provide a charge integration light sensor that will be easily interconnected to digital electronic circuits. Another object of the present invention is to provide a charge integration light sensor with an output signal indicating the intensity of the incident light and the temperature of the sensor. A further object of the present invention is to provide a load integration light sensor with an externally determined integration period. To carry out the above objects and other objects and features of the present invention, a light sensor is provided. The light sensor includes an exposed photodiode light transducer that accumulates charge in proportion to the light incident during an integration period. A sensor logic circuit determines the period of light integration before the start of integration. The accumulated charge in the light transducer exposed at the beginning of the light integration period is restored. The load accumulated by the exposed light transducer during the integration period is measured and an impulse having a width based on the accumulated load is determined.

In one embodiment of the present invention, the light sensor includes a comparator with one input connected to the exposed light transducer and the other input connected to a switched capacitor circuit. The switched capacitor circuit charges a capacitor to a fixed voltage when the switch is closed and discharges the capacitor at a constant speed when the switch is open. The sensor logic circuit closes the switch during the light integration period and opens the switch after the light integration period, thereby creating the pulse at the comparator output. In a refinement, the light sensor further includes a second comparator with an input connected to a fixed voltage and the other input connected to the switched capacitor circuit. The output of the second comparator inhibits the determined impulse output if the ramp voltage is less than the fixed voltage. In another embodiment of the present invention, the light sensor includes a photodiode light transducer protected from light. The protected light transducer accumulates charge in proportion to noise during the integration period. The sensor logic circuit restores the accumulated load in the protected light transducer at the beginning of the determined light integration period. The charge accumulated by the protected light transducer during the determined light integration period is measured and an output pulse having a width based on the difference between the load of the exposed light transducer and the load of the protected light transducer is determined. . In still another embodiment of the present invention, the light sensor has an input to receive an integration signal. Since the noise depends on the temperature of the light sensor, the output pulse can be used to indicate the temperature of the sensor. The output pulse is sent after the end of the received integration signal after a length of time based on the noise level. In other embodiments of the present invention, the period of light integration can be determined from the asserted portion of a control signal received by the sensor logic circuit or can be determined in the sensor control by cyclizing through a sequence of periods. of predetermined time. In a further embodiment of the present invention, the light sensor includes at least one additional exposed photodiode light transducer. Each additional exposed light transducer accumulates charge in proportion to light incident during an integration period at a different speed than the speed of any other exposed light transducer. The sensor logic circuit emits a pulse that has a width based on the accumulated load for each of the additional exposed light transducers. In a refining, each exposed light transducer has a different collector area. In another refinement, each exposed light transducer has an opening with a different light intake area. A package of light sensors is also provided.

The package includes a housing that has a window to receive light. The housing supports a power terminal, a ground terminal and an output terminal. Within the housing, an exposed photodiode light transducer accumulates charge in proportion to the light received through the window that 'impacts during the integration period. A light-to-voltage circuit emits a light voltage signal based on charge accumulated by the exposed light transducer. An impulse voltage circuit emits a pulse on the output terminal. The pulse width is based on the light voltage signal. A light sensor with a photodiode that overlaps a substrate is also provided. The photodiode accumulates charge generated by incident light in the photodiode in a photodiode cavity formed in a region of the substrate that underlies the photodiode. The photodiode has an intrinsic photodiode capacitance. A floating diffusion having an intrinsic floating diffusion capacitance is also formed in the substrate. The floating diffusion has a diffusion cavity formed in a region of the substrate that underlies the floating diffusion when the load is restored. The floating diffusion removes charge in the diffusion cavity when the load is restored. The load of the floating diffusion determines an output potential. A transmission aperture having an intrinsic transmission aperture capacitance is placed between the photodiode and the floating diffusion. The transmission aperture forms a transmission cavity in a region of the substrate between the region of the substrate underlying the photodiode and the region of the substrate underlying the floating diffusion. The transmission cavity has a depth less than the depth of the photodiode cavity and the diffusion cavity. When the load is restored, the charge in the cavity of the photodiode on the depth of the transmission cavity flows through the transmission cavity, through the floating diffusion and is eliminated. During a period of light integration, the charge produced by incident light on the photodiode flows through the transmission cavity and into the diffusion cavity, producing output voltage inversely proportional to the capacitance of the floating diffusion. Once the diffusion cavity is filled to the depth of the transmission cavity, the charge produced by light incident on the photodiode fills the photodiode cavity, the transmission cavity and the diffusion cavity, producing an inversely proportional output voltage. to the sum of the capacitance of the floating diffusion, the capacitance of the photodiode and the capacitance of the transmission aperture. This double capacitance provides a first sensitivity during charge accumulation in the diffusion cavity only and a second sensitivity during charge accumulation in the diffusion cavity, the transmission cavity and the photodiode cavity. The first sensitivity is greater than the second sensitivity. In one embodiment, the light sensor includes an anti-reflex aperture between the phodiode and a feed voltage diffusion. The anti-reflection aperture defines an anti-reflection cavity formed in a region of the substrate between the region of the substrate underlying the photodiode and the supply voltage diffusion. The anti-reflex cavity has a depth less than the depth of the transmission cavity. The above objects and other objects, features and advantages of the present invention are readily apparent from the following detailed description of the best way to carry out the invention when taken in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS Figure 1 is a drawing illustrating vehicle rear view mirrors that can incorporate the present invention; Figure 2 is a block diagram of an embodiment of the present invention; Figure 3 is a timing diagram illustrating integration control and sensor emission for a light sensor that can be used to implement the present invention; Figure 4 is a schematic diagram of circuitry that allows the logical attenuation circuit and a light sensor to be interconnected by a single cable carrying both integration control and sensor emission; Figure 5 is a timing diagram illustrating the operation of the circuitry of Figure 4; Figure 6 is a block diagram of a rear view mirror system with interior and exterior rear view mirrors according to embodiments of the present invention; Figure 7 is a schematic diagram illustrating one mode of the attenuation logic circuit; Figure 8 is a schematic diagram illustrating the operation of the transmission control of the electrochromic element; Figure 9 is a timing diagram illustrating transmission control of the electrochromic element; Fig. 10 is a graph indicating the attenuating reflectance as a function of the duty cycle of the attenuating control signal; Fig. 11 is a flow diagram illustrating the operation of the attenuation logic circuit according to an embodiment of the present invention; Figure 12 is a graph illustrating the binary logarithmic approximation implemented in a modality of the attenuation logic circuit; Fig. 13 is a schematic diagram illustrating the operation of a light sensor having a pulse output according to an embodiment of the present invention; Fig. 14 is a timing diagram illustrating the operation of the light sensor of Fig. 13; Fig. 15 is a schematic diagram illustrating the operation of a light sensor with noise compensation according to an embodiment of the present invention; Fig. 16 is a timing diagram illustrating the operation of the light sensor of Fig. 15; Fig. 17 is a schematic diagram of an implementation of the light sensor of Fig. 15, using photodiodes as light transducers; Figures 18-21 are block diagrams of various modalities for packaging, emission and control of the light sensor; Fig. 22 is a block diagram of a sensor logic circuit for internally determining the signal of the integration period; Figure 23 is a block diagram illustrating the use of light transducers having different effective areas to achieve an increased dynamic range, according to one embodiment of the present invention; Fig. 24 is a block diagram illustrating the use of light transducers having different apertures to achieve an increased dynamic range, according to one embodiment of the present invention; Fig. 25 is a schematic diagram illustrating different transducer capacitances for different amounts of light induced charge to achieve an increased dynamic range, according to one embodiment of the present invention; Figure 26 is a graph of the output potential as a function of the cumulative incident light for the transducer of Figure 25; Fig. 27 is a schematic diagram illustrating a photodiode transducer incorporating an anti-reflex aperture, according to an embodiment of the present invention; Fig. 28 is a drawing illustrating a housing for a light sensor, according to an embodiment of the present invention; Fig. 29 is a graph illustrating a field of view of the light sensor as a function of the distance of the light transducer from the lens; Figure 30 is a graph illustrating the optical gain of the light sensor as a function of the distance of the light transducer from the lens; Figure 31 is a graph illustrating that illustrates the frequency response of the human eye; Figure 32 is a graph illustrating the frequency response of a typical light transducer; Figure 33 is a drawing of a cavity incorporating an infrared filter, according to one embodiment of the present invention.

DETAILED DESCRIPTION OF THE DRAWINGS Referring now to Figure 1, there is shown a drawing that illustrates vehicle rearview mirrors that may incorporate the present invention. The vehicle 20 is driven by the operator 22. The operator 22 uses the interior rearview mirror 24 and one or more exterior rear view mirrors 26 to view a scene back, shown generally with the number 28. Much of the time, the operator 22 goes towards forward through the windshield 30. The eyes of the operator 22 are therefore adjusted to the ambient light 32 coming from a generally forward direction. A relatively bright light source in the back scene 28 can produce light that can be reflected from the mirrors 24, 26 and temporarily affect, distract or visually dazzle the operator 22. This relatively strong light is referred to as a reflection 34. To reduce the impact of the reflection 34 in the operator 22, the reflectance of the mirrors 24, 26 can be reduced. Before automatically attenuating the mirrors, the rearview mirror 24 could contain a prismatic reflector element that could be controlled manually by the operator 22. The automatically attenuating mirrors include a sensor for reflections 34 and, typically, for ambient light 32, and attenuate one or more mirrors 24, 26 in response to the reflex level 34. Referring now to Figure 2, a block diagram of one embodiment of the present invention is shown. An attenuation element, generally shown at 40, includes the variable transmission element 42 and the reflective surface 44. The attenuation element 40 is positioned such that the reflective surface 44 is seen through the variable transmission element 42. Attenuation element 40 exhibits a variable light reflectance in response to the attenuation element control signal 46. The front ambient light sensor 48 is positioned to receive front ambient light 32 that generally comes from the front of the vehicle 20. ambient light sensor 48 produces a discrete ambient light signal 50 which indicates the amount of ambient light 32 incident on the ambient light sensor 48 during a period of integration of ambient light. The reflex sensor 52 is positioned to detect reflex 34 which generally comes from behind the vehicle 20 and can optionally be positioned to see the reflection 34 through the variable transmission element 42. The reflex sensor 52 produces a discrete reflex signal 54 which indicates the amount of reflection 34 incident on the reflex sensor 52 during a period of reflex integration. The attenuation logic circuit 56 receives the ambient light signal 50 and determines a level of ambient light. The attenuation logic circuit 56 determines "the reflex integration period based on the ambient light level 32. The attenuation logic circuit 56 receives the reflection signal 54 and determines the reflex level 34. The attenuation logic circuit 56 emits the control signal of the attenuation element 46, adjusting the reflectance of the attenuation element 40 to reduce the reflex effects 34 perceived by the operator 22. Either the reflex sensor 52, the ambient light sensor 48 or, preferably both They are semiconductor light sensors with integral charge collection, these sensors include light transducers that convert incident light into charge, this load is collected during an integration period to produce a potential that is converted by the sensor 48, 52 into an emission The designs suitable for this application are described in U.S. Patent No. 4,916,307 entitled "Intensity Detection Circuit. of Light with Dark Current Compensation "by Nishibe et al.; and patent of E.U.A. No. 5,214,274 entitled "Arrangement of Image Sensors with Threshold Voltage Detectors and Storage Capacitors Loaded" to Yang; each of which is incorporated herein by way of reference. Preferred modes of light sensors 48, 52 are described with respect to Figures 13-33 below. One difficulty with silicon-based sensors is the difference in spectral sensitivity between silicon and the human eye. The ambient light filter 58 can be placed before or incorporated into the ambient light sensor 50. Similarly, the reflex filter 60 can be placed before or incorporated within the reflex sensor 52. Filters 58, 60 attenuate certain portions of the spectrum that can include visible light, infrared and ultraviolet radiation so that the light hitting the sensors 48, 52 is combined with the frequency response of the light transducers within the sensors 48, 52 to more closely approximate the response of the human eye and to compensating for coloration in vehicle windows such as windshield 30. The use of filters 58, 60 to compensate for the spectral sensitivity of light transducers in sensors 48, 52 is described with reference to Figures 31-33 below. The variable transmission element 42 can be implemented using a variety of devices. The attenuation can be achieved mechanically as described in the US patent. No. 3,680,951, entitled "Photoelectrically Controlled Rearview Mirror" by Jordán et al.; and patent of E.U.A. No. 4,443,057, entitled "Automatic Rearview Mirror for Automotive Vehicles" to Bauer et al.; each of which is incorporated herein by way of reference. The variable transmission element 42 can be formed using liquid crystal cells such as those described in the U.S.A. No. 4,632,509, entitled "Reflective Reflector Type Reflector" to OIM et al. , which is incorporated herein by reference. Preferably, the variable transmission element 42 is an electrochromic cell that varies the transmission in response to an applied control voltage, such as that described in the U.S.A. No. 4,902,108, entitled "Single-Chamber, Self-Erasable and Solution-Phase Electrochromic Devices, Solutions for Use in Same, and Uses of Themselves" to Byker, which is incorporated herein by reference. Many other electrochromic devices may be used to implement the attenuation element 40, some of which are mentioned in the Background Art section of this application. As will be recognized by one of ordinary skill in the art, the present invention does not depend on the type or structure of the attenuation element 40. If the attenuation element 40 includes the electrochromic variable transmission element 42, the reflective surface 44 can be incorporated into the element. variable transmission 42 or may be external to the variable transmission element 42.

Each interior rearview mirror 24 and exterior rearview mirror 26 must include an attenuation element 40 for automatic attenuation. Preferably, the interior rearview mirror 24 also includes a control logic circuit 56, light sensors 48, 52 and, if used, filters 58 and 60. Various modes for controlling the exterior rear view mirrors 26 are described with reference to the figure 6 below. Referring now to Figure 3, a timing diagram illustrating control of integration and sensor emission for a light sensor that can be used to implement the present invention is shown. The load accumulation light sensors 48, 52 exhibit increased dynamic range through varying integration periods. A control signal, generally shown by 70, is used to specify the integration period. The resulting sensor output, generally shown as 72, includes an output pulse for each integration period. The total amount of charge induced by light, which can be effectively measured, is limited. Therefore, in the presence of bright light, a short integration time is desirable to avoid saturation. However, if a short integration time is used in low light conditions, the charging signal may be lost in noise inherent in the sensor 48, 52. The control signal 70 includes a sequence of integration periods having varying lengths. In the example shown in Figure 3, the short integration pulse 74 having a short integration period 76 is generated. A semiconductor light sensor can emit a short pulse in a completely dark environment due to the noise. Therefore, any sensor output pulse 72, such as a short signal pulse 78, having a duration of less than one threshold is ignored. Subsequently, the half integration pulse 80 having an average integration period 82 is generated. The resulting average signal pulse 84 has a duration that indicates the amount of light incident on the sensor 48, 52 during the average integration period. 82. The long integration pulse 86 having a long integration period 88 is generated. If the light sensor 48, 52 is bright enough, the result will be saturation. Therefore, the long signal pulse 90 having a duration greater than a threshold is also ignored. The control signal 70 can be generated outside the light sensor 48, 52, or it can be generated by the control logic circuit inside the light sensor 48, 52. if they are generated externally, the control signal 70 and the sensor output 72 they can share a common signal line or they can use separate signal lines. Various options and modalities are described with respect to Figures 4-28 below. Referring now to Figure 4, a schematic circuit diagram is shown which allows the control logic circuit and a light sensor to be interconnected by a single cable carrying both sensitivity control and sensor emission. The light sensor 48, 52 includes a housing 100 with window 102 that admits light 104 that impinges on the exposed light transducer 106. The housing 100 admits the power terminal 108, ground terminal 110 and signal terminal 112. The use only three terminals 108, 110, 112 greatly reduces the cost of the light sensor 48, 52. A three-terminal package that can be used to implement the light sensor 48, 52 is described with respect to Figure 28 below. The light sensor 48, 52 is connected to the attenuation logic circuit 56 via the interconnection signal 114 between the signal terminal 112 in the light sensor 48, 52 and the signal terminal 116 in the attenuation logic circuit 56. As will be described below, the signal terminals 112, 116 are three-state ports that allow the interconnection signal 114 to provide both an input to the light sensor 48, 52 and an output of the light sensor 48., 52. The attenuation logic circuit 56 may include FET Ql connected between the signal terminal 116 and ground. The FET Q1 is controlled by the control line 118 connected to the base of Q1. The capacitor 120 is also connected to the signal terminal 116. Within the light sensor 48, 52, the FET Q2 is connected between the signal terminal 112 and ground. The FET Q2 is controlled by the output pulse 122 connected to the opening of Q2. A constant current source 124 is connected to the signal terminal 112 in such a way that if neither Ql nor Q2 are turned on, the interconnection signal 114 is brought to a high level. The constant current source 124 nominally provides about 0.5 mA to bring the interconnection signal 114. The input of a Schmidt trigger inverter 126 is connected to the signal terminal 112. The Schmidt trigger inverter 126 is followed by the inverters 128. and 130 in series. The output of the inverter 130 synchronizes to the multivibrator D 132. The output of the multiplexer 134 is connected to the input D of the multivibrator 132. The selected input of the multiplexer 134 is conducted by the output pulse 122 such that when the output pulse is reached. 122 is asserted, the input D of the multivibrator 134 is not asserted, and when the output pulse 122 is not asserted the input D of the multivibrator 134 is asserted. The output of the NAND opening 136 is connected to the low assertive recloser 138 of the multivibrator 132. The output of the multivibrator 132 is the integration pulse 140. The integration pulse 140 and the output of the inverter 128 are inputs to the NAND opening 136. An impulse light circuit 142 accepts the integration pulse 140 and the output of the exposed light transducer 106, and produces the output pulse 122. Modes for the pulse light circuit 142 are described with reference to FIGS. 13-17 and 23-27 below. In a preferred embodiment, the light sensor 48, 52 includes a protected light transducer 144 that does not receive light 104. The impulse light circuit 142 uses the output of the protected light transducer 144 to reduce the effects of noise on the transducer of exposed light 106. Referring now to Figure 5, a timing diagram illustrating the operation of the circuitry of Figure 4 is shown. Initially, the low assertion interconnect signal 114 is high. The state of the multivibrator 132 must be zero so that, if the state is one, both inputs to the NAND opening 208 are high, asserting the recloser 138 and forcing the state of the multivibrator 132 to zero. At time 150, the attenuation logic circuit 56 affirms the control cable 118 by turning on the transistor Ql. The interconnection signal 114 is then brought down in time 152. The output of the inverter 130 changes from a low to a high setting by adjusting the state of the multivibrator 132 to one, which causes the integration pulse 140 to be asserted in time 154. The impulse light circuit 142 begins to integrate the light 104 incident on the exposed light transducer 106. At time 156, the control cable 118 is brought down by turning off the transistor Q1. The difference between the time 156 and the time 150 is the integration period 158 requested by the attenuation logic circuit 56. Since both Ql and Q2 are off, the interconnection signal 114 is brought high by the current source 124 in the time 160. Since the output of the inverter 128 and the integration pulse 140 are both high, the recloser 138 is asserted causing the state of the multivibrator 132 to change to zero and the integration pulse 140 not being asserted at time 162. This signals the impulse light circuit 142 to stop the integration of light 104 incident on the exposed light transducer 106. At time 164, the impulse light circuit 142 affirms the output pulse 122 to initiate the emission of information. of the intensity of light. The assertion of the output pulse 122 turns on the transistor Q2, bringing the interconnection signal 144 down at time 166. This causes the inverter 130 to transmit a transition transition from low to high to zero as the state of the multivibrator 132. The impulse light circuit 142 defies the output pulse 122. at time 168. The difference between time 168 and time 164 is the period of light intensity 170 which indicates the amount of light 104 incident on the exposed light transducer 106 during integration period 158. Transistor Q2 is off when the output pulse 122 goes down at time 168. Since both transistors Q1 and Q2 are turned off, the interconnection signal 114 is brought to a high at time 172. The compensator 120 in the attenuation logic circuit 56 it detects the transitions in the interconnection signal 114 at times 166 and 172. Unlike in time between the times 172 and 166 it is used by the attenuation logic circuit 56 to determine the intensity of the light 104 received by the light sensor 48, 52. If the protected light transducer 144 is included in the light sensor 48, 52, the difference in time between the deflection of the integration pulse 140 at time 162 and the assertion of the The output pulse 122 at time 164 is due, in part, to the thermal noise in the light sensor 48, 52. This difference is expressed as the thermal noise period 174. The thermal noise period 174 can be used by the logic circuit of attenuation 56 to determine the temperature of the light sensor 48, 52 or can be used more simply to determine if the noise level in the sensor 48, 52 is too high for a reliable reading. The attenuation logic circuit 56 can deactivate the automatic attenuation of the attenuation element 40 if the temperature of the light sensor 48, 52 exceeds a preset limit. The ability of the light sensor 48, 52 to use the output of the protected light transducer 144 to generate the output pulse 122 indicating the amount of thermal noise in the light sensor 48, 52 is described with respect to FIGS. 17 below. Referring now to Figure 6, a block diagram of a rear view mirror system with interior and exterior rear view mirrors according to embodiments of the present invention is shown. The attenuation element 40 in the interior rearview mirror 24 operates as described with reference to Figure 2 above. Each exterior rearview mirror 26 includes an exterior attenuating element 180 having an outer variable transmission element 182 which functions to attenuate the light coming from the rear scene 28 both before and after being reflected from the outer reflective surface 184. The element of External attenuation 180 provides variable reflectance based on the control signal of the external attenuation element 186. The external attenuation element 180 can operate in any manner described with respect to the attenuation element 40 and, preferably, is an electrochromic mirror. The outer mirror control 188 generates a control signal from the external damping element 186. The control of the outer mirror 188 can be part of the exterior rearview mirror 26, the interior rearview mirror 24, or it can be located outside of any mirror 24, 26. Various Modes for controlling the external attenuation element 180 depend on the amount of detection and control that will be included in the exterior rearview mirror 26. In one embodiment, the attenuation logic circuit 56 in the interior rear view mirror 24 determines the control signal of the external damping element 186 based on the emission from the ambient light sensor 48 and the reflex sensor 52. The control signal of the external damping element 186 can be generated directly by the attenuation logic circuit 56 or the external mirror control 188 can generate an external attenuation element control signal 186 based on a reflectance level calculated in the attenuation logic circuit 56 and transmitted to the external mirror control 188 through a signal between mirrors 190. In another embodiment, the exterior rearview mirror 26 includes the external reflex sensor 192 positioned to receive reflex 34 from the rear scene 28 and which is operative to emit the exterior reflex signal 194 based on the amount of reflection 34 incident on the reflex sensor 192 during a period of reflex integration. Since the light sensors 48, 52 with silicon-based charge accumulation transducers 106, 144 have a lower cross-sensitivity at lower temperatures, mounting the light sensors 48, 52 outside the body of the vehicle 20 is more practical than with other types of light transducers. The attenuation logic circuit 56 uses the outer reflection signal 194 and the ambient light signal 50 to determine the reflectance level for the external attenuation element 180. Again, the control signal of the external attenuation element 186 can be generated directly by the attenuation logic circuit 56 or can be developed by the external mirror control 188 based on the level of reflectance contained in the signal between mirrors 190. The exterior reflection filter 196, similar to the reflection filter 60, can be placed before of the exterior reflex sensor 192 or integrated into the exterior reflex sensor 192 to provide the exterior reflex sensor q92 with a response closer to the response of the human eye. The signal between mirrors 190 and outer reflection signal 194 may be in the form of a pulse width modulated signal, pulse density signal, serial data stream, or digitized and communicated on an automotive busbar such as the CAN bar.

In yet another embodiment, the external reflex sensor 192 produces an external reflex signal 198 which is routed directly to the external mirror control 188. The external mirror control 188 determines the control signal of the external attenuation element 186 based on the exterior reflection signal 198 and the ambient light level 32 determined by the attenuation logic circuit 56 and sent to the exterior mirror control 188 through the signal between mirrors 190. In another embodiment, the exterior rear view mirror 26 determines the reflectance for the external attenuation element 180 independent of the reflection 34 or ambient light 32 detected by the interior rearview mirror 24. In this embodiment, the exterior rearview mirror 26 operates as described above with respect to Figure 2. In reference Now to Figure 7, a schematic diagram illustrating one mode of the attenuation logic circuit is shown. The circuit represents an effective and inexpensive implementation for automatically attenuating the interior rearview mirror 24. The attenuation logic circuit 56 uses a small and inexpensive microcontroller, indicated by Ul, such as the PI15C620A from Microchip Technology, Inc. of Chandler, Arizona. The ambient light sensor 48 communicates with the microcontroller Ul via the interconnection signal 114 connected to the microcontroller input RBO. Similarly, the reflex sensor 52 communicates with the microcontroller Ul via the separate interconnection signal 114a connected to the microcontroller input RB2. As described above with respect to Figures 4 and 5, each interconnection signal 114 carries the integration period 158 from the microcontroller Ul to the light sensor 48., 52, as well as the light intensity period 170 from the light sensor 48, 52 to the microcontroller Ul. The resistor R29 and the capacitor C4 connected between VDD and ground provide filtered energy for the light sensors 48, 52. parallel resistor R15 and diode D5 are connected between VDD and node 208. Capacitor C12 is connected between node 208 and ground. The resistor R6 connects the common node 208 to the input / MCLR of the microcontroller Ul. The components D5, R15, R6 and C12 form an activation reset circuit generally shown by 210. The power is supplied to the attenuation logic circuit 56 through the ignition line 212. The DI diode protects the inverted polarity in the ignition line 212 and diode D2 reduces the voltage derived from ignition line 212 to approximately five volts. The capacitors C2, 7 and Cll, the resistor R3 and the ferrite element El form an energy conditioning circuit generally shown by 214. The inverse line 216 is asserted when the vehicle 20 is reversed. The CIO capacitor circuit and the resistors R8, R9 and R27 form a reverse signal conduction circuit, generally shown by 218. The reverse signal conditioning circuit 218 filters through the reverse line 216 and provides protection against electrostatic discharge for the digital input terminal RB6 in the microcontroller Ul. The microcontroller Ul uses the signal on the reverse line 216 to clear the variable transmission element 42 as long as the vehicle 20 is reversed. The microcontroller Ul is synchronized by an RC oscillator formed by the resistor R2 connected between the terminal OSC1 and VDD, and the capacitor Cl connected between the terminal OSC1 and ground. The resistor R30 and LED D3 connected in series between VDD and the open consumption output RA4 of the microcontroller Ul form an indicator lamp which can be mounted on the interior rearview mirror 24 to alert the operator 22 of the operating state of the attenuation logic circuit 56. Switches SI and S2 are connected to digital inputs RB1 and RB3, respectively, of the microcontroller Ul to allow selection of control options. Referring now to Figure 8, a schematic diagram illustrating the operation of the electrochromic attenuator control is shown. A portion of the attenuation logic circuit 56 has been highlighted to more clearly illustrate the control of the electrochromic variable transmission element 42. The electrochromic variable transmission element 42 is preferably similar in operation to those described in the U.S.A. No. 4,902,108 entitled "Single-Compartment, Self-Erasing, Solution-Phase Electrochromic Devices, Solutions for Use Therein, And Uses Thereof" issued to Byker, which is incorporated herein by reference. The electrochromic variable transmission element 42 darkens in response to a control voltage applied at the input node 220. If the applied control voltage is removed, the electrochromic variable transmission element 42 will self-discharge, transmitting an increasingly high amount of light. The electrochromic variable transmission element 42 can be quickly cleaned by shortening the input node 220 to ground. Although the described application pertains particularly to automotive rear-view mirrors, it will be understood by one skilled in the art that all or part of the attenuation logic circuit 56 may be used in a wide variety of electrochromic mirror and window applications. The resistor R.17 connects the input node 220 to the emitter of the Darlington Q10 pair in the node 222. The collector of Q10 is connected to a power source through the current limiting resistor R5, typically 27 O. The base of the Darlington pair Q10 is connected to the digital output RB4 of the microcontroller Ul through the resistors Rl and R7. The base of Q10 is also connected to ground through resistor R4 and through resistor R7 and capacitor Cld. The digital output terminal RB4 is driven by the pulse output 224 in response to the pulse control 226 generated by software running in the microcontroller Ul. The pulse output 224 can produce an impulse signal such as, for example, a signal modulated in pulse width. Preferably, the pulse output 224 functions as a switch, by adjusting the output terminal RB4 either at a high voltage or at a low voltage one time during each transition period as described below in relation to FIG. 9. FIG. capacitor C16 and resistors Rl, R4 and R7 form a low-pass filter, generally shown by 227, to smooth the signal appearing in the digital output RB4. This smoothing results in a substantially constant applied control voltage at the input node 220 for a fixed desired control level. In addition, the base emitter emitter diode falls in Q10 together with the voltage divider formed between the resistor R4, and the sum of resistors R1 and R7 adjusts the operating voltage for the electrochromic variable transmission element 42. Typical values for the components are 1 kO for Rl and R4, 100 O for R7 and 100 μF for C16. With the digital output RB4 at 5 volts and the nominal current draw by the electrochromic variable transmission element 42, the input node 220 is approximately 1.2 volts. The performance of the attenuation logic circuit 56 can be improved through the feedback of the applied control voltage of the electrochromic variable transmission element 42 at the input node 220. The microcontroller Ul includes a comparison logic circuit to cause the pulse emission 224 supply a low voltage if the applied control voltage is greater than the desired control level, and to supply a high voltage otherwise. Typically, the high voltage is close to VDD and the low voltage is close to ground. This comparison can be made by comparing a digital number representing the desired level of control with the applied digitalized control voltage obtained using an analog to digital converter. As an alternative, the digital to analog converter (DAC) 228 and the digital comparator 230 are used. The DAC 228 produces a desired voltage level in the analog output. AN2 in response to the desired control level in the 232 control of the DAC supplied by software running in the Ul microcontroller. The resistor R31 is connected between the analog output AN2 and the node 234, and the resistor R26 is connected between the node 234 and ground. An input of the comparator 230, in the analog input AN3, is connected to the node 234. The other input of the comparator 230, in the analogous input .NO, is connected to the input node 220. The output of the comparator 230 indicates whether the level of The desired voltage is greater than the applied control control voltage. The values for resistors R31 and R26 are chosen such that the voltage at node 234 is within the range of applied control voltages expected at input node 220 along the desired control voltage output scale. from DAC 228. Typical values for R31 and R26 are 390 kO and 200 kO, respectively. Positive feedback is achieved by connecting resistor R24 between node 234 and node 222. Resistor R17 is used to detect the pulse current through the electrochromic variable transmission element 42 and, therefore, is typically a low value such as 10 O. Resistor R24 is typically a high value such as 1.3 MO. As the impulse current is increased through resistor R17, the voltage across resistor R17 is increased by increasing the voltage at node 234. This increase in voltage at the positive input terminal of comparator 230 has the regenerative effect of increasing the duty cycle of the pulse output 224. This regenerative effect provides a better response of the system at higher temperatures when the electrochromic variable transmission element 42 has an increased current draw along with an increase in the maximum operating voltage. The positive feedback also compensates for the effects of the internal resistors within the electrochromic variable transmission element. Referring now to FIG., a timing diagram is shown illustrating the transmission control of the electrochromic element. During the automatic attenuation operation, the software running in the microcontroller Ul is initiated at the transition points, one of which is indicated by 24Q, separated by the fixed transition period 242. The desired control level 244 indicates the desired transmission level for the electrochromic variable transmission element 42. The desired control level 244 may be an analogous value or, preferably, it is a digital number determined by the microcontroller Ul. The desired control level 244 is compared to the voltage of applied control 246 by means of comparison logic. The comparator 230 accepts the applied control voltage 246 and the desired control voltage that appears at the node 234. The comparator output 236 produces a different signal 248, which is asserted when the desired voltage level representing the control level desired 244 is greater than the applied control voltage 246. The comparator output 236 is used to generate the control signal 250 at the output RB4. If the desired control level 244 is greater than the applied control voltage 246, the digital output RB4 is switched to high. If the desired control level 244 is lower than the applied control voltage 246, the digital output RB4 is switched to low. Preferably, the low pass filter 227 filters the control signal 250 to produce the applied control voltage 246. The duration of the transition period 242 is adjusted to inhibit the fluctuation in the electrochromic element 42 which can be noticed, for example, by the operator of the vehicle 22. The transition period 242 may preferably be between two seconds and two microseconds. For the system described above with respect to Figures 7 and 8, five milliseconds can be used for the transition period 242. Referring now to Figure 10, a graph indicating the attenuating reflectance as a function of the control voltage applied is shown. . The curve 254 graphs the percentage of reflectance for the attenuation element 40, which contains the electrochromic variable transmission element 42, as an applied control voltage function 256. The curve 254 indicates a decrease in reflection from about 86% to about 8 % by increasing the applied control t voltage from about 0.2 V to about 0.9 V. Figure 10 also includes the curve 256 which illustrates the current draw as a function of the applied control voltage 256 for the typical electrochromic variable transmission element 42. . Referring again to FIG. 7, additional circuitry is provided to rapidly erase the variably transmissive electrochromic element 40. The transistor Qll is connected through the variably transmissive electrochromic element 40 to the collector at node 220 and the emitter on the ground. The base of transistor Qll is connected through resistor R23 to digital output RB7. When the digital output RB7 is asserted, the transistor Qll is turned on, acting as a switch for rapidly discharging the electrochromic variable transmission element 42. The capacitor C6 is connected between the collector and the base of the transistor Qll to reduce the electromagnetic interference created to the commute the transistor Qll. Transistor Q12 is connected between the base of transistor Q10 and ground, and is also controlled by digital output RB7. The transistor Qll is turned on with the transistor Q12 to turn off the transistor Q10 thus preventing the simultaneous attempt to darken and erase the electrochromic variable transmission element 42. The resistor R7 is placed between the capacitor C16 and the collector of the transistor Q12 to limit the discharge current from capacitor C16 through transistor Q12. Referring now to Figure 11, there is shown a flow diagram illustrating the operation of the attenuation logic circuit 56 according to an embodiment of the present invention. As will be appreciated by one skilled in the art, the operations illustrated are not necessarily sequential operations. Also, although the operations are preferably implemented by software running on the Ul microcontroller, the operations can be carried out by software, hardware or a combination of both. The present invention transcends any particular implementation, and the aspects are shown in the form of a sequential flow chart to simplify the illustration. A reading of ambient light is taken and the average ambient light is initialized in block 260. When the automatic dimming system is initially activated, the average ambient light level is initialized by taking a first reading of the front ambient light 32 using the sensor of front ambient light 48. The acquisition of a reading of ambient light and the average ambient light level are described with respect to blocks 262 and 270, respectively, below. A reading of ambient light is taken and the logarithm of the ambient light reading is in block 262. The use of the semiconductor environmental light sensor 48 with integral charge collection produces the ambient light signal 50 which has an adequate resolution over a wide range of levels of ambient light 32. As described above with respect to Figure 3, this is achieved by taking several ambient light readings 32 using different integration periods 76, 82, 88. In a preferred embodiment, they are used four separate integration periods such as, for example, 600 μs, 2.4 ms, 9.6 ms and 38.4 ms. Each of these integration periods is different by a factor of four from adjacent periods. Therefore, for example, the integration period of 2.4 ms causes the ambient light sensor 48 to be four times more sensitive to ambient light 32 than the integration with the 600 μs integration period does. Typically, the shortest integration pulse 74 is first used by the ambient light sensor 48 to produce the short signal pulse 78. The width of the short signal pulse 78 is measured by the attenuation logic circuit 56. Since the sensor of ambient light front 48 in full darkness can still develop a short signal pulse 78 having a width of less than 100 μs, a minimum threshold is set to accept the short signal pulse 78 so accurately reflecting the ambient light level 32. Typically, this threshold can be 300 μs. If the short signal pulse 78 does not exceed the threshold, the longest integration following is used by the ambient light sensor 48. If the longer integration time does not produce a suitably long signal pulse, the ambient light 32 is a an extremely low level and the mirror 24, 26 can be operated at a maximum sensitivity to the reflection 34. The use of the ambient light signal logarithm 50 allows the use of an inexpensive microcontroller such as Ul, which can have internal registers of only 8 bits and have no multiplication instructions. Since microcontrollers are binary devices, base two logarithms require fewer instructions to calculate than base ten logarithms or natural logarithms. An algorithm is now described to obtain an 8-bit binary logarithm that has the four most significant bits representing an integer part and the four least significant bits a fractional part. The 8-bit ambient light signal 50 resulting from the appropriate integration period is examined bit by bit starting with the most significant bit until the first binary one is found. The bit position that contains the first binary one becomes the entire portion of the logarithm. The four most significant bits that follow the bit position that contains the first binary one becomes the fractional portion of the logarithm. This value is increased by a sixteenth to better approximate the logarithm. An example of the approximation to the binary logarithm is now provided. The assumed ambient light signal 50 is determined as 44 (00101101 in base two). The most significant bit affirmed is bit five, so the integer portion of the resulting value is binary 0101. The next four bits that follow bit five are 0110 so the fractional part of the resulting value is 0110 for a total value of 0101.0110. After the increment, the approximation to the binary logarithm becomes 0101.0111. Referring now to Figure 12, a graph is shown illustrating the approximation to the binary logarithm according to the previous algorithm. The binary logarithm is plotted for values of N between 1 and 255. Curve 290 shows the real binary logarithm. Curve 292 shows the approximate binary logarithm. The ambient light signal 50 must be scaled to compensate for different possible integration periods. This can be achieved by adding a climbing factor to the binary logarithm of the ambient light signal 50. For example, if the longest integration time (38.4 ms) is used to measure ambient light 32, a scale factor of 0 is added. If the next longer integration time (9.6 ms) is used, a scale factor of 2 is added. If the next longer integration time (2.4 ms) is used, 4 is added. shorter integration (600 μs), is added 6. Since the largest value resulting from the approximation to the binary logarithm is 8 (1000.0000), no overflow results from adding the scale factor. Referring again to Figure 11, the logarithm of the ambient light level is compared to the day detection level in block 264. The day detection level is used to avoid attenuation of, or to erase more quickly the Attenuation element 40, during rapid transitions from dark to bright, such as if the vehicle 20 emerges from a tunnel in daylight. If the logarithm of the ambient light 32 exceeds a preset day detection level, the variable transmission element 42 is normalized to adjust the attenuation element 40 to a maximum reflectance in the block 266. The processing is then delayed in the block 268 A waiting loop is entered that has a time long enough to make the period between taking the ambient light readings equal to a constant ambient light loop delay. This period can be, for example, 400 ms. After waiting in block 268, another reading of ambient light 32 is taken in block 262. If the logarithm of ambient light 32 does not exceed the day detection level, an average is obtained in block 270. The average of the The logarithm of the ambient light level is determined in block 270. The averaging of readings converted first to the logarithm of ambient light 32 reduces the effect of a temporary bright light in front of the vehicle 20 to dramatically shift the average reading of an ambient light 32 from another dark way. An average run of the logarithm of ambient light signals 50 can be obtained from a digital low pass filter as described by Equation 1: y (n) = x (n) / 64 + 63y (nl) / 64 ( 1) where x (n) is the approximation to the most recently obtained binary logarithm of the ambient light signal 50 scaled appropriately for the integration period, and (n-1) is the previous filter output and y (n) is the Current filter output. The use of logarithms averaged with analogous light signals is described in the U.S. patent. No. 5,204,778 entitled "Control System For Automotive Rearview Mirros" issued to Bechtel, which is incorporated herein by reference. The average logarithm of the ambient light level is compared to a threshold in block 272. If the ambient light 32 is sufficiently bright, the vehicle operator 22 will not be dazzled by any reasonable amount of reflection 34, allowing the mirror 24 , 26 is adjusted to a maximum reflectance. Therefore, if the average of the logarithm of the ambient light signal 50 is not less than the threshold, the attenuation element 40 is normalized in block 266 and the wait of block 268 is executed. If the average logarithm of ambient light signals 50 is less than the threshold, reflex processing occurs starting at block 274. Typically, the threshold used for comparison in block 272 is less than the day detection level used in the comparison of block 264. The reflex integration period is determined in block 274. The integration period for the reflex sensor 52 is determined based on the ambient light signal 50. The mirror integration period is inversely proportional to the binary antilogarithm of the average of the ambient light signal logarithm 50 as shown in FIG. described by Equation 2: TG (n) = antilog2 (K? -y (n)) -K2 (2) where Te (n) is the integration period for the reflectance sensor 52 for the filter output in the sample time n, K? is a multiplicative constant and K? it is an additive constant. The constants I <;? and K2 are determined experimentally. If the average logarithm of the ambient light signal 50 is below a certain level, a maximum reflex sensitivity integration period is used. A reflex count is established in block 276. The reflex count indicates the number of reflex readings taken between the ambient light readings. The product of the reflex count and the delay of the reflex loop must be equal to the time between taking the ambient light readings. For example, the reflex count may be three and the time between taking the reflex readings may be 133 ms. A reflex reading is taken in block 278. The pulse width returning from the reflex sensor 52 as the reflection signal 54 is measured for the reflex integration period determined in block 274. The value of the attenuation element is set in block 280. Reflex signal 54 is used to determine the desired control level 244 by setting the reflectance for attenuation element 40. This can be achieved, for example, by the use of a look-up table. The precise relationship between the reflection level 34 and the setting for the variable transmission element 42 depends on factors including the construction of the mirror 24, 26, the configuration of the vehicle 20 and the preferred settings by the operator 22. The level of control The desired 244 can be used to control the variable transmission element 42 as described above with respect to Figures 7-10. A review of the reflection count is made in block 282. If the reflection count is zero, the next reading of ambient light is taken in block 262. If the reflection count is not zero, the reflection count decreases in the block 284. A wait loop is then entered into block 286. The delay period of the reflex loop is set so that the reflex readings are taken at regular and predetermined intervals. Referring now to Figure 13, there is shown a schematic diagram illustrating the operation of a light sensor having pulse emission in accordance with an embodiment of the present invention. The pulse light circuit 300 includes the exposed light transducer 106 for converting light 104 incident on the exposed light transducer 106 in charge accumulated in the light storage capacitor 304, indicated by CSL- The exposed light transducer 106 may be any device capable of converting light 104 into charge, such as the photoentrant sensor described in the US patent No. 5,471,515 entitled "Active Pixel Sensor With Intra-Pixel Charge Transfer" to E. Fossum et al. Preferably, the light transducer 106 is a photodiode such as that described below with reference to Figures 25 and 26. Except where indicated, the following description does not depend on a particular type or structure for an exposed light transducer 106.

The impulse light circuit 300 operates under the control of the sensor logic circuit 306. The sensor logic circuit 306 generates a reset signal 308 which controls the switch 310 connected between the output of the exposed light transducer 312 and VDD. The sensor logic circuit 306 also produces a sample signal 314 which controls the switch 316 between the output of the exposed light transducer 312 and the light storage capacitor 304. The voltage across the light storage capacitor 304, the light storage capacitor 318, is fed into an input of comparator 320. The other input of comparator 320 is ramp voltage 322 through ramp capacitor 324. Ramp capacitor 324 is in parallel with the current source 326 generating the IR current. The sensor logic circuit 306 further produces a ramp control signal 328 that controls the switch 330 connected between the ramp voltage 322 and VDD. The comparator 320 produces comparator output 332 based on the relative levels of the voltage of the light storage capacitor 318 and the ramp voltage 322. The sensor logic circuit 306 can generate reset signal 308, the sample signal 314 and the signal ramp control 330 based on the internally generated timing or on the externally generated integration pulse 140 as described below with reference to FIGS. 18-21. Referring now to Figure 14, a timing diagram illustrating the operation of the light sensor of Figure 13 is shown. A measurement cycle is started at time 340 when the sample signal 314 is asserted while the readjustment signal 308 is affirmed. This closes the switch 316 to charge the light storage capacitor 304 to VDD as indicated by the voltage level 342 at the voltage of the light storage capacitor 318. The reset signal 308 is then challenged at time 344, opening the switch 310 and initiating the integration period 346. During the integration period 346, the light 104 incident on the exposed light transducer 106 generates negative charge causing voltage at the slope 348 in the voltage of the light storage capacitor 318. In at time 350, the ramp control signal 328 is asserted by closing the switch 330 and charging the ramp capacitor 324 so that the ramp voltage 322 is VDD as indicated by the voltage level 352. The sample signal 314 is challenged at time 354, causing switch 316 to open, thus concluding the integration period 346. At some time 356 after time 354 and before the next cycle of measurement, the reset signal 308 must be affirmed by closing the switch 310. At time 358, the ramp control signal 328 is challenged by opening the switch 330. This causes the ramp capacitor 324 to discharge at a constant speed through the current source 326 as indicated by the declining voltage 360 at the ramp voltage 322. Initially, as indicated by the voltage level 362, the output of the comparator 332 is challenged because the ramp voltage 322 is higher that the voltage of the light storage capacitor 318. At time 364, the declining voltage 36Q in the ramp voltage 322 falls below the voltage of the light storage capacitor 318 causing the output of the comparator 332 to be asserted. The output of the comparator 322 remains affirmed until the time 366 when the ramp control signal 328 is asserted by closing the switch 330 and bringing the ramp voltage 322 to VDD. The difference between the time 366 and the time 364, indicated by the pulse duration 368, is inversely related to the amount of light 104 received by the exposed light transducer 106 during the integration period 346. Referring now to Figure 15 , a schematic diagram illustrating the operation of a light sensor with noise compensation according to an embodiment of the present invention is shown. An impulse light circuit, shown generally at 380, improves after the impulse light circuit 300 by incorporating a protected light transducer 144 and associated electronic circuits. The protected light transducer 144 preferably has the same structure as the exposed light transducer 106. However, the protected light transducer 144 receives no light 104. The charge generated by the protected light transducer 144 is therefore only a function of noise. This noise is predominantly thermal in nature. If the protected light transducer 144 has the same structure as the exposed light transducer 106, the noise signal produced by the protected light transducer 144 will closely approximate the same noise in the signal produced by the exposed light transducer 106. Subtracting The light produced by the protected light transducer 144 of the signal produced by the exposed light transducer 106, the noise effect on the light transducer 106 can be greatly reduced. The reset signal 308 controls the switch 382 connected between the output of the protected transducer 384 and VDD. The sample signal 314 controls the switch 386 connected between the output of the protected transducer 384 and the noise storage capacitor 388, indicated by CSN. The voltage across the noise storage capacitor 388, the voltage of the noise storage capacitor 390, is an input to the comparator 392. The second input to the comparator 392 is the ramp voltage 322. The output of the comparator 392, output of the noise comparator 394 and output of comparator 332, serve as inputs to exclusive OR opening 396. Exclusive OR opening 396 generates exclusive OR emission 398 which indicates the intensity of light 104. Referring now to FIG. 16, a timing diagram illustrating the operation of the light sensor of Figure 15. The pulse light circuit 380 operates in the same manner as the impulse light circuit 300 with respect to the reset signal 308, the sample signal 314 , the voltage of the light storage capacitor 318, ramp voltage 322, ramp control signal 328 and output of the comparator 332. At time 340, the sample signal 314 is asserted while the reset signal 3 08 is affirmed. Switches 382 and 386 are both closed by charging the noise storage capacitor 388 to VDD as indicated by the voltage level 410 in the noise storage capacitor voltage 390. At time 344, the reset signal 308 is challenged by opening the switch 382 and causing the voltage in decline 412 in the voltage of the noise storage capacitor 390 of the load produced by the protected light transducer 144 due to the noise. At time 354, the sample signal 314 is challenged by ending the integration period 346 for noise collection. At time 358, the ramp control signal 328 is challenged by causing the declining voltage 360 at the ramp voltage 322. Initially, as indicated by the voltage level 414, the output of the noise comparator 394 is challenged because the ramp voltage 322 is greater than the voltage of the noise storage capacitor 390. Since the output of the comparator 332 is also challenged, the output 398 that comes from the comparator 396 is challenged as indicated by the voltage level 416. In the at time 418, the ramp voltage 322 falls below the voltage level of the noise storage capacitor 390, causing the output of the noise comparator 394 to be asserted. Since the output of the noise comparator 394 and the output of the comparator 332 are different, the output 398 of the comparator 396 is asserted. At time 364, the ramp voltage 322 falls below the voltage level of the light storage capacitor 318, causing the output of the comparator 332 to be asserted. Since both the output of the noise comparator 394 and the output of the comparator 332 are now asserted, the output 398 of the exclusive OR opening 396"becomes now unchallenged." The difference between the time 364 and the time 418, the duration of the output pulse 420, has a time period proportional to the intensity of the light 104 that affects the exposed light transducer 106 minus the noise produced by the protected light transducer 144 during the integration period 346. The duration between time 418 and time 358, the duration of noise 422, is directly proportional to the amount of noise developed by the protected light transducer 144 during "the integration period 346. Since the majority of this noise is thermal noise, the noise duration 422 is indicative of the temperature of the protected light transducer 144. At time 366, the ramp control signal 328 is asserted, defying both the output of the noise comparator 394 and the output of the comparator 332. Referring now to Figure 17, a schematic diagram is shown of an implementation of the light sensor of Figure 15 using photodiodes as light transducers. The pulse light circuit 380 is implemented using the exposed photodiode 430 for the exposed light transducer 106 and the protected photodiode 432 for the protected light transducer 144. The anode of the exposed photodiode 430 is connected to ground and the cathode connected through the transistor Q20 to VDD. The base of transistor Q20 is controlled by reset signal 308. Accordingly, transistor Q20 operates as switch 310. Transistors Q21 and Q22 are connected in series between VDD and ground to form a compensator, generally shown by 434. The base of the transistor Q21 is connected to the collector of the exposed photodiode 430. The base of the charge transistor Q22 is connected to the fixed voltage VB. The output of the compensator 434 is connected through the transistor Q23 to the light storage capacitor 304. The base of the transistor Q23 is activated by the sample signal 314, allowing the transistor Q23 to function as the switch 316. The anode of the protected photodiode 432 is connected to earth and the cathode is connected to VDD through transistor Q24. The base of transistor Q24 is activated by reset signal 308 allowing transistor Q24 to operate as switch 382. Transistors Q25 and Q26 form a compensator, generally shown by 436, isolating the output of protected photodiode 432 in the same manner as the transistor. compensation 434 isolates the exposed photodiode 430. Transistor Q27 connects the output of compensator 436 to noise storage capacitor 388. The base of transistor Q27 is activated by sample signal 314, allowing transistor Q27 to operate as switch 386. Typically, the light storage capacitor 304 and the noise storage capacitor 388 are 2 pF. The ramp capacitor 324, typically 10 pF, is charged to VDD through transistor Q28. The base of the transistor Q28 is activated by the ramp control signal 328 allowing the transistor Q28 to operate as a switch 330. The ramp capacitor 324 is discharged through the current source 326 at an approximately constant current IR of 0.01 μA when transistor Q28 is off. The activation response of the sensor is improved, and the effective dynamic range of the sensor is extended, including circuitry to inhibit the output if the ramp voltage 322 drops below a preset voltage. The pulse light circuit 380 includes the comparator 438 which compares the ramp voltage 322 with the initialization voltage (V? Nrr) 440. The output of the buyer 422 is WALKED with the exclusive OR output 396 by the AND opening 444 to produce opening output AND 446. During operation, if the ramp voltage 322 is less than the initialization voltage 440, the output 446 is challenged. The use of the comparator 438 and the AND opening 444 ensures that the output 446 is not asserted regardless of the state of the impulse light circuit 380 after activation. In a preferred embodiment, the initialization voltage is 0.45 V. The sensor logic control 306 generates control signals 308, 314, 328 based on the integration pulse 140 that can be generated internally or provided from an external source. The compensator 447 receives an integration pulse 140 and produces sample control 314. A prime number of sequentially connected inverters, generally shown as inverter train 448, accepts the sample control 314 and produces readjustment control 308. A second set of connected inverters sequentially numbered with prime numbers, generally shown as the inverter train 449, accepts the reset signal 308 and produces the ramp control signal 328. The circuit shown in FIG. 17 has a resolution of at least 8 bits and a sensitivity of approximately 1 volt per lux-second. The maximum output pulse duration 420 is independent of the integration period 346 provided by the duration of the integration pulse 140. Referring now to FIGS. 18-21, various embodiments are shown for the packaging, output and control of the light sensor . Each embodiment may include impulse light circuitry as described above with respect to Figures 13-17. In Figure 18, the light sensor pack 450 accepts four terminals for supply voltage VDD, ground, sensitivity control signal 452 and output signal 454. The signal of the integration period 452 may be the integration pulse 140 used. by the pulse light circuit 380 to produce the output 398, which is sent as an output signal 454. In FIG. 19, the light sensor pack 456 requires only three terminals for VDD, ground and combined signal of sensitivity control and output 458. The combined signal 458 may be the interconnection signal 114 as described above with respect to FIGS. 4 and 5. In FIG. 20, the light sensor pack 460 admits three terminals for the signal of output 454, ground and VDD control signal and combined integration period 462. As is known in the art, the combined signal 462 can be separated into the VDD power supply voltage and the sensitivity control signal 452 by using filters. In Figure 21, the light sensor pack 464 supports three terminals for VDD, ground and output signal 454. The integration period signal 452 is generated within the light sensor pack 464 as described below with respect to the figure 22. Referring now to figure 22, a block diagram of a sensor logic circuit for internally determining the signal of the integration period is shown. The sensor logic circuit 306 may include a free trigger counter 470 activated by the internal oscillator 472. The counter 470 may have sockets, one of which is indicated by 474, connected to different counter bits. For example, a socket 474 may be connected to the n bit, the next socket 474 to the n + 2 bit, the next socket 474 connected to the n + 4 bit, and so on, with each succeeding take, thereby providing an impulse with a four period times longer than the previous take 474. The sensor control signal generator 476 controls the switch 478 to determine which socket 474 will be used to produce the integration pulse 140. Typically, the sensor control signal generator 476 sequences through each socket 474 repeatedly. The sensor control signal generator 476 then uses the integration pulse 140 to generate control signals such as reset signal 308, sample signal 314 and ramp control signal 328 as described above with respect to FIG. 17. Referring now to Figure 23, a block diagram illustrating the use of light transducers having different effective areas to achieve an increased dynamic range is shown. As an alternative to vary the integration time, or together with the variation of the integration time, exposed light transducer pairs 106 and protected light transducer 144 having different effective areas can be used. If the photodiodes 430, 432 are used as light transducers 106, 144, the effective area is the collector area of the photodiode. The exposed small light transducer 490 produces charge, which is converted into a voltage by the light circuit at voltage 492. The light circuit at voltage 492 can be implemented using switches 310, 316 and light storage capacitor 304 as described above with reference to Figure 15. The load produced by the small protected light transducer 494 is converted into voltage by the noise circuit at voltage 496. The voltage-to-noise circuit 496 can be implemented using switches 382, 386 and storage capacitor Noise 388 as described above with respect to Figure 15. The outputs of the light circuit at voltage 492 and the noise circuit at voltage 496 are converted into a pulse with a width based on the charge accumulated by the exposed light transducer. small 490 less load due to the noise integrated by the small protected light transducer 494 during an integration period by the voltage-to-impulse circuit 498. The Voltage to impulse voltage 498 can be implemented using comparators 320, 392, capacitor 324, current source 396, and opening 396 as described above with respect to FIG. 15. The average exposed light transducer 500 has a larger effective area than the effective area for the small exposed light transducer 490, resulting in increased sensitivity. For example, if the effective area of the exposed medium light transducer 500 is four times larger than the effective area of the small exposed light transducer 490, the exposed medium light transducer 500 will be four times more light sensitive 176 than the transducer of small exposed light 490. The medium protected light transducer 502 has an effective area that is equal to that of the exposed medium light transducer 500. The additional light circuit to voltage 492, noise circuit to voltage 496 and voltage circuit to pulse 498 produces an output pulse corrected for noise with a width based on light 104 incident on the exposed medium light transducer 500. Similarly, the large exposed light transducer 504 and the large protected light transducer 506 provide more enhanced sensitivity on the exposed medium light transducer 500 and the medium protected light transducer 502 having an even larger effective area. The switch 508 under the control of the sensor logic circuit 306 establishes which output of the impulse voltage circuits 498 will be used for the output signal 454. The output signal 454 can be selected based on a signal generated in the sensor logic circuit 306 or it may be based on a signal provided from outside the sensor logic circuit 306.

In an alternative embodiment, only a protected light transducer 144 is used. The output of the protected light transducer 144 is scaled before each noise circuit at voltage 496 in proportion to the variable effective areas of the exposed light transducers 106. It will be recognized by one skilled in the art that, although the examples shown in Figure 23 have three pairs of exposed light transducers 106 and protected light transducers 144, any number of pairs can be used. Referring now to Figure 24, a block diagram illustrating the use of light transducers having different openings to achieve an increased dynamic range is shown. As an alternative or in conjunction with the specification of the integration period, the exposed light transducers 106 having the same effective area can each have a different aperture admittance area to admit light 104. The variable apertures can be produced using a partial protector 520 which blocks light 104 from reaching a portion of exposed light transducer 106. Each exposed light transducer 106 produces charge converted to a voltage by a corresponding light circuit at voltage 492. The switch 522 under the control of the sensor logic circuit 306 selects which output of the light circuits at voltage 492 send to the pulse voltage circuit 498. The pulse voltage circuit 498 produces the output signal 454 compensated by the noise detected by the protected light transducer 144 and processed by the voltage-to-noise circuit 496. The sensor logic circuit 306 may select the output of the light circuits at voltage 492 based on an internally generated control signal or on a received control signal from the sensor logic circuit 306. Referring now to FIG. 25, a schematic diagram illustrating different transducer capacitances for different amounts of light-induced charge to achieve an increased dynamic range, according to one embodiment of the present invention, is shown. . A photodiode, generally shown by 530, is formed by the n-type diffusion 532 in the p-type substrate 534. The light 104 incident on the photodiode 530 generates charge 536 that can accumulate in the photodiode cavity 538 under the n-type diffusion. 532. Photodiode 530 has intrinsic photodiode capacitance CPD. The floating diffusion 540 is also formed by spreading n-type material on the substrate 534. The floating diffusion 540 is connected through the transistor Q20 to the reset voltage VRESET- The input of the transistor Q20 is connected to the reset signal 308 under the control of the sensor logic circuit 306. The floating diffusion 540 is also connected to the input of the compensator 542. The output of the compensator 542 is the output of transducer V0ut- The floating diffusion 540 defines the diffusion cavity 544 formed in a region of the substrate 534 when the Reset signal 308 is asserted. The floating diffusion 540 has an intrinsic floating diffusion capacitance CFD- The transmission input 546 is placed between the diffusion 532 and the floating diffusion 540. The transmission input 546 is maintained at a voltage VTG to form the transmission cavity 548 under the the same. The transmission cavity 548 has a depth less than the depth of the photodiode cavity 538 and the diffusion cavity 544. The transmission input 546 has an intrinsic transmission input capacitance When the reset signal 308 is asserted, carrying the floating diffusion 540 to VESET, the charge is eliminated in the diffusion cavity 544. Further, when the charge is reset in the diffusion cavity 544, any load 536 in the photodiode cavity 538 above the depth of the transmission cavity 548 flows through the transmission cavity 548, through the floating diffusion 540, and is eliminated. During a period of light integration, the reset signal 308 is challenged, causing the floa diffusion voltage 540 to float based on the amount of the load 536 in the diffusion cavity 544. When the light 104 hits the diffusion 532, the load 536 is created. Since the load 536 in the photodiode cavity 538 up to the level of the transmission cavity 548 was not eliminated by the readjustment of the load, additional load 536 produced by the incident light 104 flows from the photodiode cavity 538 through the transmission cavity 548 and into the diffusion cavity 544. At the load level 550, below the level of the transmission cavity 548, only the diffusion cavity 544 is filled with load 536. Accordingly, the voltage of the Floa diffusion 540 is inversely proportional to the floa input capacitance CFD. When sufficient charge 536 has been generated to fill the diffusion cavity 544 above the level of the transmission cavity 548 such as, for example, the level 552, the diffusion cavity 544, transmission cavity 548 and photodiode cavity 538 all Accordingly, the floa diffusion voltage 540 is now inversely proportional to the sum of the floa diffusion capacitance CFD, transmission input capacitance CTG and photodiode capacitance CPD. Referring now to Figure 26, a graph of the output potential is shown as a function of the cumulative incident light for the transducer of Figure 25. A curve, generally shown by 554, shows the transducer output V0ut as a function of the light 104 incident on the diffusion 532 and, possibly, the floa diffusion 540 during the integration period. During the deep portion 556, the load 536 is accumula in the diffusion cavity 544 alone. Since the conversion gain is based solely on the floa diffusion capacitance CFD, the photodiode 530 appears to have a high sensitivity to incident light 104. During the shallow portion 558, the load 536 is accumulated in the diffusion cavity 544, cavity transmission 548 and photodiode cavity 538. Since the conversion gain now depends on the parallel combination of capacitance CFD CTG and CPD, photodiode 530 now seems less sensitive to incident light 104. By adjus the VRESET and VTG voltages, the of angle change 559 between the deep portion 556 and the shallow portion 558 can be displaced by affec the dynamic range. For example, if the maximum voltage oscillation for the floa 540 diffusion is 1 volt; the ratio of CFD to the sum of CFD, CTG and CPD is 1: 100; and the angle of change of angle 559 is adjusted to 0.5 volts, the dynamic range of photodiode 530 is increased approximately 50 times over the dynamic range of a similar photodiode without double capacitance. Referring now to Figure 27, there is shown a schematic diagram illustrating a photodiode transducer incorporating an anti-reflective aperture. The anti-glare aperture 560 is formed between the diffusion 532 and the feed voltage diffusion 562 attached to VDD. The anti-reflective aperture 560 is attached to the anti-reflective voltage VAB. The anti-reflective aperture 560 forms an anti-reflecting cavity 564 on the substrate 534 between the photodiode cavity 538 and the source diffusion cavity 566. The anti-reflective voltage VAB is less than the input voltage cavity of the VTG transmission. 564, making the anti-reflection cavity 564 less deep than the transmission cavity 548. When the accumulated charge generated by the photodiode 530 exceeds the load level 568 equal to the depth of the anti-reflecting cavity 564, the excess load it flows below the anti-reflecting cavity 560 into the feed voltage diffusion 562 and is eliminated. The anti-reflective aperture 560 prevents the output voltage V0ut from falling below a level detectable by the comparator 320 in the pulse light circuit 380. Referring now to FIG. 28, there is shown a drawing illustrating the housing for a light sensor according to one embodiment of the present invention. The light sensor 48, 52 includes the housing 100 having the window 102 for admitting light, a beam of which is indicated by 570. The housing 100 supports the power terminal 108, ground terminal 11Q and signal terminal 112. The semiconductor frame 572, encapsulated within the housing 100, incorporates light transducers 106, 144 and associated electronic circuits as described above with reference to Figures 4-5 and 13-26. The terminals 108, 110 and 112 can be connected by wires to the frame 527, as shown by the cable 574 for the power terminal 108 and the cable "576 for the signal terminal 112, or can be directly attached to the frame 527, as shown for the ground terminal 110. Preferably, the housing 100 is of the same type used to construct three-terminal light-emitting diodes (LEDs) A preferred format is commonly referred to as the package t-? 3 4 or 5 mm The encapsulating electronic circuits in such packages are well known in the art of the manufacture of optical electronics A lens, generally shown by 578, is preferably used to focus light on the exposed light transducer 106. The lens 578 can be placed in front of the light sensor 48, 52 or, preferably, it can be incorporated in the window 102 as shown in figure 28. The lens 578 defines the field of vision- of the sensor 48, 52 and provides better sensitivity. prayed through optical gain. Referring now to Figure 29, there is shown a graph illustrating the field of view of the light sensor as a function of the distance of the light transducer from the lens. The field of view for the exposed light transducer 106 in the light sensor 48, 52 is defined as the viewing angle? made by the marginal ray 570 with respect to the optical axis 580 through the exposed light transducer 106. The medium angle field of view for the spherical lens 578 is expressed by Equation 3:? = 90-arccos. { r / R } +? 2 / nJ * sin. { bows { r / R } -arctan. { (d- (R- (R2-r2) 1 / z)) / r} } where r is the opening radius of the lens, R is the radius of curvature of the lens 578, n? is the refractive index of the material within the housing 100, n, is the refractive index outside the housing 100, d is the distance from the center of the lens 578 to the exposed light transductox 106, and? It is measured in degrees. Typically, the housing t-l3 4 100 is filled with epoxy resin and the sensor 48, 52 operates in air making the ratio of n2 to ¿of about 1.5. the curve 590 graphs the field of view of medium angle? as a function of distance d for a housing t-l3 4 having a spherical lens 578 with radius J? of 5.0 mm. As the light transducer 106 moves further away from the lens 578, the field of view decreases. Referring now to Figure 30, there is shown a graph illustrating the optical gain of the light sensor as a function of the distance of the light transducer from the lens. Assuming a paraxial approximation for the rays 570, the optical gain of the lens 578 can be estimated by considering the additional optical energy ratio collected by the light transducer 106 with the lens 578 to the optical energy collected by the light transducer 106 without the lens 578. This can be calculated by considering a light cone with a base on the surface of the lens 578 and a point at the focal point f of the lens 578. The optical gain G can then be expressed as a function of the section ratio transverse of the cone to the area of the light transducer 106 which is reduced to Equation 4: G = f2 / (í-d) 2 (4) The curve 600 shows the optical gain G as a function of the distance d for housing t- 13 having a spherical lens 578 with a radius JI of 5.0 mm and a focal length f of 15.0 mm. As the light transducer 106 moves further away from the lens 578, the optical gain increases. The distance d between the lens 578 and the light transducer 106 can be adjusted for the optimum performance of the front ambient light sensor 48 and the reflex sensor 52. The front ambient light sensor 48 should have a wide field of view but has no to be as sensitive as the reflex sensor 52. The reflex sensor 52 should have a narrower field of view but should be more sensitive and, therefore, benefit from a higher optical gain. For the lens described above with reference to Figures 29 and 30, a distance d of between 2 mm and 3 mm is suitable for the ambient light sensor 48 and a distance d of between 6 mm and 7 mm is suitable for the sensor of the reflections 62. In addition to modifying the parameters of the lens, other types of lenses such as spherical, cylindrical and the like are possible within the spirit and scope of the present invention. Referring now to Figure 31, a graph illustrating the frequency response of the human eye is shown. Curve 610 represents the relative photopic or daylight frequency response of the human eye. Curve 612 represents the relative scotopic or nocturnal frequency response of the human eye. In addition to being more sensitive to light intensity, the scotopic response 612 shifts more towards violet than the photopic response 610. To preserve night vision, which degrades rapidly when exposed to bright light, particularly in the range of the scotopic curve 612, the exposed light transducer 106 should have a frequency response similar to the scotopic curve 612. If this is not practical, the exposed light transducer 106 should at least have an attenuated infrared response. This is increasingly important as high-intensity discharge (HID) headlamps, which emit a more bluish light than incandescent or halogen headlights, are becoming increasingly popular.

Referring now to Figure 32, a graph illustrating the frequency response of a typical light transducer is shown. The relative frequency response of a typical photodiode is shown as the curve 620. When compared to the scotopic response curve 612, the frequency response of the exposed light transducer 106 contains significantly more infrared sensitivity. As described in relation to Figure 2 above, the filter 58, 60 can be placed before or incorporated into the sensor 48, 52 in such a manner that the emission of the exposed light transducer 106 more closely simulates the scotopic frequency response 612 of the human eye. Referring now to Figure 33, a drawing of a housing incorporating an infrared filter is shown, according to one embodiment of the present invention. The window 102 in the housing 100 includes the infrared filter 630, which is operative to attenuate the infrared components of light rays 570 that strike the exposed light transducer 106. The infrared filter 630 may be a hot mirror available from Optical Coating Laboratories , Inc. of Santa Rosa, California. A lens, such as that described with respect to Figures 28-30 above, can be placed in front of the infrared filter 630.

Additional filtration for the exposed light transducer 106 is described in the patent of E.U.A. No. 4,799,768 entitled "Automatic Rearview Mirror with Filtered Light Sensors" to Gahan, which is incorporated herein by reference. Although embodiments of the invention have been illustrated and described, it is not intended that these embodiments illustrate and describe all possible forms of the invention. Rather, it is intended that the following claims cover all alternative modifications and designs, and all equivalents, that fall within the spirit and scope of this invention. It is noted that in relation to this date, the best method known to the applicant to carry out the aforementioned invention, is that which is clear from the present description of the invention.

Claims (22)

CLAIMS Having described the invention as above, the content of the following claims is claimed as property:

1. A light sensor characterized in that it comprises: a photodiode light transducer exposed to light, the exposed light transducer is operative to accumulate charge in proportion to light incident during an integration period; and a sensor logic circuit in communication with the exposed light transducer, the sensor logic circuit is operative to a) determine a period of light integration before starting the integration, b) restore the accumulated load on the light transducer exposed to the start of the determined light integration period, c) measuring the accumulated load by the exposed light transducer during the determined light integration period, and d) determining an impulse having a width based on the measured and accumulated load of the exposed light transducer .

2. A light sensor according to claim 1, characterized in that it further comprises a comparator with an input connected to the exposed light transducer and the other input connected to a switched capacitor circuit, the switched capacitor circuit operable to charge a capacitor up to a fixed voltage when the switch is closed and to discharge the capacitor at a constant speed when the switch is open, where the sensor logic circuit is also operative to close the switch during the determined light integration period and open switch after the integration period of determined light, thus creating the impulse at the output of the comparator.

3. A light sensor according to claim 2, characterized in that it also comprises a second comparator with an input connected to a fixed voltage and the other input connected to the switched capacitor circuit, the output of the second comparator is operative to inhibit the output of the determined pulse. if the ramp voltage is less than the fixed voltage.

4. A light sensor according to claim 1, characterized in that it also comprises a photodiode light transducer protected from light, the protected light transducer is operative to accumulate charge in proportion to noise during the integration period, wherein the circuit The sensor logic is also operative for: restoring the accumulated load in the protected light transducer at the beginning of the determined light integration period; measure the load accumulated by the protected light transducer during the determined light integration period; and determining an output pulse having a width based on the difference between the measured and accumulated load of the exposed light transducer and the measured and accumulated load of the protected light transducer.

5. A light sensor according to claim 4, characterized in that the light sensor has an input to receive an integration signal and where the noise depends on the temperature of the light sensor, the sensor logic circuit is also operative to: determine an integration period based on the integration signal; and emitting the output pulse after a length of time that follows the end of the integration signal, the length of time is based on the noise level, the length of time thereby indicating the temperature of the light sensor.

6. A light sensor according to claim 1, characterized in that the light integration period is determined from the asserted portion of a control signal received by the sensor logic circuit.

7. A light sensor according to claim 1, characterized in that the sensor control is operative to determine each integration period by cycling through a sequence of predetermined time periods.

8. A light sensor according to claim 1, characterized in that it also comprises at least one additional exposed photodiode light transducer., each additional exposed light transducer is operative to accumulate charge in proportion to light incident during a period of integration at a speed different than the speed of any other exposed light transducer, the sensor logic circuit is operative in addition to: restoring the load accumulated in each of the at least one additional exposed light transducer at the start of the determined light integration period, measuring the accumulated load for each of the at least one additional exposed light transducer during the determined light integration period, and emitting an impulse having a width based on the measured and accumulated load for each of the at least one additional exposed light transducer.

9. A light sensor according to claim 8, characterized in that each exposed light transducer has a different collector area which thus has a different charge accumulation rate.

10. A light sensor according to claim 8, characterized in that each exposed light transducer has an aperture for admitting light incident on the sensor, each sensor aperture having a different admission area thus giving each light transducer exposed a different charge accumulation speed.

11. A light sensor according to claim 1, characterized in that the light transducer comprises: a photodiode that overlaps a substrate, the photodiode is operative to accumulate incident charge in a photodiode cavity formed in a region of the substrate underlying the photodiode, the photodiode has an intrinsic photodiode capacitance; a floating diffusion formed in the substrate, the floating diffusion has a diffusion cavity formed in a region of the substrate that underlies the floating diffusion when the load is restored, the floating diffusion is operative to eliminate charge in the diffusion cavity when the load is restored, the load of the floating diffusion determines an output potential, the floating diffusion has an intrinsic floating diffusion capacitance; and a transmission aperture between the photodiode and the floating diffusion, the transmission aperture defines a transmission cavity formed in a region of the substrate between the region of the substrate that underlies the photodiode and the region of the substrate that underlies the floating diffusion, the cavity of transmission has a depth less than the depth ^ of the photodiode cavity and the diffusion cavity, the transmission aperture has an intrinsic transmission aperture capacitance; wherein, when the charge is restored, the charge in the cavity of the photodiode on the depth of the transmission cavity flows through the transmission cavity, through the floating diffusion and is eliminated; and wherein, during the determined light integration period, the charge produced by incident light on the photodiode flows through the transmission cavity and into the diffusion cavity, producing output voltage inversely proportional to the capacitance of the floating diffusion, until the diffusion cavity is filled to the depth of the transmission cavity where the charge produced by light incident on the photodiode fills the photodiode cavity, the transmission cavity and the diffusion cavity, producing voltage output inversely proportional to the sum of the capacitance of the floating diffusion, the capacitance of the photodiode and the capacitance of the transmission aperture; thus providing a first sensitivity during the accumulation of charge in the diffusion cavity and a second sensitivity during charge accumulation in the diffusion cavity, the transmission cavity and the photodiode cavity, the first sensitivity is greater than the second sensitivity.

12. A light sensor according to claim 11, characterized in that it further comprises an anti-reflection aperture between the photodiode and a primary supply voltage diffusion, the anti-reflection aperture defines an anti-reflection cavity formed in a region of the substrate between the substrate region underlying the photodiode and the primary supply voltage diffusion, the anti-reflex cavity has a depth less than the depth of the transmission cavity.

13. A light sensor package, characterized in that it comprises: a housing having a window for receiving light, the housing admits an energy terminal, a ground terminal and an output terminal; an exposed photodiode light transducer placed inside the housing, the exposed light transducer is operative to accumulate charge in proportion to light received through the window incident on the exposed light transducer during an integration period; a light-to-voltage circuit arranged within the housing and in communication with the exposed light transducer, the light-to-voltage circuit is operative to emit a light voltage signal based on charge accumulated by the exposed light transducer; and a voltage-to-impulse circuit disposed within the housing and in communication with the voltage-to-voltage circuit, the operating voltage-to-impulse circuit for emitting a pulse on the output terminal, the pulse width is based on the voltage signal of light.

14. A pack of light sensors according to claim 13, characterized in that it further comprises: a photodiode light transducer protected from light, the protected light transducer has the same structure as the exposed light transducer, the protected light transducer operative to accumulate charge in proportion to noise during the integration period; and a voltage-to-voltage circuit arranged within the housing and in communication with the protected light transducer, the voltage-to-voltage circuit is operative to emit a noise voltage signal based on the charge accumulated by the protected light transducer; wherein the pulse voltage circuit is operative to emit a pulse at the output terminal having a pulse width based on the difference between the light voltage signal and the noise voltage signal.

15. A pack of light sensors according to claim 13, characterized in that it further comprises a sensor logic circuit disposed within the housing and in communication with the exposed light transducer and the voltage light circuit, the sensor logic circuit is operative to: determine a period of light integration; restore the accumulated load in the light transducer exposed at the start of the determined light integration period; and controlling the charge accumulation by the exposed light transducer during the determined light integration period.

16. A pack of light sensors according to claim 15, characterized in that the sensor logic circuit is operative to determine the period of integration of light from the asserted portion of a control signal received at the output terminal.

17. A pack of light sensors according to claim 15, characterized in that the sensor logic circuit is operative to determine each light integration period by cyclizing through a sequence of predetermined time periods.

18. A package of light sensors according to claim 15, characterized in that it further comprises at least one additional exposed photodiode light transducer, each additional exposed light transducer is operative to accumulate charge in proportion to light incident for a period of time. integration at a speed different than the speed of any other exposed light transducer, the sensor logic circuit is further operative to: restore the accumulated charge in the at least one additional exposed light transducer at the start of the determined light integration period, measuring the load accumulated by the at least one additional exposed light transducer during the given light integration period, and emitting an impulse having a width based on the measured and accumulated load for each of the at least one light transducer exposed additional.

19. A pack of light sensors according to claim 18, characterized in that each exposed light transducer has a different collector area which thus has a different charge accumulation rate.

20. A pack of light sensors according to claim 18, characterized in that each exposed light transducer has an aperture for admitting light incident on the sensor, each sensor aperture having a different admission area thus giving each transducer of light exposed a different charge accumulation speed.

21. A light sensor characterized in that it comprises: a photodiode that overlaps a substrate, the photodiode is operative to accumulate charge generated by incident light in the photodiode in a photodiode cavity formed in a region of the substrate underlying the photodiode, the photodiode has a capacitance of intrinsic photodiode; a floating diffusion formed in the substrate, the floating diffusion has a diffusion cavity formed in a region of the substrate that underlies the floating diffusion when the load is restored, the floating diffusion is operative to eliminate charge in the diffusion cavity when the load is restored, the load of the floating diffusion determines an output potential, the floating diffusion has an intrinsic floating diffusion capacitance; and a transmission aperture between the photodiode and the floating diffusion, a transmission cavity formed in a region of the substrate between the region of the substrate underlying the photodiode and the region of the substrate underlying the floating diffusion, the transmission cavity has a depth lower than the depth of the photodiode cavity and the diffusion cavity, the transmission aperture has an intrinsic transmission aperture capacitance; where, when the charge is restored, the charge in the cavity of the photodiode on the depth of the transmission cavity flows through the transmission cavity, through the floating diffusion and is eliminated; and wherein, during a period of light integration, the charge produced by incident light on the photodiode flows through the transmission cavity and into the diffusion cavity, producing output voltage inversely proportional to the diffusion capacitance floating, until the diffusion cavity is filled to the depth of the transmission cavity where the charge produced by light incident on the photodiode fills the photodiode cavity, the transmission cavity and the diffusion cavity, producing output voltage inversely proportional to the sum of the capacitance of the floating diffusion, the capacitance of the photodiode and the capacitance of the transmission aperture; thus providing a first sensitivity during the accumulation of charge in the diffusion cavity only and a second sensitivity during charge accumulation in the diffusion cavity, the transmission cavity and the photodiode cavity, the first sensitivity is greater than the second sensitivity.

22. A light sensor according to claim 21, characterized in that it further comprises an anti-reflection aperture between the photodiode and a primary supply voltage diffusion, the anti-reflection aperture defines an anti-reflection cavity formed in a region of the substrate between the substrate region that underlies the photodiode and the primary supply voltage diffusion, the anti-reflex cavity has a depth less than the depth of the transmission cavity. PHOTODIOD LIGHT SENSOR SUMMARY OF THE INVENTION Light sensors are described that have a wide dynamic range and are used in a variety of applications. A wide dynamic range light sensor (48, 52) includes an_ exposed photodiode light transducer (106) that accumulates charge in proportion to light (104) incident during an integration period (158). A sensor logic circuit (306) determines a period of light integration (158) before the start of integration and the load is restored. The accumulated load is measured by the exposed light transducer (106) during the light integration period (158) and an impulse (122) having a width is determined (170) based on the accumulated load. 1/24 2/24 V. CONTROL SIGNAL SENSOR OUTPUT faS And 4 fa 6/24 ? ff 8/24 FC # 244 DESIRED CONTROL LEVEL WEATHER 10/24 CURRENT (ma). N 13/24 * 14/24 ! I 15/24 fcSff 16/24 ! l? 17/24 18/24 19/24 20/24 INCIDENT LIGHT (LUX-SECONDS) #ff _ _ DISTANCE (mm) fc ^^ DISTANCE (mm) b * # 0 WAVE LENGTH (nm) v * 3 / 24/24 0 WAVE LENGTH (nm) v * á &