MXPA01002232A - Combined channel coding and space-block coding in a multi-antenna arrangement - Google Patents

Abstract

Enhanced performance is achieved by combining channel coding with the space-time coding principles. With K synchronized terminal units transmittingon N antennas to a base station having M$m(G K) receive antennas, increased system capacity and improved performance are attained by using a concatenated coding scheme where the inner code is a space-time block code and the outer code is a conventional channel error correcting code. Information symbols are first encoded using a conventional channel code, and the resulting signals are encoded using a space-time block code. At the receiver, the inner space-time block code is used to suppress interference from the other co-channel terminals and soft decisions are made about the transmitted symbols. The channel decoding that follows makes the hard decisions about the transmitted symbols. Increased data rate is achieved by, effectively, splitting the incoming data rate into multiple channels, and each channel is transmitted over its own terminal.

Description

Combined Channel Coding and Coding by Space Blocking in a Multiple Antenna Array BACKGROUND OF THE INVENTION This invention relates to wireless communication and, more particularly, to techniques for effective wireless communication in the presence of signal fading, co-channel interference, and other impairments. The physical limitations of wireless channels represent a fundamental technical challenge for reliable communication. Broadband limitations, loss of propagation, variation of time, noise, interference, and multipath signal fading make the wireless channel a narrow "conduit" where the data flow is not easily adjusted. The additional challenges come from the limitation of energy, size, and speed of the devices that are used within the portable wireless device. The use of multi-transmit antennas in remote and base stations increases the capacity of wireless channels, and information theory provides a measure of this increase. The standard process to take advantage of this capability is Hef: 126595 linear processing at the receiver, as described, for example, by J. Winters, J. Salz and RD Gitlin, in "The i pact of antenna diversity at the capacity off wirelless communication systems "IEEE Trans. Communications, Vol. 42. No. 2/3/4, pages 1740-1751, Feb / March / April 1994. The transmission diversity has been explored by Wittneben in "Base station modulation diversity for digital SIMULCAST," Proc. IEEE 'VTC, page 505-511, May 1993, and by Ses adri and Winters in "Two signaling schemes for improving the error performance of frequency-division-duplex (FDD) transmission system using transmitter antenna diversity", In terna ti onal Journal of Wirel ess Informati on Networks, Vol. 1, No. 1, 1994. The Wittneben and Seshadri and collaborators involve the process of transmission diversity from a point of view of signal processing. The space time codes combine the signal processing in the receiver with appropriate coding techniques to the multiple transmission antennas. See, for example, V. Tarokh, N. Seshadri, and A. R. Calderbank in "Space-Time Codes for High Data Rate Wireless Communication: Performance Analysis and Code Construction" IEEE Trans. Info. Theory, Vol. 44, No. 2, pp. 744-765, March 1998. The time-space process provides a significant advantage over the aforementioned technique. The specific space time codes designed for the good performance of the transmit antennas 2-4 in slowly varying signal fading environments (such as the internal transmission) and arrive within 2-3 dB of the theoretical cutting capacity. For example, cutting capacity is described, for example, J. Foschini, Jr. and MJ Gans, "On limits of wireless communication in a fading environment, when using multiple antennas," Wirel ess Personal Communi cation, Vol. 6, No. 3, page 311-335, March 1998. The broad band efficiency of the codes described in the Tarokh et al. Document is approximately 3-4 times that of the current systems. The most important contributor to improved performance is diversity, which can be considered as providing several replicas of the signal transmitted to the receiver, with some replicas that are less attenuated by signal fading. The space time codes presented in the Tarokh et al. Document provide optimized traffic between constellation size, data rate, diversity amplification, and link complexity. When the number of transmit antennas is fixed, the complexity of the decoding (measured, for example, by the number of link states in the decoder) increases exponentially with the transmission rate. This can be refined to some extent by designing space time codes with a multilevel structure and adopting multiple phase decoding, as described by Tarokh et al. For a moderate number of transmit antennas (3-6), this method provides higher data rate while reducing the decoding complexity. However, there is a penalty to be paid for simplified decoding. The multi-phase decoding is sub-optimal, due in part to the amplification in the error coefficient, and this performance penalty means that alternative solutions are needed to achieve higher data rates. To achieve very high data rates in the shortband wireless channels, several antennas are needed in both the transmitter and the receiver. Consider a wireless communication system that employs n transmitting antennas and receiving antennas, where each transmitting and receiving antenna is Rayleigh quasi-static, flat, and independent of the others. If n is fixed, then the capacity increases logarithmically with m. On the other hand, if m is fixed then the intuition suggests that it should reach a point where adding more transmit antennas will not make much difference. In fact, this can be seen in the mathematics of cutting capacity, shown by Foschini and Gans in the aforementioned document. Therefore, it turns out that in the presence of a receiving antenna, little can be obtained in terms of the cutting capacity using more than 4 transmitting antennas. A similar argument shows that if there are two receiving antennas, using 6 transmit antennas, almost all the capacity increase that can be obtained is provided. If n increases and m _ >; n, then the information theory shows that the capacity of the system is increased at least linearly as a function of n. Therefore, it makes sense to increase the number of reception and transmission antennas to obtain greater capabilities. Using multiple transmit and receive antennas, a multi-input and multiple-output system is created where the number of degrees of freedom is given by the product of the number of transmit and receive antennas. Foschini considered a system in "Layered space-time architecture for wireless communication in a fading environment when using multi-element antennas," Bell Labs Technical Journal, Vol. 1, No. 2, Fall 1996. A structure with multiple layers is proindicated. which, in principle, can achieve a firm lower union of capacity. If transmitting and n receiving antennas are used, then at the receiver, the transmitted signal of the transmitting antenna 1 is treated as the desired signal, while the signals transmitted from other transmitting antennas are treated as interference. The linear process is then used to suppress the interference signals using the receiver antennas, providing an amplification of the diversity of one. Once the signal transmitted from the antenna 1 is correctly detected, the signal transmitted from the antenna 2 is treated as the desired signal while the signals transmitted from the transmitting antennas 3,4, ..., n are treated as interference. The contribution of the signal transmitted from the antenna 1, now already detected, is subtracted from the signal received in the receiver antennas 1 to n. Then, the detection of the signal transmitted by the antenna 2 proceeds with the linear process that is applied to suppress the interference signals from the antennas 3 to n. This provides an amplification of diversity of two. This process is repeated until all the transmitted signals are detected. Clearly, the worst case of diversity in this architecture is in one. For this system, large data frames combined with the best coding technique-s are needed to achieve the lowest link in cutting capacity. In the North American Application Serial No. 09/114838 filed July 14, 1998, the claimed priority of Provisional North American Application No. 60/052689 filed on July 16, 1997, describes an arrangement that provides improved performance that is It is done using a perspective that combines the signal process in series with the channel coding. Specifically, the antennas in the transmitter are divided into small groups, and the individual space time codes are used to transmit the information from each group of antennas. At the receiver, an individual space time code is decoded by means of a linear series processing technique that suppresses the signal transmitted by other groups of antennas by treating them as interference. The contribution of the decoded signal to another received signal is then subtracted from the received signals. The results are a simple receiver structure that provides the diversity and amplification of coding with respect to the non-encoded systems with a given diversity amplification. This combination of serial processing in the receiver and the coding techniques for the multiple transmission antennas, provides a reliable communication and with a high speed of data with respect to the wireless channels. An advantage of the group interference suppression method with respect to the Foschini architecture is that the number of receiving antennas may be less than the number of transmitting antennas. In the North American Application Serial No. 09 / 149,163 filed on September 4, 1999, the claimed priority of US Provisional Application Number 60/052689 filed on July 17, 1997, describes an arrangement wherein synchronized terminal units K transmit on N antennas to a base station that has M >; K of antennas. An improvement is achieved by employing interference cancellation (IC) and maximum probability decoding (ML). More specifically, time-space blocking coding is employed in each of transmitters employing N transmitting antennas, and the signals are received in a receiver employing M receiving antennas. By scanning the structure of the code by time-space blocking, the interference transmission units Kl are canceled at the receiver, without considering the number of transmission antennas TS, when the decoding of the signals transmitted by a unit given mobile. Also described is an array where the signals of a first terminal unit are first decoded, and the resulting decoded signals are used to cancel their contribution to the signals received on the base station antennas while the signals of the terminal units K are decoded. - l remaining. The process is repeated between the remaining K-l terminal units.

BRIEF DESCRIPTION OF THE INVENTION The improved embodiment is achieved by combining the channel coding with the space-time coding principles described in applications 163. More specifically, with synchronous terminal units K transmitting on N antennas to a base station having receiving antennas M > K, increased system capacity and improved performance are achieved using a linked coding scheme where the internal code is a time-space blocking code and the outer code is a conventional channel error correction code. That is, the information symbols are first encoded using a conventional channel code. The channel code then encodes the signal using a code by space time blocking then, and transmitted over the N antennas. At the receiver, the internal space time blocking code is used to suppress interference from the other co-channel terminals and flexible decisions are made on the transmitted symbols. The decoding of the continuous channel, makes firm decisions on the transmitted symbols. The increased data rate is achieved by ppr effectively dividing the incoming data into multiple channels, and each channel is transmitted on its own terminal. Seen in another way, the information symbols of a transmission terminal are divided into parallel flows L. Flow 1 is encoded using a channel code with the ratio R_ and then encoded with a space time blocking encoder with N transmitting antennas. Advantageously, the coding rates are chosen such that Ri > R2 > • - * > R.

BRIEF DESCRIPTION OF THE DRAWINGS Figure 1 describes an arrangement that, illustratively, includes a base station 30 with four antennas, terminal units 20 with two antennas, and a terminal unit 20 with two antennas; and Figure 2 illustrates a terminal unit that divides the input signal into two streams, and each stream is transmitted over an array of two independent antennas.

DETAILED DESCRIPTION FIG. 1 illustrates a unit 10 employing a time-space-blocking coding unit 13 which is followed by a conventional constellation map and a pulse-forming circuit 16. The output of circuit 16 feeds the two antennas of transmission 11 and 12. The input symbols to the encoder by time-space blocking are divided into groups of two symbols each, and in a given symbol period, the two symbols in each group. { c_, Cz} they are transmitted simultaneously from the two antennas. The signal transmitted from the antenna 11 is ci and the rowan transmitted from the antenna 12 is c2. In the next period of the symbol, the signal -c2 * is transmitted from the antenna 11 and the signal c_ * is transmitted from the antenna 12. In the receiver 20, the signals are received by the antennas 21 and 22 and are applied to the detector 25 The channel estimators 23 and 24 operate on the incoming signal of the antennas 21 and 24, respectively, in a conventional manner to develop estimates of the channel parameters. These estimates apply to the detector 25. In the mathematical development of the algorithms described herein, it is assumed that the channel of each of the two transmit antennas remains fixed over one of the two consecutive symbol periods. That is, hi (nT) = h. { n + 1) T), i = 1,2. (1) To determine the characteristics of the channel, the transmitter conducts a calibration session, during which signals or pilot tones are transmitted. That is, the signals that are received during the calibration session that are used by the channel estimation circuits 23 and 24, which are well known circuits.

Maximum Probability. The signals received in the antenna 21 can be expressed as r_ = -hi c + hz Ci * +? ¿, (3) where r_ and r are the signals received over two consecutive symbol periods, h_ denotes the signal fading channel between the transmitting antenna 11 and the receiving antenna 21, h_ denotes the channel between the transmitting antenna 12 and the receiving antenna 21, and ?? and? _ are the noise terms, which are assumed to be the complex Gaussian probability variables with zero means and the spectral density energy N., / 2 per dimension. Defining the vectors r = [? R2 *] t, s = [c? C2]? , and TJ = [?. ,? *} ~, equations (2) and (3) can be rewritten in a matrix form as r = E-c +?, (4) where the H-channel matrix is defined as The vector? is a Gaussian probability vector complex with zero means and covariant o. I. Defining C as the group of all possible symbol pairs c =. { c, c_} , and assuming that all pairs of the symbol are equi-probable, it can easily be shown that the optimal maximum likelihood decoder (ML) is selected from C, the pair of the symbol c that minimizes the expression || r-H-S || '. This can be written as c = arg minllr - H -el (6) It was demonstrated by S. Alamouti in "Space Block Coding: A Simple Transmitter Diversity Scheme for Wireless Communications," presented to IEEE JSAC, September 1997, that the order of code diversity by Previous space time blocking is equivalent to that of a received combination of maximum ratio of two branches (MRRC). Alamouti also showed that, due to the orthogonal form of the H matrix, this decoding rule is broken down into two independent decoding rules for c- and c_. The non-certainty,? _., Of the decoded symbols q is defined as ?, = r-H »?? 2 (7) The maximum probability rule (ML) of equation (6) can be simplified by understanding that the matrix of channel H is orthogonal; that is, that H * í = (| íi? | - + | .O :: | =) I. This generates a received, modified vector, where ? = H *. This leads to c = arg rnm | r - (JA,) 2 + |? , | 2) • cj. (9) Therefore, the next thing is that using a simple linear combination, the decoding rule of equation (9) is reduced to two independent decoding rules, and much simpler for Ci and C? . By using a signaling constellation with the constellation points 2b, this reduces the number of decoding metrics that have to be calculated for ML decoding from 2 to 2 x 2. When the receiver 20 uses the receiving antennas M, the vector received in the antenna m is rm = Hm-c + 77n ,, (io; where the matrix of the Hm channel is defined as H_ = (H) LA-- ~ hlm In this case, the optimal ML decoding rule is '«SfSh -H-H 2) and the corresponding uncertainty? c, of the decoded symbols c is defined by (13) As above, in the case of the receiver antennas M, the decoding rule can be simplified by the pre-multiplexing of the signal received by H *. As indicated above, Figure 1 shows two terminal units 10 and 30, and the problem that needs to be addressed is the detection made in the base station receiver when the two terminal units transmit synchronously over the same frequency channel and of time . In the following notation, g_.? denotes the signal fading channel between the transmitting antenna 31 and the receiving antenna 21, g12 denotes the channel between the antenna 31 and the antenna 22, g2_ denotes the channel between the antenna 32 and the antenna 21, and g2 denotes the channel between antenna 32 and antenna 22. Also,. { c?, c2} Y . { s?, s2} denote the two symbols transmitted from the terminal units 10 and 30, respectively. In the receiver 20, the signals received over the two consecutive symbol periods in the receiving antenna 21, r__ and rí 2, are r "= A ,, C? + AÍ.C2 + S? A + &A + * / "(14) r12 - -A, + h" cl - g, 2 + g2fy 4- 1? ,,. (fifteen) Defining, r_ = [r11rlz *] t, c = [ci c2] r, s = [s? s2] t, and n, = [? n * 2] T equations (14) and (15) can be rewritten in the matrix form as indicated below ri = H_-c + G_-s + n_, (16) where the channel matrices H_ and d, between the transmitting units 10 and 30 and the receiving antenna 21 are given by The vector ni = [11, tj? , *] t is a complex Gaussian probability vector with zero means and covariant No I. Similarly, the signals received over the two consecutive symbol periods in the receiving antenna 22, r2 ?, and r2, are ?! = * ?? + AzA + £ 12 *. + Sns? + ?? 06) rn = -A2C2 +? 2A ~ ft? + 821 + *? 22 • (17) In a similar way, defining r_. = [r2 ?, r22 *] t and ni = [? 2? , c. *] t equations (16) and (17) can be rewritten as r2 = H2-c + G2-s + n2, (18) where the matrices of channel H._ and are given by Equations (14) and (18) can be combined to give the matrix form Cancellation of Minimum Intermediate Square Error (MMSEIC) When searching to detect and decode the signals. { c?, c_ ..}. To minimize a square error criterion, the purpose is to find a linear combination of the received signals such that the square average error in the detection of the signals is minimized. { ci, c_} . In general terms, this can be expressed by a cost error function that will be minimized, such as the function, (twenty-one) where r = [r? r_ r .. r.] r = [rn rí 2 r.i r223r. One can notice that a minimum is certainly reached when and ß are equal to 0, but that, of course, is not desired. Therefore, any of ß? or from ß2 is indicated to 1. When ß? indicated at 1, the following minimization criterion is obtained from equation (40) Jt «*, M -c«, (flft), (22) where?: «Iß.], ap.a_3.« i4 »-A] ß [«? - / 52] and rj = [rr c_] ', From this it can be seen that where 0 = [0 0 0 0] •. What is needed is to select céi so that the expected value of the expression in equation (22) is minimized. That is, select 5 ??, to minimize £. { -/,(2,)} Taking the partial derivative with respect to% and setting it to zero results in where M = HH * + -i, r is the signal to the noise ratio, I is the 4 by 4 identity matrix, hi, is the first column of H, and h2 is the second column of H. It is continued that (M- 2A *) _ 1h ?. (26) It can be shown that -. . AA-? Xhh2.hh. * MM- ~? (M-h2A * = M .- M '"i (27) l - h2M ~ l¡ which gives From the matrix structure H it can be easily verified that hi and h2 are orthogonal. Using this fact and the matrix structure M, it can be shown that ß2 = 0 (29) Now, the MMSE IC solution given by equations (29) and (30) will minimize the mean square error in ci without considering c2. Considering the alternative cost function when ß2 is indicated at 1, a similar analysis leads to the conclusion that: 3i: - M "1 !!: 32; In this case, the MMSE IC solution given by equations (31) and (32) will minimize the mean square error in c_ without considering Ci. Accordingly, from equations (29) - (32), it can easily be seen that the MMSE interference canceller for the signals of the terminal unit 10 will consist of two different groups of weights cti, and a2 for ci and c_, respectively. As expected, the weights can be obtained to decode the signals from the terminal 30 in a similar manner. Thus, the decoding of the signals from the terminal units 10 and 30 can be performed with a single subprogram of MMSE.DECODE in the decoder 25 as indicated below: (c,? f) = MMSE.DECODE (rl, r2, H "H2, G1, G2, r). { T = fr rf [h "h'2t h22 h¡j] T = first column of H [h2l -hj, hjj -h¡,] = second column of H With this subprogram, s and c can be estimated, as follows (c,?) = MMSE.DECODE (r1, r2, H, J? 2, G., C2, T) (33) (s,?) = MMSE.DECODE (r?, r2, G ,, G2. H ,, H > r) (34) Further improvement can be made using a two-stage interference cancellation process. In this two-stage process, the receiver decodes the signals from both terminals using the MMSE.DECODE subprogram described above. Assuming that the symbols of the terminal unit 10 c0 have been decoded correctly, the receiver can then perfectly cancel the contribution of the terminal unit 10 in the received signal vectors r__ and r_. The receiver then uses xi and x_., The received signal vectors after canceling the signals from the terminal unit 10, decode again the symbols s_ from the terminal unit 30 using the optimal decoding rule ML in the equation (10). ). Assuming that the symbols have been decoded correctly from the terminal unit 10, the embodiment for the terminal unit 30 will be equivalent to 2 transmit antennas and 2 receiver antennas (which is equivalent to the MRC diversity of 4 branches). The receiver then repeats the previous stage which assumes that the symbols of the terminal unit 30 §? they have been decoded using the MMSE.DECODE applet correctly. As previously, the receiver cancels the contribution of the terminal unit 30 in the received signal vectors ri and uses y: e y_, the received signal vectors then cancel the signals of the terminal unit 30, to return to decode the symbols of the unit terminal 10 6? using the optimal ML decoding rule in equation (10). Also as before, assuming that the symbols of the terminal unit 30 have been decoded correctly, the embodiment for the terminal unit 10 will be equivalent to two transmit antennas 2 and 2 receiver antennas. Allowing ?? =? c +? S and? i =? c +? S, denotes the global uncertainty for s0 and s0 and for ßi and if, respectively, the receiver compares the global non-certainty and chooses the pair (c, on the other hand) , s) yes? ,, <; ? i, and (c?, s?). The two stage interference cancellation and the ML decoding algorithm are presented later in the pseudo code II subprogram .MMSE.DECODE. (c, s) = p.DECODE. { r1) r2, Ht.H2.GpG2, r). { (c0,? M) = MMSE.DECODE (r1, r2, H1, H?, Gl, G2, r) i, = arg min (F (s)),? ? = F (s) (s ^?,.,) = MMSE.DECODE. { r "r2, G" G2.H] ,. J2.r) F (c) = | yJ-Hl-eSa + || y2-H2.cf c, = ar min (F (c)),? cl = F (c) < «C Yes:? "+?") < (? C, I +? JI1) (c, S) = (£ f) additionally (c, s) = (£ ,, §,)} With the appreciation of the above-described theoretical background, it is understood that the improved embodiment is achieved by dedicating the space-time blocking coding to the interference cancellation and ML decoding, while another coding scheme can be used to overcome the degradations caused in the channel, such as signal fading. Accordingly, each of the transmitters in Figure 1 includes a channel encoder (14 and 34, respectively) that is interposed between the input signal and the encoder by space-time blocking of the transmitter. The channel coders 14 and 24 may employ any conventional channel error correction code (e.g., a link code, or a convolution code). In receiver 20, the internal space time blocking code is decoded in element 26 and used to suppress interference from the various co-channel terminals, using the MMSE process described above. Element 26 forms the two interference cancellation vectors aj;, and a_? corresponding to some terminal, i, and the element 27 forms the two decision variables These decisions, however, are used as the flexible decisions of the transmitted information symbols, and are fed to the channel decoder 8, which is a conventional decoder which corresponds to the type of coding performed in the channel coders 14 and 34. Therefore, in the arrangement described in Figure 1, the structure of an internal coder is used for interference suppression, such that several co-channel terminals they can operate simultaneously while diversity is provided. The output of the internal space time decoder forms the input to the decoder of the external encoder, which decides on the transmitted information while providing protection against channel errors. Figure 2 presents an arrangement for increasing the speed or data output in wireless systems. In Figure 2, the information to be transmitted is demultiplexed in the element 40 in two flows. One flow is applied to the channel encoder 41, and the other flow is applied to the channel encoder 51. The output of the channel encoder 41 is applied to the encoder by space time blocking 42, and then to the pulse and map formation circuit 43 and to the antennas 44 and 45. Similarly, the output of the channel encoder 51 is applied to the encoder by space time blocking 52, and then to the pulse and map formation circuit 53 and to the antennas 54 and 45. Generalizing, the symbols The information of a transmission terminal is divided into parallel flows L. The stream 1 then encodes, using a channel code with the speed .R. and then encoded with a space-time lock encoder with the N transmit antennas.

The coding rates may be the same, but an advantage is increased when the coding rates are chosen as R ± > R2 > > RL. In this case, the symbols transmitted in stream 1 will have better immunity against channel errors as compared to the symbols transmitted in the stream u where u > 1 . It is assumed that the base station receiver is equipped with at least the receiving antennas L. The base station receiver treats each stream as a different user and uses the repetitive interference cancellation techniques described above, or some of those described in the request 163 already mentioned. Since the first stream has the lowest coding rate Rl r it has the best immunity against channel errors and the highest probability of being error-free. The receiver then uses the decoded symbol of stream 1 to subtract the contributions of the first stream in the total received signals, while the remaining L-1 flows are decoded. By decoding the remaining flows L-1, the decoder first decodes the signals of the second flow, since it has the best immunity against channel errors between the remaining flows L-1 (due to its lower speed, R2 from among the remaining flows). Then the receiver uses the decoded symbols for the second stream to cancel its contribution to the received signal. This process is repeated until all flows are decoded. It can be shown that, in this case, the output of the system is given by (36) / - i where FER_ is the error frame of the flow rate 1 It is noted that in relation to this date, the best method known by the applicant to carry out the aforementioned invention, is the conventional one for the manufacture of the objects or products to which it refers.

Claims (14)

2. Having described the invention as above, the content of the following claims is claimed as property: 1. An arrangement characterized in that it comprises; a channel code encoder responsive to an applied input signal, a space time encoder responsive to the output signal of the channel code encoder; and a modulator responsive to the space time encoder. The arrangement according to claim 1, characterized in that it additionally comprises a pulse-forming circuit and at least two antennas for transmitting a space-time coded signal created by a space time coder and modulated by the modulator.
3. 3. A transmitter characterized in that it comprises: a demultiplexer responsive to an input signal applied to develop a plurality of at least two signal streams, and a similar plurality of time encoding transmitters / channel coding, each responsive to a signal flow different from the plurality of signal flows.
4. The transmitter according to claim 3, characterized in that each of the space coding / channel coding transmitters comprises: a speed channel encoder R1 r a space time encoder sensitive to the output signal of the transmitter channel code encoder, a modulator responsive to the space time encoder, a pulse formation circuit responsive to the modulator, and at least two antennas to transmit a time-encoded signal space created by the space time encoder, modulated by the modulator, and conditioned by the pulse formation circuit.
5. The transmitter according to claim 4, characterized in that the demultiplexer develops a plurality L of signal flows, where the channel coders in the space-time coding / L-channel coding transmitters develop the speeds of? _, i = 1,2 ..., L, which are not identical to each other.
6. 6. The transmitter according to claim 4, characterized in that the demultiplexer develops a plurality L of signal flows, where the channel coders in the space-time coding / L-channel coding transmitters develop the speeds R i = 1,2 , .., L, such that _R? > R2 > > RL
7. 7. The transmitter according to claim 1, characterized in that the channel code encoder performs link coding.
8. 8. The transmitter according to claim 1, characterized in that the channel code encoder performs circumvolutional coding.
9. 9. A receiver characterized in that it comprises: a time-coded signal detector; and a decoder for decoding a channel code encoded signal that is included in the detector output signals.
10. The receiver according to claim 9, characterized in that the detector employs an MMSE IC decoder.
11. The receiver according to claim 9, characterized in that the detector employs a two-step algorithm to develop a vector of weights to cancel the interference signals from the terminals instead of a given terminal whose signal is detected.
12. 12. The receiver according to claim 11, characterized in that the two-step algorithm is: (c, s) = p.DECODE (r "r2, H1, H2, G1, GJ.r). { (cß.? ^) = MMSE.DECODE (r., r2, H..H2, G1, GJ, r) x ^ r. -H, c,, x2 = r2 -H2 cß sß = arg min (F (s)),?,., = F (s) ($ 1,? L) = MMSE.DECODE (rJ, r2, GI, Ga, H1, H2 > r) F (c) = || 1 -H1 -c) + || y2 -H2 .c | c, = argmin (F (c)),? ^ = F (C) c.C

    (c, s) = (£., § ")

    also (c, s) = (c ,, sl)}
13. 13. The receiver according to claim 9, characterized in that the decoder for decoding a channel code is a link decoder.
14. 14. The receiver according to claim 9, characterized in that the decoder for decoding a channel code is a circumvolutional decoder.

    SUMMARY OF THE INVENTION

    Improved execution is achieved by combining channel coding with the principles of time-space coding. With the synchronized terminal units K transmitting on the antennas N to the base station having the M receiving antennas $ m (GK), the increased capacity of the system and the improved execution are obtained using a linked coding scheme where the internal code is a code by space-time lock and the external code is a conventional channel error correction code. The information symbols are first encoded using a conventional channel code, and the resulting signals are encoded using a code by space-time blocking. In the receiver, the internal space-time blocking code is used to suppress the interference of the other co-channel terminals and flexible decisions are made on the transmitted symbols. The following channel decoding makes the weight decisions on the transmitted symbols. The increased data rate is achieved by, effectively, dividing the proportion of incoming data into multiple channels, and each channel is transmitted over, its own terminal.