MXPA00009996A - Differential coding and carrier recovery for multicarrier systems - Google Patents

Abstract

A method of mapping information onto at least two simultaneous carriers (202, 206, 208) having different frequencies in a multi-carrier modulation system involves the step of controlling respective parameters of the at least two carriers such that the information is differential encoded. A method of de-mapping information based on at least two simultaneous encoded carriers having different frequencies in a multi-carrier demodulation system comprises the step of recovering the information by differential decoding (142) of respective parameters of the at least two carriers. In a method of performing an echo phase offset correction in a multi-carrier demodulation system, phase shifts are differential phase decoded (142) based on a phase difference between simultaneous carriers having different frequencies. An echo phase offset is determined for each decoded phase shift by eliminating (500) phase shift uncertainties corresponding to codeable phase shifts from the decoded phase shift. The echo phase offsets are averaged (520) in order to generate an averaged offset. Finally, each decoded phase shift is corrected (524) based on the averaged offset.

Description

CORRECTION OF COMPENSATION OF THE ECO PHASE IN A MULTIPLE CARRIER DEMODULATION SYSTEM FIELD OF THE INVENTION The present invention relates to methods and apparatus for modulation and demodulation in multiple carrier modulation systems (MCM systems) and, in particular, with methods and apparatus for mapping and differential demapping of information on symbol carriers. of modulating multiple carriers in these systems. In addition, the present invention relates to methods and apparatuses for performing an echo phase compensation correction when decoding the encoded information into carriers of multiple carrier modulation symbols in multiple carrier modulation systems. BACKGROUND OF THE INVENTION The present invention relates generally to the transmission of digital data to mobile receivers over multiple path channels with time variants. More specifically, the present invention is particularly useful in multi-path environments with low channel coherence time, i.e., rapidly changing channels. In the preferred embodiments, the present invention can be applied to systems for implementing a multiple carrier modulation scheme. Multiple carrier modulation (MCM) is also known as multiplexing orthogonal frequency division (OFDM). In an MCM transmission system the binary information is represented in the form of a complex spectrum, that is, a different number of complex subcarrier symbols in the frequency domain. In the modulator, the bitstream is represented by a sequence of spectra. Using the inverse Fourier transformation (IFFT) a MCM time domain signal is produced from this sequence of spectra. Figure 7 shows a perspective of the MCM system. In 100 an MCM transmitter is shown. A description of this MCM transmitter can be found, for example in William Y. Zou, Yiyan u, "COFDM: A PERSPECTIVE", Transmission Transactions IEEE, vol. 41, No. 1, March, 1995. A data source 102 provides a serial bit stream 104 to the MCM transmitter. The serial bit stream entering 104 is applied to the bit carrier mapper 106 which produces a sequence of spectra 108 from the serial bit stream entering 104. A fast Fourier transformation (FFT) 110 is performed on the sequence of spectra 108 in order to produce a time domain MCM signal 112. The time domain signal MCM forms the useful MCM symbol of the MCM time signal. To avoid interference between the symbol (ISI) caused by multipath distortion, a unit 114 is provided to insert a guard interval of the fixed length adjacent to the MCM symbols in time. According to the preferred embodiment of the present invention, the last part of the useful MCM symbol is used as the guard interval when placing it in the front part of the useful symbol. The resulting MCM symbol is shown at 115 of Figure 7. A unit 116 is provided to add a reference symbol to each of the numbers of the MCM symbols in order to produce an MCM signal having a frame structure. By using this frame structure comprising useful symbols, guard intervals and reference symbols it is possible to retrieve useful information from the MCM signal on the receiver side. The resulting MCM signal has the structure shown at 118 of Figure 7 applied to the front end of the transmitter 120. Briefly speaking, at the front end of the transmitter 120, a digital / analog conversion and an upconversion of the MCM signal is performed. . From this, the signal MCM is transmitted through a channel 122. Next, the mode of operation of an MCM receiver 130 described with reference to Figure 7 is briefly described. The MCM signal is received at the front end of the receiver 132. At the front end of the receiver 132, the MCM signal is converted downwards and, in addition, a digital / analogue conversion of the converted down signal is performed. The downconverted MCM signal is provided for a frame synchronization unit 134. The frame synchronization unit 134 determines the location of the reference symbol in the MCM symbol. On the basis of the determination of the synchronization unit of the frame 134, the reference symbol extracted by the unit 136 extracts the frame information, ie, the reference symbol, from the MCM symbol that comes from the front end of the receiver 132. After extraction of the reference symbol, the MCM signal is applied to a guard interval removal unit 138. The result of the signal processing performed so far in the MCM receiver are the useful MCM symbols. The output of useful MCM symbols from the guard interval removal unit 138 are provided to a fast Fourier transformation unit 140 in order to provide a sequence of spectra from the useful symbols. From this, a sequence of spectra is provided to a mapper of the bit carrier 142 in which the serial bitstream is retrieved. This serial bit stream is provided to the data socket 144. As is clear from Figure 7, each of the MCM 100 transmitters must contain a device that maps the bitstream transmitted at the amplitudes and / or subcarrier phases. Additionally, in the MCM receiver 130, a device is needed for the inverse operation, i.e., recovery of the transmitted bitstream of the amplitudes and / or phases of the subcarriers. To have a better understanding of the MCM mapping schemes, it is preferable to think about the mapping as it is the assignment of one or more bits to one or more subcarrier symbols in the time plane -frequency. In the following, the term symbol or signal point is used for the complex number representing the amplitude and / or modulation of the phase of a subcarrier in the equivalent baseband. Whenever complex numbers are designated represent all symbols, the term symbol MCM is used. DESCRIPTION OF THE PREVIOUS TECHNOLOGY In principle, two methods for mapping the bit stream in the time-frequency plane are used in the previous technology: A first method is a differential mapping along the time axis. When the differential mapping is used along the time axis, one or more bits are encoded within the phase and / or amplitude turns between two subcarriers of the same center frequency in the adjacent MCM symbols. This coding scheme is shown in Figure 8. The arrows shown between the symbols of the subcarrier correspond to the information encoded in amplitude and / 9 in the phase turns between two subcarrier symbols. A system that applies this mapping scheme is defined in the European Telecommunication Standard ETS 300 401 (EU147-DAB). A system in compliance with this standard uses the Differential Quadrature Phase Twist Manipulation (DQPSK) to encode every two bits at 0, 90, 180, or 270 degrees of the phase difference between two subcarriers of the same center frequency as it is located in the adjacent symbols in time. A second method for mapping the bitstream in the time-frequency plane is a non-differential mapping. When non-differential mapping is used the information carried out on one subcarrier is independent of the information transmitted on any other subcarrier, and the other subcarrier may differ either in frequency, ie the same MCM symbol, or in time, is say, the adjacent MCM symbols. A system to apply this mapping scheme is defined in the European Telecommunication Standard ETS 300 744 (DVB-T). A system in compliance with this standard uses a Quadrature Amplitude Modulation 4.16 or 64 (QAM) to assign bits to the amplitude and phase of a subcarrier. The quality with which the multi-carrier modulated signals are transmitted can be recovered in the receiver depending on the properties of the channel. The most interesting property when transmitting the MCM signals is the time interval in which a mobile channel changes its characteristics considerably. The coherence time of the Tc channel is normally used to determine the time interval in which a mobile channel changes its characteristics considerably. Tc depends on the maximum rotation of the Doppler in the following manner: f Doppler, max = V "fport adora / C (ECUaC l On 1) with v: speed of the mobile receiver in [m / s] f carrier: frequency of the carrier of the RF signal [HZ] c:: speed of light (3.108 m / s) The coherence time of the Tc channel is often defined as Tc \ 50% = 9 or Tc | 2do Def | = 9 (Equation 2) 1 or Diffuser, max 167C J Doppler, max It is clear from the existence of more than one definition that the coherence time of the Tc channel is merely a value of a general rule for the seasonality of the channel. As explained above, the differential mapping of the time axis of the previous technology requires that the mobile channel is almost stationary during several periods of MCM symbols, ie the coherence time Tc of the required channel > > symbol period MCM. The non-differential MCM mapping of the prior technology only requires that the mobile channel be quasi-stationary during a symbol interval, ie the coherence time of the required MCM symbol period channel.

This is how the mapping schemes of the previous technology have specific disadvantages. For differential mapping in the direction of the time axis, the channel must be almost stationary, that is, the channel must not change during the transmission of two adjacent MCM symbols in time. If this requirement is not met, the changes in the induced phase of the channel and the amplitude between the MCM symbols will produce an increase in the bit error rate. With a non-differential mapping, exact knowledge of the phase of each of the subcarriers (ie, coherent reception) is needed. For channels with multiple paths, coherent reception can only be obtained if the response of the channel pulse is known. Therefore, a channel estimate has to be part of the receiver's algorithm. The channel estimate usually needs additional sequences in the transmitted waveform that does not carry information. In case of rapidly changing channels, which need to update the channel estimate in short intervals, the additional air can quickly lead to non-differential mapping insufficiency. P.H. Moose: "Differentially Encoded Multiple Frequency Modulation for Digital Communications", SIGNAL PROCESSING THEORIES AND APPLICATIONS, 18. - 21. September, 1990, pages 1807 - 1810 Amsterdam, NL, teaches a differentially encoded multiple frequency modulation for digital communications . A differential modulation of multiple frequencies is described in which the symbols are differentially coded within each of the bauds between the adjacent tones. In the receiver, after a digital Fourier transformer (DFT), a complex product is formed between the digital frequency coefficient DFT k and the complex conjugate of the DFT coefficient of the digital frequency k-1. From this, the result is multiplied by the correct terms in such a way that the bits of the differentially encoded phase are realigned to the original constellations. Thus, the constellation that follows the differential decoding must correspond to the original constellation. SUMMARY OF THE INVENTION It is an object of the present invention to provide methods and devices for performing a compensation correction of the echo phase in a multiple carrier demodulation system. According to a first aspect, the present invention provides a method for performing an echo phase compensation correction in a multiple carrier demodulation system, comprising the steps of: differential phase that encodes the turns of the phase on the basis of a phase difference between the simultaneous carriers having different frequencies; determining the compensation of the echo phase for each of the decoded phase turns by eliminating phase rotation uncertainties related to the information transmitted from the decoded phase rotation; averaging the compensations of the echo phase in order to generate averaged compensation; and correct the rotation of the decoded phase based on the averaged compensation. According to a second aspect, the present invention provides a method for performing a phase compensation correction in a multiple carrier demodulation system, comprising the steps of: differential phase that decodes the phase turns based on a difference of phase between the simultaneous carriers that have different frequencies, where the turns of the phase define the signal points in a complex plane; the signal points that previously rotate within the sector of the complex plane between -45 ° and +45 A "determination of the parameters of a straight line that approximates the location of the signal points previously rotated in the complex plane; determination of the compensation of phase based on the parameters, and correction of each decoded phase rotation based on the phase compensation According to a third aspect, the present invention provides an echo phase compensation correction device for a system of multiple carrier demodulation, comprising: a differential phase decoder for decoding the phase turns on the basis of a phase difference between the simultaneous carriers having different frequencies, elements for determining an echo phase compensation for each of the Decoded turns of the phase eliminating the phase turns related to the transmitted information for phase rotation decoded; elements to average the echo phase compensations in order to generate averaged compensation; and elements to correct each of the turns of the decoded phase on the basis of phase compensation. The present invention provides methods and devices for performing an echo phase compensation correction, suitable for multi-carrier digital transmission over rapidly changing multipath channels, comprising the differential coding of the data along the frequency axis of such that there is no need for the seasonality of the channel to exceed a multiple carrier symbol. When using the mapping process along the frequency axis, it is preferred to use a receiver algorithm that will correct the offsets of the symbol phase that can be caused by channel echoes. The mapping scheme together with the frequency axis of the modulation of the multiple carriers deliver the transmission to some extent independent of the rapid changes of the multi-path channel without introducing a large air space to support the channel estimate. Especially systems with high carrier frequencies and / or high speeds of the mobile that the receiving unit drives can benefit from this mapping scheme. Thus, the mapping scheme of a differential coding along the frequency scheme does not exhibit the two problems of the prior art system described above. The mapping scheme is more robust with respect to multi-path channels that rapidly change what happens in mobile high-frequency and / or high-speed receivers. The respective controlled parameters of the sub-conductors are the phases thereof, in such a way that the information is differentially encoded by phase. According to the mapping described above, the mapping is also differential, however, not within the direction of the time axis but within the direction of the frequency axis. Thus, the information is not contained in the phase shift between the adjacent subcarriers in the frequency. Differential mapping along the frequency axis has two advantages when purchased with the mapping schemes of the previous technology. Due to differential mapping, an estimation of the absolute phase of the subcarriers is not required. Therefore, the estimation of the channel and the related airspace are not necessary. By choosing the frequency axis as an address that differentially encodes the information bitstream, the requirement that the channel must be stationary for several MCM symbols can be eliminated. The channel only has to remain unchanged during the current MCM symbol period. Therefore, just as for the non-differential mapping this maintains that the coherence time of the required channel _ > to the symbol period MCM. The present invention provides methods and apparatus for correction of phase distortions that can be caused by channel echoes. As described above, the differential mapping in the direction of the frequency axis solves the problems related to the seasonality of the channel. However, differential mapping in the direction of the frequency axis can create a new problem. In multi-path environments, path echoes after and prior to the main path can lead to systematic phase compensations between subcarriers in the same MCM symbol. In this context, it is thought that the main trajectory is the trajectory echo with the highest energy content. The echo of the main path will determine the position of the FFT window in the receiver of an MCM system. In accordance with the present invention, the information will be contained in a phase shift between the adjacent subcarriers of the same MCM symbol. If not corrected, the compensation of the induced echo phase of the path between the two subcarriers can lead to an increase in the bit error rate. Therefore, the application of the MCM mapping scheme presented in this invention will preferably be used in combination with a correction of the systematic phase compensations of the subcarrier in the case of a multi-path channel. The phase compensation introduced can be explained from the rotation property of the Discrete Fourier Transformer (DFT): (Equation 3) with x [n] sampling of the time domain signal (0 <n <Nl) X [n] frequency domain signal transformed DFT (...) N: cyclic rotation of the DFT window in time m : length of the DFT rotation in the time domain Equation 3 shows that in a multi-path channel, the following echoes of the main path will produce phase compensation dependent on the subcarrier. After demapping the differential in the direction of the frequency axis in the receiver, a compensation of the two surrounding symbols remains. Because the induced phase compensations of the channel between the differentially demodulated symbols are systematic errors, these can be corrected by an algorithm. In the context of the following specification, the algorithms that help correct phase rotation are called Echo Phase Compensation Correction algorithms (EPOC). These two algorithms are described as preferred embodiments for the correction of phase distortions that can be caused by channel echoes. These algorithms produce sufficient detection security to map the MCM frequency axis even in channels with echoes close to the guard interval limits. In principle, an EPOC algorithm must calculate the compensation of the induced echo phase from the constellation of signal space following the differential demodulation and subsequently correct this phase compensation. BRIEF DESCRIPTION OF THE DRAWINGS Next, preferred embodiments of the present invention are explained in detail on the basis of the accompanying drawings, in which: Figure 1 shows a schematic view depicting a mapping scheme used in accordance with the invention; Figure 2 shows a functional block diagram of an embodiment of a mapping device; Figures 3A and 3B show scattered diagrams of the differential demapping output of an MCM receiver to illustrate the effect of a compensation correction of 1 to echo phase. Figure 4 shows a schematic block diagram for illustrating the position and functionality of an echo phase compensation correction unit; Figure 5 shows a schematic block diagram of an embodiment of a compensation correction device for echo phase according to the present invention; Figure 6 shows schematic views to illustrate a projection made by another embodiment of the compensation correction device for the echo phase of the present invention; Figure 7 shows a schematic block diagram of a generic multiple carrier modulation system; and Figure 8 shows a schematic view representing a differential mapping scheme of the prior technology. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS In a preferred embodiment of the present invention, it is applied to an MCM system shown in Figure 7. With respect to this MCM system, the present invention relates to the carrier mapper of bit 105 of transmitter MCM 100 and bit carrier mapper 142 of MCM receiver 130, shown with a shaded background in FIG. 7. A preferred embodiment of the inventive mapping scheme used by the carrier mapper of sample bits 106 in Figure 1. A number of MCM symbols 200 is shown in Figure 1. Each MCM symbol 200 comprises a number of subcarrier symbols 202. Arrows 204 in Figure 1 illustrate the coded information between the two symbols of the subcarrier 202. As can be seen from the arrows 204, the bit carrier mapper 106 uses a difference mapping within an MCM symbol throughout of the direction of the frequency axis. In an embodiment shown in Figure 1, the first subcarrier (k = 0) in an MCM 200 symbol is used as a reference subcarrier 206 (shaded) in such a way that the information is coded between the reference subcarrier and the first active carrier 208. The other information of an MCM 200 symbol is coded between active carriers, respectively. Thus, there is an absolute reference phase for each of the MCM symbols. According to Figure 1, this absolute phase reference is supplied through a reference symbol inserted into each of the MCM symbols (k = 0). The reference symbol may have a constant phase for all MCM symbols or a phase that varies from the MCM symbol to the MCM symbol. A variant phase can be obtained by replicating the phase from the last subcarrier of the preceding MCM symbol in time. In Figure 2 a preferred embodiment of a device for performing a differential mapping along the frequency axis is shown. With reference to Figure 2, the assembly of the MCM symbols in the frequency domain is described using a differential mapping along the frequency axis according to the present invention. Figure 2 shows the assembly of an MCM symbol with the following parameters: NFFT designates the number of complex coefficients of the discrete Fourier transformation, the number of subcarriers respectively. K designates the number of active carriers. The reference carrier is not included in the count for K. According to Figure 2, quadrature phase shift manipulation (QPSK) is used for the mapping of the bit stream in the complex symbols. However, other M-ary mapping schemes (MPSK) similar to 2-PSK, 8-PSK, 16-QAM, 16-APSK, 64-APSK, etc. are also possible. In addition, for the ease of filtering and minimizing false duplication effects some subcarriers are not used to encode the information in the device shown in Figure 2. These subcarriers, which are set to zero, constitute so-called guard bands in the upper and lower edges of the MCM signal spectrum. Also, for the ease of filtering and minimizing false duplication effects some subcarriers that are not used to encode the information in the device shown in Figure 3. These subcarriers, which are set to zero, constitute the so-called guard bands in the upper and lower edges of the MCM signal spectrum. The "Gray" (gray) mapping is used to prevent differential detection phase errors of less than 135 degrees from causing double bit errors in the receiver. The differential phase encoding the K-phases is carried out in a differential phase encoder 222. In this process step, the K pi [k] phases generated by the Gray QPSK mapper are differentially encoded. Mainly, a feedback loop 224 calculates an accumulated sum on all the phases K. As a starting point for the first calculation (k = 0) the phase of the reference carrier 226 is used. A switch 228 is provided in order to provide the absolute phase of the reference subcarrier 226 or the coded information phase in the preceding subcarrier (ie, (ie, z "1, where z" 1 denotes the unit delay operator) of the subcarrier to a coincidence point 230. At the output of the differential phase encoder 222, the information of the theta phase [k] with which the respective subcarriers will be encoded is provided In the preferred embodiments of the present invention, the subcarriers of an MCM symbol are spaced equally in the direction of the frequency axis The output of the differential phase encoder 222 is connected to a unit 232 to generate the symbols of the complex subcarrier u using the information of the theta phase [k]. Up to this point, the differentially encoded K phases are converted into complex symbols through multiplication with the f actor ß3 [2 * pi teta [] + PHi p (Equation 4) where the factor designates a factor scale and PHI designates an additional angle. The scale factor and the additional PHI angle is optional. By choosing PHI = 45 ° a constellation of DQPSK signal rotated at 45 ° can be obtained.

Finally, the assembly of an MCM symbol is performed in an assembly unit 234. An MCM symbol comprising subcarriers NFFT is assembled from the symbols of the guard band NFPT-K-1 that are "zero", a subcarrier symbol reference and the subcarrier symbols DQPSK K. This is why the assembled symbol MCM 200 is composed of complex values K containing the encoded information, two guard bands on both sides of the complex values NFFT and a subcarrier symbol of reference The MCM symbol has been assembled in a frequency domain. For transformation within the time domain, an inverse discrete Fourier transform (IDFT) of the output of the assembly unit 234 is performed through a transformer 236. In the embodiments of the present invention, the transformer 236 is adapted to perform a fast Fourier transformation (FFT). An additional method of the MCM signal in the transmitter as in the receiver is described above with reference to Figure 7. In the receiver a demapping device 142 (Figure 7) is needed to reverse the operations of the mapping device described above. with reference to Figure 3. The implementation of the mapping device is simple, and therefore, need not be described in detail in the present. However, the systematic phase turns that are contained from the echoes in the multipath environments may occur between the subcarriers in the same MCM symbol. These phase compensations can cause bit errors when the MCM symbol in the receiver is demodulated. Thus, it is preferred to use an algorithm to correct the systematic phase turns that are contained from the echoes in multipath environments. Preferred embodiments of the echo phase compensation correction algorithms are explained below with reference to FIGS. 3 to 6. In FIGS. 3A and 3B, scattered diagrams are shown at the output of a differential demapping device. of an MCM receiver. As can be seen in Figure 3A, the systematic phase turns between the subcarriers in the same MCM symbol cause a rotation of the phase shifts demodulated with respect to the axis of the complex coordinate system. In Figure 3B, the turns of the demodulated phase with respect to the axis of the complex coordinated system are shown. In Figure 3B, the demodulated phase rotates after having performed the echo phase compensation correction. Now, the positions of the signal points are substantially on the axis of the complex coordinate system. These positions correspond to the modulated phase turns of 0o, 90o, 180o and 270o, respectively. An echo phase compensation correction algorithm (EPOC algorithm) must calculate the compensation of the induced phase of the echo from the constellation of the signal space following the differential demodulation and subsequently correct this phase compensation. For purposes of illustration, one can think of the simplest possible algorithm to eliminate the phase of the symbol before calculating the element of all phases of the subcarriers. To illustrate the effect of this EPCO algorithm, reference should be made to the two scattered diagrams of the symbols of the subcarriers contained in an MCM symbol in Figures 3A and 3B. These scattered diagrams have been obtained as a result of an MCM simulation. For the simulation, a channel that can be displayed typically in simple frequency networks has been used. The echoes of this channel stretch to the limits of the guard interval MCM. The guard interval was chosen to be 25% of the duration of the MCM symbol in this case. Figure 4 depicts a block diagram to illustrate the position and functionality of an echo phase compensation correction device in an MCM receiver. The signal from an MCM transmitter is transmitted through the channel 122 (Figures 4 and 7) and is received at the front end of the receiver 132 of the MCM receiver. The signal processing between the front end of the receiver and the fast Fourier transformer 140 has been omitted in Figure 4. The output of the fast Fourier transformer is applied to the demapping machine, which performs a differential demapping along the frequency axis. The output of the demayer is the respective phase turns for the subcarriers. The phase offsets of these twists phase that are caused by echoes in environments multipath displayed through a block 400 in Figure 4 showing an example of a spread of subcarrier symbols diagram without correction echo phase compensation. The output of the demapper 142 is applied to the input of a correction device 402. The phase compensation device correction phase 402 uses an EPOC echo algorithm in order to eliminate the phase offsets in the output echo of the demapping 142. The result is shown in block 404 of FIG. 4, that is, only the encoded phase turns, 0o, 90o, 180o or 270o are present at the output of the correction device 402. The The output of the correction device 402 forms the signal for the metric calculation that is performed in order to recover the bit stream representing the transmitted information. A first embodiment of an EPOC algorithm and a device that performs the same thing as described now with reference to Figure 5. The first embodiment of the algorithm EPOC starts from the assumption that each received differentially decoded complex symbol is rotated at an angle due to echoes in the multipath channel. For the subcarriers an equal spacing in frequency is assumed since this represents a preferred embodiment. If the subcarriers are not equally spaced in frequency, a correction factor must be entered in the EPOC algorithm. Figure 5 shows correction device 402 (Figure 4) to perform the first embodiment of an EPOC algorithm. From the output of the demagover 142 containing an echo phase compensation as shown, for example, in FIG. 3A, the related phase turns to transmit the information should be discarded. At this point, the output of the demapping unit 142 is applied to a disposal unit 500. In case of a DQPSK mapping, the disposal unit can perform an operation "(.) 4". The unit 500 projects all the symbols received in the first quadrant. Therefore, the phase turns related to the transmitted information are removed from the turns of the phase representing the symbols of the subcarrier. The same effect can be achieved with an operation of a module - 4. Having removed the information related to the phases of the symbol in unit 500, the first approach to obtain an estimate will be simply to calculate the average value over all the phases of symbol of an MCM symbol. However, it is preferred to make a threshold decision before determining the average value over all phases of the symbol of an MCM symbol. Due to Rayleigh fading some of the received symbols may contribute unreliable information for the determination of the compensation of the echo phase, therefore, depending on the absolute value of a symbol, a threshold decision is made in order to determine whether the symbol should contribute to the estimation of the compensation of the phase or not. Thus, in the embodiment shown in Figure 5, a threshold decision unit 510 is included. After the unit 500 the absolute value and the argument of the differentially decoded symbol are calculated in the respective calculation units 512 and 514. Depending on the absolute value of a respective symbol, a control signal is derived. This control signal is compared to a threshold value in a decision circuit 516. If the absolute value, ie the control signal thereof, is less than a certain threshold, the decision circuit 516 replaces the value of the angle that goes to the average operation through a value equal to zero. At this end, a switch is provided in order to disconnect the output of the calculation unit from argument 514 from the input of an additional processing stage and connect the input of the additional processing stage to a unit 518 that provides an output constant of "zero".

An average 520 unit is provided in order to calculate a mean value based on the phase compensations fi determined by the symbols of the individual subcarrier of the MCM symbol as follows: v: = V fa (Equation 5) In the unit of average 520, the sum is performed on the addends K. The output of the average unit 520 is provided to hold the unit 522 which retains the output of the unit 520 K times average. The output of the holding unit 522 is connected to a phase rotation unit 524 which performs the correction of the phase compensations of the points of the complex signal K on the basis of an average value f. The rotation unit of phase 524 performs the correction of the phase compensations when making use of the following equation: = Vk. e "jf (Equation 6) In this equation, v'k designates the differentially corrected decoded symbols of the K phase for input to the soft metric calculation, where vk designates the output symbols, while a channel that is almost stationary for the duration of one of the symbols MCM can be assumed, using the average value of all the subcarriers of one of the symbols MCM will provide the correct results. 527 in order to damp the complex signal points until the average value of the phase compensations for an MCM symbol is determined. The output of the phase rotation unit 524 is applied to the additional processing step 526 to perform the soft metric calculation. With respect to the results of the correction of the echo phase compensation, reference is again made to Figures 3A and 3B. The two graphs contain a simulation including the first embodiment of the correction algorithms of the phase compensation. of echo described above. At the moment of the instantaneous taking of the scattered diagram shown in the left part of Figure 3A, obviously the channel distorts the constellation in such a way that a simple rotation of angle is a valid assumption. As shown in the right part of Figure 3B, the constellation of the signal can be rotated backwards of the axis by applying the average value determined for the rotation of the differentially detected symbols. A second embodiment of an algorithm of a phase compensation correction is described below. This second embodiment can preferably be used in relation to multiple path channels having up to two strong path echoes. The algorithm of the second embodiment is more complex than the algorithm of the first embodiment. The following is a mathematical derivation of the second embodiment of a method for compensation of the echo phase. The following assumptions can be made in order to facilitate the explanation of the second embodiment of an EPOC algorithm. In this embodiment, the guard interval of the MCM signal is assumed to be at least as long as the impulse response h [q]. q = 0, 1, ..., Qh-1 of the multipath channel. In the transmitter each of the MCM symbols is assembled using the previous frequency axis mapping. The reference symbol of the subcarrier is equal to l, that is, the phase rotation is 0 degrees. The optional phase shift PHI is equal to zero, ie the constellation of the constellation of the DQPSK signal is not rotated. Using an equation, this can be expressed as: ak 0 ak-? A? Nck (Equation 7) with k: index k = l, 2, ..., k of the active subcarrier, - -: increment symbol in the complex phase; m = 0, l, 2,3, is the number of the QPSK symbol that is derived from the coded pairs "Gray" (2-bit gray) aO = 1: symbol of the reference subcarrier In the DFT output of the receiver the variables of decision e = akHk (Equation 8) are obtained with , - "! [-] -AJ" (Equation 9) with the DFT of the impulse response of the channel h [q] in the position k. With | to. | 2 = 1 the differential demodulation efficiencies (Equation 10) For the receiver, an additional phase term f is introduced, which can be used to correct the systematic phase compensation caused by the channel. Therefore the variable of the final decision in the receiver is * í '- * i (Equation 11) As can be seen from Equation 11, the useful information a? nck is also weighted with the product e: fk. Hk. H * k_? (rotation and effective channel transfer function). This product should be valued as real for error-free detection. Considering this, it is better to choose the angle of rotation equal to the neve argument of Hk H * k- ?. To derive the algorithm for 2-path channels, the nature of Hk. H * k-? It is investid in the next section. It is assumed that the 2-path channel exhibits two echoes with an energy content that is not equal to zero, that is, at least two dominant echoes. This assumption produces the impulse response. ? [< ?] - cAÍ?] + cAb "lo] (Equation 12) with C?, c2: complex coefficients that represent the echoes of the trajectory; q0: delay of the second echo of the trajectory with respect to the first echo of the trajectory; d0 : Pulse Dirac; d0 [k] = 1 for k = 0 d0 [k] = 0 any other The channel transfer function is obtained by applying a DFT to Equation 12: (Equation 13) With equation 13 the effective transfer function for differential demodulation along the frequency axis is: (Equation 14) Assuming a 2-channel noise-free channel, it can be seen from Equation 14 that the symbols on the receiver side are located on a straight line in the event that the symbol 1 + jO has been sent (see the assumption previous). This straight line can be characterized by a point -i .2 - '*, KG + e- (Equation 15) and the vector -i- *. -ICJCJ • é (Equation 16) that determines its direction. With the above assumptions, the following geometric derivation is performed. A more suitable notation for the geometric derivation of the second embodiment of an EPOC algorithm is obtained if the real part of the complex plane is designated as x = Re. { z} , the imaginary part as y = Im. { z} , respectively, that is, z = x + jy. With this new annotation, the straight line, in which the received symbols lie in the case of a channel of two noise-free paths is f (*) - a + - b - x (Equation 17) with a - In?.,} - ^ 4 • Hc (Equation 18) Rete Y b. (Equation 19) The additional noise will scatter the symbols around the straight line given by Equations 17 through 19. In this case Equation 19 is the regression curve of a grouping of symbols.

For the geometric derivation of the second embodiment of an EPOC algorithm, the angle f of Equation 11 is chosen to be a function of the square distance of the symbol considered from the origin: fk = f? (Equation 20) Equation 20 shows that the entire signal space is distorted (torsion), however, preserving the distances from the origin. For the derivation of the algorithm of the second embodiment fk (.) Has to be determined in such a way that all the VA decision variables (without assuming noise) lie on the real axis: y (Equation 21) imlfx + if (x) ) • ej * H = 0 The additional transformations of Equation 21 lead to a quadratic equation that must be solved to obtain the solution of fk. In the case of a two-path channel, the echo phase compensation correction for a given decision variable vk is v = ck. e3 ^ (Equation 22) with (Equation 23) From these two possible solutions the quadratic equation mentioned above, Equation 23 is the solution that can not cause an additional phase shift of 180 degrees. The two graphs in Figure 15 show the projection of the EPOC algorithm of the second embodiment for a quadrant of the complex plane. Described here is the quadratic grid in the | arg (z) \ < _ p / 4 and the straight line y = f (x) a + b.x with a = -1.0 and b = 0.5 (dotted line). In case of a noise-free channel, all received symbols will lie on the straight line if 1 + jO is sent. The circle shown in the graphs determines the boundary line for the two cases of Equation 23. On the left, Figure 15 shows the situation before the projection, on the right side, Figure 15 shows the situation after applying the projection algorithm. Looking at the left side, one can see that the straight line now lies on the real axis with 2 + jO fixed at the point of the projection. Therefore, it can be concluded that the correction algorithm compensates the echo phase according to whether the second embodiment complies with the designed goal. Before the second embodiment of the EPOC algorithm can be applied, the approach line has to be determined through the received symbols, that is, parameters a and b must be estimated. For this purpose, it is assumed that the received symbols lie in the | arg (z) | < _ p / 4, if 1 + jO is sent. If other symbols than 1 + jO are sent, the module operation can be applied to project all the symbols in the desired sector. Proceeding in this way avoids the need to decide on the symbols at an earlier stage and allows averaging all the signal points of an MCM symbol 8 instead of averaging over only all the points of the signal). For the following calculation rule for the EPOC algorithm of the second embodiment, Xi is used to denote the real part of the signal point i-th and yi for the imaginary part, respectively (i = 1, 2, ..., k). All together, the K values are available for determination. By choosing the least squares method, the straight line to be determined can be obtained by minimizing (a, £ >) = arg min V (y, - [a + b • xA). ". ^ _? . '(^) irí (Equation 24) The solution for Equation 24 can be found in the literature that remains open. This is J (A - *) 'A ¿, = id a = y - x • (Equation 25) with average values 1_ K 1? x = N (Equation 26) l - l iW i-1 If necessary, a more robust estimation method can be applied. However, the exchange will have a much higher computational complexity. To avoid problems with the range in which the projection is applicable, the determination of the straight line must be separated into two parts. First, the centers of the gravity cluster move on the axes, then the signal space becomes distorted. Assuming that a and b are the original parameters of the straight line and ° c is the rotation angle, f (.) = Must be applied with the transformed parameters. b • cos (a) - without (a). i, s. -. / \\ b = 'T-. { -, a = a • sew - b • s? naj cos (a) + b • sin (ar) (Equation 27) In addition to the two EPOC algorithms explained in the previous section, you can designate different algorithms that, however, it will be more possible to exhibit at a higher degree of computational complexity. The new mapping method for Modulation of Multiple Carrier schemes presented here consists mainly of two important aspects. The differential mapping within an MCM symbol together with the direction of the frequency and the echo correction of the channel related to the phase compensation on the subcarriers on the receiver side. The advantage of this new mapping scheme is its strength with respect to multipath channels that change rapidly which can occur at high frequencies and / or at high speeds of mobile receivers.

Claims (18)

1. CLAIMS 1. A method for performing an echo phase compensation correction in a multiple carrier demodulation system, comprising the steps of: differential phase that decodes the phase turns based on a phase difference between the simultaneous carriers that have different frequencies; determination of an echo phase compensation for each of the decoded phase turns by eliminating the uncertainty of rotation of the phase in relation to the information transmitted from the rotation of the decoded phase. averaging the compensations of the echo phase in order to generate an averaged compensation, and correcting each of the decoded phase turns on the basis of averaged offsets.
2. 2. The method according to the Claim 1, wherein the step for decoding the differential phase comprises passing the differential phase by decoding the phase turns based on a phase difference between the simultaneous carriers that are adjacent in the direction of the frequency axis.
3. 3. The method according to claim 1, wherein the step of decoding the differential phase comprises the passage of the differential phase that decodes the phase turns based on the phase differences between at least three simultaneous carriers that are They also spaced in the direction of the frequency axis.
4. 4. The method according to claim 1, further comprising a step for comparing an absolute value of a symbol associated with a respective decoded phase rotation to a threshold, where the phase turns are used which are only associated therewith. symbols that have an absolute value that exceeds the threshold in the average step of the compensations of the echo phase.
5. 5. A method for performing an echo phase compensation correction in a multiple carrier demodulation system, comprising the steps of: the differential phase that decodes the phase turns based on a phase difference between the simultaneous carriers which has different frequencies, where the phase turns define the signal points in a complex plane; turn previously the signal points in the sector of the complex plane between -45 ° and +45 A "determine the parameters a, b of a straight line that approximates the location of the signal points previously rotated in the complex plane, determine the compensation of the phase on the basis of the parameters a, b; and correcting each of the decoded phase turns on the basis of phase compensation 6.
6. A method according to Claim 5, wherein the simultaneous carriers are spaced also in the direction of the frequency axis 7.
7. The method according to the Claim 5, wherein the step pair determining the parameters a, b comprises at least one square method for selecting the parameters that minimize the deviations of the signal points previously rotated from the straight line.
8. The method according to Claim 7, wherein parameters a, b are determined as follows: 2 (A -) • y, b = 'LA? 'a = y - x - b (Equation 25) 1? i? x and l - l and A (Equation 26) where x and y designate the coordinates of the points of the signal in a complex plane, i is an index from 1 to N, and K is the number of the signal points.
9. 9. The method according to the Claim 28, where the phase compensation k is determined as follows: (Equation 23) where Vk is a determinate decision variable.
10. 10. A device for the correction of echo phase compensation for a multiple carrier demodulation system, comprising: a differential phase decoder for decoding phase turns based on a phase difference between the simultaneous carriers that have different frequencies; elements for determining an echo phase compensation for each of the decoded phase turns comprising elements for eliminating the uncertainties of phase rotation related to the information transmitted from a decoded phase rotation; elements to average these echo phase compensations in order to generate averaged compensation; and elements to average the compensations of the echo phase in order to generate an averaged compensation; and 11.
11. The device according to Claim 10, wherein the decoder of the differential phase is adapted to decode the phase turns based on a phase difference between the simultaneous carriers that are adjacent in the direction of the frequency axis.
12. The device according to Claim 10 further comprises elements for comparing an absolute value of a symbol associated with a respective decoded phase rotation with a threshold, where the elements for averaging the phase compensations only use the phase turns. they have associated symbols there that have an absolute value that exceeds that threshold.
13. 13. The device according to Claim 10, wherein the differential phase decoder is adapted to decode phase turns on phase phase differences between at least three simultaneous carriers that are equally spaced in the direction of the frequency axis.
14. 14. An echo phase compensation correction device for a multiple carrier demodulation system comprising: a differential phase decoder for decoding the phase turns based on a phase difference between simultaneous carriers having different frequencies, these phase turns define signal points in a complex plane; elements to previously rotate the signal points within the sector of the complex plane between -45 ° and +45 A elements to determine the parameters, a, b of the straight line that approximates the location of the signal points in the complex plane; elements to determine a phase compensation based on parameters a, b; and elements to correct each of the decoded phase turns on the basis of phase compensation.
15. 15. The device according to the Claim 14, wherein the differential phase decoder comprises elements for decoding the phase turns of at least three simultaneous carriers that are equally spaced in the direction of the frequency axis.
16. 16. The device according to the claim 14, where these elements for determining the parameters a, b comprise elements to perform a method of fewer frames to select the parameters that minimize the deviations of the signal points previously rotated from the straight line. The device according to Claim 16, wherein the elements for determining the parameters a, b calculate the parameters a, b as follows: (Equation 25] And, (A - *) • y. b, a = y - x • b i (*. - * y 1 ? 1 ? x = - y A, y = - y y? M N Ái (Equation 26) where x and y designate the coordinates of the signal points in the complex plane, i is an index of 1 to N, and K is the number of points in the signal. The device according to Claim 17, wherein these elements for determining the phase compensation f calculate the phase compensation fk in the following manner: (Equation 23) where vk is a variable of a given decision,