MXPA00009740A - Cross coupled motor gate drive - Google Patents

Abstract

A motor system including a gate drive for driving a motor. An inverter bridge circuit selectively connects power supply link rails to a winding of the motor for energizing the winding with a motoring current. The bridge circuit has upper and lower power switches connected between the winding and the upper and lower power supply link rails, respectively. Each lower switch corresponds to one of the upper switches on the same side of the winding as the lower switch to define an arm of the bridge circuit. A control circuit generates a motor controlled signal to control the switches. A drive circuit drives the upper switches in response to the state of the corresponding lower switches, which are responsive to the motor control signal. The drive circuit includes a voltage gain element connected to each arm of the bridge circuit that is responsive to current in the respective lower switch for maintaining the corresponding upper switch in its nonconducting state. In a three phase embodiment, a quadrature axis winding corresponds to each phase winding. Each of the quadrature axis windings is in magnetic coupling relation with the rotatable assembly and positioned for generating an output signal representative of angular position of the rotatable assembly. The control circuit generates the motor control signal to control commutation of the phase windings in response to the output signals of the quadrature axis windings.

Description

ENGINE ACTIVATOR COUPLED ENGINE COUPLING BACKGROUND OF THE INVENTION This invention relates generally to engine systems and, in particular, to an improved gate actuator for a single or multi-phase electronically controlled engine. An electronically commutated motor (ECM) of the type described herein has a stator with a plurality of teeth and a rotor with permanent magnets mounted thereon. When the wound coils of wire in the teeth are energized with current, the stator and the rotor interact to produce positive or negative torque, depending on the direction of the current with respect to the polarity of the magnets. In motors of this type, an inverter electronic bridge controls the energization of the stator winding to control the direction and amount of torque produced by the motor as well as to control the speed of the rotor shaft. The inverter bridge commonly has a number of energy interrupting devices to connect the winding or the motor windings to a power supply. Generally, the devices are configured with respect to the upper and lower rails of a power supply link and have a return diode coupled to each of the devices. The application assigned to the same assignee of the present Serial Number 08/865, 135, the complete description of which is hereby incorporated by reference, shows an improved surface motor gate actuator which controls the upper energy switches of the inverter bridge. motor in response to the states of the lower power switches. This gate activator eliminates the use of high voltage gate level switches to activate the inverter bridge and reduces the trip current. As is known in the art, the switching of the motor is usually controlled as a function of the angular position of the motor rotor. The United States of America Patent Serial No. 5, 796, 194 of the same assignee as the present one, and the application Serial Number 09 / 048,946, the complete descriptions of which are hereby incorporated by reference, show an improved means for detect the position of the rotor in a single phase motor with a quadrature axis winding. Although these inventions provide improvements, additional improvements are still desired including reducing the cost of the engine. In particular, the gate activation circuit of the application Serial Number 08/865, 135 employs a power diode that responds to the ignition of the lower power switch and forces the shutter release of the upper breaker to shut off. Then, the diode provides charge current conduction of the winding while fixing the top gate to an acceptable voltage. However, this diode also blocks the flow of the return current. The unimpeded flow of the return current through the internal return diodes of the lower energy switches is desired for a complete operation. Therefore, an improved cross coupling gate activator is desired in this regard. In addition, a gate activator is desired for use at higher energy levels and for use with a multi-phase motor. With respect to the activation of a three-phase motor, there may be no conventional techniques available to detect the angular position of the motor rotor, such as detecting the counter electromotive force (EMF) in the windings. Additionally, conventional driving and regulation schemes can produce undesirable amounts of noise and torque ripple. Therefore, further improvements are also desired in the control of three-phase gate actuators including current regulation and position detection of the quadrature axis based on quadrature axis voltages. The Patent of the United States of North America Number of Series 5,552,685, U.S. Patent Serial Number 5,423,192, U.S. Patent Serial Number 4,933,584, U.S. Patent Serial Number 4,757,603 and U.S. Patent Serial Number 4,757,241, all of which are assigned to the same assignee of the present application and whose entire descriptions are incorporated herein by reference, describe electronically controlled engines. BRIEF DESCRIPTION OF THE INVENTION The invention fulfills the aforementioned needs and solves the deficiencies of the prior art by providing an improved system that includes a cross coupling gate activator to activate a motor. Briefly described, an engine system including aspects of the invention has a stationary assembly that includes at least one winding and a rotary assembly in magnetic coupling relationship to the stationary assembly. The system also includes a power supply link having upper and lower rails supplied by an energy supply. A bridge circuit selectively connects the rails to the winding to energize it with a motor current to produce an electromagnetic field to rotate the rotary assembly relative to the stationary assembly. The bridge circuit has upper and lower power switches connected between the winding and the upper and lower rails, respectively. Each lower switch corresponds to one of the upper switches on the same side of the winding as the lower switch to define an arm of the bridge circuit. Additionally, the system includes a control circuit to generate a motor control signal to control the switches. An activation circuit activates the upper switches in response to the state of the corresponding lower switches, which respond to the motor control signal. The activation circuit includes a voltage gain element connected to each arm of the bridge circuit which responds to the current in the respective lower switch to maintain the corresponding upper switch on the same arm of the bridge circuit in this non-conductive state. Generally, another form of the invention is directed to an inverter bridge to activate a motor. The motor has a stationary assembly that includes at least one winding and a rotating assembly in magnetic coupling relationship to the rotating assembly. The motor also includes a power supply link having upper and lower rails supplied by a power supply and a control circuit for generating a motor control signal to control the switching of the winding. The inverter bridge has upper and lower switches connected between the winding and the upper and lower rails, respectively. Each lower switch corresponds to one of the upper switches on the same side of the winding as the lower switch. The inverter bridge also includes an activation circuit to activate the upper switches in response to the state of the corresponding lower switches, which respond to the motor control signal. The activation circuit has a voltage gain element connected to each of the lower switches that responds to current in the respective lower switch to maintain the corresponding upper switch in its non-conductive state. In this way, the inverter bridge selectively connects the rails to the winding to energize it with a motor current to produce an electromagnetic field to rotate the rotary assembly relative to the stationary assembly. In still another embodiment of the invention, a three-phase motor system has a stationary assembly including three-phase windings and a rotary assembly in magnetic coupling relationship with the stationary assembly. A quadrature axis winding corresponds to each phase winding. Each of the quadrature axis windings is in magnetic coupling relationship with the rotary assembly and positioned to generate an output signal representative of the angular position of the rotating assembly. Additionally, the system includes a control circuit for generating a motor control signal to control the switching of the phase windings in response to the output signals of the quadrature axis windings. Alternatively, the invention may comprise several other methods and systems. Other objects and aspects will be partly evident and partly indicated below. BRIEF DESCRIPTION OF THE DRAWINGS Figure 1 is a block diagram of an engine system in accordance with a preferred embodiment of the invention. Figure 2 illustrates an inverter bridge including a preferred cross coupling gate activator for activating the motor of Figure 1. Figure 3 illustrates an inverter bridge including another preferred cross coupling gate activator for activating the motor of Figure 1. Figures 4A-4C graphically illustrate the exemplary operation of the motor of Figure 1 with respect to speed in accordance with a three-phase electronically commutated motor control based on instantaneous peak current regulation in the negative rail of the inverter bridge. Figures 5A and 5B illustrate graphically example voltages and phase currents with respect to electrical degrees of rotation for the operation of the engine of Figures 4A-4C. Figure 6 graphically illustrates two exemplary phase voltage sequences applied to the motor of Figure 1 to operate a three phase electronically commutated motor. Figures 7A-7C graphically illustrate the exemplary operation of the motor of Figure 1 with respect to the speed of conformity with the control of a three phase electronically commutated motor based on sine wave voltage regulation of the width modulation values Pulse of the phase leg in response to the current observed in the negative rail of the inverter bridge. Figures 8A-8C graphically illustrate the exemplary operation of the motor of Figure 1 with respect to the speed of conformity with the control of a three-phase electronically commutated motor based on trapezoidal voltage regulation of the pulse width modulation values of the phase leg in response to the current observed in the negative rail of the inverter bridge. Figures 9A-9C graphically illustrate voltages, currents, and phase leg workings exemplary of the engine of Figure 1 during the trapezoidal operation. Figure 10 is a flattened side view of the stator of Figure 1, which includes quadrature axis windings that extend through a plurality of stator slots. Figures 1 1 A-1 1 C graphically illustrate voltages, currents and duty cycle of phase leg exemplary of the motor of Figure 1, using the windings of the quadrature axis of Figure 10 for tracking the position of the rotor. Figures 12A-12D graphically illustrate the starting operation of the engine of Figure 1, operating in accordance with the rotor position tracking of the quadrature axis of Figures 1 1 A-1 1 C. The corresponding reference characters they indicate corresponding parts in the drawings. DETAILED DESCRIPTION OF THE PREFERRED MODALITIES With reference now to the drawings, Figure 1 shows a motor system 21 in accordance with a preferred embodiment of the present invention. The system 21 includes a motor, generally designated 23, having a stationary assembly or stator 25 and a rotating assembly, or rotor 27 in magnetic coupling relationship with the stator 25. In the embodiment described herein, the motor 23 is an electronically commutated motor. However, it should be understood that the aspects of the present invention can be applied to any electronically controlled motor or electronic machine commonly activated by an electronic control circuit. These motors include, for example, external rotor motors (that is, motors turned upside down) permanent magnet motors, variable speed and one speed motors, selectable speed motors having a plurality of speeds, brushless direct current motors, electronically commutated motors, motors. switched reluctance and induction motors. In addition, the motors can be multi-phase or single-phase motors and, in any case, these motors can have a split-phase winding or a multi-phase winding. These motors may also provide one or more finite discrete rotor speeds selected by an electrical switch or other control circuit. In a preferred embodiment of the present invention, a motor shaft 31 mechanically connects the rotor 27 to a particular device to be activated, such as a rotating component 33. For example, the rotating component 33 comprises a fan, blower, compressor or the like for use in an air conditioning and ventilation system, heating or cooling system. Although the motor 23 is particularly useful for activating a fan, it should be understood that the motor 23 can be part of a number of different systems for activating other rotating components. In addition, the rotating component 33 may include a connecting mechanism for coupling it to the shaft 31.

Preferably, a user interface, or system control 35, provides control signals from the system to a control circuit 39 via line 41. In this case, the control signals of the system take the form of motor commands representing, for example, on and off commands, desired fan speeds and the like. In response to the control signals of the system, the control circuit 39 then generates motor control signals. As represented by the block diagram of Figure 1, the control circuit 39 provides the motor control signals via the line 43 to electronically control a plurality of gate triggers 45. In turn, the gate triggers 45 provide activation signals via line 51 for switching a plurality of power switches 53, such as bipolar isolated gate transistors, bipolar junction transistors or silicon metal oxide field effect transistors. In addition to providing sufficient voltage (for example 1.5 volts) to activate the power switches 53, the gate triggers 45 also condition the signals provided by the control circuit 39 for optimum operation of the power switches 53. In a preferred embodiment of In the present invention, the control circuit 39 is represented by a microprocessor, microcontroller and / or a specific application integrated circuit (ASIC). A digital signal processor (DSP) can also be used to implement some or all of the functions of the control circuit 39.

An energy supply 57 provides high voltage direct current energy to the switches 53 via the line 59. The power switches 53 then provide power to the motor 23 via the line 61 by selectively switching the power supply 57 in connection with one or both motor windings 63 (see Figure 2) included in the stator 25. Preferably, the power switches 53 energize the motor winding 63 in at least one previously selected sequence to switch the motor 23 in response to the control circuit 39. In this case, the control circuit 39 selectively energizes the power switches 53 to control rotation in the motor 23 as a function of the motor control signals. It should be understood that the power supply 57 can also provide power to operate the control circuit 39. With further reference to Figure 1, a position sensor 65 provides the control circuit 39 with a feedback signal via the line 67 which is representative of the angular position of the rotor 27 relative to the stator 25. For example, one or a quadrature windings 69 (see Figure 10) include the position sensor 65. In general, the position signal has a previously defined angular relationship in relation to the counter electromotive force of the motor (for example in pass or up to 90 out of phase with the back electromotive force of the motor 23) that allows an estimation of the position of the rotor. In a three-phase motor, for example, the control circuit 39 can observe the phase quadrature waveforms of the motor in the windings for position detection. Other position sensors, such as one or more Hall sensors or optical sensors, may be used to provide feedback of the position of the rotor in place of or in addition to the quadrature winding 69 or back electromotive force information. The control circuit 39 preferably generates its control signals as a function of the zero crossings estimated from the emf of the winding 63. As the product of the current and the back electromotive force determines the production of torque in the motor 23, the circuit of control 39 maintains positive torque by energizing winding 63 when the back electromotive force has crossed zero in the direction that will oppose the voltage by energizing them. It is desired that the motor current crosses zero at the moment that the counter-electromotive force of the motor also crosses zero, so that the control circuit 39 preferably switches the motor 23 at a relative angle to the next zero crossing of the back electromotive force. In other words, the control circuit 39 estimates the subsequent counter electromotive force zero crossings based on the detected position of the rotor 27 and generates gate trigger signals on the line 51 to activate the power switches 53 coinciding with or in front of the zero crossings of the estimated counter electromotive force. Therefore, the control circuit 39 generates the motor control signals as a function of the detected position of the rotor 27 represented by the position signal. U.S. Patent No. 5,423, 192 of the same assignee as the present one, describes a preferred means of detecting zero crossings. In operation, the control circuit 39 implements a state machine to generate signals that define the desired switching intervals based on the control signals of the system. In response to motor control, or switching, the signals generated by the control circuit 39, the gate triggers 45 cause the power switches 53 to switch. The resulting motor current preferably matches the load torque demand as a function of a regulated current reference level. By matching the torque load with the torque produced, the motor 23 can operate at a desired speed or torque. In one embodiment, the motor control signals include a series of pulse width modulated cycles, wherein each cycle produces a corresponding switching event of the power switches 53. The current in the winding 63 produces an electromagnetic field to make rotate the rotor 27 of the motor 23. To control the speed of the rotating component 33, the system 21 preferably controls the energy delivered to the load to control the speed of the motor 23. In particular, the system 21 regulates the current in the motor 23 that in turn it regulates the torque, to obtain the desired engine speed when the load and the engine loss demand torque coincide at the desired speed.

As an example of the current regulation scheme, the control circuit 39 defines alternate on and off intervals based on instantaneous current levels in the winding 63. When the measured current reaches a predetermined peak level, the control circuit 39 commands the the power switches 53 that turn off the current for a certain time interval for example, an oscillator or an off-timer. In the alternative, the control circuit 39 performs voltage regulation using a variable duty cycle signal. The work cycle can vary from, for example, 0% to 100% where 100% corresponds to a maximum voltage. Figure 2 illustrates a preferred cross coupling gate activator scheme for activating a motor 23 in accordance with the present invention. as shown in Figure 2, an inverter bridge circuit 71 has two lower switches 75, 77 (Q05, Q06) and two upper switches 79, 81 (Q03, Q04) which represent the power switches 53. A return diode (not shown in Figure 2) is coupled in an anti-parallel relationship with each switch 75, 77, 79, 81. In this case, Figure 2 illustrates switches 75, 77, 79, 81 as field effect transistors in which the return diodes are internal to the components. As it is connected to the motor winding 63, the inverter bridge circuit 71 forms a bridge configuration H for use in driving the motor 23. For simplicity, FIG. 2 shows a motor 23 as an electronically commutated motor of a phase that it has a winding of a phase 63. However, it should be understood that the aspects of the present invention are contemplated for use with multi-phase motors. The inverter bridge circuit 71 also has an upper or positive rail 85 and a lower or negative rail 87 supplied by the power supply 57. A derived resistor, current transformer, Hall effect current sensor, integrated current sensor or another sensor or circuit known in the art can be used to detect the winding or motor current of the motor 23 for current regulation. It should be understood that the rails 85, 87 constitute an energy supply link, also indicated by the lines 59, 61 to provide power to the motor winding 63. In addition, Figure 2 illustrates a gate driver circuit, generally indicated with 89, which is associated with the right side of the inverter bridge circuit 71 (i.e., the switches 77, 81) and a gate activator circuit, generally indicated with 91, which is associated with the left side of the inverter circuit. inverter bridge 71 (i.e., switches 75, 79). Although the power switches 53, line 51 and gate actuators 45 are illustrated separately in Figure 1 for simplicity, it should be understood that the inverter bridge circuit 71 of Figure 2 includes aspects of each of these components .

As an example of the operation of the motor 23, the control circuit 39 provides control signals to activate a pair of power switches 53 (i.e., the switches 77, 79 or the switches 75, 81) each in a opposite side of the winding 63. In normal operation, the control circuit 39 selects one of the active power switches 53 to be used to control the motor current. One of the two active switches (e.g., switch 77 or switch 75) performs pulse width modulation (PWM) while the other (e.g., switch 79 or switch 81) remains in its conductive or ignition state during the switching interval ordered by the switching logic. The polarity of the back electromotive force of the motor during this time interval is opposite to the supply voltage so that the driving electromotive force to develop current in the motor 23 is the supply minus the counter electromotive force. In the illustrated mode, the control circuit 39 applies control signals to the switches 75, 77 as a function of the switching and regulating signals. For example, a pulse width modulated signal is applied to the switch 75 while a low logic level signal is applied to the switch 77, and vice versa. Although the driving time of one or both of the conductive power switches can be modulated in pulse width to control the current provided to the motor winding 63, only the lower switches 75, 77 are used for pulse width modulation in the illustrated mode . U.S. Patent No. 4, 757, 603 of the same assignee as the present one shows an exemplary pulse width modulation control of an engine. The control circuit 39 applies a voltage to the gate of the switch 77 through a resistor 95 (R 17) to control the switching of the transistor. In a preferred embodiment of the invention, the value of this lower ignition resistor 95 is selected to cause the lower breaker 77 to light relatively slowly with respect to the speed at which the upper breaker 81 is turned off. Similarly, an upper ignition resistor 97 (R 1 3), whose value is much higher than that of the resistor 95, causes the upper switch 81 to slow down more slowly than the speed at which the lower switch 77 is turned off. The gate activator circuit 89 of the inverter bridge 71 employs a voltage gain element, such as an NPN transistor 99 (Q08), to turn off the upper switch 81 when the lower switch 77 is conducting. Although connected to the respective arm of bridge circuit 71, transistor 99 is not part of the motor current path. When the switch 77 is conducting, a branch resistor 103 (R 16) carries motor winding current 63. The branch resistor 103 allows the unimpeded flow of the return current through the return diode of the lower power switch 77 and Eliminates the use of relatively large diodes of energy. This is particularly beneficial in a small-scale application.

The voltage gain element transistor N P N, which responds to a voltage drop across the branch resistor 103 when the lower switch 77 initially pulls the current, accomplishes the function of forcing the upper gate actuator to turn off for the switch 81. Although the starting current appears to be tripping current, its duration is limited (ie, it has a transient nature) and commonly occurs only at each switching of the motor activator. During the subsequent pulse width modulation operation of the lower switch 77, the initiating current is supplied by the upper return diode of the switch 81 in the form of a diode recovery current. A base resistor 105 (R15), connected to the base of the gate shutdown transistor N PN 99, limits the significant current flow requirement at the start of the lower breaker driving. This avoids an overcurrent condition in the base-emitter junction of transistor 99. In the illustrated embodiment, the gate activator circuit 91 operates in the same manner as the corresponding circuit 89. The gate activator circuit 91 includes a resistor. lower ignition 107 (R 18) and an upper ignition resistor 1 1 1 (R08) in combination with an NPN transistor 1 13 (Q07) and a resistor derived 1 15 (R1 1). In this case, a resistor 1 17 (R10) is connected to the base of the gate shutdown transistor N PN 1 13. In accordance with the switching strategy of the present invention, the direct switching of only the lower breakers 75, 77 commutes and regulates the motor current. The states of the upper switches 79, 81 automatically complement the states of the lower switches 75, 77, respectively. By automatically controlling the upper gate actuator, the combination of transistor 99 and resistors 95, 97, 103, 105 of gate driver circuit 89 minimizes the tripping currents that occur if both switches 77, 81 are driving at the same time . Also, the combination of the transistor 1 13 and the resistors 107, 1 1, 1 15, 1 17 reduce the tripping currents in the switches 75, 79 in a manner similar to the activation circuit 89. With additional reference to the activator circuit of gate 89, when the voltage drop across the derived resistor 103 exceeds the threshold to turn on the transistor N PN 99, the resistor 103 informs the transistor 99 of the current pulled by the respective lower power switch 77 through the resistor base 105. A positive feedback retroalum resistor 121 (R 14) maintains the base current activation to transistor 99 when the respective respective lower power switch 77 is turned on. The NPN 99 gate shutdown transistor pulls the low voltage that is supplied through the gate resistor 97 when the bias conditions at its base terminal produce adequate collector current. Figure 2 also illustrates preferred load pumping circuits for turning on the upper switch 81 when the lower switch 77 is turned off. Preferably, a capacitor 123 (C06) charges + 15 volts via a high voltage diode 125 (D07) when the switch 77 is conducting. This forms a supply of "charge pump" or "fast capacitor" to produce the required polarization for the upper energy switch 81. In other words, the voltage across the capacitor 123 turns on the upper switch 81 when the lower switch 77 goes off. A resistor 129 (R 12) limits the passage of current in the logic power supply when the capacitor 123 is being charged by the on state of the respective lower power switch 77. With respect to the left side of the inverter bridge circuit 71 , the digital circuit 91 operates in a manner similar to the digital circuit 89. In this case, the transistor N PN 1 13 is turned on when the current passing through the base resistor 1 17 through the lower switch 75 causes a voltage drop through the derived resistor 1 15 to exceed the threshold to turn on the transistor 1 13. A positive feedback resistor 131 (R09) keeps the base current activation to transistor 1 13 when the respective respective lower power switch 75 is turned on and transistor 1 13 pulls the low voltage that is supplied through the gate resistor 1 1 1 when the polarization conditions at its base terminal produce an adequate collector current. Additionally, the activation circuit 91 includes load pumping circuits for turning on the upper switch 79 when the lower switch 75 is turned off.

Preferably, a capacitor 133 (C05) charges + 15 volts via a high voltage diode 135 (D06) when the switch 75 is conducting to turn on the upper switch 79 when the lower switch 75 is turned off. A resistor 139 (R07) limits the passage of current in the logic power supply when the capacitor 133 is being charged by the on state of the respective lower power switch 75. It should be understood that the two resistors 129 and 139 are they can be replaced by a single resistor placed between the logic power voltage VCC and the connected anodes of the diodes 125, 135. Therefore, for each arm of the bridge circuit 71 (ie, the lower and upper switches on the same side of the load) the state of the upper switch (i.e., switch 79 or switch 81) depends on the state of its corresponding lower switch (i.e., switch 75 or switch 77, respectively). In this way, the present invention provides a "hands-free" control of the upper switches 79, 81. The gate activator circuits 89, 91 of the bridge circuit 71 also reduce the opportunity for non-transient tripping currents to be present by causing the upper switch of each arm to turn on at a slower speed than the speed at which it is turned off its corresponding lower switch. In Figure 2, the transistors that include the switches 53, each have a gate electrode, a source electrode and a leak electrode. These transistors also include internal return diodes. Preferably, the lower ignition resistors 95, 107 are connected to the gate electrodes of the lower switches 77, 75, respectively, and the upper ignition resistors 97, 1 1 1, are connected to the gate electrodes of the upper switches. 81, 79, respectively. The combination of the capacitor 123 and the diode 125, as well as the combination of the capacitor 133 and the diode 135, constitute load pumping circuits for turning on the respective upper switch 81, 79 in response to the corresponding lower switch 77, 75 in the same arm of the bridge circuit 71 off. As illustrated, the resistor 129 and the diode 125 are connected between the upper ignition resistor 97 and the + 15 volt source and the capacitor 123 is connected between the resistor 129 and the leakage electrode of the corresponding lower switch 77. From the Similarly, the resistor 139 and the diode 135 are connected between the upper ignition resistor 1 11 and the + 15 volt source, and the capacitor 1 33 is connected between the resistor 139 and the leakage electrode of the corresponding lower switch 75. In addition, the activation circuits 89, 91 include the voltage gain elements, the transistors 99, 1 13, connected between the gate electrode of the respective upper switches 81, 79 and the leakage electrode of their corresponding lower switches 77, 75. The speeds at which the switches 53 of the gate activator inverter bridge 71 are turned on and off also provide advantages over circuits that can be you conventional. Particularly, the ignition speeds of the upper switches 81, 79 are slower than the shutdown speeds of the lower switches 77, 75, respectively. This means that the switch 81 will turn on after the switch 77 goes off and that the switch 79 will turn on after the switch 75 goes off. Therefore, the gate triggers included by the bridge circuit 71 minimize non-manageable trip currents without requiring a time gap between the control signals applied to the switches 75, 77. In contrast to the conventional gate actuators, the bridge circuit 71 does not include the use of a timer or the like to produce time slots between the activation signals applied to the power switches 53, that is, "dead time" to reduce the tripping current. For these reasons, the present invention is particularly suitable for controlling the speed or torque of the motor by modulating the lower switches 75, 77 of the bridge circuit 71 by pulse width. As an example, the upper switch 81 remains continuously on when the lower switch 77 is turned off, that is, when a logic level low signal is applied to the switch gate 77. Meanwhile, the lower switch 75 can be modulated in width of pulse for control purposes. Since the on state of the upper switch 81 depends on the state of the lower switch 77, instead of the on or off state of the lower switch 75, the switch 81 remains on and off unnecessarily during the pulse width modulation of the switch 75. Referring now to Figure 3, another preferred embodiment of the present invention includes a gate activator circuit 143 which employs a comparator 145 (U9B) as its voltage gain element. For simplicity, Figure 3 shows a leg of a phase having a lower power switch 147 (Q8) and an upper power switch 149 (Q 19). As described below, a dual comparator circuit, which includes a comparator 151 (U9A) in addition to the comparator 145, provides sub-voltage protection of the upper switch 149. A derived resistor 153 (R 1 17) measures the current in the lower switch 147 to effect a change of state by the comparator of the voltage gain element 145, which is operated through a resistor 155 (R 1 16) connected to its reversing terminal (pin 6). In this embodiment, a series resistor network 157 (R 1 18, R 1 19, R 120) constitutes a positive feedback resistor. The output (leg 7) of the comparator 145 pushes the gate of the upper power switch 149 down when the voltage across the derived resistor 153 (reflected through the resistor network 157) exceeds the reference voltage at the non-inverting input ( leg 5) of the comparator 145. According to the invention, a zener diode 161 (D26) and a voltage divider 163 (R 1 13, R 1 14) establish the reference voltage.

By using the comparator 145 as the voltage gain element, the gate activator circuit 143 does not require a direct voltage drop of the diode to be activated. Therefore, circuit 143 allows a lower trigger threshold voltage across lead 153, which reduces the insertion loss for the circuit. In particular, a lower activation threshold (e.g., approximately 50 mV) allows the use of a current-derived resistor of lower value 153 (e.g., approximately 50 m). In contrast, the threshold for switching the transistor 99, 13 is approximately 0.8 to 1.2 V. As a result, the present invention presented by the gate activator circuit 143 of Figure 3 is particularly suitable for applications at relatively high energy levels. higher (for example, operating current of approximately 2.25 A instead of approximately 0.2 A in a lower energy application). For example, the insertion loss is approximately 2.25 W with an activation level of one volt while the insertion loss is approximately 0.25 W in a lower energy application with the lower activation level. For higher current operation, the derived resistor 153 can be scaled downward to minimize insertion loss. With further reference to Figure 3, a resistor 165 (R 101) limits the current produced by the comparator 145, 151, when it discharges the gate of the upper breaker 149. A resistor 167 (R109) pushes the gate up (turning the gate on). switch 149) when allowed by comparators 145, 151 and a resistor 171 (R 1 12) produces current to zener 161. The comparator 151 provides protection of the sub-voltage gate trigger when the voltage of the capacitor 173 (C30) falls below a threshold voltage (eg, approximately 9 V) established by a resistor network 175 (R 1 15, R 1). 1 1, R 109, R! 21). A resistor 179 (R 1 1 1) provides hysteresis so that the capacitor 173 is reloaded at a nominal level of approximately 12.7 V before re-activating the operation of the upper gate. A high-voltage diode 181 (D25) and capacitor 173 provide the above-described charge pumping supply with respect to the cross coupling gate activator 71 of Figure 2. In the alternative, gate actuators 71, 143 employ windings of isolated transformer for voltage source instead of charging pumping supplies. In a preferred embodiment of the present invention, the capacitor 173 has a larger size than the capacitors 123, 133 of FIG. 2 because the comparators 145, 151 and the additional network of the gate activator circuit 143 utilize a higher total bias current. . As shown in Figure 3, an NPN transistor 183 (Q16) with an activation resistor 185 (R102) activates the lower gate switch 147 in response to control motor signals from the control circuit 39. This gain element of relatively simple voltage, transistor 183, allows a 5V source, such as a microcomputer or DSP, to activate circuit 143. As mentioned above, Figure 3 shows only one phase leg for simplicity. Combined with a central intake power supply, the gate activator circuit 143 can maintain the operation of a phase. However, the gate activator circuit 143 is preferably double for one phase H bridge operation or tripled for three phase operation. The application of the cross coupling gate activator schemes of Figures 2 and 3 to electronically commutated three phase motor applications is complicated by the intrinsic electrical conduction 180 of the concept. For single-phase electronically commutated motor applications, as well as for three-phase variable speed motor activators and electronically commutated three-phase sine wave motor activators (flow and sinusoidal distributed windings), 180 driving is a standard procedure. However, for three-phase electronically commutated motor applications, 120 conduction accommodates back electromotive force position detection and allows current regulation by a derived resistor placed on the negative rail 87. In addition, the conduit 120 prevents complications in the circulation current that would otherwise occur during driving 180. Therefore, a three-phase electronically commutated motor activator is desired which provides the benefits of a cross coupling gate activator.

In general, the application of cross coupling gate activating circuits produces a driving scheme 180. An example of the direct application of these circuits to activate the motor 23 includes the use of a resistor derived from current detection in the negative rail 87 to control the motor current. In this example, the control of the current pulse width modulation is established by turning on all the lower energy switches of the inverter bridge when it is observed that the current exceeds a desired threshold. Turning off one of the lower power switches after an appropriate interval (set by an off timer or an asynchronous pulse width modulator timer) resets the source current. It should be understood that it may be appropriate during some intervals to turn off two lower switches in a three phase application. Although this approach works perfectly for higher current levels, significant issues develop at lower current levels. Figures 4A-4C illustrate the exemplary operation of the motor 23 with a single resistor derived in the negative rail 87 to activate the ignition of all the lower switches (for example, the switch 147 corresponding to each phase). As described above, turning on the lower switches of the power switches 53 provides a means to limit the current in a three phase electronically commutated motor control. Figure 4A depicts in a graph the torque versus speed and the regulated peak current level and demonstrates the desired characteristic of having motor torque controlled by the peak current over a wide speed range. Figure 4B graphizes the efficiency against velocity and regulated peak current and demonstrates relatively efficient operation above 0.5 A regulated peak. Figure 4C shows the maximum phase amperes against the speed and the peak regulated amps. Although the observed maximum phase amperes are reduced in correspondence to the regulated peak value of 2 A to 1 A, attempts to regulate below 1 A do not significantly reduce the maximum phase amps observed below 1 A. Referring now to the Figures 5A and 5B, example currents and phase voltages, respectively with respect to rotation (in electrical degrees) are shown at a regulated peak level of 0.25 A. FIG. 5A shows phase currents and trapezoidal phase counter electromotive force for the windings of the 63 motor and basic inverter voltage states against electrical degrees. Figure 5A also shows the currents that do not respond to the effort of regulating below the level of one ampere. Figure 5B adds the instantaneous and average torque and current derived to the data of Figure 5A. It can be seen from the graph of Figure 5B that one of the phase currents corresponds to the regulation of the derivative current at any time but the other two phases are circulating current between themselves out of sight and control of the current regulator .

Referring now to Figure 6, the present invention provides improved gate activator response for three phase electronically commutated motor applications by establishing phase voltage regulation of the type usually employed for variable speed induction motor activators (VS VS ). An appropriate sequence of phase voltages applied to the three phase electronically commutated motor 23 produces currents that satisfy the general requirements of efficiency and controlled torque. Figure 6 illustrates two alternative phase-pulse pulse width modulation reference sequences for the generation of appropriate phase voltages. , Each sequence corresponds to only one leg of the activation of three phases. The legs of two The phases not shown can be derived directly from the specific sequence shown in Figure 6 by consecutively changing the sequence by 120 (2094 radians). The sine wave and trapezoidal wave sequences are derived from the shape of the voltage that is presented to the motor phase to the neutral of the Y-connected windings 63 of the motor 23. These voltages are shown referenced to the inverter negative rail 87 with the requirement that the phase with the most negative instantaneous value is set to zero with reference to negative rail 87. As shown, both sequences are zero for the range of about 3.67 to 5.76 radians (a range of about 2/3 radians). By setting the voltage of one of the three phases to zero, the operation according to Figure 6 provides numerous advantages. For example, the voltage available to activate the motor 23 is maximized, the efficiency is improved due to the total amount of pulse width modulation, the switching is minimized and individual phase leg shunts are placed between the negative rail 87 and the lower power switch / return diode (e.g., switch 147) provides information about the full motor phase current during the interval in which the phase is set at zero volts. In Figure 6, the first sequence is based on a trapezoidal voltage waveform presented to the motor phase to neutral and the second sequence is based on a voltage sine wave presented to the motor phase to neutral. The sequences are distinguished by the marked characteristic of the trapezoidal shape against the rounded shape of the sine wave. For example, in variable speed induction motor actuators, the shape of the sine wave is preferred because of the need to minimize harmonics in an induction motor driver to maximize efficiency. In an electronically commutated motor (especially with non-sinusoidal counter electromotive force waveforms) there is no similar advantage in terms of minimizing harmonics with respect to efficiency. Figures 7A-7C and Figures 8A-8C illustrate three concurrent planes of exemplary operation of the engine in accordance with the two-phase voltage sequences of Figure 6. In particular, Figures 7A-7C show the operation of the engine 23 when it is regulated based on a sine wave voltage duty cycle and Figures 8A-8C show the operation of the motor 23 when it is regulated based on a trapezoidal phase voltage. Figures 7A and 8A illustrate the torque of the motor with respect to the motor speed and the regulated nominal current levels. Figures 7B and 8B show in graph the efficiency against the speed and the regulated level of nominal current and figures 7C and 8C show in graph the maximum observed phase current against the speed and the regulated level of nominal current. In contrast to Figures 4A-4C, the exemplary planes of Figures 7A-7C and 8A-8C are not based on the instantaneous regulation of the peak current on the negative rail 87. Instead, these drawings illustrate the operation of the Motor by adjusting the pulse width modulation values of the phase leg in response to the current observed in the negative rail 87. In essence, this provides a delayed servo response, which introduces implicit circuit stability in these systems. A comparison of Figures 7A-7C with Figures 8A-8C reveals that the less complicated trapezoidal shape is not significantly different from the sinusoidal wave sequence. In a preferred embodiment, the control circuit 39 (e.g., a digital signal processor) implements one of the voltage regulation approaches of Figures 7A-7C and Figures 8A-8C to operate the motor 23. The Figures 9A-9C illustrate exemplary phase-leg voltages, currents and duty cycle during trapezoidal operation.

By reference, each of the planes has the counter electromotive force voltages of the three phases plotted with respect to the rotation of the rotor 27 (in electric degrees). Figure 9A shows that the motor 23 generates, for example, approximately 20 Oz Ft of torque. Figure 9A also shows the filtered peak values of derived current plotted against a target value of 0.8 A (the Y axis on the right). Figure 9B includes the duty cycle per unit for each graded phase leg and Figure 9C graphs the phase currents. For improved efficiency and low cost, the three phase electronically commutated motors are designed with outgoing pole construction. This construction uses the outgoing stator phase poles to return their flow through the other two projections and is generally referred to as a "consistent pole design". There are half stator projections that the number of phase times poles in this construction. For example, a pole pole stator of three 12-pole phases has 18 projections. U.S. Patent Serial Number 5, 796, 194 of the same assignee as herein, discloses a single-phase motor having a quadrature axis winding extending through a notch in each of the motor stator teeth. The direction in which the quadrature axis winding is wound alternately for the adjacent teeth. In other words, the quadrature axis winding for a single-phase pole motor is placed at the center of each projection. As a pole design with sequential three-phase motor also means that each of the stator phase poles are wound in the same direction, this initially suggests a dilemma in the placement of the quadrature windings 69. As mentioned previously, conventional trapezoidal phase voltage strategies for three-phase motors require tracking of rotor position. For example, three Hall sensors in proximity to the edge of the rotor magnet can be used to estimate the rotation angle by elapsed time in reference to the previous switching interval. Likewise, three optical switches and a shaft-mounted shutter can also be used to estimate the angle of rotation. In accordance with the present invention, the quadrature shaft windings 69 provide rotor position feedback. Figure 10 is a top view in flattened section of the motor 23. In this case, the motor 23 is a three-phase motor having main windings 63 designated A, B and C and a coil for detecting the position (i.e. quadrature axis winding 69) corresponding to each phase. The main and quadrature axis windings 63, 69 are electromagnetically affected by the flux of the rotor magnet 27. However, the quadrature axis windings 69 are not electromagnetically affected by the stator flux 25. This allows the windings of quadrature 69 detect the position of the rotor 27 without being affected by currents in the main windings 63. Both winding groups are electromagnetically affected by the flow of the rotor 27 because the direction of flow ao of the rotor 27 due to each rotor magnet depends on the polarity of the rotor magnet adjacent to a respective portion of the stator 25. In one embodiment, the stator 25 has a plurality of teeth 189, each having three notches. The quadrature shaft windings 69 are wound between the central notches of the teeth 189, as shown in Figure 10. In accordance with the present invention, the motor 23 returns the respective quadrature axis winding 69 through the groove. between the other two phases. As shown in Figure 10, the crossed end for each of the quadrature axis winding phases is placed in the center of the stator projections. Figure 10 also shows the point end for each phase returning between the windings 69 of the other two opposite phases. The final termination of windings 69 returns to a common connection (not shown) to complete a Y-connected configuration for quadrature windings 69. In operation, the three quadrature windings connected in Y 69 develop a reference series of voltages. These phase quadrature voltages provide information for estimating the angular position of the rotor 27 based on the elapsed time with reference to a pre-switching interval. In addition, the quadrature winding voltages can be integrated to achieve a series in the phase of voltages to use them as reference for the regulation of each of the phase currents. In one embodiment, the control circuit 39 constitutes an integrator. As an example, the control circuit 39 provides the estimate of the angular position of the rotor 27 in combination with an advanced commutation angle calculated based on the principles described in the U.S. Patent of North America Serial Number 5, 796, 194 of the same assignee as the present one. In this example, a trapezoidal voltage sequence presented to the motor phases produces results like those shown in Figures 9A-9C. Conveniently, the present invention employs the information obtained from the quadrature windings 69 to form the motor current in addition to the position detection. This form of regulating current can reduce the amount of torque harmonics normally seen for driving schemes 120. In a preferred embodiment of the present invention, the control circuit 39 implements low-pass filtering of the quadrature voltages (either by conventional analog circuits or computationally by a digital signal processor, for example) so that the reference voltages are in phase with the counter electromotive force of the motor. By having voltages representative of the counter electromotive force phase voltages, the motor system 21 can regulate phase currents at a scaled value of the phase voltages. Conveniently, phase reference voltages allow controlling currents through the critical sequences between the rotating states and also provide automatic advanced angle compensation. As an alternative to the isolated current measurements, separate phase shunts connecting the negative rail 87 to the three lower power switches (eg switch 147) and their corresponding return diodes present the correct phase current when the lower power switch or return diode is driving. However, the taps have no current present when the upper power switches (for example, switch 149) are turned on. In the cross coupling gate activator scheme described herein, the correct current can be observed each time the lower power switch is controlled to be on. For this reason, the control circuit 39 (e.g., a digital signal processor) provides a preferred means for implementing a control based on regulating the current to match a scaled value of the filtered quadrature winding voltages. low step In a digital signal processor, the pulse width modulation period is restricted to less than 100% duty cycle for any phase leg. At the start of each pulse width modulation cycle, samples of the derived currents are recorded for a time when all the lower energy switches are active. For a low-cost digital signal processor, each phase sample may require approximately three microseconds. Therefore, the total sample time is approximately nine microseconds for the three phases or approximately six microseconds for two phases. As one of the lower energy switches of the three phases is selected to remain on during the pulse width modulation cycle, it would not have to be measured during the interval in which all the lower switches are on. After taking samples of the current, the control circuit 39 preferably initiates the pulse width modulation period. In one embodiment, the control circuit 39 selects one of the three phase legs to have the respective lower power switch on during the pulse width modulation range. The control circuit 39 makes this selection by determining which phase current requires the most negative correction compared to f a goal established by the quadrature voltages filtered in low pass scaled. Turning on the remaining upper energy switches that require positive phase current correction initiates the pulse width modulation period. During the interval in which the upper switches are conducting, the respective phase currents in the legs that have the upper switches conducting can be estimated if they are not measured. Estimating the current uses the time in the pulse width modulation cycle, the phase inductance (increased by the phase-to-phase winding coupling), the estimated voltage for the phase back-electromotive force by the quadrature voltage and the rail voltage assumed or estimated. For greater accuracy, the voltage measured at the center connection of the Y-connected main windings can also be incorporated. In general, work cycles can be restricted to discrete duty cycle selections as high as, for example, 45%. A rapid and uncontrolled increase in current can be observed by shunting the selected phase leg to have the lower power switch turned on during the pulse width modulation period, and if it exceeds the desired regulated value, it can be used to finish the driving of all the upper energy switches. Referring now to FIGS. 1 1 A-1 1 C, exemplary phase leg voltages, currents and duty cycle are shown. By reference, each of the planes has the counter electromotive force voltages of the three phases graphically represented with respect to the rotation of the rotor 27 (in electric degrees). Figure 1 1 A shows that the motor 23 generates, for example, approximately 50 Oz Ft of torque. In this case, the control circuit 39 monitors the current via three branches so that the filtered peak values are not graphically represented. Figure 1 1 B includes the work cycle per unit for each phase leg plotted and Figure 1 1 C graphs the phase currents. Preferably, the control circuit 39 determines the duty cycles based on the regulation of the estimated phase current at a given value of the selected field voltages in low-level conditions. However, the operation of the motor 23 based on position detection by the quadrature windings 69, does not start by itself. At start-up, the quadrature winding voltages are zero. Therefore, any current value scaled to the quadrature voltages filtered in low pass will also be zero. A preferred starting method includes a sequence of six states of objectives or goals of positive current, zero value, and negative. In this case, the control circuit 39 controls the communication between states by a sum of Volts Seconds determined by a sequence of the estimated phase volts derived from the quadrature voltages. An exemplary sequence is: 0, 1, 1, 0, -1, - 1 for states 0 to 5 for phase 1 (or phase A). Phase 2 (or phase B) is two advanced states of phase 1 in the same sequence, and phase 3 (or phase C) is two advanced states of phase 2. In this case, the sequence represents a series of coefficients by which the desired peak current levels are multiplied. The estimated phase voltages are derived from the quadrature winding voltages filtered as follows: First, an RC divider has the following divider relationship for the stable state. In complex arithmetic form for the fundamental: Ve = Vq (-jXc) / (R-jXc) Where Ve is through the capacitor, and Vq is the quadrature voltage. Discarding the angles for the magnitude of the dividing relation and dividing between Xc and Vq: Ve / Vq = 1 / V ((R / Xc)? 2 + 1) Substituting 1 / wC for Xc: Ve / Vq = 1 / V ( (RC w)? 2+ 1) Where w = 2 Poles RPM / 120 Substituting Vq = Vph / ny = RC where n is the ratio of effective turns phase winding / quadrature coil: Ve n / Vph = 1 / V (()? 2 + 1) Solving for Vph: Vph = Ve n V ((w)? 2 + 1) The control circuit 39 then uses this estimate of the phase voltage for the selected phase to add the Volts Seconds to commutation. Only one Volt sum is allowed Second above zero with negative sum below zero inhibited.

The control circuit 39 determines the selected phase for sum of Volts Seconds by the state assigned to phase 1 in the previous sequence of current goals. The following sequence for states 0 to 5 (selected by the state assigned to phase 1) determines the estimated phase voltage to sum, with the sum direction determined by the sign of the phase number in this sequence: - 1, 2, -3, 1, -2, 3. In the alternative, the control circuit 39 directly adds the quadrature voltages filtered in low pass for Volts Seconds when w is small at start-up: Vph = Ve nk Where k is drift experimentally for acceptable start-up operation. If the switching does not take place by adding Volts Seconds in an appropriate time limit, the control circuit 39 preferably employs forced switching. The start algorithm persists only at a predetermined speed, such as 100 RPM, after which the operation passes the current regulation to a scaled value of the quadrature voltage filtered in low pass. Figures 12A-12D illustrate the exemplary operation of the engine 23 starting in this manner. Figure 12A includes the counter electromotive force versus the estimate of the back electromotive force derived from the low pass filtered quadrature voltages. As an example, the low pass filter has a time constant of 0.1 seconds, which corresponds to 100 RPM for a three-phase, 12-pole motor. As you can see from this graph, the angle of these estimates is acceptable. Although not illustrated in Figure 12A, the control circuit 39 preferably provides digital or analog derivative calculations to set the output of the low pass filter to expedite the centering of the phase voltage estimates. When setting to values determined by a centering algorithm, the control circuit 39 (e.g., a digital signal processor) preferably performs a pure integration to achieve the desired phase voltage estimates. Figure 12B shows the simulated phase currents at start-up and demonstrates the intervention of a peak current limit based on observation of the derivation of the selected phase leg to have the lower power switch turned on. The asymmetry in the starting currents reflects the asymmetries in the estimated phase voltage, which can be substantially corrected by the means described above. Figure 12C shows an inversion and correction in the direction of rotation at start-up. This reversal can occur due to an "out of position" start. The inversion is seen in the negative RPM from 0 to 0.055 seconds and the negative angle from 0 to 0.09 seconds. Acceleration variations starting at approximately 0.12 seconds are attributed to the previously mentioned current asymmetries. Figure 12D illustrates only the first 0.03 seconds while the boot algorithm is active. A parameter of mV Seconds is shown being added to produce the first state change at 0.0215 seconds, and the second at 0.023 seconds. These planes are based on a four-wire connection to the three-phase quadrature windings 69. For three-wire connection (the central connection of the three phases connected in Y is omitted) three equal resistors are used to sum to a central point artificial. The voltages that exist from this summed center point to the three terminals may have significant instantaneous errors as estimates of the Y-connected source voltages. However, the filtered low-pass voltages derived from these approximate voltages are generally inappreciable from those directly derived from the quadrature voltages connected in Y. In view of the above, it will be noted that the different objects of the invention are achieved and other convenient results are obtained as well. Since various changes can be made to the above methods and constructions without departing from the scope of the invention, it is intended that all matter contained in the above description or shown in the accompanying drawings be construed as illustrative and not in a sense of limitation. .

Claims (27)

1. CLAIMS l. A motor system comprising: a rotating assembly; a stationary assembly in magnetic coupling relation with the rotary assembly, said stationary assembly includes at least one winding; a power supply link having a top rail and a bottom rail provided by a power supply; a bridge circuit including a series of upper energy switches connected between the winding and the upper rail and a series of lower power switches connected between the winding and the lower rail, each of said lower energy switches corresponds to one of such upper energy switches on the same side of the winding as the lower switch, said upper and lower power switches are connected to the same end of the winding defining an arm of the bridge circuit, said power switches each having a conductive state and a non-conducting state wherein the state of each upper switch is determined by the state of its corresponding lower switch in the same arm of the bridge circuit; a control circuit for generating a motor control signal for controlling the switches, said lower energy switches receiving and responding to said motor control signal; and an activation circuit that responds to the state of the lower power switches to activate the corresponding upper power switches to selectively connect the rails of the power supply link to the winding where the winding is energized with a motor current for producing an electromagnetic field for rotating the rotary assembly relative to the stationary assembly, said activation circuit includes a voltage gain element connected to each arm of the bridge circuit, such voltage gain elements respond to the current in the respective lower switch to maintain the corresponding upper switch in the same arm of the bridge circuit in its non-conductive state.
2. 2. The motor system of claim 1, further comprising a resistive branch on each arm of the bridge circuit, each of said resistive branches is connected between the winding and the respective lower switch for detecting the current in the lower switch.
3. The motor system of claim 2, wherein each of the voltage gain elements is connected between the winding and the respective resistive shunt and responds to a threshold voltage across the resistive shunt to make the switch respective top is made non-conductive.
4. The motor system of claim 1, wherein the upper breakers each have a gate electrode for determining their status and wherein the voltage gain elements are each connected to the respective upper breaker so that the gate element The voltage gain pushes the gate electrode of the respective low upper switch when the current in the respective lower switch exceeds a threshold level thereby causing the respective upper switch to become non-conductive.
5. The motor system of claim 1, wherein the voltage gain element comprises a transistor connected to each of the upper energy switches, such transistors each having a conductive state and a non-conductive state, and wherein the upper energy switches respond to the conductive states of the transistors to become non-conducting.
6. The motor system of claim 5, wherein the transistors each have a base electrode for determining their state and additionally comprise a base resistor connected between the respective resistive shunt and the base electrode of each transistor to limit the current of the base electrode.
7. 7. The engine system of claim 6, further comprising a positive feedback resistor connected between the upper rail and the base electrode of each transistor to maintain the base current activation to the transistor when the respective lower switch is conducting.
8. The motor system of claim 1, wherein the voltage gain element comprises a comparator circuit connected to each of the upper switches, such comparator circuits each having a low level salt state and an output state of high level, and where the upper switches respond to the low level output states of the comparator circuits to become non-conductive.
9. 9. The motor system of claim 8, wherein the comparator circuits each have an input for determining their status relative to a reference level and additionally comprise a resistor connected between the respective resistive bypass and the input of each comparator circuit to limit the input current.
10. The motor system of claim 8, wherein the comparator circuit comprises a pair of comparators connected together at their outputs to provide sub voltage protection to the bridge circuit. eleven .
11. The motor system of claim 1, wherein the lower switches each have a return diode coupled thereto and wherein the voltage gain elements each respond to the diode recovery current for the respective lower switch for keep the corresponding upper switch in its non-conductive state.
12. The motor system of claim 1, wherein the winding comprises a winding of a phase and wherein the bridge circuit comprises a bridge H having two upper switches and two lower switches for selectively connecting the winding of a phase to the power supply link.
13. The motor system of claim 1, wherein the winding comprises one of three interconnected phase windings and wherein the bridge circuit comprises a complete inverter bridge having at least three upper switches and at least three lower switches to selectively connect the three phase windings to the power supply link.
14. The motor system of claim 13, wherein the motor control signal defines switching intervals during which one of the lower switches and one of the upper switches of different arms of the bridge circuit are conducting to connect the winding to the upper and lower rails of the power supply link, said switching intervals are defined as an angular position function of the rotary assembly, and additionally comprise a quadrature axis winding corresponding to each of the three phase windings, each one such quadrature axis windings is in magnetic coupling relationship with the rotary assembly and positioned to generate an output signal representative of the angular position of the rotating assembly.
15. 15. The motor system of claim 14, further comprising an integrator for phase retarding the output signals from the quadrature axis windings, such integrated output signals are representative of back-electromotive force in the respective phase windings, and wherein the control circuit responds to the integrated output signals to generate the motor control signal to regulate the motor current in each of the three phase windings.
16. 16. The motor system of claim 14, wherein the stationary assembly includes a plurality of stator teeth defining slots therebetween, said phase windings being wound sequentially on the teeth so that one of the Three phase windings are placed in each slotEach of said teeth has a notch located centrally on its face and where the quadrature axis windings are sequentially placed in the notches of the stator teeth.
17. The motor system of claim 16, wherein the quadrature axis winding corresponding to one of the three phase windings is placed in the slot in which the other two phase windings are placed.
18. The motor system of claim 14, wherein the quadrature axis windings comprise a Y-connected circuit.
19. The motor system of claim 14, wherein the motor control signal generated by the circuit of control implements a start sequence to start the motor in the absence of the output signals of the quadrature axis windings at the start, said start sequence defines a sequence of target current levels in the phase windings based on a pair of desired start torque.
20. The motor system of claim 1, further comprising an axle in activation relation with the rotary assembly for activating a rotating component. twenty-one .
21. A gate actuator for activating an engine, said motor has a rotary assembly and a stationary assembly in magnetic coupling relation thereto, said stationary assembly includes at least one winding, such motor also has a power supply link that includes a upper rail and a lower rail supplied by a power supply and having a control circuit for generating a motor control signal for controlling the switching of the winding, said gate actuator comprises: a series of upper power switches connected between the winding and the top rail; a series of lower power switches connected between the winding and the lower rail, such lower switches receive and respond to the control signal of the motor, each of said lower switches correspond to one of said upper switches on the same side of the winding the lower switch, said switches each having a conductive state and a non-conductive state; and an activation circuit that responds to the state of the lower switches to activate the corresponding upper switches to selectively connect the rails of the power supply link to the winding where the winding is energized to produce an electromagnetic field to rotate the rotary assembly with relation to the stationary assembly, said activation circuit includes a voltage gain element connected to each of the lower switches, such voltage gain elements each responding to current in the respective lower switch to maintain the corresponding upper switch in the same arm of the bridge circuit in its non-conductive state.
22. 22. A three-phase motor system comprising: a rotating assembly; a stationary assembly in magnetic coupling relation to the rotary assembly, said stationary assembly includes three phase windings; a quadrature axis winding corresponding to each phase winding, each of said quadrature axis windings is in magnetic coupling relationship with the rotary assembly and positioned to generate an output signal representative of the angular position of the rotary assembly; and a control circuit that receives and responds to the output signals of the quadrature axis windings to generate a motor control signal to control the switching of the phase windings.
23. 23. The motor system of claim 22, wherein the stationary assembly includes a plurality of stator teeth defining grooves therebetween, said phase windings being wound sequentially on the teeth so that one of the three Phase windings are placed on each tooth and two of the three phase windings are placed in each slot, each or not of such teeth has a notch located centrally on its face, and where the quadrature axis windings are positioned so sequential in the notches of the stator teeth.
24. The motor system of claim 23, wherein the winding of the quadrature axis corresponding to one of the three phase windings is placed in the slot in which the other two phase windings are placed.
25. 25. The motor system of claim 22, wherein the winding of the quadrature axis comprises a circuit connected in Y.
26. 26. The motor system of claim 22, further comprising an integrator for delaying in phase the output signals of the quadrature axis windings, such integrated output signals are representative of counter electromotive force in the respective windings of phase, and wherein the control circuit responds to the integrated output signals to generate the motor control signal to regulate the motor current in each of the three phase windings. The motor system of claim 22, wherein the control signal of the motor generated by the control circuit implements a start sequence to start the motor in the absence of the output signals of the quadrature axis windings in the start, said start sequence defines a sequence of target current levels in the phase windings based on a desired starting torque. RESU MEN A motor system that includes a gate activator to activate a motor. An inverter bridge circuit selectively connects energy supply link rails to a motor winding to energize the winding with a motor current. The bridge circuit has upper and lower switches connected between the winding and the upper and lower power supply link rails, respectively. Each lower switch corresponds to one of the upper switches on the same side of the winding as the lower switch to define one arm of the bridge circuit. A control circuit generates a controlled signal from the motor to control the switches. An activation circuit activates the upper switches in response to the state of the corresponding lower switches, which respond to the motor control signal. The activation circuit includes a voltage gain element connected to each arm of the bridge circuit which responds to the current in the respective lower switch to maintain the corresponding upper switch in its non-conductive state. In a three phase mode, a quadrature axis winding corresponds to each phase winding. Each of the quadrature axis windings is in magnetic pick-up relationship with the rotating assembly and positioned to generate an output signal representative of the angular position of the rotating assembly. The control circuit generates the motor control signal to control the switching of the phase windings in response to the output signals of the quadrature axis windings.