KR920002044B1 - Power apparatus - Google Patents

Abstract

No content.

Description

Power supply for load

1 is a view showing an example of a test winding.

2 is a circuit diagram showing an example of an annular core test apparatus.

3A and 3B are equivalent circuit diagrams of a FIG. 2 device.

4 is an output characteristic diagram of the device of FIG.

5 is a circuit diagram showing a first embodiment of the annular core test apparatus of the present invention.

6 is an equivalent circuit diagram of the FIG. 5 apparatus.

7 is a waveform diagram showing the oscillation generated by the apparatus of FIG.

8 and 9 are schematic diagrams showing the basic principle of the present invention.

10 and 11 are circuit diagrams showing second and third embodiments of the power supply of the present invention, respectively.

12 is an output characteristic diagram of the device of FIGS. 10 and 11;

* Explanation of symbols for main parts of the drawings

1 core 2 power supply

5: amplifier 6: feedback resistor

7: phase adjustment circuit 71: capacitor

72: Resistor

The present invention relates to a negative feedback power supply for a load. For example, the present invention is used in a device for testing the alternating current (AC) magnetization characteristics of an annular core or a disconnected core wire, a load power supply of a rectifier circuit, or a distorted current load power supply.

In order to carry out tests of the alternating magnetization characteristics of iron losses such as excitation currents, overcurrent losses and hysteresis losses of a large number of cyclic cores, it is necessary to wind the windings around the core each time the test is performed. In order to simplify this winding operation, splice contact can be used to reduce the deviation of the winding. However, in the actual design of a transformer or the like, the winding DC resistance is made as small as possible to make the winding as large as possible, and the number of windings is very large. do. However, since the space for installing the splice contact is limited when the splice contact is used, the thickness of the winding is thin and the number of turns is small. As a result, the test excitation current flowing in the primary winding must be increased, but the resistance of the winding and the contact is greatly affected.

In a test apparatus for a ring core having a primary winding and a secondary winding according to the prior art, an input AC power supply and an AC ammeter are connected to the primary winding, and an AC voltmeter is connected to the secondary winding. Also, the power meter is connected between the primary winding and the secondary winding. In this device, the measurement of the primary current I _{p and} the measurement of iron loss are carried out around a saturated magnetic flux in which one of the alternating current (AC) magnetization characteristics is to be tested, and the measurement of iron loss is performed by a power meter.

However, the same change in DC resistance of the primary winding is caused by heat generated by the excitation current, and the change in contact DC resistance of the splice contact is caused by the state of the contact. In addition, the variation of the equivalent direct current (DC) resistance of the primary winding is caused by the saturation characteristics of the core. Therefore, the output voltage of the secondary winding varies greatly. In addition, near the saturation magnetic flux density of core, the instantaneous equivalent alternating current resistance decreases due to the abnormal current generated locally during one cycle of alternating current, and the spiral shape of the output voltage generated in the secondary winding is greatly distorted. These fluctuations and distortions cause the magnetic flux in the core to fluctuate and become distorted, which makes it unreasonable as a core test that should be essentially a correct alternating flux. Therefore, in order to make the output voltage of the secondary winding constant, even if the voltage generated in the secondary winding is adjusted by adjusting the voltage of the AC power source in the core, distortion remains in the magnetic flux waveform, that is, the output voltage, and at the same time, the peak of the excitation current The current portion is suppressed, making accurate measurement impossible.

Therefore, the object of the present invention is to eliminate the winding work by using the contact type contact, and even if the DC resistance value, the equivalent AC resistance value of the primary winding and the DC resistance value by the contact change, it is unnecessary to adjust the input voltage for each core. The present invention provides an annular core test apparatus capable of accurate and stable measurement without distortion of the magnetic flux waveform.

Another object of the present invention is to provide a power supply device for the load of the rectifier circuit and the load of the distortion current load.

First, a power supply device for annular core, that is, an apparatus for testing the annular core will be described.

As shown in FIG. 1, in order to test the annular core 1, the test windings W ₁ and W ₂ are wound around the core 1, and when the test is completed, their windings are removed. In order to omit such a complicated winding operation, it is possible to facilitate the detachment of the windings W ₁ and W ₂ by the contact type contacts CW ₁ and CW ₂ . However, in an actual design of a transformer or the like, the winding is made as large as possible to make the winding DC resistance small, and the winding number is also wound very much. Therefore, the current is small, and the voltage drop and heat generation due to the winding low phase component are sufficiently small. However, in the case of FIG. 1, the thickness of the windings is small and the number of turns does not have to be small because restrictions are placed on the arrangement spaces of the contact-type contacts CW ₁ and CW ₂ . The test excitation current flowing in the primary winding W ₁ must be large, but the resistance of the windings W ₁ , W ₂ and the contacts CW ₁ , CW ₂ is greatly affected. In Fig. ₁ , the upper and lower contacts of the contact type contacts CW ₁ and CW ₂ are separated from each other when the core 1 is inserted therein, and are connected at the time of test (measurement), thereby winding the windings W ₁ and W ₂ . The work becomes unnecessary.

2 is a circuit diagram showing a test apparatus for the annular core _{1 having the} test windings W ₁ and W ₂ . In FIG. 2, 2 is an input AC power source, 3 is an AC current gauge, 4 is an AC voltmeter, P ₁ and P ₂ are terminals for connection to the primary winding W _1, and S ₁ and S ₂ are connected to the secondary winding W ₂ . Terminal for connection. This circuit measures the primary current I _p near the saturation magnetic flux density, which is one of the core magnetization characteristics tests, and the iron loss accordingly. In addition, iron loss is measured by the power meter W.

An equivalent circuit thereof for easily understanding the circuit of FIG. 2 is shown in FIG. 3A, and a more simplified equivalent circuit is shown in FIG. 3B. 3A and 3B, the power meter W is omitted. In Fig. 3a, T is an ideal transformer, R _p is the equivalent direct current resistance of primary winding W ₁ , R _c is the equivalent contact DC resistance of contactor contact CW ₁ , and R _zp is equivalent winding of primary winding W ₁ . Resistance. In addition, the resistance of the secondary winding W ₂ and the contact CW ₂ shall be negligible. Here, the equivalent DC resistance R _p and the equivalent contact DC resistance R _c are unstable. For example, the resistance R _p is varied from 1 to 1.2 kV by the heat generated by the excitation current I _p , and the contact DC resistance R _c is varied from 0 to 0.8 kPa depending on the state of the contact CW ₁ .

In addition, the equivalent alternating current resistance R _zp also varies by 0.5 dB at infinity due to the saturation characteristic of the core 1. Therefore, when the excitation current I _p is applied to the primary winding W ₁ with the voltage of the AC power supply 2 being 1 V, the output voltage Es generated in the secondary winding W ₂ is R = 1 Ω, R _c = 0 Ω, R _zp = It is 1V at ∞, 0.2V at R _p = 1.2 kV, R _c = 0.8 kV, and R _zp = 0.5 kV, and the output voltage E _s fluctuates greatly. In addition, near the saturation magnetic flux density of core 1, an abnormal current generated locally in one cycle causes a momentary drop in the equivalent AC resistance R _zp , and thus a waveform of the output voltage E _s generated in the secondary winding W ₂ . Is greatly distorted. These fluctuations and distortions cause the magnetic flux in the core 1 to fluctuate and become distorted, which is extremely unreasonable in the test of the core 1 which should be a constant correct alternating magnetic flux. Accordingly, in order to make the output voltage E _s constant, distortion remains in the voltage E _s generated in the secondary winding W ₂ even when the voltage of the AC power supply 2 is adjusted for each core. The peak current portion of _p was suppressed and no accurate measurement was possible.

5 is a circuit diagram showing the first embodiment of the annular core test apparatus of the present invention, in which an analog negative feedback loop circuit is added to the elements of FIG. That is, the phase adjusting circuit 7 formed by the amplifier 5, the feedback resistor 6 and the capacitor 72 and the resistor 71 is added to the element of FIG. 2, and FIG. 6 is equivalent to the apparatus of FIG. It is a circuit diagram.

Here, assuming that there is no negative feedback circuit, amplification of the amplifier 5 having an output impedance of 0 is F and 100, and the voltage of the AC power supply 2 is 1V x 1/100 = 0.01V. Is substantially the same as the circuit of FIG. 2, and in this case, therefore, the fluctuations of the equivalent DC resistance R _p , the equivalent contact resistance R _c and the equivalent AC resistance R _{z p} are the same as those in FIG. 3A, and the output voltage E _s is also Varies from 1 to 0.2V "When an amplification factor A 'end, the input voltage (= 0.01V), the ratio of the output voltage E _s of the AC power supply (2) a full-amplification and A is a variation in A' = 100~20.

Next, a case in which the negative feedback circuit according to the present invention is considered will be described. If the feedback rate F by the feedback resistor 6 is set to 0.99, the output voltage E _s is

Where E ₁ is the output voltage of the AC power supply 2, in this case 1V. Therefore, the maximum output voltage E _s is

Minimum output voltage E _s

to be.

Thus, the fluctuation of the output voltage E _s becomes small compared with the case of FIG. 3A (FIG. 3B). Therefore, even if the DC resistance value by the contact, the equivalent AC resistance value of the primary winding W ₁ and the DC resistance value fluctuate, adjustment of the input voltage is practically unnecessary, and as a result, the magnetic flux wave in the core 1, namely, the output voltage. There is also little distortion of E _s . In addition, when the amplification of the amplifier 5 is made larger, the stability of the output voltage E _s is improved. In addition, when the feedback rate F is increased, oscillation may occur due to leakage reactance of the winding, but such oscillation may cause the resistor 71 and / or the capacitor 72 of the phase adjustment circuit 7 to be in a constant range. It can be prevented by encouraging until.

In the circuit of FIG. 5 with an analog negative feedback circuit, an accurate measurement can be obtained because of an oriented silicon steel core having a magnetic flux density of about 17000 gauss. In addition, when the highly distorted excitation I _p is supplied to the primary winding W ₁ to produce a saturated magnetic flux density of about 18000 to 20,000 gauss in the core 1, the return F and the amplification A 'are highly accurate and stable. It must be increased to obtain a measurement. In this case, however, oscillation always occurs. That is, the amplification factor A 'is greater than 1, and the phase potential amount of the negative feedback lepu circuit is jointly generated by the capacitance component in the negative feedback loop circuit of a relatively high frequency band, and additionally by the phase adjusting circuit 7 It is difficult to fully adjust the amount of phase potential. It is difficult to adjust this amount of phase potential even when an inductance component is added to the negative feedback loop due to the winding and the leakage inductance therebetween.

Thus, even if an analog negative feedback loop circuit is used in the core 1 having a relatively high magnetic flux density, the output voltage E _s of the secondary winding W ₂ is distorted as it is and is unstable.

That is, in the circuit of FIG. 5, after the oscillation condition is satisfied in the analog negative feedback loop circuit, the positive feedback control at a specific frequency of about 10 KHz to 1 MHz is limited by differential noise and so on, and thus the input voltage used in the core 1 The amplification of P is rapidly increased as shown in FIG.

8 and 9 showing the digital negative feedback control according to the present invention, the waveform to be negative feedback is stored and delayed by one cycle of the reference sine wave to prevent the negative feedback of the negative feedback loop circuit from occurring. Feedback control is performed using a delayed waveform. That is, as shown in FIG. 8, the voltage error is stored between the input sinusoidal voltage and the output voltage and delayed for period A, and the input sinusoidal voltage is corrected by the voltage error delayed for period B. Thus, as shown in FIG. 9, if the amount of phase potential of the negative feedback loop circuit is greater than 180 degrees, the positive feedback phenomenon is separated during every cycle of the reference sine wave so that no oscillation is formed.

In FIG. 10 showing a second embodiment of the present invention which realizes the principles of FIGS. 8 and 9, an apparatus for testing an annular core is used.

Referring to the digital negative feedback loop circuit in FIG. 10, 1001 is a control unit (central processing unit), 1002 is a frequency switch, 1003 is a voltage setting switch formed by a digital switch, and 1004 is a reference for generating a sinusoidal voltage E ₀ . A sine wave issuing circuit (ROM) is shown, and 1005 represents a voltage adjusting circuit (multiplier) for adjusting the output of the reference sine wave generated in the reference sine wave generating circuit 1004. Here, the frequency of the reference sine wave voltage E ₀ of the reference sine wave generation circuit 1004 is changed by the control unit 1001 according to the state of the frequency switch 1002. That is, the control unit 1001 generates a clock signal having a frequency in accordance with the state of the frequency switch 1002, and this clock is applied to the reference sine wave generating circuit 1004. In addition, the controller 1001 generates a voltage α according to the output of the voltage setting switch 1003. Therefore, the voltage adjusting circuit 1005 multiplies each digital value of the reference sine wave voltage E ₀ by the voltage ratio α to generate a new reference sine wave voltage E ' ₀ .

In addition, 1006 and 1007 represent delay memories for adjusting the total delay time corresponding to one cycle of the reference sine wave voltage E ₀ (E ′). Each delay memory 1006 and 1007 may be constituted by a bidirectional memory capable of simultaneously performing a write operation and a read operation. That is, the delay memories 1006 and 1007 are connected to the address counters 1006a and 1007a, respectively, and are thus operated by writing to the addresses WA of the delay memories 1006 and 1007 and decoding the address RAs of the delay memories 1006 and 1007, respectively. By the address counters 1006a and 1007a. In this case, the delay time of the delay memory 1006 1007 is determined by (RA-WA) × T when the RA is an address to be read, the WA is an address to be written, and T is a period of a clock signal supplied from the controller 1001. .

The subtraction circuit 1008 converts the output voltage E _s from the end of the secondary winding W ₂ via the analog / digital converter 1012 to convert the digital voltage between the reference sine wave voltage E ' ₀ and the analog / digital (A / D). Calculate the error. The digital / analog converter 1010 performs digital / analog conversion in accordance with the output of the addition circuit 1009, and thus the power amplifier 1011 supplies the input current I _p to the primary winding W ₁ in accordance with the analog voltage of the digital / analog converter 1010.

Thus, the output voltage E _s of the secondary winding W ₂ is negative feedback by the digital negative feedback loop circuit formed by the elements 1006, 1008, 1007 and 1012. In addition, there is a delay of one cycle of the reference sine wave voltage E ₀ to be fed back. Therefore, the output voltage E ₀ is recognized as shown in FIG. 12 even when the magnetic flux density of the core 1 is extremely large.

In FIG. 11 showing the third embodiment of the present invention, the delay memory 1101, the address counter 1101a and the addition circuit 1102 are added to the elements in FIG. In this case, the three delay memories 1006, 1007 and 1101 adjust the overall delay time corresponding to the cycle of the reference sine wave voltage E ₀ (E ₀ ′). Therefore, in Fig. 11, the voltage error E 'to be fed back is summed and added to the reference sine wave voltage E ₀ ' to significantly increase the feedback rate F. For example, if the feedback rate F of the first cycle is 0.9, the feedback rate F of the second cycle is 0.99. In addition, the feedback rate F of the third cycle is 0.999 and the feedback rate F of the fourth cycle is 0.9999.

The second and third embodiments of FIGS. 10 and 11 can be merged with the first embodiment of FIG. That is, the analog negative feedback loop circuit formed by elements 6 and 7 of FIG. 5 is added to the circuits of FIGS. 10 and 11 as indicated by the dotted lines. In this case, the power amplifier 1011 also serves as the differential amplifier 5 of FIG. Thus, it is possible to fully calibrate the output voltage E _s even at higher speeds.

In addition, in the above-mentioned embodiment, the primary winding W ₁ and the secondary winding W ₂ may have a form other than the connector type. It is also possible to carry out a highly accurate and stable test (measurement) even when the winding is one. In addition, the arrangement and installation of the contacts CW ₁ and CW ₂ can be freely determined. Moreover, the number of primary windings W ₁ need not be the same as that of secondary windings W ₂ . The invention can also be used in devices for testing separate cores and the invention can also be used in other power supplies, such as power supplies for loads of rectifier circuits or power supply windows for distorted current loads.

As described above, the present invention has the effect that the core winding operation in the core test apparatus is unnecessary, and the accurate and stable test (measurement) can be performed without adjusting the input voltage even when the magnetic flux density of the core is extremely large. Will be.

Claims (4)

A reference sine wave voltage (E ₀ ′) generating means (1004, 1005), means 1008 connected to the output of the reference sine wave voltage generating means and a load and detecting a voltage error (ΔE) therebetween; Means for delaying the voltage error by one cycle of the reference sine wave voltage (1006, 1007) and the reference sine wave generating means and the delay means, connected to a voltage error detecting means, to determine the delay voltage error as the reference. A means 1009 for adding to a sinusoidal voltage, and means 1010 and 1011 connected between said adding means and a load for supplying an input current I _p to said load. Device. The power supply device for a load according to claim 1, wherein a negative feedback loop is connected between an output of the load and an input current supply means. _2. An input current supply means according to claim 1, wherein the load is composed of a core 1 having a primary winding W ₁ and a secondary winding W ₂ , and supplies an input current to the primary winding W ₁ . And an output voltage detection means for detecting a voltage between both ends of the secondary winding (W ₂ ). 2. The apparatus as claimed in claim 1, further comprising an adding means (1102) connected to said voltage error detecting means and said delay means for adding said delay voltage error [Delta] E 'to the voltage error [Delta] E. Power supply for the load.