KR830000399B1 - A color television receiving system that utilizes multiple modes inferred from high frequency calibration to reduce color mismatch. - Google Patents

Abstract

No content.

Description

A color television receiving system that utilizes multiple modes inferred from high frequency calibration to reduce color mismatch.

Figures 1a and 1b illustrate a functional diagram of a single mode operation of inferred high frequency calibration circuits that are automatically driven or stopped.

Figures 2a to 2c are schematic diagrams illustrating the functional operation of a simple multi-mode inferred from high-frequency calibration and decision circuits.

FIG. 3 is a configuration diagram of a first embodiment of a receiving system using multiple modes derived from a high-frequency calibration circuit according to the present invention; FIG.

FIG. 4 is a schematic diagram showing the characteristics of the important circuit elements used in the other embodiments described herein and in the embodiment of FIG. 3; FIG.

FIG. 5 shows suitable waveforms and passbands for the mode decision; FIG.

Figures 6-10 are schematic diagrams illustrating AC mode decision circuits useful in multiple modes deduced in a high frequency calibration system.

Figures 11a-c are schematic diagrams illustrating the operating functions of the mode decision circuits that respond constantly to monochromatic signal components.

12-13 are diagrams illustrating decision circuits for non-synchronous band receivers being processed as multiple-mode calibration and for multi-band multiple-mode calibration.

14-14 are diagrams illustrating other embodiments of receive systems using multiple mode calibration circuits in accordance with various aspects of the present invention;

Color television transmission systems used worldwide are driven or based on the signaling specifications originally prescribed in the United States by the International Television Systems Association (NTSC). These systems, referred to herein as NTSC type systems, include the NTSC system used in the United States and the well known PAL and SECAM systems used abroad. These systems utilize composite color television signals that include a wideband monochromatic signal and a plurality of chrominance signals (sometimes referred to as color differential signals).

In general, the broadband monochromatic signals represented by the symbol Y 'are a combination of three primary color signals, i.e., red, blue, and green, which have been previously calibrated by the power gamma characteristic of conventional display tubes.

The presence of pre-calibrated in the element of the signal can conveniently be displayed by displaying the signal as a prime ('). A monochrome code is typically denoted Y '= ΣA _c C' = A _r R '+ A _g G' + A _l B ', where C' represents a gamma correction primary color signal, A _c , A _r , A _g and A _l denote the nominal comparative brightness coefficients for the respective primary colors, and R ', G' and B 'represent the gamma corrected color signals for the respective basic red, green and blue colors.

The monochromatic signal Y 'defined herein must not be confused with the color matrix name Y, which is the corresponding combination of uncorrected primary color signals, nor should it be considered equal to the gamma corrected brightness signals. Because in monochromatic signals, this is a calibrated independent variable and not a mixture.

That is, Y '= ΣA _c C' does not exclusively represent the general gamma correction index Y = ΣA _c C.

Chrominance signals in a typical NTSC type system include a signal indicating the difference between a gamma corrected primary color signal and a monochromatic signal or a linear combination of these color chrominance signals. To illustrate, color differential signals can generally be represented by the equation (C'-Y ') _L , where C' represents a constant gamma corrected primary color. Also, the symbol L below indicates that the chroma signals are typically reduced relative to the Y 'signal to be transmitted in the wide bandwidth, which represents the reduced bandwidth at the receiver.

Typical NTSC transmission systems are designed to transmit a linear combination of Y 'of the full-wave bandwidth and a chroma signal of the reduced bandwidth. In the United States, for example, (R'-Y ') (B'-Y') and (G'-Y ') are transmitted in linear composite chrominance signals designed as L' chrominance signals and Q 'chrominance signals. The linear composite chrominance signal is measured as the peculiar phase of the chrominance subcarrier known as the color sketch. Thus, for example, the I 'and Q' signals define the respective color slaughter axes.

While the I 'and Q' signals have distinct bandwidths that are significantly narrower than the Y 'signals, the excess of the relatively wide bandwidth I' signal is lost in most receivers designed to operate in the same band.

Conventional receivers have used varying grades of intermediate insertion bandwidth I 'signals that are either equivalent chromaticity for all axes or transmitted as a single sideband component and added.

Other receivers also use a simplified approximation to the nominal I 'passband or use a broadband equal-bandwidth system. To reduce the transient chromaticity time, these receivers adopted the error chromaticity component in the nominal single-axis I 'component, which was variably proportional to these error components between the I' and Q 'channels.

The equilibrium state with the circuit arrangement relating to the improved chromaticity of the present invention will be described herein and will also be described herein for the purpose of processing I 'and Q' chrominance signals of the same bandwidths for processing first and then not the same bandwidths The apparatus and method will be described.

Typical NTSC receivers demodulate and matrix received chroma signals to (C'-Y ') _L , which is a number of reduced bandwidth chroma signals.

The receiver then effectively adds the monochromatic signal Y 'to the respective chroma signals to drive a plurality of signals, each containing a high frequency component combined with the low frequency components C' _L of the primary color signals generated by the color camera .

The low frequency primary color components are sometimes referred to as wideband color signals.

Makin used high-frequency monochromatic signal component Y _'H is a general high level signal mixed in, because this would be able to only transmit or only indicate the particular combination of a high-frequency primary color components.

It has been observed too many times that conventional NTSC type receiving systems exhibit visible color mismatches during viewing, especially in areas where one lightness shifts to another lightness, or within a single color or a sharp transition from one color to another. Also, in the case where conventional NTSC type receivers compare visibility errors and a large number of polarity transitions and high inaccuracies within the region of the color variation when all the primary color signals are compared against a reference presentation with visible bandwidth mismatches with a bandwidth that is comparable to Y ' Lt; / RTI >

These mismatches are clearly observed in the latest receivers with resolution (resolution) and brightness error, chromatic blot, local destabilization, and blurred light.

In the transition phase of the single primary color, the high frequency component may be much smaller than the low frequency component.

In the transition steps of several primary colors, the high frequency components may be inaccurate amplitudes to provide even a single color metric such as brightness Y. [ Moreover, due to the conversion of the brightness ΔY is ΣA _c ΔC said ΔY 'is ΣA _c ΔC' is also such a polarity having a high-frequency component of the visible brightness Y reproduced on sometimes stage high-frequency components of, Y ', so that confusion. This mismatch occurs on steps where? Y 'is a single polarity and? Y is of a different polarity. In the step of transitioning from the first primary color of one region to the second primary color of the horizontally adjacent region in the case where the third primary color is small or absent, the conventional color television set receiver has the following characteristics: (1) amplitude in the high- (2) the opposite polarity in the high frequency component of one primary color, and (3) the rectified high frequency components in the other primary color in the high frequency part of the step, producing unsaturation and quasi low frequency components therein. And the apparent saturation color and the observed pseudo-darkness, which is usually known as the lightness error, in the transition between the different color of the color metric supplementary component.

Conventional color television awards in the area of definite color saturation may result in (1) inaccurate due to inadequate high-frequency components of one or more strong primary-color colors, or (2) excessive high frequencies in one or more weak primary colors Will typically exhibit over demodulation and process flow due to signal components. A simple example is a single saturated primary color.

In this case, the transmitted high frequency signal useful for the unique primary color from the high frequency signal Y 'is very small while the same Y' signal components are overdone in the other primary color colors.

Moreover, in general, similar inconsistencies often occur and are always visible when the local color is clearly out of white. These mismatches appear to be clearly visible to the color television set as resolution, brightness error, chromaticity error, and sometimes local oversaturation and quasi-low frequency error. Color metric errors, which generally appear in the form quoted in normal television images, occur differently and commonly in different parts of the frequency spectrum and also occur in other parts of the image.

In conventional receivers that receive the transmitted signal in the monochromatic signal Y 'portion, if the color phases are generated including the shifted single side wave I' components, the transmitted signal side major wave I ' And is clearly drawn into the spectral range from what can be seen by the application of smoothness and test pattern. This color mismatch is particularly noticeable near the vertical sides. Using a multi-line signal synthesis technique, there is a tendency to transition indirectly to other patterns, such as sides of other slopes, or to provide a pattern at the sub-high frequencies of magnetic field ratios.

It will be readily appreciated that the image of NTSC-type signals and the conventional receiving method have already been widely recognized to result in the above-described color mismatch, so that they do not have satisfactory results without the desired receiver calibration circuit in the prior art.

In order to reduce these distortions, the prior art has largely been a drawback by using the Y 'signal on the transmission instead of the original brightness method, such as a gamma corrected Y signal. Therefore, this method has included various other methods of pre-calibrating the transmitted monochromatic signal and a method of converting the signal to Y 'to Y with inverse gamma power.

All these methods,

(1) they do not normally perform the desired color correction,

(2) they reduced the quality of the image in other respects

(3) In many cases, they have not been substantially used because they are not fully compounded.

Specific problems with inadequate high frequency components have been identified, but the proposed solution has, on the other hand, made the quality of the image worse. For example, the receivers of the prior art utilize the increased gain in the high mixed region of monochromatic signals. However, this method does not provide polarity correction, can not provide the required differential comparative amplitude in the independent primary colors, and improves the quality of the displayed image because it increases rectification and coloration. It was also aimed at demodulating common mixed high-frequency components by the ratio of squares of gamma-fixed brightness evaluated by Y 'provision. This method also does not provide polarity corrections and differential comparative amplitudes, and will be used to increase the transconductance and critical commutation, such as pseudo-high frequency signal lights.

According to the present invention, a color television reception system of an NTSC-type signal is provided with a signal processing circuit and an inrushing high-frequency component which is visible in a region of color transition and in a region of color saturation including a high-frequency component, Thereby providing a method for reducing color mismatch. These mismatches may be reduced by effective correction or by chromatic or primary color signals having high frequency chrominance signals or color components driven from the signal information that are useful in NTSC type signals and found to be useful within their specifications . More specifically, in the sharp color transition region and in the region of high color saturation, the high frequency chrominance components can be divided into the high frequency portions of the received monochromatic signals by the difference due to the counting of each logarithm.

The receiving system automatically determines when fixed rods approximate a given time or near the frequency range of the received signal and when it is appropriate to automatically activate a calibration circuit to provide the appropriate one containing high frequency calibration signals And a high frequency calibration circuit for responding to the received NTSC type.

The crosstalk of single side color components into the generally displayed region of the Y 'channel is suppressed by the use of reasoning components driven from the I' component deduced on the transition stage to effectively neutralize the transmitted crosstalk. Leaks of Y 'into the chromaticity, i.e., reddish, are reduced in the receiver according to the invention by using a narrow chromaticity passband and by using the chromaticity components or the like used here to reduce the color spots.

(1) theory of mixed high frequency components by inference

Contrary to the prior art which contemplates the problem of correcting the above-mentioned color misalignment, in the present invention it has been seen that the use of a monochromatic image signal Y 'is sufficiently possible and that such color mismatches are not due to the form of the Y' But mainly due to a significant reduction in the bandwidth of the chroma signal at the transmitter, at least not due to a reduction in the conventional same-band receiver. Accordingly, in the present invention, under the object and condition of specializing this concept, the contained high frequency chrominance components can be driven from NTSC type signals to compensate for the reduced bandwidth chrominance or low frequency primary color signals so that visibility mismatches Can be additionally reduced.

These contained components may be generated in one of the design (C _"H -Y _'H) contained in the high frequency color component and the corresponding deduced probably been generated high-frequency color component directly to C" _H in this respect. By efficiently or relatively imposing these chromaticity components on the chromaticity or color channels, the very high R " _H , G" _H , and B " _H different color from the other color in the highly colored region of the image, This solution to the problem of correcting the desired color mismatch has the following important advantages.

1. The currently transmitted signals can be used without any conversion,

2. Easily attachable when designing television receivers. It is therefore very simple to include these in existing devices and systems.

The features of the present invention are as follows.

Local to the appropriate inference of reconstructing the high frequency components have been received video signal Y with respect to the relevant portions of the NTSC-type signal, _{i.e., ΣA c (C "H -Y} ' to maintain the fidelity to be _H) = O.

This feature becomes even more important because only Y < 1 > of the transmitted signal components supply sufficient broadband of the primary color signals.

This fact is based on the nonlinear color space determined by Y ', R'-Y' and B'-Y 'measured separately along the world's major contractions. In this interval, the primary colors R ', G' and B 'each indicate the specified direction, and each has a positive image on the Y' axis. And, since only Y 'is transmitted with sufficient bandwidth, it is only the transmitted components that provide an important way of the primary color signals beyond the sufficient bandwidth of light. The fidelity of the received Y 'signal is a prerequisite for faithfulness of important color signals generated only in the camera. It is also a feature of the present invention to provide such polarity and amplitude such that the inferred composite high frequency components for the significant portions of the NTSC type signals will certainly show less of the above-mentioned color mismatch.

More specifically, the inferred high-frequency signals typically reduce the difference between R ', G' and B 'generated by the receiver, and the differences between the broadband R', G 'and B' . A major advantage of the present invention is that the signal components deleted by the synthesized high signal components and the signal components deleted in the same band signal processed by suppression of the single sideband I 'components do not need to be deleted in the image .

Appropriate and specified suppressed signal methods and circuit components of the inputs are perceived and utilized. And the inferred high frequency chrominance components can be differentially combined or driven to effective recombination to effectively reduce visibility color mismatches including those mentioned above. Bandwidth limitation of the NTSC color system driving in the present invention the two color channel, that is, causing the (R'-Y ') _H and (B'-Y') of the missing high-frequency components of the _H or I 'and Q _{H' H} do.

In conventional receivers, only low frequency color coordinates are transposed for gamma correction and imaging. Receiver coordinates are generally _{C 'R = (C'-Y} ') L + Y 'L + Y' H = C 'L + Y' being represented by _H transmitter color coordinates C _'T = (C'-Y is represented as a _{') L + Y' L +} Y 'H = C' L + C 'H. It will be appreciated that the missing signal information in conventional transmissions that are equal to or not equal to the receiving bandwidth of the NTSC signals is at a high frequency chromaticity.

If the missing chrominance components (C'-Y ') _H can be significantly restored at the receiver, the image combinations observed in the color receiver using the shifted monochromatic composite high frequency signal will be additionally eliminated.

In accordance with the principles of the present invention, the lost high frequency components (C'-Y ') _H are deduced from the signal information available at the receiver, and the circuit elements are separately and separately provided to provide the differential high- is provided to generate a high-frequency component of the inference is designed to be used independently _{(C "H -Y 'H)} . the high frequency components are mixed in the image does not have the same color in each of the screening with the inference signal telrebijyon In order to eliminate the high characteristics, the high frequency signal components also have the same result as the resultant synthesized high frequency signal components.

A further advantage of the present invention is that when the separable frequency time portions of the video signal are limited to one or another of the determined color high frequency components and determined gauge instructions and regulation, the effective inferred chroma high frequency components suitable for the respective portions of the video signal Independently generated and combined in it.

Yet another advantage of the present invention is that the inferred components of I 'can be used to eliminate or reduce such interference when there are defined and limited signal conditions of the visible interference I', Y '.

(2) The theory of multi-mode operation.

In the present invention, we have found a number of factors indicative of the possibility of multiple-mode calibration. These factors include the fact that there is a possibility of other types of calibration that can vary from one region of the received signal to another, as well as the difference in calibration signals required to correct other portions of the received signal, But also signal information present in usefully received signal components that are suitable for providing a drawback to other types of possibilities. These decisions can be utilized as a basis for selective control along multiple calibration modes.

For example, a method for providing inferred high frequency calibration signals of a color transition region is different from a method for providing a calibration signal in the region of high frequency and high color saturation. While each theory involves counting high frequency calibration signals from the high frequency portion of the received monochromatic signal, the coefficient factors that process the algebra and process the circuit are different.

In addition, the contents of typical color television images change sufficiently when a plurality of other calibrations are ideal.

In an embodiment, the content of the color television image is to be driven strongly in the components resulting from the color steps, and the inferred step calibration will benefit the atmosphere.

In another embodiment, the inferred independent high frequency correction is usually beneficial since the step content of the image may be visually ignored. Within other areas, two types of calibration can be beneficial at the same time within the temporal and spatial locations of the displayed image.

In view of the trend of these factors, it is an object of the present invention to provide a color television reception system having an inferred multi-mode high-frequency calibration circuit. Such circuitry typically includes a plurality of calibration mode circuits responsive to the received telvision signal to process the high frequency calibration signals deduced to different modes and a plurality of calibration mode circuits responsive to the received frequency A mode decision circuit responsive to the received signal to automatically determine in accordance with the calibration modes and a decision circuit responsive to the output of the decision circuit for automatically operating the circuit generating the inferred high frequency calibration in the fixed mode determined to be ideal And a mode control circuit.

(3) Inferred step High-frequency calibration mode (Mood A)

There are six calibration modes related to the present invention. These six ranges are: (1) inferred step high frequency calibration mode (2) independent high frequency calibration mode (3) inferred multiband high frequency mode (mode C): where modes A and B are the receiver Can be taken independently in response to the divided items of useful frequency-time signal components.

(5) Inferred Separated High-Frequency Modes: Here, the inferred constant-chrominance high-frequency is the frequency of Y ', which occurs at the same frequency and time interval, It is generated from a separate part of the high frequency (mode E)

(6) Here, the signal is a neutralization mode (mode F) in which a jump address given as a step signal is given and addressed.

식=

제1(b)식이다.The inferred step high frequency correction mode (signal A) generally includes signal processing methods and circuits for reducing visible color mismatches in the region of distinct color contrast or for processing inferred reconstructed high frequency signal components. In a color television reception system that uses colorimetry deduced from high frequency signal components in the area of the title color transition, which is filed in the United States of America, the present inventor has proposed a method for processing such calibration signals as inferred high frequency chrominance components Circuit. An important form here is

Expression =

This is the first equation (b).

Where Y ' _L is the low frequency portion of Y' with composite frequency-shifting characteristics, and therefore the transition responds to the chrominance component (C'-Y ') _L stretched to the bandwidth. Also, Y ' _H is the portion of Y' that is compensated for Y ' _L.

For simple in-band reception, Y '= Y' _H + Y ' _L and (C'-Y') _L depend on passive chromaticity sides. I ', Q', and inferred high frequency chrominances processed in response to Y '= Y' _LI + Y ' _HI = Y' _LQ + Y ' _HQ can be obtained with the same metric equation as the low frequency chromaticity .

It can be seen that the high frequency step components deduced from equation (a) are effectively counted from Y ' _H by the factor N _S that occurs during the maintenance phase of the comparative transformations.

The high frequency step components deduced from the first equation (a) are effectively counted as Y ' _H by the coefficient N _S represented by a constant ratio of the comparative transitions in the steps.

It can also be seen that the high frequency step components deduced from the first equation (b) are effectively counted as a non-conversion of the associated low frequency chrominance according to the waveform ratio of the components from the monochrome Y 'signal.

Also, iteratively and likewise, the inferred high frequency waveforms including the step calibration are processed as the composite primary color high frequency component C &_quot;

The signal C _'L represents the low-frequency component of a certain color. Chrominance components is made easier from the expression (C'-Y ') it is compared with the received and synthesized monochrome signal Y' filter portion Y 'sum of _L.

In each case, the chromaticity is effectively applied. The counting in the expressions (a) and (b) can be compared with the expressions in the expressions (a) and (b).

The calibration indicated by equation (2) (a) includes polarity because it can determine the polarity of the non-numerator or denominator that defines (1 + N _s ).

The derivation of the re-synthesized high frequency in the region of sharp color transitions stems from the fact that the ratio of the lost high frequency chrominance to the high frequency lightness in this region is proportional to the ratio of the derivation function of the corresponding useful low frequency component. This relationship can be assumed to be the result of a step transition chromaticity signal vector that properly determines the colored pores to maintain direction.

The operation that effectively adds to the composite color signals of the inferred chromaticity components corresponding to the algebraic expression can be utilized as a normally-on mode or a mode selection operation only for a specific time. The above signals are utilized in a normally-on state where the mode decision circuitry does not have a particular counter signal state here, or if the step correction determines a mode that is very appropriate for the signal content. In the latter case, all directional corrections in the color space are obtained for the sine wave form in the frequency domain (C'-Y ') _L and Y' _L is superimposed at the frequency of Y ' _H.

An epoch operation mode determination circuit may be utilized to determine the step specifically indicated for the desired epoch.

(4) Independent High Frequency Calibration Mode (Mode B)

In general, the independent high frequency calibration mode (mode B) is a signal processing method for processing inferred resynthesized high frequency signal components for reducing visibility color mismatch for other components than step transitions where high frequency colors are independent of low frequency color And circuitry.

The areas of special relationship for each high frequency are areas of high frequency color saturation. Recently, in a color television reception system using a high frequency signal component deduced to reduce color mismatch in the region of high frequency color saturation, which is the second time application to the United States of America, the inventor has described a method and a circuit for processing an independently deduced high frequency Respectively. In the important formula of the invention described in the above application, the lost high frequency chrominance components (C _" -Y ' _H ) can be obtained from the following equation.

Here too, the inferred high-frequency components are counted from Y ' _H but also by another factor N _I defined by the third equation. Repeatedly, the inferred high frequency components can be processed as inferred high frequency primary color components of the following equation.

The factor N _I does not have to be constant for each high frequency. Moreover, the factor (4 + N _I ) of equation (4) is always positive because of the ratio of the two signals of the common polarity.

This mode can be utilized as " normal-on " while the mode decision circuit does not have a particular counter signal condition or each high frequency calibration determines whether it is very suitable for signal capacity. In this case, there is a tendency to provide an increased approximation in response to the sine waveform component of the frequency overlap region of the modes Y ' _L and Y' _H. These modes are also used only when individual high frequencies are specifically indicated during a particular time attack.

(5) Multi-band operation (mode C)

Which consist of one or the other specified in different parts of the third alternating-mode spectrum useful for minimizing visible spots by limited bandwidths of useful chrominance components, and two cases of independent high frequencies. As described herein, the particular mode decision circuitry has concurrent high frequencies, each with a particular separation region, so that specially corresponding inferred high frequency chrominances are generated simultaneously for each portion, And the Y 'signal with multiple components of time. This multi-band operation is also used to receive the non-synchronous bands I 'and Q' of the decision processing appropriate for each axis.

In all modal present here, Y 'fidelity for _H limits ΣA _c C' _H = Y 'when the _H becomes compensation is always obtained, since the Y' fidelity for _L are the first and second, 3 and 4 equations or equivalent algebraic expressions are suitably applied.

(6) Reconstruction Mode (Mod D)

For example, the use of a fourth surrogate mode during a fork at a time when it is a chromaticity or color step is obtained from a reconstruction of the chromaticity or color phase.

This technique is based on the concept that the function determined by the ratio of Y 'high frequency to the time ratio of Y' _L low frequency is definitely constant and has a reconstruction characteristic at any stage. Thus, the partially reconstructed chromaticity phase high-frequency signal (H ' _HS - Y' _H ) is derived in the middle as follows.

or

Where the nominal term is a function with useful and widely used properties such as the ratio of the Y 'high frequency to the time ratio of the Y' low frequency transform, as defined below.

Also in this case, to the state transition from one color to another color, the color non-component (or a corresponding color to the non-component) containing having properties, such as after the d / dtY _'H versus Y' _L corresponding to the ratio Are used for the facts expressed in the first and second equations that are multiplexed by the waveform.

This waveform for a constant step bandwidth has a priori form with known amplitude, polarity and shape. However, as indicated above, the gist of the present invention resides in determining the effective bandwidth of a stage when one exists. Therefore, it is preferred to _{first have} a collection bandwidth automatically selected from a plurality of such bandwidths to provide a signal component representation of the received value &_lt; RTI ID = 0.0 > ₁ applied by the mode decision circuit described herein, and to provide a suitable form of reference waveform, It is possible to recapture or reconstruct the time. So that by the method of the circuitry described hereafter it is possible to effectively reproduce the composite color signals including the inverse of the Y 'low frequency and the chrominance low frequency conversion ratio and the component reproduction effectively added to the manufacture of the Y' high frequency Do.

(7) The inferred high-frequency (mode E)

The fifth surrogate mode uses the common nominal ratio of component representations of the signals coming out of the portion of the Y 'signal as described above. Processing in this form thereafter makes it possible to define a circuit arrangement and method for the separation of Y 'high frequencies into portions contributing to independent high frequency Y' _H and to portions contributing to the step components Y ' _HS .

here

Y " _HS = [ρ ' ₁ (nominal)] × d / dt (Y' _L ) Actual

And Y ' _HI = Y' _H - Y ' _HS .

Here the circuit device and the method described enables the application of the concept to which may be consistently separate parts of the signal Y _'H.

Moreover, in the application of the reference principle, it is possible to process and separate intelligent signal components which are comparable to the processing of the remainder after the subtraction from the synthesized Y 'signal. The inferred step high frequency calibration components and the inferred independent high frequency calibration components can be extracted from Y " _HS and Y ' _HI, respectively, and combined with chromaticity or color signals.

(8) A neutralization mode (mode F) for restricting crosstalk from I 'to Y'

A sixth mode suitable for improving the reproducibility of a color (or black and white) television receiver signal is to effectively nullify a single sideband I 'component which reliably stretches to a portion of the frequency band produced by a higher frequency component than the Y' The present invention relates to a circuit device and a method as described in the present invention. This crosstalk is due to increased visibility due to the use of gamma corrected signals and the combination of basic nonlinearity images.

When a combination of multiple linear signals is used by the application of the dog delay technique, it will be conveniently converted into a pattern without removing crosstalk components.

이고 추론된 I' 고주파보다 I' 저주파의 변환비의 현존함에 의하여 확인되는 단계가 단일 측대 색조 스펙트럼에서 나타내어지는 바와 같이 포함되어질 수 있는 것처럼 교정성젼이 조건의 모우드 결정회로에 지시를 제공한다.Portion of the Y _'H according to the techniques of the present invention, that is, the low-frequency part Y is, for example _HL

And the calibration template provides an indication to the conditional decision circuitry as if the step identified by the presence of the conversion ratio of the I 'low frequency than the inferred I' high frequency could be included as shown in the single sideband hue spectrum.

Then a comparable component of opposite polarity may be generated in the receiver by the circuit devices and methods specified herein to reliably suppress crosstalk. Since single-sideband components are partially shown due to non-linear processing, they can be reliably reduced by a simple approximation to neutralization.

(9) Operating area and determination

The operating modes are due to selective operation as indicated herein with respect to mode decision and control such that circuit devices and methods configure or operate these mode decisions and controls.

비의 존재에 의해 추론됐었다.The determination of whether the associated step is present at a particular time has the same appearance as the amplitude and polarity characteristics and the nonzero ratio change of the comparable chromaticity component

It was inferred by the existence of rain.

Such a presence can be measured in various ways, where various methods are employed by means of a clearly related waveform, such as a first component recurrence ratio of Y ' _H and a second component recurrence ratio of d / dt Y' _L , Or specific means of such a waveform by means of an exact difference. Theories and methods of these means will be explained as follows.

The decision to do not do individual high frequency are present and are present at the same time, the color components (C'-Y ') that is proportional to the ratio Y that must exist within the continuous amplitude comparison for showing how the _L and clearly becomes independent high frequency' _H The measurement is based on.

가 국부적인 그리고 영이 아닌 색도(C'-Y')_L의 동시적인 존재에 따라서 사용되어진다.Moreover, it requires that d / dt (C'-Y ') _L be simultaneously and relatively small, since the independent high frequencies are high frequencies in local color or local color, such as are compared to the transition between two colors. However, since only the Y 'signal is extensively useful at the receiver without bandwidth limitation, the quoted ratio derived from the component of Y'

Is used according to the simultaneous presence of local and non-zero chromaticity (C'-Y ') _L.

The same decision method can be used for independent high frequencies as if it were for a stage, not for a particular other criterion. It may also be applied as a method based on the signal reproduction of Y ' _H , as in the method for d / dt Y' _L of signal reproduction.

In the case of independent high frequencies this method can be used to exceed the threshold. When a direct linearly processed waveform ratio is used, this method will be exceeded (or very close) at zero amplitude of the signal Y ' _H. Wherein the first (C _"H -Y _'H) is gamyeo to zero by the base number and the amplitude threshold, the second device indicates that may lead to very small spikes (spike) during those intervals. Spikes and very short input here The signals are easily removed by a number of conventional circuit devices.

The circuits that perform these methods are called < RTI ID = 0.0 > mode decision circuits, < / RTI >

The outputs of these circuits are nominal mode control signals and for convenience, they are symbolically reproduced by B ₁ for control of the symbol β ₁ for phase step control and the independent high frequency (but respectively applied).

Here, β = 1 indicates an on state or a possible condition, and β = 0 indicates an off state or an impossible condition. Single and multiple controls will be described herein.

As shown, these crystal processes are driven for one or more axes, or for band operation, either unequal or unequal. Corresponding high frequencies are used or evaluated for the complementary synthesis band at the corresponding Y 'low frequencies, or the high frequencies are divided into a number of parts such as Y' _H1 and Y ' _H2 .

In all cases, a clear distinction is made between time delays that can be compensated in the conventional way and time propagation due to non-ideal signal superposition. And circuit devices and methods are shown to suppress propagation of the waveform of time. The modality decision and the appropriate time are shown in Table 1 by the following symbols.

β _s → 1. For acceptable or acceptable steps

(1 -? _S )? 1. For unacceptable steps

[[beta _s ]] - > 1. For acceptable or precise time steps

β ₁ → 1. For independent high frequencies that can be accepted or received

(1 -? ₁ )? 1. For unacceptable independent high frequencies

The particular mode decision circuits disclosed in the foregoing coarse theoretical description and in the specification, and the mode decision circuits describing in more detail the illustrated circuit, include those defined in Table 1. [ These circuits are shown to be effectively reproducible with respect to signal ratings and / or by having circuitry and methods for signal normalization and signal comparison.

Y'_L그리고 선택 성분 Y"_H와 Y"_L들은 주파수 성분 Y'_H1Y'_H2에서 분리되어지는 것으로 구성되어지며 이들의 비는 다음과 같다.The waveforms that are component representations of the above states are as follows. Y ' _H , Y' _L and

Y ' _L and the optional components Y " _H and Y" _L are separated from the frequency component Y' _H1 Y ' _H2 , and their ratios are as follows.

그리고

로 나타내지는 신호는 Y'_H와 Y"_H중간의 통과대역을 갖는다고 알려졌다.

And

The signal was reported as having the Y 'and Y _{H "H} middle of the pass band represented by.

In this case, several waveforms will be useful at the same time and this will be explained.

For the associated Y ' _L , the nominal (step) epoch is used here to represent the TS, and the TS will vary by non-identical Q' and I 'band systems.

The presence or absence of minimum time propagation will be fully described in accordance with the embodiments herein and will be represented in simplified form in Table 1. [

[Table 1]

10. Mode decision to control generation of inferred high frequency

The reproducibility description is shown in Table 2, which is a mode decision to control the inferred high frequency generation according to the technique of the present invention.

Substantially inferred high frequencies are simply denoted by N _s Y ' _H or N ₂ Y' _H unless the other form is a related primary color such as another specified type.

In this table, the symbols β _S and β _I or (1-β _s ) and (1-β _I ) are used to represent single-mode decisions. The double bracketed symbol [[β _s ]] is used to represent the correct time gate of the appropriate period T _s . Symbols [β _S ] and [β ₁ ] in single brackets are used to represent time-delay-array isochronous control according to the following equation:

[β _S ] = [π _K β _SK ] [π _J (1 - β _IJ )]

And [β _I ] = [π _J β _IJ ] [π _K (1 - β _SK )] where π _K or π _J indicates what is indicated by the sign K or J. They are not necessary for an independent combination.

The various modes of operation will be described herein in a pre-specified fashion. The types shown in Table 2 are as follows.

1. Step algebra running all the time

2. Time-gated step operation during ideal epoch T _S

3. Independent mod algebra that operates all the time

4. Instantaneous independent high frequency operation of step algebraic operation alternately during all cycles of a generic waveform with compatible phase operation

5. At the stage of the period T _S , during the port, a special keyed (alternately, stepwise algebraic instantaneous independent high frequency operation

6. Instantaneous phase with independent logarithms operating alternately during all periods of a general bipolar waveform with independent operation

7. The periodicity as indicated by the signal information is operated in single phase algebraic operation

8. Single independent algebraic instantaneous operation with periodic stop as indicated by the signal information

9. Simultaneous operation of insulated and redundant insulated

10. Described for the Y ' _H signal and the axes with independent modes and step modes and multiple Y' _H auxiliary parts operating as controlled by simultaneous decision equalization of time-domain for each auxiliary part Multiple simultaneous operation

11. The step corresponding to the keyed step in the porch T _S at the time with the split decision to control the output signals sufficiently corresponding to the input signals and the auxiliary parts of the high frequencies

12. Reconstructed steps of Y 'high frequencies, including splitting decisions etc., for the respective separable part

13. Separate processing where Y ' _H is separated into concurrent portions assigned to step high frequencies or independent high frequencies

14. A reconstructed step of inferred chrominance high frequency having controlled separated parts and reconstructed separation

15. The inferred transition step used to make the crosstalk neutralization of the I 'single side band with Y' as described by Y ' _HI for the magnitude of the gain

16 with respect to the magnitude of the gain (but Y ', including _H2 optionally extended) Y' is inferred to be used for "in I 'is that described by Y _H1 to the single-band crosstalk neutralize the reconstructed phase I'

17. An example of a control signal application having input signals that are more controlled than the output signals or number factors to provide normal operation,

18. A method as in claim 6, wherein the input signals are more controllable than the output signals or number factors to provide normal operation.

19. An example of a control signal application in which input signals are more controlled than output signals or number factors to provide general operation as described in the preceding paragraph

If non-equal-bandwidth chroma processing is used, these forms represent the appropriate I ' _L and Q' _L and represent the bandwidths involved in the decision processing for the respective axes.

Some of these forms may be combined in accordance with the teachings of the present invention.

It is possible to perform the comparative operation by a plurality of operations shown in Table 2. [ For example, here is the following.

_{{N S [β S] +} N I [β I]} Y 'H + Y' H = {N S [β S] Y 'H + N I [β I] Y' H} + Y 'H = ( _{1N S) [β S] Y} 'H + N I [β I] Y' H = N S [β S] Y 'H + (1 + N I) [β I] Y' H

(Or controlled) to add the estimated chrominance input to the signal from another generator that can represent the actual sum of Y ' _H or Y' _H and the estimated chromaticity at a certain time, as if properly controlled, It is obviously possible to add chromaticity from the chromaticity.

They do not represent other modes of operation, since the resulting composite color signals substantially provide the same ones as those defined in Table 2. [ Other combinations are derived algebraically. That is to say, Table 2 shows a particular form of displayed color images, or in accordance with the teachings of the present invention, by substantially defining the special forms of the actual composite color signals provided for collection and presentation, namely R ", G" and B " So that the controlled inference chrominance component provides the effectively added signals.

[Table 2]

This is suitable for considering the response of the waveforms appearing in the chromaticity passband when the important computational schemes for the independent mode deduced by the step mode or the chromaticity high frequency persist in the idle mode. These pass bands cut out the minimum time vortices and stepped amplitude of transients. And have cut offs in common. For these components appearing in the Y ' _L and compensatory Y' _H signals, the step mode has the effect of smoothing or fitting a substantial passband response while providing sufficient transition correction. An independent mode composed of its basic operation of independent spectrum appearing at a stand-alone Y " _H has sufficient correction on pure primary colors, but only partially corrected if there is a colored residue in a multicolor background.

The quiescent on modes are useful as described herein. This step mode is keyed by [[β _s ]] for stop independent mode operation. By switching on the independent high-frequency to high-frequency switch on a stand-modal in geujeom to the step of stopping the high-frequency operation, and the (C'-Y ') _L and Y' _L curvature components in the selected frequency near the upper end of the band It is possible to provide a decision and control system that is capable of maintaining a stepped state in a stalled state.

Consider the effective chroma passband having the form of a nominal frequency of f ₁ hereinafter defined as F _L (f) for f≤f _1. Then Y ' _L has the same passband form and Y' _H is an addition. I.e. for f≤f ₁ will have a single component that is _{(1-F L (f)} . F = f ₀ from the left that _{F L (f) F LO,} W O = Let 's put that the 2πf _0.

이면

이고 ρ₁₁은 주파수 f상에 독립 상수인

에 비례한다.For various control methods defined by ρ ₁₁ , (ρ ₁ ) and ρ ₂ as described in various parts of this specification, common mode decision parameters not limited to β ₁ are important for the Y ' _L passband And can be set to maintain the step mode utilized for the curvature component of the portion and to sweep the independent algebraic expression for high-frequency components. For example, if

If

And ρ ₁₁ is an independent constant on the frequency f

.

For a limited bandwidth (ρ ₁ ), the mode is maintained for the entire low-pass band chromaticity to prevent overlap at regular frequencies. Also, a moderately filtered deformation of the signal (r ₁₁ ) or (r ₂ ) can be designed to make the transition point close to f ₁ . Nonlinear processing is substantially rectangular and provides switching transitions without conventional non-ideal delay and instantaneous effects provided for accurate transitions.

The use of the co-linear phase or non-overlapping passbands of the components in the numerator and denominator for Y 'coming out of the mode decision controls is such that the curvature components are contained in Y' _H of the compensating amplitudes of the overlap region of Y ' _L If there is a tendency to limit use. This processing method tends to prevent the quadrature relationship between the denominator and the molecule depending on whether the linear processing or the nonlinear processing is applied in the mode decision system.

"Considering the single primary color having a low frequency sine wave defined as ₁ sin W _O t C" C _"1 = C" ₁₀ + C at the transmitter _{1L = F LO C '1 sin} W O t, Y' L = A ₁ F _LO sin W _O t and Y ' _H = (1F _LO ) A ₁ sin W _O t. Easily if _{C 'IL + Y' H =} C '10 + C' 1 sinw o t (F LO + A 1 (1 + F LO) that is a sine wave amplitude chrominance channel response is attenuated increasingly by smaller or amplitude The broadband chrominance signal receivers use at least a large amount of leaking energy from Y 'into the chromaticity channel. There is a tendency to take out. As an inferred step equation

가 되며 이러한 이득은 축적되어진다. 이것은 효과적 통과 대역 응답이 단계과도 현상에 확실히 응답함이 없고, 시간와류 및 시간지연이 없이 실질적으로 스퀘어오프(squaredoff)된다는 것을 알수 있게 한다.

And these gains are accumulated. This makes it possible to see that the effective passband response does not respond positively to the step transient and is substantially squared off without time vorticity and time delay.

If the sinusoidal waveform consists of a number of primary color components, the same things are relatively visible.

The independent algebraic expression for a single primary color is

가 된다.

.

If there is no zero-killer value of the other primary colors, the fixation is not complete, but it has little effect.

With television receiving systems having an improved solution as compared to current receiving systems, the present invention reduces the need for portions of chromaticity that are constrained by crosstalk. Therefore, the present invention can reduce the color of the object caused by the band component gkf. Leaning was a major grave issue of the current TV reception system. Leakage is generated from the use of the band gkf, where the Y 'signal is generated in the band simultaneously with the chromaticity signal.

The crosstalk from the Y 'signal to the chroma channel produces a pseudo chroma which is rendered more visible by non-linear processing on the image. Colorless, high-frequency resolution (resolution) images will be described in detail, such as a stain. Color-free high-resolution images have residual brightness due to commutation due to gamma characterization and, if the image is moving, this provides a low-frequency light clarity that is clearly visible to people's eyes.

In addition to providing improved operation at the receiver, the present invention provides interference at the transmitter by further reducing the interference by constraining the I ' bandwidth to the double-side Q ' bandwidth and by constricting the Y ' So that it is possible to perform better transmission.

13. Relationship to color primaries

The ideal calibration for transforming color metric primary colors for use in a color television signal presentation other than the primary colors used in the NTSC type transmitter is as follows.

1) Induction C ' ₁ , C' ₂ , C ' ₃ 2) Induction C ₁ , C ₂ , C ₃

3) Matrix C _a , C _b , C _c 4) Re-application of gamma to obtain C ' _a , C' _b , C ' _c ,

Several approximations are commonly used for simple calibration algebra.

The present invention is not non-monochrome combined with these approximate values and by providing more accurate signals C " ₁ , C " ₂ , C " ₃ to all monochromatic signal bandwidths fw, .

BRIEF DESCRIPTION OF THE DRAWINGS The above and other features and advantages of the present invention will be more fully understood by reference to the following description, taken in conjunction with the accompanying drawings, in which: FIG. For ease of reference, the same elements are designated by the same reference numerals in the drawings.

(A) Inferred high frequency calibration circuit that is automatically turned on and off. Figs. 1a and 1b

Inferred high frequency generators that are automatically driven and stopped are described in the application filed concurrently with the applicant referred to herein, and a detailed description of their operation follows.

Referring to the drawings, Figures 1a and 1b are schematic diagrams that illustrate the operation of inferred high frequency calibration circuits that automatically turn on or off.

Figure 1a illustrates the operation of an inferred high frequency step calibration generator 10 that is automatically switched between an operating mode and a non-operating mode via an electronic switch 11 responsive to a step mode decision circuit 12. The step-mode decision circuit 12 includes a detection circuit device 13 and a switching signal generator 14.

The detector 13 responsive to the components of the received television signal may select one of the characteristics of the phase of the contradiction or coincidence that has an operation compared to the chrominance high frequencies substantially inferred to those described by the inferred step algebra, And generates an output signal Zs due to one or more signals. Here, the value of the output signal Zs is an indication of the degree of measurement. The switching signal generator 13 cuts this detector output signal to a dual switching control signal with? S = 1 or 0 corresponding to the presence or non-presence of a degree indicative of the capability of the inferred step high frequency logarithmic operation.

Here, the double switching control signal is defined by the above-mentioned threshold Zs.

These control signals are applied to the switch 11 for substantially activating or stopping the step calibration generator 10 according to the magnitude value.

This type of automatic drive calibration circuit is described in more detail in the applicant's co-pending application entitled " Color Television Receiving System Utilizing Inferred High Frequency Components to Reduce Color Mismatch in Color Transition Region "

1b illustrates the operation of an inferred independent high frequency calibration circuit 20 that can be similarly switched between an operating mode and a non-operating mode via a switch responsive to an independent high frequency mode decision circuit 22. [

The independent high-frequency-mode decision circuit 22 includes a detection circuit device 23 and a switch signal generator 24. The detector 23 responsive to the components of the received television signal provides an output signal Z _I. The value of the output signal Z _I represents an indication of the presence or absence of an independent high-frequency components of the received signal appropriately in the Mode Control.

The switching signal generator 24 generates a 2 < nd > switched seventh signal < _{RTI ID} = 0.0 > ₁ < / RTI _> = 1 or 0 as specified by a predetermined threshold Z _I corresponding to the presence or absence of the presence of the inferred independent high- Converts the detector output signal.

These control signals are applied to the switch 21 in accordance with the operation of the inferred independent high frequency calibration circuit 20 or the stop.

This type of dynamically operating independent high frequency calibration circuit is described in more detail in FIG. 11 of a color television receiving system utilizing inferred high frequency signal components to reduce color mismatch within the title filed high frequency color saturation region of the applicant .

(B) In-band receiving system using simple multiple mode derived from high-frequency calibration circuit (2, 3, and 4) Figures 2a and 2b illustrate simple multiplexing inferred in a high frequency calibration circuit according to one embodiment of the present invention. And a functional description of the mode operation.

The multiple-mode calibration circuit includes an inferred high-frequency step calibration circuit 10, an inferred high-frequency independent high-frequency calibration circuit 20, a mode control switch 31 for switching between the calibration circuits 10 and 20, and a switching control signal generator 34.

The detection circuitry accepts the components of the received signal and provides one or more output signals indicative of the likelihood of the degree of operation described above in the effective generation and synthesis of the chromaticity high frequency inferred according to either phase algebra or independent logarithm do. The switch signal generator 34 accepts the output of the detector 33 and converts it into a switch control signal for operation via the switch 31 to either the independent high frequency generator 20 or the step high frequency generator 10 according to a preselected mode control method.

For example, as shown in FIG. 2a, the switch 31 basically provides an independent high frequency calibration circuit 20 as long as the mode decision circuit 32 provides a switching signal value of? S = 1 or [? S] Can be driven.

Further, as shown in FIG. 2b, the switch 31 can basically drive the one-stage high-frequency calibration circuit 10 in which the mode decision circuit 32 provides the switch signal of the value? ₁ = 1.

Figure 2c is a functional diagram illustrating the operation of the mode decision and control system utilizing a number of mode decision circuits to increase safety.

A band separator, not shown here, is coupled to a synthesis mode decision circuit 32 for providing the form of a majority mode selection signal as? S? _I (1-? S) (1-? _I ) or [? S] RTI ID = 0.0 > signals. ≪ / RTI > The various mode decision signals may be temporarily constrained to an ideal time level by appropriate time delay elements (not shown). The signals from the element 32 are combined in the decision element 32 while simultaneously providing the forms of the plurality of mode decision signals? _S and? ₁ to control the switching required to the mode control circuit 34.

Still another control can be obtained by simultaneously requesting another mode decision signals < RTI ID = _0.0 >#sc< / RTI > Element 36 accepts low frequency chrominance signals and any Y 'signal as an input. If at least one component nonconversion exceeds a pre-selected threshold, the threshold is applied to determine for beta _IC . The parametric choice allows the use of a threshold within a 1, 2 or 3 dimensional color space for β _IC and a 1, 2 or 3 dimensional color space color conversion for βsc. The control signal? _IC is applied to the threshold of [? _I ] at the same time, and the control signal [? _S ] is applied to the threshold of [? _SC ] at the same time. Thus, Figure 2c includes circuit devices and methods for determining and responding to the components that emanate from low frequency chromaticity and which respond in common to components that emanate from the broadband Y 'signal. Examples of the mode decision and control circuits will be described in more detail below.

(C) Receiving system using multiple modes derived from high frequency calibration.

FIG. 3 is a detailed example of a receiving system using multiple modes inferred from high frequency calibration, a form described in connection with the second aspect. The processing of the color television receiver which supplies the reduced bandwidth chroma signals (here referred to as the low frequency chrominance) to the independent high frequency control signal generator 4 of the deduced independent high frequency generation circuit and to the step high frequency control signal generator 40 of the deduced step high frequency control circuit (Not shown) are shown in FIG.

Also, during the water phase, the components of the monochrome video signal Y 'are supplied to the control signal generator 40, the band distributor 41, and the mode decision circuit 32 through the band splitter 41.

The details of the mode decision circuits 2c, 5a, b, 6, 7, 8, 9 and 10 are described herein in detail. The form of the mode decision circuit shown in element 32 of FIG. 3 is based on the circuit of FIG. 9.

The input signals described or shown in each case will be applied as if they were described as appropriate comparison times. The mode coupling circuit 32 shown here is a form of a broadband decision circuit according to the technique of the present invention.

로부터 구동되어지거나, 혹은 이것의 재현인 제2신호성분들을 포함한다. 2개의 성분들간에 관계하는 신호들은 작동모우드를 위하여 광대역 결정정도를 결정하도록 사용된다.This is due to the special form to be used with a first signal component which is driven from the Y _'H

Or second signal components that are reproductions thereof. Signals related to the two components are used to determine the degree of broadband decision for the operating mode.

Other forms are used to have a particular benefit in the described place.

의 적당한 특별한 형태들은 처리수신기 부분으로부터 공급된다.Y ' _H driven from the input signals

Suitable special forms of < / RTI >

The various special elements shown in Figure 3 are defined in detail in Figure 4.

As illustrated in FIG. 3, the band splitter 42 splits the monochromatic signal Y 'into at least two components, and provides a compensated high frequency portion Y' _H (referred to as Y 'high frequency) and reduced bandwidth chromaticity signals , And divides the comparable low frequency component Y ' _L (referred to as Y' low frequency) in the frequency domain and bandwidth.

Because of the above reasons, in this particular embodiment the place of the band divider claim little faster than the time Y 'components are designed as general reproduction _{Y "H (tT D / 2} ) of _H Y' _H in the example, and the frequency overlap Y _'L in order to reduce the modal provides the decision circuit 32 in the Y _'H in bandwidth.

The inferred step-mode fixed-control generator 40 generates an inferred high-frequency chromaticity step calibration component (C " _H -Y ') that provides both time non-conversion of low frequency chrominance and Y' high frequency, all separated by the monochromatic low- ). ≪ / RTI >

By expressing the above expression,

Ns as calibrated by the excitation formula can be referred to as a step mode calibration signal.

The inferred high-frequency step-mode calibration circuit includes a band splitter 42, a step-mode calibration control signal generator 40, and a modulator 43.

The control signal generator 40 receives the low frequency chrominance from the processing receiver circuit and the Y 'low frequency from the band divider 42 and generates a control signal having the following form.

Specifically, the monochromatic low frequency signals are applied to the time difference sync 44 (A) and the resulting non-converted signal is subtracted from the denominator (A) of the non-circuit 46 (A) Lt; / RTI >

The low frequency chromaticities are applied to the differential sync 44 (B) and the resulting chroma non-converted signal is applied to the numerator input of the non-circuit 46 (A) via the (B)

The output of this non-circuit is the inferred step high-frequency control signal Ns as defined above.

The control signal from the generator 40 provides the inferred high frequency step calibration components by applying the Y 'high frequencies provided by the band divider 42 to the demodulator 43 when activated by the switch 31 via the switch 31.

The components provide the conventional C _"H -Y _'H and 48 in the signal Y' _H, Y 'and _L (C' Y ') as _L and the combination doeeojim _{C" = C "H + C} " L designed reasoning do.

According to the conventional matrix method, other forms are used to generate three color components according to the conventional matrix equation ΣAc (C "-Y ') = 0, which is used for the matrix of reasoning low frequency chromaticities known as the received low frequency chromaticity 2 may be utilized to the matrix from the side of the color _{(C "H -Y 'H)} .

The included independent high frequency modulating circuit includes a band spreader 42, an independent high frequency calibration control signal generator 41, and a demodulator 43.

The monochromatic low frequencies from the band splitter 42 are applied to the denominator input terminal of the noncircuit 46 (B) through the threshold A at 45 (B).

And is applied to the molecular terminals of the non-circuit through the type B which is critical to the low-frequency chromaticity 47 (B) from the receiver.

인 출력 독립 고주파 제어신호는 복조기 43의 스위치 31를 통하여 인가되어진다. 독립 고주파 제어신호 발생기가 스위치 31에 의하여 가동된다면 복조기 43은

에 의하여 규정된 추론된 독립 고주파 색도 교정 성분 (C"_H-Y'_H)들을 제공한다.

The output independent high frequency control signal is applied through the switch 31 of the demodulator 43. If the independent high frequency control signal generator is operated by the switch 31, the demodulator 43

(C _" -Y ' _H ) of the estimated independent high-frequency chromaticity correction components (C &_quot;

These calibration components are combined with the signals presented in the summing circuit 48 as described above relatively.

As described herein, the form of the mode decision circuit 32 includes a signal comparator and neutralizer 33 and a display class and control signal generator 34.

The circuitry 33, which may be responsive to the components representing the steps or independent high frequencies, receives a portion of the reproducible monochromatic signal of Y 'high and low frequencies and compares their relative values to determine the expected And provides a comparison signal indicative of operation.

In this particular embodiment, the integrated unipolar method Z _H of the monochromatic high frequency signal is compared to the monochromatic low frequency signal of the non-circuit 46 (C), the time-conversion unipolar method Z _L.

Particularly, the monochromatic low-frequency signal from the band splitter 42 is applied to the car recorder 44 (C), and consequently the low-frequency ratio of the converted signal is due to the positive pole of the square circuit 43, And is applied to the molecular input terminal.

Solid high-frequency signal Y _"H is a time interval beyond the T _D becomes applied to the time integrator 55 to integrate their squares the integrated output is applied to the denominator terminal of the non-circuit 46 (C). Nominal of the integrator is T _D12 Since it has a time delay, the Y " _H signal, which is a time line, is used before the Y ' _H signal.

This result is (C'-Y) _L and Y 'leads to the time of concurrent adult integrator output, such as _H.

Output Z _I of the non-circuit 46 (C) provides a method for measuring the capacity of an independent high-frequency of the received signal.

Signal Z _I is exceeds the threshold level Z _I fixed, and a switch control signal β _I and Z _I of the reference value is applied to the younggap switch control signal β _I The mode switch control signal generator 34 for generating a time less than the threshold level Loses.

The signal beta _I is applied to an electronic mode switch 31 which provides a demodulator 43 with a compensation control signal of the form & lt _; RTI ID = 0.0 & gt _; {beta _I N _I + (1-beta _I )

When β _I is invariant, the control signal corresponding to a clear independent high-frequency detection is equal to N _I, and when β _I is zero, it is equal to Ns.

Therefore, by means of the selection switch 31, the system normally has the step calibration mode of operation in a stop mode of operation so as to keep the step calibration mode in the normal state.

However, during periods when Z _I exceeds the selected threshold, β _I is close to 1 and the switch transitions the operation of the independent high-frequency calibration mode.

This operation was described in mode 6 of table 2. Also, by using an additional output on element 34, operation can be transformed to mode 4.

Switch control signal (βs), (1-βs ), (β I), (1-β I) [[βs]] or [βs] and [β _I] determine the different modal described herein which to provide circuit May provide operation of modes 4, 5, 6, 9, 10, and 11 (single band) of Table 2.

In the receiving system of FIG. 3, the mode decision circuit 32 performs the broadband mode decision control.

The low-frequency chroma received component supplied to 47 (B) or the signal indicating the non-conversion of the low-frequency chroma supplied to element 47 (A) by applying the thresholds 47 (A) and 47 Exceeds the selected threshold for the reasonably inferred high frequency chrominities utilized.

(D) Examples include transfer characteristics of the circuit elements (FIG. 4)

FIG. 4 illustrates the transfer characteristics of the important circuit components used in the embodiment of FIG. 3.

For example, in FIG. 4a, a subtraction circuit for measuring the input at time t and a time difference (t +? T) for a delay? T compared with a small comparison of input signal bandwidths, The transfer characteristic of the differential circuit 44 will be described.

This follows the usual mathematical decisions in induction. The description of the transfer characteristics of type (A) and type (B) criticality 45 and 47, respectively, is provided in Figures 4b and 4c.

The (A) -type element of FIG. 4b represents a generated dipole or polarity-sensing transducer represented by an index A and including a threshold that is determined at a location by the parameter A _O.

This translates to a polarity at the limiting amplitude (A _O ) below the threshold, and is close to the equilibrium of the input to the linear strain and the output at one of the two poles.

The (B) type element of FIG. 4c represents a bipolar or polarity sensing transducer having a criticality defined by (B ₀ ) and a form defined by an index (B).

The output below the threshold is small or zero, and for the two polarities above the threshold, the output approaches the linear equivalent of the input to the output.

The transfer characteristic of the non-circuit 46 is shown in Figure 4d.

The transfer characteristic of the prescribed switch control generator 34 as described in FIG. 4e provides a monopole monotonic converter having a slope and a threshold, with the output value being zero or 1, as shown in the equation .

4f shows an exemplary form of a basic linear filter applied for circuit calibration or functional definition of a band splitter 42 as used herein.

The Y 'signal represents a ground line 50 connected to a tap.

Synthesis of terminal outputs from 51 are from Y 'to the differential amplifier 52 to Y' provides the _L and in the comparability delay Y 'is driven from a suitable tap from the delay line 50 Y provides' _L is compensatory Y' _H signal Is subtracted.

The taps of the surrogate are centered on each other and their sum may easily provide other required signals driven from Y '.

Figure 4g shows an apparatus for effectively improving the device 55.

The same tap delay line provides an input to the squaring circuit 53, the output of which is summed at 54 to approximate the particular integral shown. The desired gain constant is provided at 54.

Square law signal processing is described in circuit 55 and is not limited in operation.

(E) Appropriate waveforms of the mode crystal (FIG. 5)

In order to explain the operation and structure of the mode decision signal processing circuit, it is highly desirable that they consider a time waveform suitable for the morphology determination as shown in various signal processing circuits.

To illustrate, it is highly desirable to test arbitrary waveforms just as waveforms appear during a fork in a step transition or during the presence of a dominant independent high frequency.

In general, the waveforms for a step are a representative waveform (for a constant signal bandwidth) or a representative waveform that sufficiently represents the polarity, shape, and amplitude.

These are described in Section 5a. The waveforms relating to the measurement of the independent high frequency of the broadband Y 'signal can take many forms and this is essentially a criticality that always excludes or exceeds the zero crossing point when dominant high- Which is very useful for making measurements in excess of.

Thus, as shown in Figure 5b, examples of independent high frequency waveforms are easily reproduced by the characteristics shown for exemplary waveforms at particular circuit points.

5a shows two frequency passbands, denoted 1 and 2, which show the four column groups of the time waveform appearing during the phase transition.

The column shows basic waveforms numbered 3-7.

Column II shows waveforms 8-10 appearing from different frequency filters.

Ⅲ-line shows a 11-degree determination processing waveforms useful modal and Ⅳ-line and shows a waveform of 15 to 18 suitable for calibration of the various parts of the Y _'H band.

The various portions will be described in more detail with reference to FIG. 12, and FIG. 12 shows suitable passband forms for the separation of the passbands into at least two frequency regions. These illustrate configurations with minimal time vortices and these characteristics are illustrated in waveforms in Figure 5a.

If the reference to the frequency pass band shown in the band-pass 5a1 degrees is Y _'L pass band close to zero in the frequency f _1, and as described, and can be seen that the approximation with a cosine square waveform.

The Y 'passband as described is similar in shape, but the actual high bandwidth is configured as a low time distribution (lowtimerdisperdispersion). And Y _'H is Y' and Y 'are shown as difference of the band between the _L, also bandwidth signal of the illustrated ones mediated Y' are ₁ = Y 'is Y ₁ + and surrogate substantially similar to the squared frequency response of the form' _L .

The passage 5a degrees band 2, Y 'the frequency of the _L passing band, and (a) Y' _L-band and does not overlap the (b) Y additional tonggu limited Y _"H signal" than _L frequency not to have a sharp slope The third pass band of Y ' _{H shown} by the dotted line corresponds to the limitation of (a) only.

Referring to the types of column I, waveform 3 represents the component Y ' _L during the forward transition and waveform 4 represents d / dt (Y' _L ) during the same step.

Waveform (5) "denotes a _H, also narrow-band Gene component Y 'bipolar transition components alternatively Y shows the _HI.

Waveform (6) represents a d / dt (Y _'H) and d / dt (Y' _HI) 7 shows the waveform ^{d 2 / dt 2 (Y '} L).

The waveforms in this column and in the remaining columns of Figure 5a show during the transition epoch Ts of the step transition. Because the Y ' _H and Y' _HI tend to go towards zero at the beginning and end of the transition epoch, the time gate driving the basic inferred step high frequency fixed signal to be passed for the Ts period does not need to have a critical time requirement It is important to know that this is applied to timing or keying to define a specific portion of the Y ' _H center of gravity gradient.

Referring to the waveforms in column II, the alternating Y ' _H waveform 8 has an amplitude greater than Y' _H in the transition region near the step.

Waveform 9 corresponding to Y &_quot; _H is limited to epoch Ts by tilt limitation.

Waveform 10 for Y "&_quot; _H occurs for longer than Ts and continues to warp.

The waveforms with characteristics of 8 appear in the monochromatic signal channels of the receiver using the edge enhancement of the monochromatic signal channel.

In addition to these facts, there is no difficulty in using basic logarithms for signal processing, and they are here comparable enough to the mode decision circuitry.

Also, additional components in 8 tend to be suppressed when a reduced band signal such as Y ' _H1 is used to make the mode decision.

The major feature of the step-mode decision circuits is that they respond beyond the epoch of the same stage as Ts, but also over a very long period of time.

Those occupied for a minimum amount of time are ideal.

The difference in absolute time delay also exists within the process as shown.

The mode decision circuits illustrated here in response to broadband Y ' signals, or variations or combinations thereof, or modulated logarithms derived in accordance with independent mode logarithms or step modes, The decision circuits for driving decision information have certain special characteristics.

As shown by the circuit types shown here, these mode decision circuits perform signal processing operations that include signal comparisons and inefficiencies that are effectively induced by signal or display equations.

These circuits may use one or more of the non-signal waveforms referenced in the mode decision waveforms and during the step transition they have the forms as shown in column III.

Waveforms ₁₁ are designed for ρ ₁₁ and are illustrated for two different bandwidths of Y ' _H useful for determining. Waveform 12 is denoted ρ ₂ and is similarly shown for two bandwidths.

The waveforms 13 are obtained by using the signal components of the waveform 9 shown in FIG.

Y 'illustrated waveforms for two bandwidth of 14 _H are represented by ρ _1.

The subscript A in the denominator indicates a proposed, but not absolutely necessary, treatment method that allows the ratio to be zero, but the denominator value is not zero.

It can be easily understood that the waveforms 11, 12, 13, and 14 are not due to the direction or polarity of the step. The step transition of the bandwidth can be sufficiently specified in the ratio waveform of the following characteristics.

(1) Form (2) Polarity

(3) Amplitude (4) Time

The presence of the step can therefore be determined from the separable part of the broadband Y 'signal and can be ascertained by the simultaneous presence of chromatic components or color-supply ratios.

Ⅳ-line shows a waveform suitable for crystal phase exists in the pass band corresponding to the signal _{Y 'H2 = Y' H -Y} 'H1. These are suitable for mode determination and control for a number of signal components as described with respect to FIG. 12.

Waveform 15 shows the basic waveform for the step as described in 5.

Then, 16, 17 and 18 reproduce the non-waveform when the distorted waveforms ρ ₁₁ · ρ ₂ and ρ ₁ are driven from Y ' _H instead of Y' _H or Y ' _H1 .

These waveforms are configured at the same time as the frequency. These entities are evaluated as circuit devices compared to those used for column III waveforms.

The determination of the presence or absence of independent high frequencies may be based on the measurement of the ratio proportional to Y ' _H actually contained in the comparison amplitudes for the indicated independent high frequencies, Y ') _L. &_{Lt; / RTI} >

Furthermore, it is preferable that d / dt (C'-Y ') _L is relatively small, since independent high frequencies are high frequency in a color or a certain polar color as compared to the transition between two colors.

이 사용되거나 또는 ρ₁₁또는 극부의 동시존재 ρ₁₁과 같은 존재상태의 적당한 방법으로서 사용되어질수 있으며, 그리고 0이 아닌 색도(C'-Y')_L은 제어와 임계확인에 자동적으로 사용되어진다.However, since only the Y 'signal is useful at the receiver without bandwidth limitation,

Can be used as a suitable method of existence such as ρ ₁₁ or the coexistence of ρ ₁₁ of poles, and non-zero chromaticity (C'-Y ') _L is automatically used for control and thresholding .

Therefore, the same types of decision circuits used in determining the suitability of the step mode algebraic operation can be used to control the independent mode algebraic operation, but are not subject to certain other thresholds.

This allows the use of a method based on the representation signal of Y ' _H , like a signal method driven from d / dt Y' _L. This does not interfere with the use of inferred-step mode control and other signal measurements at the same time, or at the same time, in the deduced independent high-frequency generator.

For example, it is not ideal if the independent mode Y ' _H is large compared to Y' _L.

Accordingly, the form of the inferred independent high frequency is controlled by the one or more low frequency chroma for Y _'H ratio.

Also synthetic methods based on a number of components driven from Y 'may be used.

To illustrate, if measurements are shown to be distinct and present from the high frequency content measurement step of the locally separable part, this measurement can be used to enable independent high frequencies.

In the case of independent high frequencies, a typical measurement can be expected to exceed a threshold. A direct linearized waveform ratio is used here.

This method may also be exceeded, except for the amplitude of the signal point Y _'H.

Exceptions are the first (C _"H -Y _'H), and a zero by the basic number is somewhat important because it indicates the amplitude threshold devices are very small spikes and little impact during these intervals to the second.

Where these spikes are easily removed by conventional circuit devices.

And since these spikes are so short, the switching elements do not respond at all.

5b shows the output waveforms for the multiple mode decision circuits of the successive figures.

In FIG. 5b, the arbitrarily located time scale is provided to indicate the epochable duration of the time period Ts.

The curves (a) and (b) show possible spikes of the dual level translators 71 (A) and 71 (B) of FIG.

They are sandwiched between times. Because the appropriate components driven from Y ' _H are in the rectangle as described. Short sparks are easily removed by a conventional short pulse rejector or phase width discriminator.

The curves (c), (d) and (e) are processed by the dual level devices 91 (A) and 91 (B) of 96 and 8 of FIG. 9 in the presence of independent high frequencies or in the absence of high frequencies As shown in Fig.

Curves f and g are the same as in FIG. 10 and show the outputs from devices 101 and 102.

Therefore, the components of Y 'have properties suitable for step mode control and for inductive operation mode control of inferred lightness high frequency.

The circuit arrangements and methods shown here illustrate the use of perceptible characteristics that, when signal conditions are determined, ensure that a particular signal mode is present and provide a useful circuit method.

(F) Broadband mode decision circuit 6th to 10th

Useful broadband mode decision circuits, such as the device 32 of FIG. 2c and in other embodiments described herein, are shown in FIGS. 6, 7, 8, 9 and 10. FIG.

These circuits are a method for driving in connection with various embodiments described herein and based on the form of control signals as been described in Table 2 βs, 1-βs, β I, 1-β I or [[βs]] And the device, respectively.

6, which performs the function of the elements 32 of the circuits 60 and 61, the additional element 67 and the optional element 68 provide an additional form of the element 32. In FIG.

In FIG. 7, circuits 60, 70 and 80 drive device 32. Element 32 is also comprised of element 103 and element 60 shown in FIG. 8, element shown in FIG. 9, elements 104 and 105, element 10 and element 60 in FIG.

The waveforms to illustrate the associated performance of broadband mode decision systems of Figs. 6, 7, 8, 9 and 10 were reduced to small groups.

With respect to Figures 6 and 7, the same waveforms were used to compare the circuit.

The embodiment of FIG. 6 can reliably detect the time and existence of the phases characterized by the rho ₁ component of the basic modal logarithm.

The shown squared differential circuit is one form of measurement that becomes zero only when a selected waveform is present at the calibration time.

Therefore, it rejects noise or independent high frequencies in the crystal [[βs]] and recognizes the phase.

When ρ ₁ has its characteristic shape, amplitude and polarity, a signal ratio having a characteristic shape, amplitude and polarity at the time specified by the step generation is relatively used.

D, or an integrated monopole differential detector 61, is _one of the waveforms ρ ₁ , (ρ ₁ ), ρ ₁₁ and ρ ₂ or other related unique ones that are relatively used. However, there are differences in other circuits that are somewhat less restrictive or nearly similar in signal measurement.

Described in FIG. 7, the circuit 70 can be used in place of the elements 61 of FIG. 6 or in place of the elements 61, 67 and 68 of FIG. 6 to be described in detail in the following. In this operating mode, the circuit 70 can provide either a stepped mode stop signal of the general form 1 -? S or an independent mode driven signal of the general form? _I , or each of these, or one of them. Where ρ ₁₁ and (ρ ₁ ) are of the type described above and ρ ₁ is shown as a secondary or simultaneous waveform.

Vertical circuit 80 may utilize the same waveform to provide [[beta s]]. Therefore, the selection of p ₁₁ in FIG. 6 is an operation for simplification as compared to FIG. 7.

로 나타나는 제2성분과 Y'_H로부터 구동되는 제1성분에 응답하는 그러나 표 2에서 설명된 이들에 비교되는 작동의 모우드를 위하여 추론된 색도모우드 제어들의 응용을 나타내는 방법을 제공하도록 이들로 부터 구동되어지는 다른 처리기를 도시한다.Figs. 8, 9 and 10

To provide a method representing the application of inferred chroma mode controls for a mode of operation in response to a second component represented by _< RTI ID = 0.0 _>Y'H< / RTI ≪ / RTI >

And FIG. 6 shows a mode decision circuit for generation of step time keying signals of the type [[beta] s], which is one of the above-mentioned forms. This includes signal comparison and signaling circuitry 60 responsive to the components of the received monochromatic video signal to process one or more reference comparison signals that appear as generic waveforms having characteristic waveforms of characteristics in the step.

An evaluation circuit 61 responsive to this comparison signal is utilized to determine the presence or absence of these particular characteristics and ideally determines the keying or disabling of the deduced independent high frequency calibration circuit or the independent step calibration circuit in accordance with the present invention.

In the particular decision circuit of FIG. 6, the received monochromatic signal Y 'in the receiver (not shown) processing portion is separated into a low frequency component representation of Y' _L and a high frequency component representation of Y ' _{H in} the band divider 42.

의 재현성 저주파성분은 시간에서 와류되지 않으며 D.C성분을 포함하지 않은 Y'_L로부터 구동되는 기동(起動)신호를 포함한다. 이들 요구에 의한 신호들은 전술된

과

또는

로서 표시도는 구동성 교류신호들을 포함하며 여기서 M이 이상적인 낮은 정수이다.44 (A)

Reproducible low frequency component includes a start signal that is not vortexed in time and is driven from Y ' _L that does not include a DC component. The signals according to these requests are

and

or

As shown in the figure, includes a dynamic AC signal, where M is an ideal low integer.

Y' 또는

를 포함한다.The Y ' _H signal without the DC component is similarly limited. Signals due to these demands

Y 'or

.

Here Y ' _H is replaced by another limited signal to avoid overlapping the frequency with the Y' _H or Y ' _L bands, although a pair of pairs is used, but low order derivative functions are used.

In the case where Y ' _H is not limited, it is ideal that the phase characteristics become mutually linear as in the case of the illustrated case.

The step determination bandwidth for Y ' _L is not limited to exactly compare to chromaticity (C'-Y') _L.

The Y 'high and low frequencies are then applied to the comparison and standard circuit 60 to process one or more standardized comparison signals.

는 원색결정 신호처럼 처리되어지며,

은 제한적으로 구적부가 신호처럼 사용된다.In the illustrated embodiment

Is processed like a primary color decision signal,

Lt; / RTI > is used as a quadrature addition signal.

In the other circuit in which rho ₁ is used as the signal or the primary color decision signal, the non-overlapping passband shown also in Figs. 5a and 5b provides the ideal (rho ₁ ).

은 A형 임계 45를 통하여 비회로 46(A)와 46(B)들의 분모입력에 인가된다.In the circuit 60, the signal from the differential circuit 44 (A)

Are applied to the denominator inputs of the non-circuits 46 (A) and 46 (B) through the A-type threshold 45.

The Y 'high frequencies in one shunt are applied to the molecular terminals of the noncircuit 46 (A) through the differential sync 44 (B) in the primary color circuit.

Likewise, the Y 'high frequencies are applied to the molecular terminals of the noncircuit 46 (A) through the differential sync 44 (B) in the primary color circuit.

The outputs of each of the non-circuits 46 (A) and 46 (B) are the normalized comparison signals ρ ₁₁ and ρ ₁ .

ρ ₁₁ is applied to the evaluation circuit 61.

The evaluation circuit shown is a display evaluation circuit which responds to an average comparison signal for confirming the occurrence time of waveform characteristics of a stage. The particular evaluation circuit shown includes a summing and squaring circuit 65 that allows the functioning of elements 53 and 54 in the 4g diagram, a source of a plurality of subtraction circuits 64 and a voltage source 63, and a tapered delay line Sum-squared differential detector. The delay line 62 has a standard total delay time corresponding to the epoch at the time of the transition with respect to the associated chromatic bandwidth.

Comparing the waveform in operation ₁₁ is ρ (f ₁ / f _w) is applied to the delay line 62 gradually tapered to be separated by a time differential is small compared to the less than Ts Ts, and provides a number of distinct output.

These multiple outputs are subtracted from the respective reference level amplitude characteristics of the steps supplied by the reference voltage supply 63 of the multiple differential circuit 64, and their respective squared sums are calculated in circuit 65.

If the total squared output of the circuit 65 of the step is located in the middle of the delay line close to zero, this state is the keyed time gate to provide a step time pulse designed as an accurate time delay of Ts, [[beta _SH ] 66 are driven. The double parentheses represent time-driven signals compared to the long term format of the permissible range.

Where the first subscripts are in parentheses to indicate the reference waveform and the additions in Table 2 are located outside of the insert or parenthesis to indicate the relationship of the particular frequency bands.

The precise centering of the drive gate does not limit the operation of the inferred high frequency step correction circuit.

Because the inferred monochromatization as described above also leads to zero at the beginning and end of the high frequency period Ts. However, the triggering of the waveform generator requires more time accuracy.

Optionally, a similar display evaluation _circuit 67 is utilized to provide simultaneous verification by evaluation of other waveforms such as ρ ₁ . This circuit provides a second stage time pulse designed to [[beta _S1 ]] and the simultaneous requirement of the thickness pulse is generated by an optional AND gate 68 generating a concurrent step time pulse designed [[[beta] _SH [beta] _S1 ] .

A sixth-degree time waveform of the scale 1 and 2 are shown the "" Delayed Y in _(H and _H positions of the key time gate _H -Y comparison time and, C) "in the output _H 'Y.

The time delay can be reduced by using a potential reduction of the crystal species by using a short delay line of the period Ts / M. Where M is greater than the others. In the limit, there is only one effective time sample (Sample), and the delay is made as small as disappearing.

This approach is applied to FIG. 8, which is a display evaluation circuit with substantially no time delay.

In addition, these circuits are free from the time vortices for which all the crystal processing is limited at the time of the Porch Ts.

The decision circuit of FIG. 6 also has another feature that the sum of the differential gains in the severe temperature noise of the lowest signal levels is sufficient brightness to automatically turn off the inferred broadband chroma signals.

FIG. 7 shows other mode decision circuits including substantially delay-free evaluation circuits 70 and 80 and the signal comparison and invalidation circuit 60 described in connection with FIG.

The equalized comparison signals ρ ₁₁ and ρ ₁ from circuit 60 are applied to evaluation circuits 70 and 80, respectively.

An evaluation signal 70, characterized by an amplitude window form, receives at least one comparison signal, such as rho ll, and evaluates the signal for either a phase or a polarity amplitude having a phase by being applied to the amplitude window 71 (A) .

This window provides a zero output for the input amplitudes in the region where the reference output signal for the input amplitudes and the step absence are not combined in the excess region of the amplitude region that is step-wise combined. The pulse width discrimination opportunity 72 (A) is provided to compensate for the very narrow spikes of the amplitude window output and the resulting output signal for the ρ ₁₂ channel is designed as (1-β _S11 ) or (β _I11 ) Is a stop signal that is an indication of the opposite condition.

Thus, a second comparison signal, such as rho ₁ , is provided simultaneously by the discriminator 72 (B) and the amplitude window 71 (B) to provide a second stage stop signal designed to (1-? _S1 ). The two stop signals may be applied to an AND gate 73 which is selected to provide a combined simultaneous step stop signal 1- (1- _S11 ) (1-? _S1 ) or (? _I11 ), (? _I1 ). Or one of the outputs of the circuit 70, that is, [1- (1-B _S11 )], by using a fully compensating circuit.

An evaluation _circuit 80, characterized by a continuous vertical form, is used to generate a pair of comparisons, such as ρ ₁₁ and ρ _1, to produce an operating signal such as [[β _S ]] when the amplitudes of ρ ₁₁ and ρ ₁ are within the illustrated amplitude window Signal. As shown for reference in time diagrams 1,2,3 and amplitude windows 81 (A) and 81 (B), ρ ₁₁ and ρ ₁ are only time Y ' _H and corresponding C " _H -Y' _The temporal error is not critical because _H and _H are simultaneously in the amplitude region for the phase transition shown near zero.

To illustrate, ρ ₁₁ and ρ ₁ are individually applied to the amplitude windows 81 (A) and 81 (B) selected to provide the primary output only when the signals have step characteristics with respect to the illustrated synchronizing amplitudes. The amplitude window outputs are applied to an AND gate 82 which provides an output to the simultaneous input signals.

Comparing one or both of the input signals to the reference waveforms controlled by the AND gate 82 is used to confirm the presence of the step. To illustrate, the AND gate output is used to drive reference waveform generators 83 (A) and 83 (B) to generate the same waveforms as the ρ ₁₁ and ρ ₁ stage characteristic waveforms shown in time charts 1 and 2. The received respective signals are compared to these reference waveforms in comparators 84 (A) and 84 (B), respectively, and each comparator output is applied to an identification circuit 85 for ascertaining the synchronous detection of the phase characteristic waveforms Loses. In this simple form, circuit 85 may include a simple AND gate.

The output of the AND gate 82 also responds to the output of the verify circuit 85 and also the start gate 86 which raises the phase width to Ts applied to the verify gate 87. The output of the confirmation gate 87 is a step time and a drive pulse [[beta] s] having a long period such as the waveform to be checked.

Alternatively, the continuous vertical circuit may be a limited number of the same type of detector circuit 61 used in FIG. 6 to drive the mode decision circuits based on various numbers of signal sample points and with various time delays from 0 to Ts It is connected to multiple sample circuits. To illustrate, a pair of amplitude windows 81 (A) and 81 (B) and an AND gate 82 are formed by the sum of the squared differential thresholds of the form shown in FIG. 6, the delay line, the multitap and the short Lt; / RTI >

The step signal is easily known because it has a known shape, amplitude, and loop polarity. Independent high frequencies can have many shapes and can be made simplest by a mode decision circuit by other excellent characteristics. However, special forms of independent high frequencies can be identified by using null type wave form recitation units and methods, such as the description of element 60 of FIG. 6 and the description of elements 7 and 7.

Thus, the waveform that the Short (伸張) of the circuit is independent high frequency can be achieved by using a calibration independent high frequency that the circuit for the waveform be within the amplitude area characteristics of the system, such as or a reproduction method and ρ _1, but Y ' The components of _H are of opposite polarity in these phase quadrants. The control signal [[beta] _I ] is preferentially generated and applied.

FIG. 8 shows the thresholds 91 (A) and 91 (B) for determining whether the integrated value exceeds the threshold level indication of the independent high frequency or less than the selectable threshold level indication of the stepable entity, and the integrated monopolar And a nonlinear integrator 94 for obtaining a measurement.

의 형태를 갖는 출력신호를 제공하기 위한 자승 및 합산회로 93에 인가되어진다. 여기서 X_K는 K^tb탭에서의 신호이고 W_K는 진폭평량계수(amplitude weighting coefficient)이다.To illustrate, the integrator includes an increasingly shrinking delay line 92 for receiving a comparison signal such as rho ₁₁ or (rho ₁ ) and for providing multiple K outputs in a short time interval. Multiple outputs are output

Summing and summation circuit 93 to provide an output signal having the form < RTI ID = 0.0 > of: < / RTI > Where X _K is the signal at the K ^tb tap and W _K is the amplitude weighting coefficient.

If the time interval of successive delay line outputs is less than 1.5 times the maximum signal frequency, then this sum is close to the integral value. Nonlinear translators that are different from this trait are also used. The output of the squaring and summation circuit 93 is applied to a threshold 91A which is an amplitude window. Amplitudes smaller than the preselected value are compared with the step and provide the drive signal? S or (1-? _I ). The output of the circuit 93 is applied to the compensating amplitude window 91B. The output signal from this window is inconsistent with the step but can be used as a step stop signal (1 -? S) or as an independent high frequency drive signal (? _I ) indicating the condition compared to an independent high frequency.

FIG. 9 shows another evaluation circuit using a pair of non-linear integrals 94A and 94B for non-linear integration of the Y 'component prior to signal comparison and neutralization of the non-circuit 95.

In this example, the conversion ratios of the Y " high frequencies and the Y 'low frequency are integrated and compared. The output of the non-circuit 95 determines whether this exceeds the empirically preselected value, The amplitude window provides the stop signal 1 -? S when the integral ratio exceeds the preselected level in this example. The delay window of the nonlinear integrators 94A and 94B, For example, the integrator of Y " _H may have a longer delayed epoch than a fork at the delay of d / dt (Y ' _L ) as shown in FIG. 3 .

FIG. 10 shows another evaluation circuit including a filter 100 closely coupled with the limited area translator 102 and a compensating critical amplitude window 101. Operation of such a circuit is in the example shown in the impulse response and the time chart which each shows the filter output for the ρ step ₁₈ (1), (2) and (3) of ρ _step characteristic waveform and pilta 100 ρ ₁ input signal and waveforms.

As described above, the filter 100 then provides a strong positive output pulse to the central region in the Ts period that is sideways from the pulling and supporting negative sides. The evaluation circuits shown in Figs. 8, 9 and 10 have a delay or eddy current, but the circuits of Figs. 6 and 7 are not the same. With appropriate delay adjustments for concurrency, various forms can be synthesized to provide synthesis mode decisions to conventional switching circuits. The waveforms at the key advantage of these circuits, suitable for unipolar high frequencies, are illustrated in FIG. 5b.

(G) A mode control system (11 degrees) using several simultaneous mode decision signals.

11 (A), 11 (B), and 11 (C) illustrate the operation of a mode decision system that utilizes components driven from chroma low frequency and multiple signal components driven from a wideband monochromatic signal to provide increased decision capability. Fig. Element 36 shows the multiple-dimensional color space threshold suitable for use in the mode decision and control system shown in FIG. 2 c to generate the simultaneous mode decision signal? Sc and? _IC .

Device 36 has a pair of non-linear matrix 36 (A) and 36 (B) input of the matrix is a chroma signal (C _'1 -Y') _L and (C _'2 -C'), as illustrated and solid _L The low-frequency signals Y ' _L are supplied as shown in FIG. As shown, the respective input signals applied to 36 (B) are supplied through differential circuits 44 (B) and 44 (A). Matrix 36 (B) processes these non-transformed input signals into nonlinear synthesis represented by a multi-dimensional contour within chromatic non-space.

For example, a matrix circuit can process a complex nonlinear combination that appears as a sum of weighed squares that appears as a low-order outline or a more complex outline. Similarly, the matrix 36 (A) treats the input signals as a nonlinear combination that appears as a multi-dimensional contour of chromaticity space. The output of the matrix 36 (A) is supplied to a B-type threshold device 34 which generates an independent mode decision signal? _IC . Similarly, the output of the matrix 36 (B) is processed by the? -Type threshold device 34 to generate a step mode decision signal? Sc.

As shown in Figs. 11a and 11b, the control signal? Sc is supplied to the simultaneous decision gate 35 (A) simultaneously with the control signal? S so as to provide the simultaneous step mode decision control signal? S] [? Sc]. Similarly, the control signal? _IC is equivalent to the simultaneous decision gate 35 (A) 35 (B) simultaneously with the control signal? _I to provide the ideal simultaneous independent mode decision control signal [? _I ] [? S]. As shown in FIG. 11b, the control signal? Sc is a compensated output (? _I ) supplied to the third input to the gate 35 (B) providing the simultaneous independent mode decision signals? _I ,? _IC , May be selectively processed by the switch 31 to provide a signal (1 - [beta] _SC ).

(H) the same and non-identical band processing system comprising a plurality of frequency or simultaneous frequency time components. FIG. 12 shows a pass band and FIG. 13 shows an equivalent or non- For a non-synchronous band processing system, a basic circuit configuration suitable for using these passbands is shown.

FIG. 12 shows chromaticity signals and various frequencies and bands of Y 'suitable for operation of multiple mode fixed circuits according to the present invention.

The passband application of I illustrates simple passband processing and pass bands suitable for the case of the same band receiver processing.

Where the receiver processes the chrominance signals I 'and Q' or (C ₁ '-Y') _L and (C'-Y ') _L and the monochromatic low frequency signals are all the same as the monochromatic video signal Y' It is the reduced bandwidth.

The monochromatic high frequency signal Y ' _H is configured as a difference between the Y' band and the Y ' _L band.

Passband Diagram II shows suitable passbands in the case of multi-band calibration processing and same-band receiver processing. Where the chroma signals and the monochromatic low frequency signals are substantially reduced bandwidths as compared to the monochromatic signal Y ', but the monochromatic high frequency signals are divided into a plurality of portions, shown here as Y' _H1 and Y ' _H2 .

A mode decision circuit is provided for analyzing each of these separated monochromatic high frequency signals and the inferred high frequency fixed signals can be counted from these respective parts in the most suitable calibration mode for the respective parts.

Therefore, at appropriate times, the inferred high frequency calibration components in one mode can be counted from Y ' _H1 , the inferred components of the other modes can be counted from Y' _H2 , and the combined calibration components can be simultaneously Are added to the signal being presented to satisfy the requirements.

Passband Table Ⅲ shows a suitable pass band when the band receiver processing is not the same as the "single Y in _{L 'H} Y for each axis.

Where the chroma signals processed by the receiver are signals of unequal bandwidth, but each is definitely smaller than the bandwidth of Y '.

Passband diagram IV shows suitable passbands in the case of unequal band processing as a number of Y ' _H components for each axis.

Here, for each color slaughter, the Y 'high frequencies are separated into two parts Y' _H1 , 1 or Y ' _H2 , 1 and Y' _H2 , 2 or Y ' _H2 .

Separated mode crystals are configured for each axis for each high frequency monochrome signal portion and separate properly reasoned high frequency mode calibration signals are controlled for each axis from each high frequency portion.

FIG. 13 illustrates the use of a mode decision and control preparation embodying the present invention in chromaticity, so that the derived Y 'crystal band comprises a plurality of frequencies and axes related to the component.

Figure 13a shows the basic elements of the modality decision and control device, as exemplified in the passband diagram II, including multi-band calibration of the same band receiver processing. To be described, the band separation filter 42 is used for determining a calibration mode close to the high frequencies in the first band and for applying a decision signal to the first mode control circuit 34 (A) and for applying a decision signal to the first mode decision circuit 32 ), A first band of Y 'high frequencies represented by Y' _H1 , and Y 'low frequencies.

In other words, in a number of selective modal and in response to the decision circuit 32 (A) control circuit 34 (A) is a (not shown), a circuit for generating an inference high-frequency coefficients from the Y _'H1 to switch to the appropriate modal .

Similarly, the signals appearing at the Y 'low frequency and the second bands of the Y' high frequency represented by Y ' _H2 are applied to the second mode decision circuit 32 (B) and the output decision signals for the band 2 are applied to the mode control circuit 34 Become human.

The control circuit 34 (B) switches the second multimode high frequency generator inferred to the displayed mode to selectively control the inferred high frequencies for band 2, and the inferred high frequencies for bands 1 and 2 are used for signaling .

Figure 13b shows a suitable mode decision and control device in the case described in the passband diagram III which includes control of a single high frequency band for each axis in the case of unequal band receiver processing.

The depicted circuit includes a plurality of comparison and comparison circuits for comparison with those described in connection with < RTI ID = 0.0 > 13a < / RTI > with the exception of providing separation determination and control circuits for the respective axes, Lt; / RTI > Shaft 1, that is, Y, _L1 and Y 'signals appear as a solid low-frequency signal and a high frequency for the _H1 are' is applied to the modal determination circuit 32 (A) for the axis-axis 1 and axis 2, i.e., Q 'I for shaft The substitute signals are applied to the decision circuit 32 (B).

The high frequencies separated and deduced within the respective determined approximation modes are synthesized by them and controlled by the respective axes.

FIG. 13c shows the mode decision and control circuits for the case shown in passband diagram IV, including multi-band calibration of unequal bandwidth receiver processing. Here, the separation mode decision circuit 32 (A) via 32 (D) is provided for each of the monochromatic high frequency signal bands of the respective axes, that is, the mode decision circuit 32 (B) 2 Receive appropriate signals to determine the proper calibration mode for axis 1. The separately derived high frequency calibration signals are each controlled in a suitable mode for the bands of the axes, and inferred from the received signals from the respective bands for separate axes (not shown) in the combinational circuit, The high frequency calibration signals are substantially combined with the received low frequency chrominances for these axes.

FIG. 13d shows a selective crosscheck form between the high frequencies of the second band and the high frequency of the first band. To illustrate, if there are independent high frequencies, they will only exist within band 2. However, if there are no high frequencies in band 1, they will never have high frequencies in band 2 and the cross-chet decision circuit is useful for this purpose.

(I) a mode control receiver with switching of low frequency components at an inferred high frequency oscillator input (FIG. 14)

FIG. 14 is a configuration diagram of an embodiment of a receiving system according to the present invention in which control from element 32 and a mode decision system is applied to the inferred high frequency oscillator to control monochromatic low frequency signals and chromaticity at the input.

In this receiver, the inferred multiple-mode high-frequency calibration components are processed from two chromaticity axes, and the calibrated signal for the third axis is processed by matrixing. To illustrate, the time-differential monochromatic low-frequency signals of the monochromatic low-frequency signals are provided in a switchable manner through a switch 31 (B) to the denominator inputs of the respective non-circuits 46 (A) and 31 (A). Similarly, for the first color slaves, the low frequency chrominances and the time differential low frequency chrominances are provided switchably through the switch 46 (A) to the molecular input of the non-channel 46 (B), and a substitute for the second chrominance axes Signals are provided to the molecular inputs of the non-circuit 46 (B) via switch 31 (C).

Each of the switches 31 (A), 31 (B) and 31 (C) is controlled by a mode decision circuit 32 shown without delay in accordance with the theory described above with respect to devices 60 and 70 of FIG.

In the element 32 of FIG. 14, the signal Y ' _L is differentiated in the element 44 and applied to the denominator input of the non-element 46 through the A- Signal Y 'is _H is applied to the input of the molecular non-circuit 46 through the device 44. The output of 46 has a single polarity in the square element 30 and a double level in element 34.

The control signal beta _I11 is applied to switch from the stopping phase mode to the independent mode when? _I1 is basic.

Y is a "short spike as 0 St. Croix epoch are in the _H switch 31 (A), 31 (B ) and 31 (C) is applied to suppress the element Y being 43 by _'H is 0 at this time. The mode decision signal beta _I11 = 0 allows the switches 31 (A), 31 (B) and 31 (C) to connect the non-converted signals to the non-circuits 46 (A) and 46 (B) Enabling non-circuit outputs that are fixed control signals and control signals N ₁ , N ₂ for axes ₁ , ₂ .

On the other hand, it is determined that the independent high frequency mode should operate and the mode decision signal beta _I11 = 1 enables the moments of each signal to be applied to the non-circuits providing the respective axis control signals for independent high frequency calibration.

Such a control system is represented as follows.

(1) [d / dt ( C'-Y ') L] (1-β I11) + (C'-Y') L (β I11) and

(2) [d'dt (Y 'L)] (1-β I11) + Y' L (β I11)

The illustrated mode decision circuits are also utilized to provide βs1-βs.β _I (1-β _I ), [βs], [β _I ] or [(βs)] as shown before.

들은 저주파 성분고 주파 성분 그리고, 동시에 스위치 제어신호를 제공하도록 대역 분할기 42에서 지연 보상된다.The components Y ' _H and

Are delay compensated in a band divider 42 to provide a low frequency component high frequency component and, at the same time, a switch control signal.

The inferred high frequency chromaticity correction components for the determined modal axes 1 and 2 are counted from the Y 'signals of the modulators 43 (A) and 43 (B), respectively. And combined with the reduced bandwidth chroma signals in summing circuits 48 (A) and 48 (B) to sum to provide corrected chrominance signals for the two chroma tones.

The signals calibrated by the third color slaughter are processed by matrixing the two calibration signals of the matrix circuit 140 in accordance with a request to maintain the fidelity of the received monochromatic signal. This is generated first by satisfying the relation of ΣAc (C _H "-Y ' _H ) = 0.

This is obtained by a certain component proportional to the nominal value defined by the listed logarithms. The use of substantial equivalence suppresses the over-modulation and commutation that is present in the mixed-high frequency receiver and maintains the fidelity of Y '. The resulting signals are converted to conventional summation and presentation circuits (not shown).

The limited selectivity gain controller 141 is shown by being located in the _Y'H channel to adjust the amplitudes of the inferred high frequency components in response to the control voltage Ec. This fact is used, for example, to reduce this when noise disturbance is extremely severe.

Element 47, consisting of a type B threshold device with a selective threshold level, applies a simultaneous chromaticity or chromaticity ratio that limits the chromaticity component deduced on each axis. Independent, selective thresholds are obtained by the use of mullet-type 47 elements prior to switch 31, which may be selectively different, respectively, by the gain control elements.

Figure 14 shows the same band system. For non-identical systems, each of the Y ' _H and Y' _L signals is used for each axis, and the dual sides of the drive are used in accordance with Figure 13b. The divided high frequencies are coupled by element radiation and used according to the control of FIGS. 13A and 13C. Of the optional double gate device 31 (B) and can be written in place of the switch 31 (C), and they always Y 'or _H (C "-Y _{H' H)} or step (C" -Y _{H 'H)} independence One for each axis, and the respective axes are indicated for best application by the mode determination system.

The particular mode decision circuit 32 utilizes a pair of signal ratios appearing as Y 'high frequency and d / dt (Y') low frequencies, respectively, as a result of non-decentralized linear filtering. During the phase, the ratios all belong to the first amplitude region and belong to the second amplitude region during independent high frequencies, with the exception of the very small spike or pulse.

Remains within the amplitude range of monochrome combined with the operation of the step mode algebra as described earlier in this specification in the presence of a strong high frequency.

의 재현성 신호들을 수신한다.(J) Inferred multi-modal high-frequency calibration system with switching control of a wide-band monochromatic signal and a narrow-band monochromatic signal (FIG. 15) FIG. 15 is a block diagram of another embodiment of a receiving system with inferred multi- Fig. The inferred high frequency calibration components in this embodiment are processed by modulating the processed chroma processed signal from the monochromatic signal. To illustrate, each of the inferred high frequency calibration components in the two chromaticity axes is provided in the modulators 43A, 43B. The mode decision circuit used here is of the type already described in detail above. For purposes of illustration, the device 32 includes Y ' _H from the band divider 42 and the processing circuit 44A,

Of reproducibility signals.

The output of the B circuit 46A is supplied to the molecular input of the B circuit 46C through the B type threshold 45. [

Display is not limited to the illustrated form of evaluation circuits [[βs]] or 1 - [[βs]] or βs and (1-βs) or β _I (1-β _I) or [βs] and [β _I] has one or more may have the form of a 61, 70 or 103 to service. The characterization process of p ₁₁ and the continuous display evaluation are shown as any of the circuits described in connection with the used (6), (7), (8), (9), (10).

The compensating output pairs (outputs) from the element 32 of FIG. 15 are denoted by [beta s] and [beta _I ], and are represented by a single key, and instead of driving or stopping the independent modal logarithms, Drive or stop the logarithm. Unequal I 'and Q' band processing, illustrated in the same-band system, utilizes the use of separate Y ' _L1 , Y' _L Q and Y ' _{H 1} Y' _H Q components and use of the mode decision circuit of FIG. 13b Lt; / RTI >

The chromaticity and chromaticity ratios of the transformed signals pass through the gates 31A and 31B respectively and are applied to the inputs of one of the summing circuit 48A and the modulator 43A . The ratio of the Y 'low frequency and the Y' high frequency of the non-circuit 46A and the change of the converted Y 'low frequency and the Y' high frequency ratio of the non-circuit 46B are alternately changed through the gates 31C and 31 (D) (48B) and the other input terminal of the modulator 43A.

형태의 변조기인 (43A)로 부터의 추론된 독립 고주파(C"_H-Y')를 만든다.When the mode decision circuit 32 provides a mode decision signal indicating the operation of the independent mode logarithm, the gates 31B and 31F change the ratio of the Y 'low frequency to the Y' high frequency and the chromaticity ratio of the conversion channel to Stop. Gate (31A) and (31C) are Y 'low-band Y' sikimyeo respectively drive the high-frequency ratio and the color channel as a result this is _{(C "H -Y 'H)} = (C' 1 -Y ') L

(C &_quot; _H- Y ') from the modulator 43A of the form.

형태의 추론된 단계 고주파 고정 선분(C"_H-Y'_H)이다.If the modal determination circuit 32 provides the modal determination signal indicating the operation of the step modal number gates (31A) and (31C) are chromaticity sikimyeo stop the channel and Y _'H / Y' _L-channel and the gate (31B) And 31D drive the chrominance channel and the Y ' _H d / dt (Y' _L ) channel. The result is (C "-Y ' _H ) = d / dt (C ₁ -Y') _L

(C _" -Y ' _H ). ≪ / _RTI >

Corresponding fixed line segments for the second chromaticity axes are processed in a modulator 43B using gates 43E and 43F in a similar manner. The summing elements 48A and 48B convert and receive the components supplied to the elements 43A and 43B at least equal to those defined by the central core logarithms.

The inferred calibration signals for each chroma channels are individually assembled with the low frequency chroma for the channels in the summation circuits 48C and 48D and the two corrected chroma signals are combined into three calibrated color chromaticity chromaticity signals Gt; matrices < / RTI >

The use of conventional matrixing coefficients provides three components R "-Y ', G"-Y' and B "-Y 'as shown. Matrixing of the inferred chrominance components is accomplished by the addition of Y" _H Ensure sex. The use of inferred chroma components that feature an inferred step mode or inferred independent mode maintains Y ' _L fidelity by preventing over-modulation commutation, saturation, and pseudo-low frequency.

In the final process, each color behavior signal is combined with a monochromatic signal in each summing circuit 48 to provide primary color signals for presentation 151.

(K) Inferred multi-mode multi-band high-frequency receiver system (Fig. 16)

Figure 16 is a block diagram of a receiving system with multiple-mode multi-band calibration in accordance with the present invention. The solid high-frequency signals are filtered into at least two frequency components Y _'H1 and Y' _H2 by the band divider 42. The mode decision are respectively Y _'H1 and Y _H2 of the modal determination circuit (32A) and (32B) deulnae crystals, and the high frequency corrected component are deduced respectively from the coefficients of each band Y 'and Y _{H1' H2} for slaughter color modulator (43A) through (43D).

For simplicity, FIG. 16 shows a mode decision circuit of the form without addition delay. Also, one of the mode decision systems described with reference to Figures 6, 7, 8, 9, 10 and 13 can be used for proper delay compensation (not shown).

Referring to FIG. 16, the inferred calibration segments, which are fixed components respectively counted from axis 1, Y ' _H1 and Y' _H2 , are received on axis 1 of summing circuit 48A to provide a synthesized calibrated chrominance signal on axis 1 Is added simultaneously with the low-frequency chromaticity. Likewise, a combined calibrated chroma signal for axis 2 is provided at summing circuit 48B.

These two calibrated synthesized signals are provided to a matrix circuit (not shown) to provide chromaticity signals or calibrated color actuation signals, each of which is combined with Y 'of the summing elements 48 to provide the primary color signals of the presentation 140).

The mode decision and control system operates in the same manner as described above in connection with FIG. 15 and in accordance with FIG. 13A. The mode decision circuits 32A and 32B determine whether or not a chrominance signal or chrominance signal is applied to one input terminal for each modulator 43 (A) through (C) (31) which selectively controls or determines whether the ratio of the Y 'low frequency to the Y' high frequency component to the Y 'low frequency, or the ratio of the Y' high frequency component to the Y 'low frequency, It can be one.

The extension of the non-identical I ', Q' curves and bands can be achieved as described in and described herein in FIG. 13c.

While the single control lead is described as being out of each element 32A and 32B to provide a daisy operation in a step or standalone mode, the dual controls [beta s] and [beta _I ] May be usefully used to provide driving.

(L) Multi-mode deformation axis processor (Fig. 17)

FIG. 17 illustrates a receiving system with inferred multiple mode distortion axis high frequency calibration circuitry according to the present invention. Herein, the inferred step high frequencies for the first (1) and second (2) chromaticity axes in the quadrature are processed in respective modulators 43A and the inferred independent high frequencies for the (1) and (2) And processed by respective modulators 43B.

In the processing of the inferred step high frequencies, a plurality of chromaticity signals defining quadrature chromaticity axes from the vortex 150 are applied to each of the actuation circuits 44A to provide output chromaticity non-transformed signals. In other words, these output signals are supplied to the root-sum-squared circuit device 170 in order to process the amplitude signals of the same chroma ratio to the squared root of the input chrominance non- Is applied to the input terminal. The respective chroma non-converted signals are applied to the molecular input terminals of the respective non-circuit 46A and the output signals of the circuit 170 are applied to the denominator inputs of the respective non-circuit. The outputs of the respective non-circuits 46A are chromatic non-converted signals that are averaged with respect to the non-converted signals on the modified chrominance non-transform axis to provide instantaneous sine and cosine shapes.

The monochromatic signal Y 'is applied to the band splitter 42 and in the band splitter 42 the signal is divided into a compensating high frequency portion Y' _H and a low frequency portion Y ' _L which is compared to chroma signals.

이 최소진폭을 초과하는 경우 온 상태로 작동되어진다. 이러한 특징은 선택적이며 그리고 실시예 들중 어느것내에도 포함되어질 수 있다.The Y 'high frequencies are applied to the molecular input terminal of the non-circuit 46B, and the Y' low frequencies are differentiated in the differential circuit 44B. And the resulting Y 'low frequency non-converted signal is applied to the denominator input terminal of the non-circuit 46B through the A-type threshold 45. The resulting threshold ratio of the Y 'low frequency non-conversion to the Y' high frequency is applied to the modulator 43A along the output of the circuit 170 via the gate 31 to provide an output providing signal. The gate 31 passes through the B-type threshold 34

If it exceeds the minimum amplitude, it is turned on. This feature is optional and may be included in any of the embodiments.

In other words, the provided signals are used to modulate the average chroma ratio signals from the non-circuit 46A in the respective modulator 43A so that they can provide the respective inferred high frequency step calibration components.

In processing the inferred independent high frequencies, the circuits are used to allow the same processing steps in the modulators in 43B, with the exception of all differential synchronization and discrimination steps being omitted.

A mode decision based on various signals representing a comparison of the Y 'low frequency versus the Y' high frequency as described above may be valid for the device 32 through the gates 31A, 31B. Described, if the step calibration mode is deemed appropriate, the gate 31A is driven and the gate 31B is stopped. At this time, only the step calibration modulator 43A may have additional summation metrics and inferred high frequency outputs for presentation.

The embodiment of FIG. 17 effectively processes the video components carried on chromaticity subcarriers.

(M) Receiver receiving multi-mode high frequency calibration inferred on subcarriers (FIG. 18)

Y _"HS signal Y 'to subtracted from the signal becomes _H is applied to the adder circuit 191. Therefore, write compensation of the output signal Y to an independent high-frequency content of the received signal' _HI = Y _'H -Y" provides _HS .

In other words, the _Y'HI signal is applied to the inferred independent high frequency generator 20 in which the inferred independent high frequencies (C " _HI- Y ') are processed according to the following equation.

The inferred step high frequencies and the deduced independent high frequencies for each chromaticity axis are added to the reduced bandwidth chroma signals for the axes in the summing circuits (not shown) and the resulting corrected signals are processed for presentation Loses.

If the decision circuit 32 determines that the step is not present, the source 190 and the inferred step generator 10 are substantially stopped. At this instant, only the inferred independent high frequencies counted from Y ' _H are added to the chroma low frequencies and they are stopped independently.

Optionally Y _"HS is similar to the Y _'HI from the element 20, 191, from 190 to reconfirm the decision to the control) it can be returned to the modal determination circuit 32. The time been appropriately identification Delays (not shown) will be processed as required.

(O) Control circuit and method for reconfiguring a bandwidth signal (19b) 19c 19d

19b shows a Y " _HS re-generator such as the source 190 of the receiving system of FIG. 19a.

로 나타나는 공칭[[ρ₁]] 파형을 제공하여 그리고 이러한 신호를 멀디플러(111)의 한 입력단자에 인가한다. 이러한 파형은 도표(1)과 (2)의 Ts에 연관하여 설명되어질 것이다. 이러한 파형은 단계 발생기(도시되지 않음)의 출력을 분할하는 그리고 시간 변별된 저주파 대 고주파의 비를 취하는 적당한 대역에 의하여 선형적으로 발생되어 질 수 있다. 실제 Y'_L신호는 차동회로(44)에 의하여 시간 변별되어지며 그리고 평균지연(194)를 통하여 멀더플러(111)의 다른 입력에 인가되어진다. 멀더플러(111)의 출력은 신호의 단계순차에 보조적인 Y' 고주파들의 재발생 부분 또는 재구성 부분을 나타낸다.To be described, a step mode decision circuit (not shown) supplies a step mode decision signal [[beta] s] bound to the time epoch Ts of the detected step. In response to [[beta s]], the keyed waveform generator 193 generates a keyed

And applies this signal to one input terminal of the multidipulator 111. The signal [&_lt; [rho ₁ ] > Such a waveform will be described in connection with Ts in Tables (1) and (2). This waveform can be generated linearly by a suitable band dividing the output of the step generator (not shown) and taking the time-varying low-to-high frequency ratio. Actual Y _'L signal is the discrimination time by the differential circuit 44 and via a delay mean (194) is applied to the other input of Mulder plug 111. The output of Mullerpler 111 represents the regenerated or reconstructed portion of the Y 'high frequencies that are complementary to the step sequence of the signal.

This may be limited to the step-time epoch by the optional gate 195, which is only valid for the time epoch of the step and is responsive to [[beta] s]. Also, a relatively defined [[rho ₁ ]] is suitable as an output.

Figure 19c shows an inferred step that can be used like the inferred step generator 10 in another embodiment, such as the elements 190 or 20 of Figures 19a and 19b, The animation is shown.

Here, the time-limited step mode decision signal [[beta] s] is received by the keyed generator 193. [[rho ₁ ]] The output signal is applied to one input terminal of the Muldler. Chromaticity low frequencies are applied to the differential sync 44 and the resulting chroma low frequency non-signal is applied to the other input of the MullerDrawler 111 via the selective equalization delay 194. The Mullerplanner output is an inferred time-limited high-frequency step calibration component of the form [[C " _H -Y ' _H ]].

_{[[C "H -Y 'H} ]] = [[ρ 1]] Nominal d / dt (C'-Y' only suitable for a) _L This real time epochs of the phase, and optionally Further responsive to the [[βs]] by the gate 195 is limited to the step time epoch. As an example, if _{(C'-Y ') L =} I' L is inferred I 'is generated during step _H will be.

A chromaticity input provided to the modulator 43 in accordance with the outputs of the waveform generators 196A and 196 that are keyed to elements 42,32 and 31 of Figure 19d and an optional correction time gating device 195, Lt; RTI ID = 0.0 > and / or < / RTI >

가 단계와 비교할 수 없는 것이 아니라 Y'_H신호가 구적 혹은 역위상(逆位相) 같은 단계로부터 실제로 다른 비를 갖는 독립 고주파들의 성분들을 구동하도록 사용되어진다. 그러므로 제어된 형태의 진폭, 극성, 시간과 대역폭의 재구성되어 추론된 색도를 사용하는 것은 독립 고주파들의 선택된 성분들에 역시 인가되어질 수 있다.This circuit

Is not incompatible with the step, but the Y ' _H signal is used to drive the components of the independent high frequencies that actually have different ratios from the phase, such as quadrature or anti-phase (anti-phase). Therefore, using the reconstructed and deduced chromaticity of the controlled form of amplitude, polarity, time and bandwidth can also be applied to selected components of independent high frequencies.

(P) Crosstalk neutralization (20 °)

Also according to another embodiment of the present invention, the crosstalk from the I 'single sideband signal on the subcarrier to the Y' signal is substantially neutralized in the signal region close to the chroma stages. Alternatively, such a method may extend to specifically recognized components of independent high frequencies.

Referring to the frequency response curves (1), (2), and (3) of FIG. 20, the curve 1 shows the areas of the frequency spectrum having a clear signal side band chromaticity due to the double side band chromaticity and the I ' Respectively. Curve 2 illustrates the low frequency portion of the spectrum for which Q 'and I' are treated as unequal band components and the corresponding comparable Y ' _LQ and Y' _LI components are also identical. Curve 3 shows possible inferred isolated I 'high frequency components utilizing the same band signal processing.

Since the compensation on the steps has been described here, the components counted from Y ' _HI are denoted by I' _S1 and the components denoted by I ' _S2 are counted from Y' _H2 . The combination of crosstalk which is the visibility of I 'to Y' is due to I ' _S1 . In this same band embodiment, the receiver circuits 150 described above are generated in a mode controlled by Y ' ₁ Q' _L , I ' _LQ and by (32) and (34) as described above. The element 32 has the same function as the inferred crosstalk detector. The band splitter 42 feeds all components with a suitably time delay. All the I 'and Q' high frequency components from (10) and (20) are fed to summing circuits 48 for use as I " _H and Q" _H. These are combined with I ' _LQ and C' _L in the matrix and combined in region 140 to provide R "-Y ', G"-Y' and B "-Y as described above. The component I ' _S1 is supplied to the neutralization oscillator 204, which is translated to the gate 31 when keyed by the mode control 34.

역전되어지며 I's₁에 의하여 변조되어진다. (200)의 출력은 에스에스비(SSB) 필터(203)에서 여과된 단일측 대역이며 그리고(48)에서 중화 I'부반송파 누화에 지연된 Y'와 조합되어진다.The subcarrier reference for the receiver, if it is present, is shifted by the phase shifter 201 to a polarity 180

Inverted and modulated by I's ₁ . The output of filter 200 is a single sideband filtered at SSB filter 203 and combined with Y 'delayed to neutralized I' subcarrier crosstalk at (48).

The inferred independent high frequency components are processed in a manner as described herein, and the various inferred components are combined into the composite calibration components in the summing circuits 48. [ The components I ' _S1 are components that have been generated or synthesized directly. The preceding bandwidth according to the above described technique is selected for neutralization that is only partially necessary for large metering. In another form of this embodiment, the component Y ' _H2 supplied to the deduced high frequency generators 10 and 20 is driven from the neutralizing component of Y'. In all other schemes, the multiple-mode calibration circuit operates in the same manner as the embodiments described so far.

Visible crosstalk on colored edges or stripe edges in colors with I 'is usually increased in a substantial visible area by rectification due to non-linear display. Therefore, although the partial neutralization substantially reduces the visibility of the residual patterns, the methods provided herein are reduced to reduce the crosstalk of the single sideband I 'components into the Y' channel for color and monochrome signal television receivers, .

This improvement is also applied for independent perceptible components of independent high frequencies.

Although the present invention has been described in connection with only a portion of certain embodiments, it is to be understood that these are only a few of the many embodiments that may utilize the theory of the present invention.

For example, while the embodiments described herein have been described using standard NTSC standard terminology used in the United States, the theory of the present invention is equally applicable to NTSC driving systems such as the well-known PAL or SECAM systems Must understand. Although the operation theory of the present invention has been described with an emphasis on a conventional home television receiver, the same signal processing methods and circuits can be reduced in comparison with the bandwidths of the dichroic signals and the bandwidth of the monochromatic video signal, It should be appreciated that any stage of signal transmission and reception that may be used may be used to increase the quality of image reproduction. Thus, the expression " tuller television receiving system " as used herein means to include or define " color television signals " at any stage prior to image presentation. The present invention relates to a television recording and reproducing apparatus and a large screen theater screening apparatus Cable television systems, and wireless receivers such as home television receivers.

Signal processing circuits such as video detectors and chrominance subcarrier synchronous detectors and video signal matrixing circuits that are important in this receiving system are well known in the art and can be used in conjunction with a broadband monochromatic signal for processing in accordance with the principles of the present invention, Bandwidth chroma input signals.

It is therefore a well-known fact that numerous and varied apparatus may be made by those skilled in the art within the spirit and scope of the invention.

Claims (1)

A composite color television signal including a broadband monochrome image signal and a plurality of narrow band color signals constrained to a relatively low frequency is used and as a result an undesirable skew A reception circuit means for generating a broadband monochrome signal and a plurality of bandwidth reduction color signals in response to the composite television signal described above; Modulated pseudo high-frequency correction circuit means for effectively correcting each color signal in a plurality of signal operation control mode states giving a correction component, and at least one of the above-mentioned one or more signal operation control One or more modal control signals to be derived from the pseudo high frequency correction of the mod In response to the mode control signal and to enable or disable the operation of the multimode pseudo high frequency correction circuit in each of the plurality of signal operation control modes in response to the mode control signal. A multichannel pseudo signal correction means as an od control means.

Images