KR800000933B1 - Transiont suppression in television video systems - Google Patents

Abstract

An apparatus for processing television video signals compries a source of television video signals, first means of deriving a first signal including signal components in a frequency range between zero hertz and a frequency f, second means having a peak amplitude response at a frequency between zero hertz and f and a relatively lower amplitude resopnse af frequencies of zero hertz and approximately f, thresholding means for inhibiting amplitude excursions of a second signal above a predetemined threshold level, and output means coupled with the first means and the thresholding means of combining the amplitude inhibited second signal and the first signal.

Description

Transient Noise Suppressor in Television Video Systems

1 is a schematic diagram showing a general arrangement of a color television receiver using an embodiment of the present invention.

FIG. 2 is a graph showing waveforms for various times related to the embodiment shown in FIG.

3 is a graph showing various amplitude versus frequency transfer characteristics related to the embodiment shown in FIG.

4 is a schematic diagram showing a circuit arrangement for implementing the embodiment shown in FIG.

5 is a schematic diagram showing another embodiment of the present invention.

6 is a graph showing various amplitude versus frequency transfer characteristics related to the embodiment shown in FIG.

The present invention relates to an apparatus for suppressing or reducing noise in a television video signal processing system, and more particularly, to an apparatus for suppressing or reducing excessive noise in a television video signal processing system having a device for improving a screen quality. It is about.

Suppressing undesired transient noise in a television video signal processing system is a television video signal using a circuit that makes relatively high frequency light signal components, for example, relatively high frequency light signal components, or peaks, especially to improve the sharpness of a reproduced image. It has long been a problem in processing systems.

The occurrence of excessive noise in the brightness signal portion of the video signal is indicated by spots on the reproduced image.

The transient noise has an amplitude corresponding to various gray levels of the reproduced image. Transient noise with an amplitude near white is very undesirable because it results in white spots that viewers can easily identify. Excessive noise is undesirable for a signal corresponding to a white or black area of a reproduced image. For example, transient noise with an amplitude close to the white level in the signal corresponding to the black background results in visible spots. In white signals, white transients cause spot defocusing, excessive beam current, and signal rectification in the image tube used to reproduce the image.

Transient noise results from power fluctuations. For example, a bandpass filter circuit in the intermediate frequency (I.F.) portion of a television receiver rings in response to an impulse signal. That is, energy boost is generated in the center region of the bandpass. The carrier of the video signal components is I.F. Since it is off-set from the center frequency of the bandpass filter circuit, the ringing and carrier waves can be beaten to form transient noise in the upper frequency range of the brightness signal component in the 1.8 MHz peripheral region. This transient noise is very undesirable because it is in the frequency range of the brightness signal which can be selectively amplified to improve the sharpness of the reproduced image. Therefore, suppression of excessive noise is difficult to perform without adversely affecting the screen quality.

Transient noise includes I.F. synchronous detectors, rather than envelope detectors, because the envelope detector rectifies the composite signal envelope to provide transients in one direction, typically in the black direction. Extreme in receivers using amplifier systems. The synchronous detector is designed to produce white and black spots. Play ringing symmetrically. Although both white and black spots are disturbed, it has been found that black spots are less disturbed than white spots for the viewer.

Various circuits are known for noise suppression. Circuits for "clipping off" are known that cut out signal and noise portions above a predetermined threshold. One such circuit is described in US Pat. No. 2,834,884 entitled "Electromagnetic Amplitude Clipping Circuit" to R. Double U. Sonnenfel.

Also known are circuits for generating an auxiliary signal from a main signal having an appropriate amplitude and time delay for noise in the main signal when combined with the main signal. This circuit, in which the noise-canceling signal is generated from the components of the television video signal within the predetermined frequency range, is described in US Patent No. 2,854,508 entitled "Circuit Device for Disturbance Signal Separation Television Receiver" issued to P. J. H. and Jansen. Is described in US Pat. No. 2,885,474 entitled "Circuit Arrangements for Television Receivers for Filtering Noise Signals". Circuits are also known as "spatters" or "noise inverters" that essentially invert the signal portion above a threshold that is intended to suppress transient noise. One such circuit is described in US Pat. No. 3,862,361 entitled "Video Amplifier Circuit Used with Registered Detectors" to J. B. and George.

It is desirable that the portion of the television video signal processor directly connected at the time of noise suppression does not adversely affect the operation of the other portions of the signal processor. As mentioned above, the noise suppression circuit is preferably not disturbed by portions of the signal processing apparatus arranged so that the image is clear by emphasizing relatively high frequency components of the brightness signal. The noise suppression circuit preferably does not operate unnecessarily on signal components, i.e., relatively low frequency light signal component signals having relatively few noise transients. In this way, undesirable nonlinearities can be prevented from occurring in the gray range of an image associated with relatively low frequency brightness components.

Desirable amplitude or phase characteristics (or both) as a function of frequency do not produce a phase nonlinear or phase curve by a device in which delay signals generated at terminals, also called taps, along a device such as a delay line are synthesized in a predetermined way. It is known that it can be formed without. Such devices, often referred to as “overturning” or “lateral filters”, are described in US Patent No. 2,263,376 entitled “Electromagnetic Filters,” generally granted to A.D.B. "The Horizontal Filters," written by Esch-Ye-Kalman, published in Bulletin No. 28, pp. 302-310, and the IRE for Broadcast and Television Receivers, July 1955. The paper, “Selectivity and Transient Synthesis,” by R. Double and Sonelfeld, presented in Newsletter Volume BTR-1, Volume 3, pages 1-8, and March, 1960, in the Bell Systems Technical Magazine. It is also described in a paper entitled "The Transverse Crushers for Television Circuits" by Al-Bui, Sperry and Dei-Shrrenian, presented in Vol. 39, No. 2, pp. 405-422.

Lateral isolators have generally been widely applied in the field of signal processing. For example, it has been found that such devices are useful for horizontal and vertical aperture beam calibration as described in US Patent No. 2,759,044 entitled "Horizontal and Vertical Beam Opening Calibration" issued to B.M.Oliver.

In US Patent Application No. 486,241 entitled “Television Signal Processing Device” by Joseph Peter Bingham, co-pending, transverse equalizers useful for television signal processing devices emphasize the amplitudes of the relatively high frequency components of the brightness signal portions of the video signal. It has been described as reducing the chromaticity and amplitude of speech parts of the signal.

The present invention relates to a device useful for suppressing the generated transient noise. According to the present invention, a narrowband device responsive to a television video signal provides a relatively narrowband high attenuation at DC and frequency f when it is desired to attenuate video signals such as chroma and voice subcarrier frequencies, and relatively high frequency of the brightness signal. If it is desired to attenuate video signals, such as in a range, provide a peak amplitude at a frequency between O and f. The threshold suppresses the amplitude portion of the narrowband signal above a predetermined threshold. Broadband devices responsive to television signals provide a wideband signal that includes signal components in the frequency range between DC and f. The wideband signal is synthesized with a narrowband signal that is amplitude suppressed to provide a relatively unnoticed output signal in which chromatic or speech signals are attenuated and relatively high frequency brightness signals are emphasized.

According to another feature of the invention, a signal delay device responsive to television signals comprises a plurality of terminals or taps for generating delay signals. At least one of the delayed signals is used to derive a wideband signal. The other two of the delay signals having an average delay time approximately equal to the delay time of the first delay signal are combined with the wideband signal to derive the narrowband signal.

Referring to FIG. 1, a general arrangement of color television receivers using the present invention comprises a composite video signal including chromaticity, brightness, voice and synchronization signals by means of a suitable intermediate frequency circuit (not shown) and a detection circuit (not shown). The responder comprises a signal processing stage 12 for generating radio frequency (RF) television signals received by the antenna to generate. The signal processing stage 12 comprises, for example, a synchronous detector in the form of a heightened carrier or red detector known in the art. A heightened carrier-type synchronous detector suitable for use in signal processing circuit 12 is described in US Pat. No. 3,812,289 entitled "Television Receivers Using Synchronous Video Detectors" issued to J. Arbins.

The output of the signal processing stage 12 is coupled to the chroma channel 14, the brightness channel 16, the channel 18 for processing the sync pulses, and the channel (not shown) for processing the audio signals.

Chromaticity channel 14 includes chromaticity processing stage 20 which acts to derive color signals, e.g., R-Y, B-Y and G-Y chrominance signals, from the chromaticity portion of the composite video signal.

The brightness signal processing stage 22 is included in the brightness channel 16 and appears on the brightness channel 16 such as chroma and audio signals while emphasizing or peaking the amplitudes of the relatively high frequency components of the brightness signals to improve the transient response and fine resolution of the television receiver. It serves to reduce undesirable signals. Brightness signal processing stage 22 includes an apparatus for reducing or suppressing relatively high frequency components having amplitudes above a threshold predetermined to suppress transient interference or noise. In addition, the brightness signal processing stage 22 also serves to assimilate the time delay of signals processed in the chroma channel 14 and the brightness channel 16.

The output of the brightness signal processing stage 22 is coupled to a brightness amplifier 24 which acts to amplify or process the output signals of the brightness signal processing stage 22 to produce the output signal Y of the brightness channel 16.

The Y output signals of brightness channel 16 and the R-Y, G-Y and B-Y chrominance output signals of chromaticity channel 14 are coupled to kinescope drivers 26 which are matched to form R, G and B color signals. R, G and B color signals drive kinescope 28. Optionally, the brightness and color chrominance signals may match within kinescope 28 as is known.

The contrast control stage 30 is coupled to the brightness amplifier 24 to control the amplitude of the brightness signals to control the contrast of the image generated by the kinescope 28. The contrast control stage 30 may be coupled to the chroma processing stage 20 to control the saturation of the image generated by the kinescope 28 by controlling the amplitude of the chroma signal. The brightness control stage 32 is coupled to the brightness amplifier 24 to control the components of the brightness signals to control the brightness of the image generated by the kinescope 28. Suitable contrast and luminance control devices are described in US Pat. No. 3,804,981 entitled "Luminance Control" to J. Avins.

Channel 18 includes a synchronous separator 34 that serves to separate horizontal and vertical synchronization pulses from the video signal. The synchronization pulses are coupled to the deflection circuit 36 from the synchronous separator 34. The deflection circuit 36 is coupled to the kinescope 28 and also to the high voltage stage 38 to control the deflection and sweep of the electron beam within the kinescope 28. The deflection circuit 36 generates an erase signal coupled to the brightness amplifier 24 to suppress the output of the brightness amplifier 24 during the horizontal and vertical retrace periods to ensure the kinescope 28 is blocked during each of these periods.

The general circuit arrangement shown in FIG. 1 is, for example, 1973 C-8, RCA Color Television Service Data (CTC-68) published by Indiana, Indianapolis, R.C.Coporation. Type receivers are suitable for use with color television receivers of the type described in.

The brightness signal processing stage 22 includes a signal delay device 10 shown as a delay line in response to the video signal Vi, and a plurality of terminals or tabs 112a, 112b, 112c and 112d coupled to the delay device 110 at successive points. The combination of signal delay device 110 and taps 112a, 112b, 112c and 112d is sometimes called a tap delay line. Although the delay device 110 is shown as an inductive or coil form of the delay line, it can be formed by any other device suitable for delaying the video signal, such as a charge coupled device (CCDV) or an array of charge transfer devices. Although tabs 112a, 112b, 112c, and 112d may appear to be directly connected to delay line 110, they are coupled to the delay device in other suitable ways provided for signal coupling, such as by capacitive coupling or the like.

Taps 112a, 112b, 112c and 112d are the time associated with V _i by the respective time intervals T _D , T _D + T ₁ , T _D + T ₁ + T ₂ and T _D + T ₁ + T ₂ + T ₃ . Coupled to delay line 110 in an isolated interval to generate respective delayed video signals a, b, c and d delayed by. Portion 116 of delay line 110 with a time delay interval T _D is provided in front of tab 112a and is selected to equalize the time delays of the processed signals in the brightness and chroma channels of the receiver. In order to equalize the time delay of the processed signals in the chroma and lightness channels, the sum of T _D , T ₁ and 1/2 (T ₂ ) equals the difference between the time delays of the processed signals in the chromaticity and lightness channels. It is preferable. It should also be noted that the signal resulting from synthesizing signals generated at taps symmetrically located around a given point of the delay line can be thought of as having a time delay equal to the average of the time delays of the synthesized signals. Therefore, if tabs 112a, 112b, 112c and 112d are symmetrically disposed near the point of the weighted delay line 110 between tabs 112a and 112d, it is derived by synthesizing the signals generated at tabs 112a, 112b, 112c and 112d. The output signal will have a time delay equal to the time delay that needs to equalize the time delay of the processed signals in the chroma and brightness channels.

Delayed signals b and c are coupled to a summation circuit 118 where they are logarithmically summed to produce a relatively wideband signal V _b . As can be seen, the bandwidth of the brightness signal processing stage 12 is mainly determined by V _b . The summation circuit 118 may be formed by a suitable circuit for algebraic summation of signals such as operational amplifiers, resistance matrices, and the like. Summing circuit 118 acts to correct (ie, load) the amplitudes of b and c before adding them.

Delay signals a and d and V _b are coupled to a summation circuit 120 which acts to algebraically subtract a and b from V _b to produce a relatively narrowband signal V _p . The summation circuit 120 is formed in the same manner as the summation circuit 118 and includes a device for correcting (ie, loading) the amplitudes of a, d and V _b before subtraction. As can be seen, the peaking characteristic of the signal processing stage 22 is mainly determined by V _p .

The output signal V _p of the summing circuit 120 is coupled to the variable gain device 122 which serves to modify the amplitude of V _p to generate the signal PV _p . Here, P is a gain (or attenuation) factor of the variable gain device 122. For example, the variable gain device 122 formed by the variable gain amplifier is arranged to create a gain range extending from a value less than 1 to a value greater than 1 in response to the control signal generated by the peaking control circuit 124. The circuit 124 may be formed of a suitable device for providing a control signal, for example in response to a manual adjustment. Optionally, circuit 124 may be arranged to derive a control signal from a portion of the composite video signal exhibiting picture quality as described in US Pat. No. 516,491 filed by Joseph Peter Tubing.

The output PV _p of the device 122 is coupled to a threshold circuit 126 which serves to reduce or suppress a portion of PV _p above a threshold that is intended to suppress transient noise. Threshold circuit 126 includes, for example, a large-lymphing and limiting circuit to provide a modified form of PV _p through which transients above a predetermined threshold are cut off. A suitable large-lymph circuit is shown as part of the circuitry of the brightness signal processing stage 22 in FIG. Threshold circuit 126 includes a noise converter or spotter circuit, for example, to provide a modification of PV _p that essentially transforms transients above a predetermined threshold. Such a circuit is described in detail in the aforementioned Georges patent.

2 shows the operation of circuit 126 for various forms. 2 shows, for example, portions corresponding to white-black transient 214a and black-white transient 216a, and portions corresponding to transient noise that may be generated by ringing in the IF portion of the signal processing stage 12 of FIG. A graphical representation 212a of PV _p including 218a is shown. In FIG. 2, the signal portion above the time axis corresponds to the transient toward white and the signal portion below the time axis corresponds to the transient toward black. It should be noted that PV _p is a relatively narrowband signal containing only the relatively high frequency components of the brightness signal. The operation of signal processing stage 126 to provide a relatively high frequency narrow band amplitude versus frequency transfer characteristic associated with PV _p will be described next with reference to FIG.

Waveform 212b corresponds to the amplitude suppression form of PV _p provided when threshold circuit 126 in FIG. 1 is arranged as a crimping circuit. The portion of waveform 212b above the predetermined threshold 220b is cut to form flat portion 222b. Only the portion of PV _p above the predetermined threshold value 220b is changed.

Waveform 212c corresponds to the amplitude suppression form of PV _p provided when threshold circuit 126 in FIG. 1 is arranged as a noise converter or spotter circuit. Portions of waveform 212c above predetermined threshold 220c are transformed in essence. Only portions of PV _p above the predetermined threshold 220c are changed.

Although the operation of the threshold circuit 126 is described as suppressing signal portions of PV _p extending beyond the predetermined threshold in the white direction, this may be arranged to suppress signal portions extending below the predetermined threshold in the black direction, It may be arranged to suppress excessive noise in two directions, white and black.

Transient suppression forms of PV _p and V _b are combined in a summation circuit 128. Circuit 128 is similar to summing circuits 118 and 120 and acts to algebraically add transient suppression forms of PV _p and V _b to issue output signal V ₀ of brightness signal processing stage 22.

The predetermined threshold of the threshold circuit 126 can be controlled according to the amplitude of PV _p . For this purpose, the control signal generated by the peaking control stage 124 can be coupled to the threshold circuit 126 to control this threshold directly in relation to the amplitude of PV _p , as indicated by dashed line 130. For this purpose, the signal induced by the peaking control stage 124 is preferably a DC signal. Likewise, as indicated by dashed line 132, the control signal induced by contrast control stage 30 (preferably in the form of DC) may be coupled to threshold circuit 126 to control its threshold directly in relation to the amplitude of brightness signal Y.

In order to easily understand the operation of the brightness signal processing stage 22 of FIG. 1, an amplitude-to-frequency transfer characteristic of a device such as a tap delay line will be described. The amplitude versus frequency propagation characteristic of the delay line portion that provides the time delay t to the applied signal can be expressed as a function of frequency, i.e., an exponentially changing coefficient as e- ^jwt . Where e is the natural logarithm base and w is the frequency unit. By algebraically adding two such signals generated at each tap (i.e., + t and -t) located symmetrically around the reference point, the amplitude versus frequency transfer characteristics associated with the generated signal vary as a cosine function.

In one example, the tabs 112a, 112b, 112c and 112d are symmetrically positioned around the point of the delay line 110 in the middle between the tabs 112a and 112d and the time intervals T ₁ , T ₂ and T ₃ are equal to 1 / 2f. Can be assumed. Where f is the frequency of signal component V _i , which may be undesirable in brightness channel 16 of FIG. 1. For example, f is a signal frequency within a range of frequencies that includes chromaticity and speech subcarriers. In particular, according to US standards, f is a color subcarrier, i.e., 3.58 MHz or a voice cross-carrier frequency, i.e., 4.5 MHz. Also assume, for example, that the summation circuit 118 is arranged to correct the amplitudes of the delay signals b and c with a half load. The summation circuit 120 is arranged to correct the amplitudes of the delayed signals a and d with a half load and the relatively wideband signal V _b with a load of one.

In general, it is preferable that the interval delay signals a and d be separated in time by the predetermined interval NT / 2. Where N is the integrator and t is the inverse of the frequency f. The preferred range of N includes integrators between 2 and 5. In the above example, N was chosen to be 3. Different values of N may be useful for other special meltings.

As values according to the above examples, the signals V _b , V _p , PV _p and V ₀ are related to the delay signals a, b, c and d according to the following equation.

3 is a graph showing amplitude versus frequency transfer characteristics associated with signals 1/2 (a + b), V _b , PV _p and V ₀ . The transfer characteristic associated with V _b is a cosine function with a repetition rate of 4f, while the transfer characteristic associated with 1/2 (a + d) is a cosine function with a 4 / 3f repetition rate. In a frequency range extending from DC (i.e., zero frequency) to f, V _b is a relatively broadband. That is, the range of its bandwidth is determined by the delay time between delay signals b and c. The propagation characteristics associated with V _p are relatively narrowband with amplitude equal to zero at DC and f, peak amplitude at 2 / 3f, and zero amplitude at DC and f. The peak amplitude position of the transfer characteristic with respect to V _p is determined by the time delay between the delay signals a and d. The transfer characteristics associated with V ₀ are relatively emphasized or peaked at 2 / 3f and relatively attenuated at f.

Since the amplitude of the transmission characteristic associated with V _p is zero at DC and f, adjusting p while controlling the amplitude of the transmission characteristic in the vicinity of 2 / 3f does not affect the amplitude of the transmission characteristic at DC or f. This adds V _b and PV _p (equation 4) so that the amplitude of V ₀ formed can be adjusted to within 2 / 3f without affecting the amplitude (1) of the transfer characteristic at DC or the amplitude (zero) at f. It is preferable because of that. This feature in DC, that is, the amplitude of the transfer characteristic from the DC component of V ₀ are related to the luminance characteristics of the reproduced picture from V ₀ is the amplitude of the transfer characteristic at f are related to the attenuation of the undesired component of V ₀ Remarkable

The amplitude transient of V ₀ includes both preshoot and overshoot (see FIG. 2). These preshoots and overshoots are, for example, because the playback image appears just before the white-black transition and appears whiter than in the initial image, and during the period immediately after the transition, the playback image appears more black than the initial image. It serves to emphasize the hue and degree in the image formed from V ₀ . In addition, phase-to-frequency propagation is related to preshoot and overshoot. For example, the linear phase relative frequency transfer characteristic corresponds to the same form of preshoot and overshoot. The preshoot and overshoot are determined by the signal formed by the sum of the delay signals a and d. Therefore, even if the loads of a and d are selected to be equal before being summed in the summation circuit 120, and the time intervals T ₁ and T ₃ are selected to be the same, they result in linear phase-to-frequency transfer characteristics.

Brightness signal processing signal 22 may be modified to produce unequal preshoots and overshoots to compensate for phase-to-frequency non-appropriation in other parts of the video signal processing system.

As already noted, the threshold circuit 126 of FIG. 1 serves to suppress an amplitude portion PV _p above or below a threshold level that is intended to suppress transient noise. The threshold circuit 126 is arranged to operate on high frequency signals in a relatively narrow frequency range without low frequency components. Therefore, the operation of circuit 126 does not affect the relatively low frequency components. Circuit 126 does not unnecessarily produce nonlinearities that can interfere with the gray scale of the image associated with the relatively low frequency signals.

It should also be noted that if P accidentally increases beyond the allowable descent, circuit 126 suppresses the final overshoot and preshoot extending beyond the predetermined threshold. Excessive backback preshoots or overshoots tend to cause spot defocusing associated with undesirable excessive beam currents.

Because of the inherent nonuniform time delays of PV _p and V _b , it is noted that V _b and PV _p are of the proper time relationship and proper polarity when they are synthesized to form V ₀ without the need for an additional signal delay circuit. Should be.

The selection of time intervals T ₁ , T ₂ and T ₃ (i.e. reciprocal of the color carrier frequency 3.58 MHz) equivalent to 140 nanoseconds (10 ^-9 seconds) to the United States standard carries the amplitude-to-frequency transmission associated with V _0. This is advantageous because the characteristic has a peak amplitude at a relatively high frequency around 3.58 MHz, approximately 2/3 x 3.58 MHz (ie 2.4 MHz), while providing efficient 3.58 MHz trapping. However, other choices of T ₁ , T ₂ and T ₃ may be used. For example, it is preferable to select T ₂ equal to 110 nanoseconds and T ₁ and T ₃ equal to 140 nanoseconds. In this case, the amplitude-to-frequency transfer characteristic associated with V ₀ has a zero value at about 4.1 MHz, while the peak amplitude at about 2/3 × 3.58 MHz (ie 2.4 MHz). Therefore, the signal processing apparatus of FIG. 1 can be modified such that the frequency components within the respective chromaticity and speech subcarriers of the video signal are relatively reduced and the amplitude of the relatively high frequency components of the brightness signal is relatively increased.

FIG. 4 shows the brightness signal processing stage 22 of FIG. 1, wherein a portion included in the dotted line 400 is suitable for being configured as an integrated circuit. Circuits having a resistance value as shown in FIG. 4 are arranged to provide a signal consistent with the example used in the circuit operation description of FIG. The circuit of FIG. 4 may be modified to apply to other applications.

In FIG. 4, delay line 410 is selected to equalize delay times of signals processed in chroma channel 14 and brightness channel 16 of FIG. In addition, the delay line 410 is a delay time T _D , T _D + T ₁ , T _D + T ₁ + T ₂ and T _D + T ₁ + T ₂ + to generate delay signals a, b, c and d, respectively. It has an output impedance nearly equal to the characteristic impedance of the delay line 410 so as to delay the input video signal Vi between the continuous taps 412a, 412b, 412c and 412d by a time equivalent to T ₃ . Delay line 410 terminates at impedance 408 with a value selected that is approximately equal to the value of the characteristic impedance of delay line 410 to minimize tip reflection.

Taps 412a and 412d are coupled to two inputs of a differential amplifier 414 consisting of NPN transistors 411 and 418, respectively, where delayed signals a and b are ½ (a + d) at the junction of resistors 420 and 422 of differential amplifier 414. Loaded and logarithmically added to derive The input impedance of the differential amplifier 414 is made relatively high compared to the characteristic impedance of the delay line 410 by appropriately selecting the values of the emitter resistors of the transistors 411 and 418.

Tabs 412b and 412c constitute a circuit for loading and algebraically adding signals b and c coupled via resistors 424 and 425 to resistors 424 and 425 respectively to the base of transistor 416 arranged in a common collector configuration. The resistors 424 and 426 are selected to have a relatively large value compared to the characteristic impedance of the delay line 410 so as not to load the delay line 410. The signal generated at the emitter of transistor 416 is equal to 1/2 (b + c), ie V _b .

1/2 (b + c) is generated in the same way as 1/2 (a + d), but is described as occurring in a summation circuit comprising transistors 416 and resistors 424 and 426 to preserve the integrated circuit input terminals. It should be noted that there is.

), 즉 PV_p를 유도하도록 1/2(b+c)로부터 감산된다. 여기서 PV_p자동증폭기 432의 이득이다.Signals 1/2 (b + c) and 1/2 (a + d) are coupled to the input of differential amplifier 432, respectively, via emitter follower stages, which include NPN transistors 428 and 430, respectively. Differential amplifier 432 includes NPN transistors 436 and 434. Where 1/2 (a + d) is the collector of transistor 434

), I.e. subtract from 1/2 (b + c) to derive PV _p . This is the gain of the PV _p auto amplifier 432.

The gain P of the differential amplifier 432 can be adjusted by adjusting the voltage at the peaking control terminal of the peaking control circuit including the NPN transistors 438, 448, 450. This adjustment is consistent with setting the gain of the variable gain device 122 in FIG. This peaking control circuit is coupled to the emitter and collector circuits of transistor 434 in such a way that the gain of differential amplifier 432 can be adjusted in response to the peaking control voltage without changing the DC voltage at the output of differential amplifier 432. That is, the current supplied by the collector of transistor 448 is proportional to keep the DC component of the output signal of differential amplifier 432 constant despite variations in peaking control voltage.

The output signal PV _p of the differential amplifier 432 is coupled to the emitter of the PNP transistor 452 which together with the variable resistor 456 and resistors 454 and 458 form a clipping circuit for suppressing transient noise by removing portions of PV _p above a predetermined threshold. The predetermined threshold is determined by controlling the position of the arm of the variable resistor 456 to set the voltage at the base of the transistor 452. If the amplitude of PV _p exceeds the voltage at the base of transistor 452 by about 0.6V, then the base-emitter junction of transistor 452 is conductive. As a result, the amplitude of PV _p is limited to the same value as about 0.6 V plus the voltage at the base of transistor 452.

Although the clipping circuit of FIG. 4 is arranged to remove PV _p portions above the predetermined threshold in the forward direction, ie, white transient noise, the clipping circuit can be modified to remove both black and white transient noise or black transient noise. It should be noted that you can. It should be noted that other suitable circuitry for suppressing transient noise above a predetermined threshold, such as a noise inverter or spotter circuit, may be used in place of this clipping circuit. In addition, a DC peaking or contrast control voltage can be coupled to transistor 452 such that a predetermined threshold is controlled in accordance with these control voltages.

A modification of PV _p is coupled to the base of NPN transistor 440, which together with the series resistances 442, 444 and 446 form an emitter follower circuit. The signal 1/2 (b + c) set at the emitter of NPN transistor 428 is coupled to the junction of resistors 444 and 446 through resistor 459. Here this signal is algebraically added to the modification of PV _p to derive the output signal V ₀ .

FIG. 5 shows a schematic diagram of another embodiment suitable for use in place of the brightness signal processing stage 22 of FIG. The embodiment of FIG. 5 is somewhat similar to that of less complexity than the brightness signal processing stage 22 of FIG. 1 but with fewer taps of delay lines being used in FIG.

The brightness signal processing stage of FIG. 5 is related to Vi 'by the delay line 510 corresponding to the input video signal Vi and the respective time intervals TD' T _D '+ T ₁ ' and T _D '+ T ₁ ' + T ₂ '. And a plurality of terminals or tabs 512a, 512b and 512c coupled to the delay line 510 at discrete intervals to generate respective delay signals a ', b' and c 'delayed with time. Delay line 510 includes part 516 before tab 512 with a time delay interval T _D ′ selected for the other parts of delay line 510 to equalize the delay times of the signals processed in the brightness and chroma channels of the receiver of FIG. do.

Delay signals a ', b' and c 'are coupled to summing circuit 518, where a' and c 'are logarithmically subtracted from b' to form a relatively narrowband signal V _p '. The output of the summing circuit 518 V _p 'is P'V _p in accordance with a control signal generated by the feeder control circuit 524 King' is coupled to the variable gain device 522 that modify the amplitude V _p 'to form. Where P 'is the gain of the device 522. The output P′V _p ′ of the variable gain device 522 is coupled to a threshold circuit 526 which serves to suppress a portion of P′V _p ′ above a threshold that is intended to suppress impulse noise. A predetermined threshold in the threshold circuit 526 may be controlled by the amplitude of the amplitude of the brightness understanding P'V _p 'as shown by dotted line 530 or call control contrast control stage indicated by a broken line signal. The modifications of P'V _p 'and b' are combined in the summation circuit 528 and these are added algebraically to form the output signal V ₀ '.

The operation of the brightness signal processing circuit of FIG. 5 will be described as an example in the case where T ₁ ′ and T ₂ ′ are selected to be equal to a delay time of 1 / f ′. Where f 'is the frequency of the signal component of Vi' such as chroma or audio subcarrier, which is undesirably output to brightness channel 16 in FIG. Again, for example, the summation circuit 518 is arranged to load these components by 1/2, 1 and 1/2 respectively before combining the components a ', c' and b '. As these values V _p ′, _P ′ V _p ′ and V ₀ ′ relate to delay signals a ′, b ′ and c ′ according to the following equations.

It is generally known that the separated delay signals a 'and c' are preferably separated in time by a predetermined interval N'T '/ 2. Where N 'is an integer and T' is the inverse of the frequency f '. The preferred range of N 'includes an integer between 2 and 5. In the above example, N 'was selected as four. Other values of N 'are useful in other specific applications.

FIG. 6 is a graph showing amplitude versus frequency transfer characteristics associated with b ', 1/2 (a' + c '), V _p ', P'V _p 'and V ₀ '. The transmission characteristic associated with b 'is flat. That is relatively broadband. The transfer characteristic associated with 1/2 (a '+ c') is a cosine function with a minimum amplitude and a regeneration ratio of f 'at integer multiples of 1 / 2f'. Transfer characteristic related to V _p 'and _p P'V "is a relatively narrow band with peak amplitude at integer multiples of the" amplitude of zero and 1 / 2f in integer multiples of DC and "f. The propagation characteristics associated with V0 'have a peak amplitude at integer multiples of 1 / 2f' and have an integer multiple of f 'and an amplitude equal to 1 at DC. In FIG. 6, for example, p 'is selected to have a value of 0.5 as a result of V ₀ ' having a peak amplitude equal to two.

It is noted that V ₀ 'is relatively emphasized at 1 / 2f' and is reduced relatively at f '. Therefore, if f 'is chosen to be equal to the color-carrier frequency of the United States standard, i.e. 3.58 MHz, the signal near the color carrier (i.e. the chroma signal) is relatively attenuated while the signal near 1.8 MHz (i.e. the relatively high frequency brightness). Signal) is highlighted or pig prior to comparison. Since the amplitude at 3.58 MHz is 1 in this example, it is preferable to provide a filter circuit or a trap before or after the signal processing circuit of FIG. 5 to attenuate the chroma signal.

In addition, the amplitude transients of V ₀ ′ include, for example, white-black transients, such that the image played back from V ₀ ′ is blacker in the initial scene immediately after the transition and more white in the initial scene just before the transition. It will include preshoots and overshoots that act to emphasize amplitude transients. These preshoots and overshoots are controlled by the delay signals a 'and c'. Preshoots and overshoots are related to the phase linearity of the brightness signal processing circuit of FIG. Therefore, although delay signals a 'and c' are selected to provide the same preshoot and overshoot exhibiting linear phase frequency propagation characteristics, the delay signals a 'and c' are different in different parts of the receiver of FIG. It may be chosen to provide unequal preshoots and overshoots to compensate for phase nonlinearity.

Adjustment of V _0, p for controlling the amplitude of the peak part of the transmission characteristics associated with "does not affect the amplitude of the transfer characteristic related to the ₀ V in 'DC or f. Therefore, reduction in the undesired portion of the "brightness nature or related to V _0, V0 is not influenced by the adjustment of the P '.

Since P'V _p 'is a relatively narrowband signal that occupies a relatively high frequency range, relatively low frequency components are not affected by the amplitude suppression action of the critical circuit 526 of FIG. Therefore, circuit 526 does not adversely affect the gray scale portion of the image reproduced from V ₀ related to the relatively low frequency component of V6.

Circuit 526 serves to prevent spot disconnection associated with excessive beam currents occurring in response to excessively large preshoots and overshoots when P 'accidentally increases above tolerance.

In addition, the relatively wideband signal b 'provided by the lightness signal processing circuit of FIG. 5 and the relatively narrowband signal P'V _p ' in a modified form have a proper polarity and are identical in time so that they are easily combined to form V ₀ '. Can be.

Although the invention has been described with reference to specific embodiments, various modifications and changes can be made within the scope of the invention. For example, although wideband and narrowband signals are provided by the respective parts of the cross filter, other devices for providing signals having amplitude and frequency characteristics similar to those associated with wideband and narrowband signals may be used. It may be.

Claims (1)

As described herein and shown in the figures, a television video signal source, a broadband device responsive to the video signals to derive a relatively wideband signal comprising a signal component within a frequency range between zero Hz and frequency f, and zero; A television video signal having a relatively low amplitude responsive at a frequency of Hz and nearly f and a peak amplitude responsive at a frequency between zero Hz and f and comprising a narrowband device responsive to the video signal to derive a relatively narrowband signal. A processing apparatus comprising, in particular, a threshold device responsive to said narrowband signal to suppress amplitude portions of said narrowband signal above a predetermined threshold level, an output signal having a relatively emphasized high frequency signal component and a relatively suppressed noise component; Synthesize the wideband signal with the amplitude suppressed narrowband signal to generate a Transient noise suppression apparatus in a telrebijyon video system that is characterized by output apparatus for.

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