KR20250008492A - Method and apparatus for designing signals immune to phase distortion - Google Patents

Abstract

A signal design technique immune to phase distortion is provided. The present invention relates to a method of a transmitter, comprising: a step of modulating a source bit stream to be transmitted to a receiver to generate a first symbol block; a step of inserting at least one pilot into the first symbol block to generate a second symbol block; a step of cumulatively linearly encoding the second symbol block to generate a third symbol block; a step of performing a real-number phase transform on the third symbol block to generate a first signal converted into a phase value; a step of adding a CP (cyclic prefix) to the first signal to generate a second signal; and a step of wirelessly transmitting the second signal to the receiver.

Description

{METHOD AND APPARATUS FOR DESIGNING SIGNALS IMMUNE TO PHASE DISTORTION}

The present disclosure relates to a signal design technique immune to phase distortion, and more particularly, to a signal design technique immune to phase distortion that has strong resistance to phase distortion caused by hardware damage, Doppler shift, etc.

With the development of information and communication technology, various wireless communication technologies can be developed. Representative wireless communication technologies include LTE (long term evolution), NR (new radio), and 6G (6th Generation), which are specified in the 3GPP (3rd generation partnership project) standard. LTE can be one of the 4G (4th Generation) wireless communication technologies, and NR can be one of the 5G (5th Generation) wireless communication technologies.

In order to process wireless data that has been rapidly increasing since the commercialization of 4G communication systems (e.g., communication systems supporting LTE), a 5G communication system (e.g., communication systems supporting NR) that uses a higher frequency band (e.g., frequency bands higher than 6 GHz) than the frequency band of the 4G communication system as well as the frequency band of the 4G communication system (e.g., frequency bands lower than 6 GHz) can be considered. The 5G communication system can support eMBB (enhanced Mobile BroadBand), URLLC (Ultra-Reliable and Low Latency Communication), and mMTC (massive Machine Type Communication).

Meanwhile, most conventional wireless communication systems may be based on the cyclic prefix (CP)-orthogonal frequency division multiplexing (OFDM) transmission scheme, which can be robust to multipath fading and have excellent frequency efficiency. However, the CP-OFDM transmission scheme has a high peak-to-average power ratio (PAPR), which may require high linearity of the amplifier, thereby increasing power consumption. In addition, the CP-OFDM transmission scheme may not be inherently robust to phase distortion caused by hardware impairments such as carrier frequency offset (CFO) and phase noise (PhN), and to phase distortion caused by high-speed mobility such as Doppler shift or Doppler spread. Accordingly, wireless communication systems may require a technology robust to phase distortion.

The purpose of the present disclosure to solve the above problems is to provide a signal design method and device that are immune to phase distortion and exhibit high resistance to phase distortion.

In order to achieve the above object, a signal design method immune to phase distortion according to a first embodiment of the present disclosure is provided as a method of a transmitter, comprising: a step of modulating a source bit stream to be transmitted to a receiver to generate a first symbol block; a step of inserting at least one pilot into the first symbol block to generate a second symbol block; a step of cumulatively linearly encoding the second symbol block to generate a third symbol block; a step of performing a real-valued phase transform on the third symbol block to generate a first signal converted into a phase value; a step of adding a CP (cyclic prefix) to the first signal to generate a second signal; and a step of wirelessly transmitting the second signal to the receiver.

In this case, in the step of generating a first symbol block by modulating a source bit stream to be transmitted to the receiver, the transmitter may generate the first symbol block from the source bit stream using any one of bipolar amplitude shift keying (BASK) modulation, phase shift keying (PSK) modulation, quadrature and amplitude modulation (QAM) modulation, Gaussian filtered minimum-shift keying (GMSK) modulation, or differential phase-shift keying (DPSK) modulation.

Here, in the step of inserting at least one pilot into the first symbol block, the transmitter can insert a first pilot into a preceding time domain of the first symbol block and a second pilot into a succeeding time domain of the first symbol block.

Here, the last time domain samples of the first pilot and the first time domain samples of the second pilot can have a power value of zero.

In this case, in the step of inserting at least one pilot into the first symbol block, the transmitter may insert a third pilot into the middle of the first symbol block.

Here, the first time domain samples and the last time domain samples of the third pilot can have a power value of zero.

Here, the step of generating a third symbol block by performing cumulative linear encoding on the second symbol block may include the step of dividing the second symbol block into first partial symbol blocks based on a cyclic period of a time-domain sample unit; and the step of performing differential encoding on each of the first partial symbol blocks to generate the cumulative linearly encoded third symbol block.

In this case, in the step of generating the cumulative linearly encoded third symbol block by performing differential encoding on each of the first partial symbol blocks, the transmitter may perform differential encoding after adding the first sample value of the previous first partial symbol block to the first sample of each of the first partial symbol blocks.

Here, the step of generating a third symbol block by performing cumulative linear encoding on the second symbol block may include the step of dividing the first symbol block within the second symbol block into second partial symbol blocks; and the step of performing differential encoding on each of the second partial symbol blocks to generate the cumulative linearly encoded third symbol block.

Here, the step of generating a third symbol block by cumulatively linearly encoding the second symbol block may include the step of differentially encoding the second symbol block to generate the cumulatively linearly encoded third symbol block.

Meanwhile, a method for designing a signal immune to phase distortion according to a second embodiment of the present disclosure for achieving the above object may include, as a method of a receiver, the steps of: converting a reception signal received from a transmitter into a baseband signal; removing a CP from the baseband signal; equalizing a wireless channel with respect to the reception signal from which the CP has been removed; performing a time phase transform on the reception signal from which the wireless channel has been equalized; performing cumulative linear decoding on the time phase transformed signal; and demodulating the cumulative linear decoded signal to generate a source bit stream.

Here, the step of equalizing a wireless channel for the received signal from which the CP has been removed may include the step of converting a time domain pilot signal of the received signal from which the CP has been removed into a frequency domain pilot signal; the step of converting a time domain symbol signal of the received signal from which the CP has been removed into a frequency domain symbol signal; the step of estimating a first wireless channel from the frequency domain pilot signal; the step of estimating a second wireless channel from the frequency domain symbol signal; and the step of equalizing the second wireless channel using the first wireless channel.

Here, the step of performing cumulative linear decoding on the time phase-shifted signal may include the step of dividing the time phase-shifted signal into first partial symbol blocks based on a cyclic period of a time domain sample unit; and the step of performing differential decoding on each of the first partial symbol blocks to generate the cumulative linear decoded signal.

Here, the step of performing cumulative linear decoding on the time phase shifted signal may include the steps of: extracting a symbol block of the time phase shifted signal; dividing the symbol block into second partial symbol blocks; and performing differential decoding on each of the second partial symbol blocks to generate the cumulative linear decoded signal.

Here, the step of performing cumulative linear decoding on the time phase shifted signal may include the step of extracting a symbol block from the time phase shifted signal; and the step of differentially decoding the symbol block to generate the cumulative linear decoded signal.

Here, the step of demodulating the accumulated linearly decoded signal to generate a source bit stream may include the steps of demodulating the accumulated linearly decoded signal to generate a first symbol block; removing a pilot from the first symbol block to generate a second symbol block; and performing symbol demapping on the second symbol block to generate the source bit stream.

Meanwhile, a signal design device immune to phase distortion according to a third embodiment of the present disclosure for achieving the above purpose includes a processor as a transmitter, wherein the processor can cause the transmitter to modulate a source bit stream to be transmitted to a receiver to generate a first symbol block; insert at least one pilot into the first symbol block to generate a second symbol block; perform cumulative linear encoding on the second symbol block to generate a first signal converted into a phase value; perform real-number phase transform on the third symbol block to generate a second signal by adding a CP (cyclic prefix) to the first signal; and wirelessly transmit the second signal to the receiver.

Here, in the step of inserting at least one pilot into the first symbol block, the processor can cause the transmitter to insert a first pilot into a preceding time domain of the first symbol block and to insert a second pilot into a succeeding time domain of the first symbol block.

In this case, in the step of inserting at least one pilot into the first symbol block, the processor may cause the transmitter to insert a third pilot into the middle of the first symbol block.

Here, in the step of generating a third symbol block by performing cumulative linear encoding on the second symbol block, the processor can cause the transmitter to divide the second symbol block into first partial symbol blocks based on a cyclic period of a time-domain sample unit; and perform differential encoding on each of the first partial symbol blocks to generate the cumulative linearly encoded third symbol block.

According to the present disclosure, a wireless communication system can eliminate central symmetric mapping of the same data without reducing frequency efficiency. In addition, according to the present disclosure, a wireless communication system can effectively allocate pilots for channel estimation. In addition, according to the present disclosure, a wireless communication system can weight the result values of each phase-shifted time domain sample to an extent that quantization error does not become a problem, so that it can have high resistance to phase distortion.

Figure 1 is a conceptual diagram illustrating a first embodiment of a communication system.
FIG. 2 is a block diagram illustrating a first embodiment of a communication node constituting a communication system.
FIG. 3 is a block diagram showing a first embodiment of a signal design device immune to phase distortion.
FIG. 4 is a flowchart illustrating a first embodiment of a signal design method that is immune to phase distortion performed at a transmitter.
Figure 5a is a conceptual diagram showing the signal constellation of 2-BASK modulation.
Figure 5b is a conceptual diagram showing the signal constellation of 4-BASK modulation.
Figure 5c is a conceptual diagram showing the signal constellation of 8-BASK modulation.
Fig. 6a is a conceptual diagram showing a first embodiment of a signal constellation of 2-PSK modulation.
Fig. 6b is a conceptual diagram showing a second embodiment of the signal constellation of 2-PSK modulation.
Figure 6c is a conceptual diagram showing the signal constellation of 4-PSK modulation.
Figure 6d is a conceptual diagram showing the signal constellation of 8-PSK modulation.
Figure 7a is a conceptual diagram showing the signal constellation of 4-QAM modulation.
Figure 7b is a conceptual diagram showing the signal constellation of 16-QAM modulation.
Figure 8 is a conceptual diagram showing a first embodiment of a pilot insertion process performed in a pilot insertion unit.
Figure 9 is a conceptual diagram of the CP addition process of the CP addition unit.
FIG. 10 is a flowchart illustrating a first embodiment of a signal design method that is immune to phase distortion performed at a receiver.
Figure 11a is a conceptual diagram of the 2-BASK demapping process performed in a 2-BASK symbol demapper.
Figure 11b is a conceptual diagram of the 4-BASK demapping process performed in a 4-BASK symbol demapper.
Figure 11c is a conceptual diagram illustrating the 8-BASK demapping process performed in an 8-BASK symbol demapper.
Figure 12 is a conceptual diagram showing the 8-PSK demapping process performed in an 8-PSK symbol demapper.
Figure 13 is a conceptual diagram illustrating the 16-QAM demapping process performed in a 16-QAM symbol demapper.
FIG. 14 is a flowchart illustrating a second embodiment of a signal design method that is immune to phase distortion performed at a transmitter.
Figure 15 is a conceptual diagram showing a second embodiment of a pilot insertion process performed in a pilot insertion unit.
FIG. 16 is a flowchart illustrating a second embodiment of a signal design method that is immune to phase distortion performed at a receiver.
Fig. 17 is a block diagram showing a second embodiment of a signal design device immune to phase distortion.
Fig. 18 is a flowchart showing a third embodiment of a signal design method that is immune to phase distortion performed at a transmitter.
FIG. 19 is a flowchart illustrating a third embodiment of a signal design method that is immune to phase distortion performed at a receiver.
FIG. 20 is a flowchart illustrating a fourth embodiment of a signal design method that is immune to phase distortion performed at a transmitter.
FIG. 21 is a flowchart illustrating a fourth embodiment of a signal design method that is immune to phase distortion performed at a receiver.
Fig. 22 is a block diagram showing a third embodiment of a signal design device immune to phase distortion.

The present disclosure may have various modifications and various embodiments, and specific embodiments are illustrated in the drawings and described in detail. However, this is not intended to limit the present disclosure to specific embodiments, but should be understood to include all modifications, equivalents, or substitutes included in the spirit and technical scope of the present disclosure.

The terms first, second, etc. may be used to describe various components, but the components should not be limited by the terms. The terms are only used to distinguish one component from another. For example, without departing from the scope of the present disclosure, the first component could be referred to as the second component, and similarly, the second component could also be referred to as the first component. The term and/or includes any combination of a plurality of related described items or any item among a plurality of related described items.

In embodiments of the present disclosure, “at least one of A and B” can mean “at least one of A or B” or “at least one of combinations of one or more of A and B.” Furthermore, in embodiments of the present disclosure, “at least one of A and B” can mean “at least one of A or B” or “at least one of combinations of one or more of A and B.”

When it is said that a component is "connected" or "connected" to another component, it should be understood that it may be directly connected or connected to that other component, but that there may be other components in between. On the other hand, when it is said that a component is "directly connected" or "directly connected" to another component, it should be understood that there are no other components in between.

The terminology used in this disclosure is only used to describe specific embodiments and is not intended to be limiting of the present disclosure. The singular expression includes the plural expression unless the context clearly indicates otherwise. In this disclosure, it should be understood that the terms "comprises" or "has" and the like are intended to specify that a feature, number, step, operation, component, part or combination thereof described in the specification is present, but do not exclude in advance the possibility of the presence or addition of one or more other features, numbers, steps, operations, components, parts or combinations thereof.

Unless otherwise defined, all terms used herein, including technical or scientific terms, have the same meaning as commonly understood by one of ordinary skill in the art to which this disclosure belongs. Terms defined in commonly used dictionaries should be interpreted as having a meaning consistent with the meaning they have in the context of the relevant art, and will not be interpreted in an idealized or overly formal sense unless expressly defined in this disclosure.

Hereinafter, with reference to the attached drawings, preferred embodiments of the present disclosure will be described in more detail. In order to facilitate an overall understanding in describing the present disclosure, the same reference numerals are used for the same components in the drawings, and redundant descriptions of the same components are omitted.

Figure 1 is a conceptual diagram illustrating a first embodiment of a communication system.

Referring to FIG. 1, a communication system (100) may include a plurality of communication nodes (110-1, 110-2, 110-3, 120-1, 120-2, 130-1, 130-2, 130-3, 130-4, 130-5, 130-6). Here, the communication system may be referred to as a “communication network.” Each of the plurality of communication nodes may support at least one communication protocol. For example, each of the plurality of communication nodes may support a communication protocol based on CDMA (code division multiple access), a communication protocol based on WCDMA (wideband CDMA), a communication protocol based on TDMA (time division multiple access), a communication protocol based on FDMA (frequency division multiple access), a communication protocol based on OFDM (orthogonal frequency division multiplexing), a communication protocol based on OFDMA (orthogonal frequency division multiple access), a communication protocol based on SC (single carrier)-FDMA, a communication protocol based on NOMA (non-orthogonal multiple access), a communication protocol based on SDMA (space division multiple access), etc. Each of the plurality of communication nodes may have the following structure.

FIG. 2 is a block diagram illustrating a first embodiment of a communication node constituting a communication system.

Referring to FIG. 2, a communication node (200) may include at least one processor (210), a memory (220), and a transceiver (230) that is connected to a network and performs communication. In addition, the communication node (200) may further include an input interface device (240), an output interface device (250), a storage device (260), etc. Each component included in the communication node (200) may be connected by a bus (270) and may communicate with each other. However, each component included in the communication node (200) may be connected through an individual interface or an individual bus centered around the processor (210), rather than a common bus (270). For example, the processor (210) may be connected to at least one of the memory (220), the transceiver (230), the input interface device (240), the output interface device (250), and the storage device (260) through a dedicated interface.

The processor (210) can execute a program command stored in at least one of the memory (220) and the storage device (260). The processor (210) may mean a central processing unit (CPU), a graphics processing unit (GPU), or a dedicated processor on which methods according to embodiments of the present disclosure are performed. Each of the memory (220) and the storage device (260) may be configured with at least one of a volatile storage medium and a nonvolatile storage medium. For example, the memory (220) may be configured with at least one of a read only memory (ROM) and a random access memory (RAM).

Referring again to FIG. 1, the communication system (100) may include a plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) and a plurality of user equipment (UEs) (130-1, 130-2, 130-3, 130-4, 130-5, 130-6). Each of the first base station (110-1), the second base station (110-2), and the third base station (110-3) may form a macro cell. Each of the fourth base station (120-1) and the fifth base station (120-2) may form a small cell. The fourth base station (120-1), the third UE (130-3), and the fourth UE (130-4) may be within the coverage of the first base station (110-1). The second UE (130-2), the fourth UE (130-4), and the fifth UE (130-5) may be within the coverage of the second base station (110-2). The fifth base station (120-2), the fourth UE (130-4), the fifth UE (130-5), and the sixth UE (130-6) may be within the coverage of the third base station (110-3). The first UE (130-1) may be within the coverage of the fourth base station (120-1). The sixth UE (130-6) may be within the coverage of the fifth base station (120-2).

Here, each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) may be referred to as a NodeB, an evolved NodeB, a base transceiver station (BTS), a radio base station, a radio transceiver, an access point, an access node, a road side unit (RSU), a digital unit (DU), a cloud digital unit (CDU), a radio remote head (RRH), a radio unit (RU), a transmission point (TP), a transmission and reception point (TRP), a relay node, etc. Each of the plurality of UEs (130-1, 130-2, 130-3, 130-4, 130-5, 130-6) may be referred to as a terminal, an access terminal, a mobile terminal, a station, a subscriber station, a mobile station, a portable subscriber station, a node, a device, etc.

Each of the plurality of communication nodes (110-1, 110-2, 110-3, 120-1, 120-2, 130-1, 130-2, 130-3, 130-4, 130-5, 130-6) can support cellular communication (e.g., long term evolution (LTE), advanced (LTE-A) specified in the 3rd generation partnership project (3GPP) standard, etc.). Each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) can operate in a different frequency band or can operate in the same frequency band. Each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) can be connected to each other via an ideal backhaul or a non-ideal backhaul, and can exchange information with each other via the ideal backhaul or the non-ideal backhaul. Each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) can be connected to a core network (not shown) via an ideal backhaul or a non-ideal backhaul. Each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) can transmit a signal received from the core network to the corresponding UE (130-1, 130-2, 130-3, 130-4, 130-5, 130-6), and can transmit a signal received from the corresponding UE (130-1, 130-2, 130-3, 130-4, 130-5, 130-6) to the core network.

Each of the multiple base stations (110-1, 110-2, 110-3, 120-1, 120-2) can support OFDMA-based downlink transmission and SC-FDMA-based uplink transmission. In addition, each of the plurality of base stations (110-1, 110-2, 110-3, 120-1, 120-2) can support MIMO (multiple input multiple output) transmission (e.g., single user (SU)-MIMO, multi user (MU)-MIMO, massive MIMO, etc.), CoMP (coordinated multipoint) transmission, carrier aggregation transmission, transmission in an unlicensed band, device to device (D2D) communication (or, ProSe (proximity services), etc.). Here, each of the plurality of UEs (130-1, 130-2, 130-3, 130-4, 130-5, 130-6 ...). The operation corresponding to 120-2) and the operation supported by the base station (110-1, 110-2, 110-3, 120-1, 120-2) can be performed.

Meanwhile, most conventional wireless communication systems can be based on CP(cyclic prefix)-OFDM(orthogonal frequency division multiplexing) transmission method, which can be robust to multipath fading and have excellent frequency efficiency. Such wireless communication systems can be, for example, 4G(fourth generation) LTE(long-term evolution) and 5G(fifth generation) NR(new radio). However, CP-OFDM transmission method has a high PAPR(peak-to-average power ratio), which can require high linearity of amplifier, which can increase power consumption.

Therefore, wireless communication systems can use SC (single carrier) transmission methods that have almost no or very low PAPR. However, SC transmission methods can be vulnerable to multipath fading and have lower spectral efficiency than CP-OFDM transmission methods, so they may be wireless transmission methods that are rarely adopted in wireless communication standard systems.

In addition, CP-OFDM transmission scheme may not be inherently robust to phase distortion caused by hardware impairments such as carrier frequency offset (CFO) and phase noise (PhN) and phase distortion caused by high-speed mobility such as Doppler shift or Doppler spread. Accordingly, wireless communication system may require application of techniques robust to phase distortion.

Recently, a FM(frequency-modulated)-OFDM transmission method based on OFDM, which has inherent characteristics of being robust to high PAPR problem and phase distortion, has been published in an academic journal. This FM-OFDM method can map data centrally symmetrically around DC(direct current) in the frequency domain. And, the FM-OFDM method can convert data centrally symmetrically mapped around DC into the time domain to generate a signal having only real components.

Afterwards, the FM-OFDM method can frequency modulate signals that only have real components, convert them into phases, and transmit them by attaching CPs. By doing so, the FM-OFDM method can make the PAPR zero and effectively eliminate phase distortions caused by FM performance.

However, the FM-OFDM transmission scheme may reduce the given spectral efficiency by more than half due to the centrally symmetric mapping of the same data in the frequency domain. In addition, since the FM-OFDM transmission scheme performs phase shift after performing FM in the time domain, the estimation performance may be poor due to the effects of phase distortion and multipath wireless channels when pilots for channel estimation are allocated in the frequency domain. In addition, the FM-OFDM transmission scheme may set the weights too low to ensure that the time-domain sample-by-sample result values of the phase-shifted FM performance are formed between [-π,π) or (-π,π).

This FM-OFDM transmission scheme may have degraded data restoration performance depending on low weight values. In addition, the FM-OFDM transmission scheme may have quantization errors depending on low weight values, which may degrade data restoration performance. To reduce the negative effects of these low weight values, the bit width of the ADC (analog-to-digital converter) can be set to a large value when implementing the FM-OFDM transmission scheme. As a result, the FM-OFDM transmission scheme increases implementation complexity, which may increase power consumption. In addition, since the FM-OFDM transmission scheme performs phase shift after performing FM in the time domain, there may be a problem with the orthogonality of OFDM. Accordingly, the FM-OFDM transmission scheme cannot allocate a data subcarrier using FM-OFDM directly adjacent to a data subcarrier using CP-OFDM in the frequency domain.

In other words, the FM-OFDM transmission scheme may not be compatible with the CP-OFDM transmission scheme (Frequency Division Multiplexing, FDM), and thus may not achieve the original spectral efficiency of OFDM. Therefore, in order to solve this problem, a mobile communication system may need to develop a wireless transmission scheme that does not reduce spectral efficiency by eliminating the central symmetric mapping of the same data, can effectively allocate pilots for channel estimation, and has high resistance to phase distortion that minimizes quantization error in the resulting values for each phase-shifted time-domain sample.

The present disclosure provides a signal design, a wireless transmission method, a new waveform and a communication device that are highly resistant to phase distortion, can efficiently allocate pilots, and can average out errors while minimizing quantization errors, without reducing frequency efficiency to the maximum extent possible to solve the problems of the prior art described above.

In the present disclosure, a wireless device may be referred to as a mobile station (MS). And, a device that transmits a signal to an MS or a device that receives a signal from an MS may be referred to as a transmission and reception point (TRP). A device that manages a TRP may be referred to as a base station (BS). An area managed by a BS may be referred to as a cell. Specific methods, procedures, and devices of the present disclosure may be described based on the configurations of such MS, TRP, BS, and cell.

In addition, in the present disclosure, a BS may control multiple TRPs, but may control only one TRP. In addition, multiple MSs or only one MS may exist within a cell controlled by the BS. The MS may perform uplink wireless transmission to the TRP. And, the TRP may perform downlink wireless transmission to the MS.

FIG. 3 is a block diagram showing a first embodiment of a signal design device immune to phase distortion.

Referring to FIG. 3, the transmitter may include a M-BASK (bipolar amplitude shift keying) symbol mapper (301), a pilot insertion unit (302), a phase encoder (303), a real phase converter (304), a CP adder (305), and/or a baseband filter (306). The receiver may include a baseband filter (311), a CP remover (312), a time-frequency converter (313), a frequency domain equalizer (315), a channel estimator (315), a frequency-time converter (316), a symbol extractor (317), a time-phase converter (318), a phase decoding unit (319), a symbol detector (320), and/or an M-BASK symbol demapper (321).

The transmitter can be a TRP, and the receiver can be a terminal. The transmitter can use a PR-SC (phase distortion-resistant single carrier) transmission scheme based on serrated encoding (i.e., a gear-shaped) encoding. Here, the PR-SC transmission scheme based on serrated encoding can have high resistance to phase distortion, and the PAPR can be zero.

FIG. 4 is a flowchart illustrating a first embodiment of a signal design method that is immune to phase distortion performed at a transmitter.

Referring to FIG. 4, the TRP can generate a source bit stream to be transmitted to the terminal (S401). Accordingly, the M-BASK symbol mapper of the TRP can perform M-BASK modulation on the generated source bit stream to generate a symbol stream or source symbol block composed of elements of n=0,1,…, _N1 having only integer components (S402). Here, M and _N1 can be positive integers. And, the M-BASK symbol mapper can be an example of a symbol mapper.

Figure 5a is a conceptual diagram showing the signal constellation of 2-BASK modulation.

Referring to Fig. 5a, the 2-BASK symbol mapper can receive one bit and output it by mapping, for example, the received bit 0 to -1 and the received bit 1 to 1. The absolute maximum size A ₂ of the output of the 2-BASK symbol mapper can be A ₂ =1.

Figure 5b is a conceptual diagram showing the signal constellation of 4-BASK modulation.

Referring to Fig. 5b, the 4-BASK symbol mapper can receive two bits and output them by mapping, for example, the received bit 00 to -1, the received bit 01 to -2, the received bit 10 to 1, and the received bit 11 to 2. The absolute maximum size A ₄ of the output of the 4-BASK symbol mapper can be A ₄ =2.

Figure 5c is a conceptual diagram showing the signal constellation of 8-BASK modulation.

Referring to Fig. 5c, the 8-BASK symbol mapper can receive three bits and output them by mapping, for example, the received bit 000 to -1, the received bit 001 to -2, the received bit 010 to -3, the received bit 011 to -4, the received bit 100 to 1, the received bit 101 to 2, the received bit 110 to 3, and the received bit 111 to 4. The absolute maximum size A ₈ of the output of the 8-BASK symbol mapper can be A ₈ =4.

As described above, the M-BASK mapping rule can be extended, and thus a detailed description thereof is omitted. Here, the integer signal constellation M-BASK mapping rules of the above bit/bits may be an example, and mapping rules of all other possible modulations may be included in the scope of the present disclosure. As an example, another modulation mapping rule may be an M-PSK (phase shift keying) modulation mapping rule. An M-PSK mapper may perform M-PSK modulation on a generated source bit string to generate a symbol string or symbol block composed of elements of n=0,1,…, _N1 having only a phase component.

Fig. 6a is a conceptual diagram showing a first embodiment of a signal constellation of 2-PSK modulation.

Referring to Figure 6a, the 2-PSK symbol mapper receives one bit and, as an example, receives bit 0. , the received bit is 1 can be output by mapping to . The absolute maximum size A ₂ of the phase component of the output of the 2-PSK symbol mapper excluding the π value can be A ₂ = 1. Here, 2-PSK can be BPSK (binary phase shift keying).

Figure 6b is a conceptual diagram showing a second embodiment of the signal constellation of 2-PSK modulation.

Referring to Figure 6b, the 2-PSK symbol mapper receives one bit and, as an example, receives bit 0. , the received bit is 1 can be output by mapping to . The absolute maximum size A ₂ of the phase component of the output of the 2-PSK symbol mapper excluding the π value can be A ₂ = 1/2.

Figure 6c is a conceptual diagram showing the signal constellation of 4-PSK modulation.

Referring to Figure 6c, the 4-PSK symbol mapper receives two bits, for example, the received bit 00. , the received bit 01 , the received bit 10 , the received bit 11 can be output by mapping to . The absolute maximum size A ₄ excluding the π value of the phase component of the output of the 4-PSK symbol mapper can be A ₄ = 3/4. Here, 4-PSK can be QPSK (quadrature phase shift keying) or 4-QAM (quadrature and amplitude modulation).

Figure 6d is a conceptual diagram showing the signal constellation of 8-PSK modulation.

Referring to Figure 6d, the 8-PSK symbol mapper receives three bits and, as an example, receives bit 000. , the received bit 001 , the received bit 010 , the received bit 011 , the received bit is 100 , the received bit is 101 , received bit 110 , the received bit 111 can be output by mapping to . The absolute maximum size A ₈ excluding the π value of the phase component of the output of the 8-PSK symbol mapper can be A ₈ = 1. This mapping rule and the value of A _M can be one example, and all other possible mapping rules and values of A _M can be included in the scope of the present disclosure. Other modulation mapping rules can be M-QAM modulation mapping rules.

The M-QAM mapper can perform M-QAM modulation on the generated source bit stream to generate a symbol stream or symbol block consisting of elements of n=0,1,…,N ₁ that have only phase components. Since 2-QAM is identical to the BPSK described above, its description can be omitted. Since 4-QAM is identical to the QPSK described above, its description can be omitted.

Figure 7a is a conceptual diagram showing the signal constellation of 4-QAM modulation.

Referring to Figure 7a, the 4-QAM symbol mapper receives two bits, for example, the received bit 00. , the received bit 01 , the received bit 10 , the received bit 11 can be output by mapping to . The absolute maximum size A ₄ excluding the π value of the phase component of the output of the 4-QAM symbol mapper can be A ₄ = 3/4. Here, 4-QAM can be QPSK or 4-PSK.

Figure 7b is a conceptual diagram showing the signal constellation of 16-QAM modulation.

Referring to Figure 7b, the 16-QAM symbol mapper receives four bits and, as an example, receives bit 0000. , the received bit is 0001 , the received bit 0101 , the received bit is 0100 , the received bit is 0010 , the received bit 0011 , the received bit 0111 , the received bit 0110 , the received bit is 1000 , the received bit is 1001 , received bit 1101 , received bit 1100 , the received bit is 1010 , the received bit 1011 , the received bit is 1111 , received bit 1110 can be output by mapping to . The absolute maximum size A ₁₆ excluding the π value of the phase component of the output of the 16-QAM symbol mapper can be A ₁₆ = 1.

The values of elements of n=0,1,…,N ₁ that have only the size component above are the element values after the real-to-phase converting process (in other words, The size component may be multiplied by the shape). In addition, the value of the absolute maximum size A _M of the output of the mapping rule and the symbol mapper above is only one example, and all other possible mapping rules and values of A _M may be included in the scope of the present disclosure. Here, M may be a positive integer. Other modulation mapping rules, such as GMSK (Gaussian filtered Minimum-Shift Keying), DPSK (differential phase-shift keying), etc., may also be included in the scope of the present disclosure. In addition, the receiver may perform demodulation and restoration in consideration of the characteristics of the modulation methods mentioned above.

Again, referring to FIG. 4, the pilot insertion unit can insert pilots (e.g., pilot 1 and pilot 2) before and after the source symbol block of the signal output from the M-BASK mapper (S403). The pilots inserted in this way can simply and effectively restore or equalize the nonlinear wireless channel within each symbol of the source symbol block at the receiver.

Figure 8 is a conceptual diagram showing a first embodiment of a pilot insertion process performed in a pilot insertion unit.

Referring to FIG. 8, the pilot insertion unit can insert pilot 1 in front of the source symbol block generated by the M-BASK mapper. Then, the pilot insertion unit can insert pilot 2 in the back of the source symbol block of N ₁ bits generated by the M-BASK mapper. Here, pilot 1 and pilot 2 can be P bits. Here, P can be a positive integer. At this time, pilot 1 and pilot 2 can be the same. Or, pilot 1 and pilot 2 can be different. Accordingly, the output x(n) of the pilot insertion unit can be composed of N-1 bits. Here, N can be a positive integer. And, N can be 2P+N _1. N can be 0, 1,…, N-1.

The pilot insertion unit can assign a power value of zero to the last time-domain samples of pilot 1 and the first time-domain samples of pilot 2 to minimize interference caused by multipath when performing estimation of a wireless channel and frequency-domain equalizing at a receiver. N ₁ can denote the length of a time-domain sample unit of a symbol block, and P can denote the length of the time-domain sample units of pilot 1 and pilot 2. Pilot 1 and pilot 2 have the same length, but may not be limited thereto. The pilot length can be set in consideration of the length of the symbol block, frequency efficiency, etc. In addition, the pilot insertion unit can configure only pilot 1 for reasons such as improved frequency efficiency. Alternatively, the pilot insertion unit can configure only pilot 2 for reasons such as improved frequency efficiency. The pilot insertion process as described above can be performed after performing a transitive encoding process.

Again, referring to FIG. 4, the phase encoding unit can perform phase encoding on the M-BASK modulated symbol block signal into which the pilot is inserted (S404). Here, the phase may be a sawtooth shape. At this time, the output q(n) of the phase encoding unit can be modeled as in the following mathematical expression 1 or mathematical expression 2. Here, in the case of applying the M-PSK modulation method mentioned above, the phase encoding unit can take only the value excluding π of the phase component for each constellation point and apply it to x(·) modeled in mathematical expression 1 or mathematical expression 2. In addition, in the case of applying the M-QAM modulation method mentioned above, the phase encoding unit can similarly take only the value excluding π of the phase component for each constellation point and apply the taken value to x(·) modeled by mathematical expression 1 or mathematical expression 2.

In mathematical expressions 1 and 2, n is a positive real number, and can be n=Bn'+n"=0,1,…,N-1. Here, n' is a positive real number. , and n" can be a positive real number such that 0≤n"<B. In equations 1 and 2, min(1,n') can be a function that compares the sizes of 1 and n' and selects the smaller one. can be a function that takes the integer part of y plus 1 when y is a positive real number with a decimal point, and takes the integer part of y as it is when y does not have a decimal point. B can be a cyclic period of the time domain sample unit. Here, B can be a positive real number.

The floating encoding unit may perform floating encoding including the pilot as in mathematical expression 1 as one embodiment. Alternatively, the floating encoding unit may perform floating encoding only on the source symbol block while leaving the pilot as in mathematical expression 2 as another embodiment. In this case, it may be efficient if P is constrained to be a multiple of B (for example, P=B, 2B, 3B,…). Of course, P may not be a multiple of B. For the convenience of all descriptions of the technology of the present disclosure, P may be set to P=B.

The forward encoding unit can perform forward encoding on a source symbol block without including a pilot as in mathematical expression 2. At this time, since the first sample to be forward encoded may be susceptible to hardware damage, the last sample(s) of the preceding pilot 1 known to both the transmitter and receiver can be taken as the first sample. In addition, the forward encoding unit can perform forward encoding in ascending order by applying x(0) or x(P) as the initial value in mathematical expressions 1 and 2. In contrast, the forward encoding unit can perform forward encoding in descending order by applying the last sample.

Such a rolling encoding may be a method of performing a new differential encoding for every cyclic period B of a rolling time-domain sample unit. In other words, the rolling encoding unit may divide a M-BASK modulated symbol block with a pilot inserted into it into N/B time-domain first partial symbol blocks according to mathematical expression 1. Then, the rolling encoding unit may perform differential encoding on each of the first partial symbol blocks. At this time, the rolling encoding unit may perform differential encoding after adding a first sample value of a previous first partial symbol block to the first sample of the first partial symbol blocks.

In contrast, the floating-point encoding unit can divide the source symbol block into N ₁ /B time domains from the M-BASK modulated symbol block into second partial symbol blocks according to Equation 2. Then, the floating-point encoding unit can perform differential encoding on each of the second partial symbol blocks. At this time, the floating-point encoding unit can perform differential encoding after adding the first sample value of the previous second partial symbol block to the first sample of the second partial symbol blocks. Here, the floating-point encoding can be accumulation linear encoding.

The real-to-phase converting unit can convert the phase-encoded signal into a phase value z(n) as in Equation 3 or Equation 4 through a real-to-phase converting process (S405). In the case of applying the M-PSK and M-QAM modulation methods mentioned above, the real-phase converting unit can follow the real-phase converting process in the case of applying the M-BASK modulation method. Here, Equation 3 may be performed on the entire phase-encoded signal. Equation 4 may be performed on the real-phase converting process on a portion corresponding to a source symbol block in the phase-encoded signal.

Here, n can be 0,1,…,N-1. Eun Yang's mistake It can be, can be. θ is a positive real number. It can be. α is a positive real number that can converge to 0.

Here, Eun Yang's mistake It can be, can be. θ is a positive real number. It can be. α is a positive real number that can converge to 0.

In Equations 3 and 4, the maximum value of q(n) can be A _M (B+1) due to the sawtooth encoding. The reciprocal of A _M (B+1), which is the maximum value of q(n), can be θ. If q(n) of time-domain sample n is weighted by θ, This could be it. For example, it exists in (-1,1) or [-1,1). , and if the positive α converges to 0, the ambiguity occurring in the demodulation process at the receiver can be eliminated. The above weighted value is much larger than the weighted value of the conventional FM-OFDM transmission method, and can significantly reduce the quantization error that causes performance degradation during actual implementation.

Here, if the transmitter performs a step-by-step encoding and real-phase conversion process on the source bits, it can be robust to various phase distortions (e.g., phase distortions due to carrier frequency offset (CFO), PhN, Doppler shift, Doppler spread, and others) while minimizing implementation-side quantization errors, etc.

When applying the M-BASK modulation method to the real phase-shifted signal z(n), the CP appended part can generate a CP-added signal z'(n) by adding CP (S406).

Figure 9 is a conceptual diagram of the CP addition process of the CP addition unit.

Referring to Fig. 9, the CP appending unit can generate a CP-added signal z'(n) by adding a CP when applying the modulation method of M-BASK to the real phase-shifted signal z(n). Here, the CP can be Nc bits, and the real phase-shifted signal can be N bits. Accordingly, the signal added with the CP can be N+N _C bits. N _C can be a positive real number.

Again, referring to FIG. 4, the TRP can perform digital baseband filtering by applying a size value A _{c,t corresponding to the transmission power for achieving a desired communication according to the call quality of the TRP and the terminal to the CP-added real phase-shifted signal and then passing the signal through a digital baseband filter (S407). Here, A c ,t} can be a real number. Through such digital baseband filtering, the TRP can band-pass filter the CP-added real phase-shifted signal by a desired bandwidth. Thereafter, the TRP can wirelessly transmit the band-pass-filtered CP-added real phase-shifted signal through the antenna. Here, the CP period can be identical to the pilot 2 period embedded in the symbol period. Alternatively, the CP period can be smaller than the pilot 2 period embedded in the symbol period.

In case of applying the above-mentioned M-PSK modulation method and M-QAM modulation method, TRP can perform CP addition process and baseband filtering process in the same way as in case of applying the modulation method of M-BASK. In addition, in case of applying the above-mentioned M-PSK modulation method, TRP can apply the size value A _c,t corresponding to the same transmission power to the symbols within the symbol block for each symbol block, as in case of applying the modulation method of M-BASK. In contrast, in case of applying the M-QAM modulation method, TRP can apply different size values A _c,t to each symbol within the symbol block for each symbol block, as shown in FIG. 7a and FIG. 7b. A signal transmitted wirelessly in TRP can pass through a wireless fading channel. A terminal can receive a signal passing through a wireless fading channel from TRP.

FIG. 10 is a flowchart illustrating a first embodiment of a signal design method that is immune to phase distortion performed at a receiver.

Referring to FIG. 10, the terminal can perform analog bandpass filtering for a desired band by applying a bandpass filter to a signal received from a TRP. In addition, the terminal can convert the received signal into a baseband signal and perform digital baseband filtering using a digital baseband filter for a desired amount (S1001). Through this process, the terminal can add AWGN (additive white Gaussian noise) to the received signal.

Afterwards, the terminal can remove the CP to remove the ISI (inter-symbol interference) or ICI (inter-carrier interference) from the complex signal (in other words, a signal having both real and complex components) that has passed through the baseband filter through the CP removal unit (S1002). The complex signal r(n) from which the CP has been removed in such a CP removal unit can be as shown in the following mathematical expression 5.

In equation 5, φ _l (n) is a real number. can be. f _c can be a carrier frequency, τ _l can be a delay value of channel path l, and β _l (n) can be a phase value of a channel coefficient. A _c,r can be a real number representing the received power. This received power can be a magnitude value reflecting the path loss, etc. when the magnitude value A _c,t corresponding to the transmission power in TRP is received at the terminal. φ _d,l (n) can be a Doppler component occurring in channel path l, φ _c (n) can be a CFO component occurring in channel path l, φ _p (n) can be a PhN component occurring in channel path l, and φ _m (n) is a phase value of the channel path l. Other hardware damage caused by may be a phase distortion component that may occur randomly. In Equation 5, w(n) may mean AWGN added to the terminal. In Equation 5, each of the phase components φ _d,l (n), φ _c (n), and φ _p (n) may generally have a very high correlation between values in adjacent n. φ _m (n) may have a random characteristic between values in adjacent n, but the influence of this component on the overall performance may not be that large.

In the present disclosure, TRP can perform on-the-fly encoding by taking these points into consideration. In other words, on-the-fly encoding can be performed at the transmitter so as to obtain the difference between values at adjacent n to remove Doppler, CFO, and phase components at the receiver, and to average the φ _m (n) component. On-the-fly decoding at the receiver related to this can be described in detail below.

The terminal can effectively estimate a wireless fading channel, convert a time domain signal into a frequency domain signal for recovering (or equalizing) the channel, estimate and recover the signal in the frequency domain, and convert the recovered signal back into a time domain signal.

To this end, the terminal may be equipped with a time-frequency conversion unit, a channel estimation unit, a frequency-domain equalization unit, a frequency-time conversion unit, etc. Here, the time-frequency conversion unit may perform a time-frequency conversion process for converting a time-domain signal from which a CP has been removed into a frequency-domain signal. In addition, the channel estimation unit may perform a channel estimation process for estimating a channel from a frequency-domain signal. The frequency-domain equalization unit may perform a frequency-domain equalization process for restoring a wireless channel using the frequency-domain signal and channel estimation. The frequency-time conversion unit may perform a frequency-time conversion process for converting a frequency-domain signal from which a wireless channel has been restored back into a time-domain signal.

The wireless channel environment experienced by the TRP and the terminal may only have a LoS (Line-of-Sight) path. Then, the time frequency conversion process, the channel estimation process, the frequency domain equalization process, and the frequency time conversion process may not be required in the terminal. The wireless channel environment experienced by the TRP and the terminal may also have an NLoS (Non-Line-of-Sight) path. Then, the terminal may cause the received signal to go through the processing process of the four functional blocks mentioned above in order to overcome the performance degradation due to the multipath effect as shown in Equation 5. The processing process of these functional blocks may be as follows.

First, the time-frequency conversion unit can convert a signal from the time domain to the frequency domain (S1003). This time-frequency conversion unit can extract a time-domain reception signal corresponding to a pilot component from a reception signal from which CP has been removed, as in mathematical expression 6.

The time-frequency transform unit can perform an N-point DFT (discrete Fourier transform) or FFT (fast Fourier transform) on the extracted signal of Equation 6 to transform it into a signal R _p (k) of a pilot component in the frequency domain as in Equation 7.

In mathematical expression 7, W _p (k) may mean a noise component, H _p (k) may mean a frequency response, and Y _p (k) may mean a frequency domain signal corresponding to a pilot component converted from a known time domain pilot signal y _p (n) expressed by mathematical expression 8 shown in Fig. 7 into a frequency domain, and may be as shown in mathematical expression 9.

In addition, the time-frequency converter can extract a time-domain reception signal corresponding to a symbol component from a reception signal from which CP has been removed, as in mathematical expression 10.

In mathematical expression 10, a time-domain transmission signal corresponding to a time-domain reception signal r _s (n) corresponding to a symbol component may be y _s (n). A time-frequency conversion unit may perform an N-point DFT or FFT on the extracted signal of mathematical expression 10 to convert it into a signal R _s (k) of a frequency-domain symbol component as in mathematical expression 11.

In mathematical expression 11, W _s (k) may denote a noise component, H _s (k) may denote a frequency response, and Y _s (k) may denote a frequency domain signal converted from the time domain symbol signal y _s (n) sent from the transmitter as shown in Fig. 3 into a frequency domain.

The channel estimation unit of the terminal can estimate the wireless channel in the frequency domain (S1004). This channel estimation unit uses Y _p (k) of Equation 9, which is known to both the transmitter and receiver, to estimate the frequency response of the wireless channel for the signal R _p (k) of the pilot component of Equation 7 in the frequency domain using least square (LS), minimum mean squared error (MMSE), discrete Fourier transform (DFT), or any other possible channel estimation method. can be obtained.

The frequency domain equalizer of the terminal can perform frequency domain equalization by compensating for the wireless channel in the frequency domain (S1005). The channel estimation unit performs a channel estimation process to compensate for the frequency response H _s (k) reflected in the symbol component shown in Equation 11. can be estimated. The frequency domain equalizer converts the time domain signal into the frequency domain signal as in Equations 7 and 11, so that H _p (k) in Equation 7 is a value almost similar to H _s (k) in Equation 11, in other words, Considering this, it is estimated as above. Estimated value of frequency response of mathematical expression 11 It can be assumed that.

And, the frequency domain equalizer can be equalized in all possible ways for Equation 11. However, the frequency domain equalizer can perform a cyclic shift on H _p (k) to make it almost similar to H _s (k) by performing a transformation different from Equations 10 to 11. In addition, the frequency domain equalizer can perform a transformation different from Equations 10 to 11 by rotating an arbitrary phase of H _p (k) to make it almost similar to H _s (k). Alternatively, the frequency domain equalizer can perform a transformation different from Equations 10 to 11 when converting H _p (k) from the time domain to the frequency domain to make it almost similar to H _s (k). In this way, Any possible method of doing so may be included within the scope of the present disclosure.

The frequency-time transform unit is a signal equalized in the frequency domain. By performing N-point IDFT (inverse discrete Fourier transform) or IFFT (inverse fast Fourier transform), the time domain signal as in Equation 12 is generated. can be converted into (S1006).

Here, φ may denote a phase error component that may occur during frequency domain equalization. In the above, for the convenience of explanation, it is assumed that the time synchronization is perfect, but this is generally not the case, and the scope of the present disclosure can be applied without loss of generality under such imperfect time synchronization conditions.

Meanwhile, it can be assumed that the terminal has performed TRP-based encoding on symbol blocks in a signal including a pilot as in Equation 2. In addition, the terminal can set P=B=2, _N1 =12, N=16. Then, the terminal can use the symbol extraction unit to convert the signal into the time domain of Equation 12. If symbol extraction is performed to extract only symbol components, it can be expressed as in mathematical expression 13 (S1007).

The time phase conversion part of the terminal extracts the symbol signal of mathematical expression 13. By performing time phase transformation on the signal, the time phase is transformed as in mathematical expression 14. can be output (S1008).

Here, n can be n=2,3,…,13. silver can be. n' can be 0≤n'≤7, and n" can be 0≤n"≤2.

The terminal's on-chip decoding unit converts the signal into the phase component of mathematical expression 14. By performing the on-chip decoding in the above, the on-chip decoded signal is obtained as in Equation 15. (i.e., the restored M-BASK symbol) can be estimated (S1009).

Such floating-point decoding may perform differential decoding anew for every cyclic period B of a floating-point time-domain sample unit. In other words, the floating-point decoding unit may divide a signal converted into a phase component of Equation 14 into N/B time-domain partial symbol blocks. Then, the floating-point decoding unit may perform differential decoding on each of the partial symbol blocks. At this time, the floating-point decoding unit may perform differential decoding after subtracting a first sample value of a previous first partial symbol block from the first sample of the first partial symbol blocks. Here, the floating-point decoding may be accumulation linear decoding.

The M-BASK symbol demapper of the terminal can perform M-BASK demapping on a signal decoded in a stationary decoding unit or on a signal detected in a symbol detection unit (S1010). The terminal can generate a recovered source bit stream through this demapping process.

Figure 11a is a conceptual diagram of the 2-BASK demapping process performed in a 2-BASK symbol demapper.

Referring to Figure 11a, the 2-BASK symbol demapper receives the signal If this decision bound is greater than or equal to 0, it can be judged as bit 1. In contrast, the 2-BASK symbol demapper If this judgment boundary is less than 0, it can be judged as 0.

Figure 11b is a conceptual diagram of the 4-BASK demapping process performed in a 4-BASK symbol demapper.

Referring to Figure 11b, the 4-BASK symbol demapper receives the signal If this judgment boundary is greater than or equal to 1.5, it can be judged as bit 11. In addition, the 4-BASK symbol demapper If this judgment boundary is greater than or equal to 0 and less than 1.5, it can be judged as bit 10. In addition, the 4-BASK symbol demapper If this judgment boundary is greater than or equal to -1.5 and less than 0, it can be judged as bit 00. The 4-BASK symbol demapper If it is less than this judgment boundary of -1.5, it can be judged as 01.

Figure 11c is a conceptual diagram illustrating the 8-BASK demapping process performed in an 8-BASK symbol demapper.

Referring to Figure 11c, the 8-BASK symbol demapper receives the signal If it is greater than or equal to the decision boundary of 3.5, it can be determined as bit 111, if it is greater than or equal to the decision boundary of 2.5 and less than 3.5, it can be determined as bit 110, if it is greater than or equal to the decision boundary of 1.5 and less than 2.5, it can be determined as bit 101, if it is greater than or equal to the decision boundary of 0 and less than 1.5, it can be determined as bit 100, if it is greater than or equal to the decision boundary of -1.5 and less than 0, it can be determined as bit 000, if it is greater than or equal to the decision boundary of -2.5 and less than -1.5, it can be determined as bit 001, if it is greater than or equal to the decision boundary of -3.5 and less than -2.5, it can be determined as bit 010, and if it is less than the decision boundary of -3.5, it can be determined as bit 011. Through the same process as above, the terminal can perform the M-BASK demapping process of M>4.

Figure 12 is a conceptual diagram showing the 8-PSK demapping process performed in an 8-PSK symbol demapper.

Referring to Figure 12, the 8-PSK symbol demapper receives the signal The symbol can be demapped by using the phase value between signal constellation points as the decision boundary.

Figure 13 is a conceptual diagram illustrating the 16-QAM demapping process performed in a 16-QAM symbol demapper.

Referring to Figure 13, the 16-QAM symbol demapper receives the signal The symbols can be demapped by using the straight lines on the real and imaginary axes between the signal constellation points as the decision boundaries.

In the above symbol extraction process, time phase conversion process, and floating-point decoding process, if it is assumed that the terminal has performed floating-point encoding on the symbol block including the pilot of Equation 1, the symbol extraction may not be performed first. Instead, the terminal may perform time phase conversion on the frequency-time converted signal and use it as an input to the floating-point decoding unit. Then, the floating-point decoding unit may perform floating-point decoding mentioned in Equation 15 on the time-phase converted signal and then perform a symbol extraction process of taking only the symbol components excluding the pilot components. Thereafter, the terminal may perform M-BASK symbol demapping on the extracted symbols.

Meanwhile, the symbol detector can obtain the interference component of Equation 14 by subtracting the estimated M-BASK symbol decoded temporally from the output signal of the frequency-time converter. At this time, the symbol detector can also obtain the interference component of Equation 14 by processing the estimated M-BASK symbol decoded temporally and subtracting it from the output signal of the frequency-time converter. The terminal can subtract the interference component calculated in this way from the output signal of the frequency-time converter again to remove the interference, and then perform the symbol extraction process, time phase conversion process, and temporal decoding process mentioned above once or repeatedly.

In addition, even if the positions of the symbol extractor and the time phase transformor are swapped, the consistency of the technology proposed in the present disclosure can be maintained. In other words, the time phase transformor can process the signal first, and then the symbol extractor can extract the symbol component excluding the pilot component. In the PR-SC transmission method based on the overlay encoding described with reference to the above Fig. 4, the transmitter can apply the M-PSK or M-QAM modulation method other than the M-BASK modulation method. Accordingly, the receiver can apply the M-PSK or M-QAM demapping process other than the M-BASK demapping process.

FIG. 14 is a flowchart illustrating a second embodiment of a signal design method that is immune to phase distortion performed at a transmitter.

Referring to Fig. 14, TRP can generate a source bit stream to be transmitted to a terminal (S1401). Accordingly, the M-BASK symbol mapper of TRP can perform M-BASK modulation on the generated source bit stream to generate a source symbol stream or source symbol block composed of elements of n=0,1,…,N ₁ having only integer components (S1402). Here, M and N ₁ can be positive integers.

Here, the integer signal constellation M-BASK mapping rules of the above bit/bits may be one example, and mapping rules of all other possible modulations may be included in the scope of the present disclosure. For example, another modulation mapping rule may be an M-PSK modulation mapping rule. The M-PSK mapper may perform M-PSK modulation on the generated source bit string to generate a source symbol string or source symbol block composed of elements of n=0,1,…,N ₁ having only a phase component. Another modulation mapping rule may be an M-QAM modulation mapping rule.

The M-QAM mapper can perform M-QAM modulation on the generated source bitstream to generate a source symbol stream or source symbol block consisting of elements of n=0,1,…, _N1 having only a phase component. Other modulation mapping rules such as GMSK and DPSK may also be included in the scope of the present disclosure. In addition, the receiver can perform demodulation and restoration by considering the characteristics of the modulation methods mentioned above.

The pilot insertion unit can insert a pilot into the middle of a source symbol block of a signal output from the M-BASK mapper (S1403). The pilot inserted in this way can simply and effectively restore or equalize a nonlinear wireless channel within each symbol of the symbol block at the receiver.

Figure 15 is a conceptual diagram showing a second embodiment of a pilot insertion process performed in a pilot insertion unit.

Referring to FIG. 15, the pilot insertion unit can insert a pilot into the middle of the source symbol blocks generated by the M-BASK mapper. In this way, the pilot insertion unit can insert a pilot into the middle of the source symbol blocks of N ₁ bits generated by the M-BASK mapper. Here, the pilot can be P bits. Here, P can be a positive integer. Accordingly, the output x(n) of the pilot insertion unit can be composed of N-1 bits. Here, N can be a positive integer. And, N can be P+2N _1. N can be 0,1,…,N-1.

The pilot insertion unit can assign a zero power value to the first and last time domain samples of the pilot to minimize interference caused by multipath when performing estimation of a wireless channel and frequency domain equalization at the receiver. N ₁ can mean the length of a time domain sample unit of a symbol block, and P can mean the length of a time domain sample unit of the pilot. The pilot length can be set by considering the length of the symbol block, frequency efficiency, etc.

Again, referring to FIG. 14, the step-by-step encoding unit can perform step-by-step encoding on the M-BASK modulated symbol block signal with the pilot inserted (S1404). At this time, the output q(n) of the step-by-step encoding unit can be modeled as in the following mathematical expression 16 or mathematical expression 17. Here, in the case of applying the M-PSK modulation method mentioned above, the step-by-step encoding unit can take only the value excluding π of the phase component for each constellation point and apply it to x(·) modeled in the mathematical expression 16 or mathematical expression 17. In addition, in the case of applying the M-QAM modulation method mentioned above, the step-by-step encoding unit can similarly take only the value excluding π of the phase component for each constellation point and apply the taken value to x(·) modeled by the mathematical expression 16 or mathematical expression 17.

In Equation 16, n is a positive real number, which can be n=Bn'+n"=0,1,…,N-1. Here, Eun Yang's mistake It can be, is a positive real number, 0≤n"<B.

In mathematical expression 17, n is a positive real number, and can be n=Bn'+n"=0,1,…,N-1. Here, n' is a positive real number. , and n" may be a positive real number such that 0≤n"<B.

In equations 16 and 17, min(1,n') can be a function that compares the sizes of 1 and n' and selects the smaller one. can be a function that takes the integer part of y plus 1 when y is a positive real number with a decimal point, and takes the integer part of y as it is when y does not have a decimal point. B can be a cyclic period of the time domain sample unit. Here, B can be a positive real number.

The floating encoding unit may perform floating encoding including the pilot as in Equation 16 as one embodiment. Alternatively, the floating encoding unit may perform floating encoding only on the source symbol blocks while leaving the pilot as in Equation 17 as another embodiment. In this case, it may be efficient if P is constrained to be a multiple of B (for example, P=B, 2B, 3B, ...). Of course, P may not be a multiple of B. For the convenience of all descriptions of the technology of the present disclosure, P may be set to P=2B.

In addition, when the overhead encoding unit performs overhead encoding without including a pilot as in Equation 17, the first sample to be overhead encoded may be susceptible to hardware damage, so the last sample(s) of the preceding pilot known to both the transmitter and receiver may be taken as the first sample.

Such a rolling encoding may be a method of performing a new differential encoding for every cyclic period B of a rolling time domain sample unit. In other words, the rolling encoding unit may divide a M-BASK modulated symbol block with a pilot inserted into it into N/B time domain first partial symbol blocks according to mathematical expression 16. Then, the rolling encoding unit may perform differential encoding on each of the first partial symbol blocks. At this time, the rolling encoding unit may perform differential encoding after adding a first sample value of a previous first partial symbol block to the first sample of the first partial symbol blocks.

In contrast, the floating-point encoding unit can divide the source symbol block into N ₁ /B time domains from the M-BASK modulated symbol block into second partial symbol blocks according to Equation 17. Then, the floating-point encoding unit can perform differential encoding on each of the second partial symbol blocks. At this time, the floating-point encoding unit can perform differential encoding after adding the first sample value of the previous second partial symbol block to the first sample of the second partial symbol blocks. Here, the floating-point encoding can be cumulative linear encoding.

Meanwhile, the output q(n) of the phase encoding unit can be modeled as in the following mathematical expression 18. Here, in the case of applying the M-PSK modulation method mentioned above, the phase encoding unit can take only the value excluding π of the phase component for each constellation point and apply it to x(·) modeled in mathematical expression 18. In addition, in the case of applying the M-QAM modulation method mentioned above, the phase encoding unit can similarly take only the value excluding π of the phase component for each constellation point and apply the taken value to x(·) modeled by mathematical expression 18.

In mathematical expression 18, n is a positive real number, and can be n=Bn'+n"=0,1,…,N-1. Here, n' is a positive real number. , and n" can be a positive real number such that 0≤n"<B. In mathematical expression 18, min(1,n') can be a function that compares the sizes of 1 and n' and selects the smaller one. can be a function that takes the integer part of y plus 1 when y is a positive real number with a decimal point, and takes the integer part of y as it is when y does not have a decimal point. B can be a cyclic period of a time-domain sample unit in the shape of a gear. Here, B can be a positive real number.

In another embodiment, the floating-point encoding unit can perform floating-point encoding only on symbol blocks while leaving the pilot as in Equation 18. In this case, it can be efficient if p is constrained to be a multiple of B (for example, P=B,2B,3B,...). Of course, P may not be a multiple of B. For the convenience of all descriptions of the technology of the present disclosure, P may be set to P=2B.

In contrast, the step-by-step encoding unit can divide the source symbol block into N ₁ /B time domains from the M-BASK modulated symbol block into second partial symbol blocks according to Equation 18. Then, the step-by-step encoding unit can perform differential encoding on each of the second partial symbol blocks. At this time, the step-by-step encoding unit can perform differential encoding after adding the first sample value of the previous second partial symbol block to the first sample of the second partial symbol blocks. Here, the step-by-step encoding can be cumulative linear encoding. Equations 17 and 18 can have a difference in the starting value when performing the second differential encoding.

The real-to-phase converting unit can convert a signal encoded in a time domain into a phase value z(n) as in Equation 19 or Equation 20 through a real-to-phase converting process (S1405). In the case of applying the above-mentioned M-PSK and M-QAM modulation methods, the real-to-phase converting unit can follow the real-phase converting process in the case of applying the modulation method of M-BASK.

Here, n can be 0,1,…,N-1. Eun Yang's mistake It can be, can be. θ is a positive real number. It can be. α is a positive real number that can converge to 0.

Here, Eun Yang's mistake It can be, can be. θ is a positive real number. It can be. α is a positive real number that can converge to 0.

In Equations 19 and 20, the maximum value of q(n) can be A _M (B+1) due to the gear-shaped encoding. The reciprocal of A _M (B+1), which is the maximum value of q(n), can be . If q(n) of time-domain sample n is weighted by θ, This could be it. For example, θ exists in (-1,1) or [-1,1). If the positive α converges to 0, the ambiguity occurring in the demodulation process at the receiver can be eliminated. The above weighted value is much larger than the weighted value of the conventional FM-OFDM transmission method, and can significantly reduce the quantization error that causes performance degradation in actual implementation.

Here, if the transmitter performs a phase-shift encoding and real-time phase conversion process on the source bits, it can be robust to various phase distortions (e.g., phase distortions due to CFO, PhN, Doppler shift, Doppler spread, etc.) while minimizing implementation-side quantization errors, etc.

When applying the modulation method of M-BASK to the real phase-shifted signal z(n), the CP appended unit can generate the CP-added signal z'(n) by adding the CP (S1406). The TRP can perform digital baseband filtering by applying a size value A _c,t corresponding to the transmission power for enabling a desired communication according to the call quality of the TRP and the terminal to the CP-added real phase-shifted signal and then passing the signal through a digital baseband filter (S1407). Here, A _c,t can be a real number. Through such digital baseband filtering, the TRP can band-pass filter the CP-added real phase-shifted signal by a desired bandwidth. Thereafter, the TRP can wirelessly transmit the band-pass-filtered CP-added real phase-shifted signal through the antenna. The signal wirelessly transmitted from the TRP can pass through a wireless fading channel. The terminal can receive the signal that has passed through the wireless fading channel from the TRP.

FIG. 16 is a flowchart illustrating a second embodiment of a signal design method that is immune to phase distortion performed at a receiver.

Referring to FIG. 16, the terminal can perform analog bandpass filtering for a desired band by applying a bandpass filter to a signal received from the TRP. In addition, the terminal can convert the received signal into a baseband signal and perform digital baseband filtering using a digital baseband filter for a desired amount (S1601). Through this process, the terminal can add AWGN to the received signal.

Afterwards, the terminal can remove CP to remove ISI or ICI from a complex signal (in other words, a signal having both real and complex components) that has passed through a baseband filter through a CP removal unit (S1602). The complex signal r(n) from which CP has been removed in such a CP removal unit can be as shown in mathematical expression 5.

In equation 5, φ _l (n) is a real number. can be. f _c can be a carrier frequency, τ _l can be a delay value of channel path l, and β _l (n) can be a phase value of a channel coefficient. A _c,r can be a real number representing the received power. This received power can be a magnitude value reflecting the path loss, etc. when the magnitude value A _c,t corresponding to the transmit power in TRP is received at the terminal. φ _d,l (n) can be a Doppler component occurring in channel path l, φ _c (n) can be a CFO component occurring in channel path l, φ _p (n) can be a PhN component occurring in channel path l, and φ _m (n) can be a component caused by other hardware damage occurring in channel path l. may be a phase distortion component that may occur randomly. In Equation 5, w(n) may mean AWGN added to the terminal. In Equation 5, each of the phase components φ _d,l (n), φ _c (n), and φ _p (n) may generally have a very high correlation between values in adjacent n. φ _m (n) may have a random characteristic between values in adjacent n, but the influence of this component on the overall performance may not be that large.

In the present disclosure, TRP can perform on-the-fly encoding by taking these points into consideration. In other words, on-the-fly encoding can be performed at the transmitter so as to obtain the difference between values at adjacent n to remove Doppler, CFO, and phase components at the receiver, and to average the φ _m (n) component. On-the-fly decoding at the receiver related to this can be described in detail below.

The terminal can effectively estimate a wireless fading channel, convert a time domain signal into a frequency domain signal for recovery (or equalization), estimate and recover in the frequency domain, and convert the recovered signal back into a time domain signal.

To this end, the terminal may be provided with a time-frequency conversion unit, a channel estimation unit, a frequency-domain equalization unit, a frequency-time conversion unit, etc. Here, the time-frequency conversion unit may perform a time-frequency conversion process for converting a time-domain signal from which a CP has been removed into a frequency-domain signal. In addition, the channel estimation unit may perform a channel estimation process for estimating a channel from a frequency-domain signal. The frequency-domain equalization unit may perform a frequency-domain equalization process for restoring a wireless channel using the frequency-domain signal and channel estimation. The frequency-time conversion unit may perform a frequency-time conversion process for converting the channel-restored frequency-domain signal back into a time-domain signal. The wireless channel environment experienced by the TRP and the terminal may only include a LoS path. Then, the time-frequency conversion process, the channel estimation process, the frequency-domain equalization process, and the frequency-time conversion process may not be required in the terminal.

The wireless channel environment experienced by the TRP and the terminal may also have an NLoS path. Then, the terminal may process the received signal through the processing of the four functional blocks mentioned above in order to overcome the performance degradation due to the multipath effect as shown in Equation 5. The processing of these functional blocks may be as follows.

First, the time-frequency conversion unit can convert a signal from the time domain to the frequency domain (S1603). This time-frequency conversion unit can extract a time-domain reception signal corresponding to a pilot component from a reception signal from which CP has been removed, as in mathematical expression 21.

The time-frequency transform unit can perform an N-point DFT or FFT on the extracted signal of Equation 21 to transform it into a signal R _p (k) of a pilot component in the frequency domain as in Equation 22.

In mathematical expression 22, W _p (k) may denote a noise component, H _p (k) may denote a frequency response, and Y _p (k) may denote a frequency domain pilot signal corresponding to a pilot component converted from a known time domain pilot signal y _p (n) expressed by mathematical expression 23 shown in FIG. 15 into a frequency domain, and may be as shown in mathematical expression 24.

In addition, the time-frequency conversion unit can extract a time-domain reception signal corresponding to a symbol component from a reception signal from which CP has been removed, as in mathematical expression 25.

In mathematical expression 25, the time domain transmission signal corresponding to the time domain reception signal r _s (n) corresponding to the symbol component may be y _s (n). The time-frequency conversion unit may perform an N-point DFT or FFT on the extracted signal of mathematical expression 25 to convert it into a signal R _s (k) of a frequency domain symbol component as in mathematical expression 26.

In mathematical expression 26, W _s (k) may denote a noise component, H _s (k) may denote a frequency response, and Y _s (k) may denote a frequency domain signal converted from the time domain symbol signal y _s (n) sent from the transmitter into the frequency domain.

The channel estimation unit of the terminal can estimate the wireless channel in the frequency domain (S1604). This channel estimation unit uses Y _p (k) of Equation 24, which is known to both the transmitter and receiver, to estimate the frequency response of the wireless channel for the signal R _p (k) of the pilot component of Equation 22 in the frequency domain using LS, MMSE, DFT or any other possible channel estimation methods. can be obtained.

The frequency domain equalizer of the terminal can perform frequency domain equalization by compensating for the wireless channel in the frequency domain (S1605). The channel estimation unit performs a channel estimation process to compensate for the frequency response H _s (k) reflected in the symbol component shown in Equation 26. can be estimated. The frequency domain equalizer converts the time domain signal into the frequency domain signal as in Equations 22 and 26, so that H _p (k) in Equation 22 is a value almost similar to H _s (k) in Equation 26, in other words, Considering this, it is estimated as above. The estimated value of the frequency response of Equation 26 It can be assumed that the frequency domain equalizer is can be equalized in all possible ways for Equation 26. However, the frequency domain equalizer can perform a transformation other than Equations 25 to 26 by cyclic shifting H _p (k) to make it almost similar to H _s (k). In addition, the frequency domain equalizer can perform a transformation other than Equations 25 to 26 by rotating an arbitrary phase of H _p (k) to make it almost similar to H _s (k). Alternatively, the frequency domain equalizer can perform a transformation other than Equations 25 to 26 when converting H _p (k) from the time domain to the frequency domain to make it almost similar to H _s (k). In this way, Any possible method of doing so may be included within the scope of the present disclosure.

The frequency-time transform unit is a signal equalized in the frequency domain. Perform N-point IDFT or IFFT on the time domain signal as in Equation 27. can be converted into (S1606).

Here, φ may denote a phase error component that may occur during frequency domain equalization. In the above, for the convenience of explanation, it is assumed that the time synchronization is perfect, but this is generally not the case, and the scope of the present disclosure can be applied without loss of generality under such imperfect time synchronization conditions.

Meanwhile, it can be assumed that the terminal has performed TRP-based encoding on symbol blocks in a signal including a pilot as in Equations 17 and 18. In addition, the terminal can set B=2, P=2B, _N1 =6, and N=16. Then, the terminal can use the symbol extraction unit to convert the signal into the time domain of Equation 27. If symbol extraction is performed to extract only symbol components, it can be expressed as in mathematical expression 28 (S1607).

The time phase conversion part of the terminal is the symbol extracted signal of mathematical expression 28. By performing time phase transformation on the signal, the time phase is transformed as in mathematical expression 29. can be output (S1608). Alternatively, the time phase conversion unit of the terminal can output the symbol extracted signal of mathematical expression 28. By performing time phase transformation on the signal, the time phase is transformed as in mathematical expression 30. can output.

Here, n can be n=2n'+n", and n=0,1,…,5. silver It can be. The terminal's on-board decoding unit converts the signal into the phase component of mathematical expression 29. By performing the on-chip decoding in the above, the on-chip decoded signal is obtained as in mathematical expression 31. (i.e., the restored M-BASK symbol) can be estimated (S1609). Alternatively, the terminal's on-board decoding unit can estimate the signal converted into the phase component of mathematical expression 30. By performing the on-chip decoding in the above, the on-chip decoded signal is obtained as in mathematical expression 32. (i.e., the restored M-BASK symbol) can be estimated.

The M-BASK symbol demapper of the terminal can perform M-BASK demapping on a signal decoded in the on-chip decoding unit or on a signal detected in the symbol detection unit (S1610). The terminal can generate a recovered bit string through this demapping process.

In the above symbol extraction process, time phase conversion process, and floating-point decoding process, if it is assumed that the terminal has performed floating-point encoding on the symbol block including the pilot of Equation 16, the symbol extraction may not be performed first. Instead, the terminal may perform time phase conversion on the frequency-time converted signal and use it as an input to the floating-point decoding unit. Then, the floating-point decoding unit may perform floating-point decoding mentioned in Equation 29 on the time-phase converted signal and then perform a symbol extraction process of taking only the symbol components excluding the pilot components. Thereafter, the terminal may perform M-BASK symbol demapping on the extracted symbols.

The symbol detector can obtain the interference components of Equations 29 and 30 by subtracting the estimated M-BASK symbol decoded temporally from the output signal of the frequency-time converter. At this time, the symbol detector can also obtain the interference components of Equations 29 and 30 by processing the estimated M-BASK symbol decoded temporally and subtracting it from the output signal of the frequency-time converter. The terminal can subtract the interference component calculated in this way from the output signal of the frequency-time converter again to remove the interference, and then perform the symbol extraction process, time phase conversion process, and temporal decoding process mentioned above once or repeatedly.

In addition, even if the positions of the symbol extractor and the time phase transformor are swapped, the consistency of the technology proposed in the present disclosure can be maintained. In other words, the time phase transformor can process the signal first, and then the symbol extractor can extract the symbol component excluding the pilot component. In the PR-SC transmission method based on the overlay encoding described with reference to the above Fig. 14, the transmitter can apply the M-PSK or M-QAM modulation method other than the M-BASK modulation method. Accordingly, the receiver can apply the M-PSK or M-QAM demapping process other than the M-BASK demapping process.

Figure 17 is a block diagram showing a second embodiment of a signal design device immune to phase distortion.

Referring to FIG. 17, the transmitter may include an M-BASK symbol mapper (1701), a pilot insertion unit (1702), a differential encoder (1703), a real phase converter (1704), a CP addition unit (1705), and/or a baseband filter (1706). The receiver may include a baseband filter (1711), a CP removal unit (1712), a time-frequency converter (1713), a frequency-domain equalizer (1715), a channel estimation unit (1715), a frequency-time converter (1716), a symbol extractor (1717), a time-phase converter (1718), a differential decoder (1719), a symbol detector (1720), and/or an M-BASK symbol demapper (1721).

The transmitter may be a TRP, and the receiver may be a terminal. The transmitter may use a phase distortion-resistant single carrier (PR-SC) transmission method based on differential encoding. Here, the PR-SC transmission method based on differential encoding may have high resistance to phase distortion, and the PAPR may be zero.

Fig. 18 is a flowchart showing a third embodiment of a signal design method that is immune to phase distortion performed at a transmitter.

Referring to FIG. 18, the TRP can generate a source bit stream to be transmitted to the terminal (S1801). Accordingly, the M-BASK symbol mapper of the TRP can generate a source symbol stream or a source symbol block composed of elements of n=0,1,…, _N1 having only integer components by performing M-BASK modulation on the generated source bit stream (S1802). Here, M and _N1 can be positive integers. Here, the above-mentioned bit/bit integer signal constellation M-BASK mapping rules can be one example, and mapping rules of all other possible modulations can be included in the scope of the present disclosure. As an example, another modulation mapping rule can be an M-PSK modulation mapping rule. The M-PSK mapper can generate a symbol stream or a symbol block composed of elements of n=0,1,…, _N1 having only phase components by performing M-PSK modulation on the generated source bit stream. Other modulation mapping rules may be M-QAM modulation mapping rules. The M-QAM mapper can perform M-QAM modulation on the generated source bitstream to generate a symbol stream or symbol block consisting of elements of n=0,1,…,N ₁ that have only phase components.

The pilot insertion unit can insert pilots (e.g., pilot 1 and pilot 2) before and after the symbol block of the signal output from the M-BASK mapper (S1803). The pilots inserted in this way can simply and effectively restore or equalize the nonlinear wireless channel within each symbol of the symbol block at the receiver.

When the pilot insertion unit performs estimation of a wireless channel and frequency domain equalization at the receiver, in order to minimize interference caused by multipath, the pilot insertion unit can assign zero to the last time domain samples of pilot 1 and the first time domain samples of pilot 2. N ₁ may denote the length of a time domain sample unit of a symbol block, and P may denote the length of the time domain sample units of pilot 1 and pilot 2. Pilot 1 and pilot 2 may have the same length, but may not be limited thereto. The pilot length may be set in consideration of the length of the symbol block, frequency efficiency, etc. In addition, the pilot insertion unit may configure only pilot 1 for reasons such as improved frequency efficiency. Alternatively, the pilot insertion unit may configure only pilot 2 for reasons such as improved frequency efficiency. The pilot insertion process as described above may be performed after performing a transitive encoding process.

The differential encoding unit can perform differential encoding on an M-BASK modulated symbol block signal with a pilot inserted (S1804). At this time, the output q(n) of the differential encoding unit can be modeled as in the following mathematical expression 33 or mathematical expression 34.

In mathematical expression 33, n is a positive real number, and n=0,1,…,N-1.

In mathematical expression 34, n can be a positive real number. Here, n' can be a positive real number. As an example, the differential encoding unit may perform the step-by-step encoding including the pilot as in mathematical expression 33. Alternatively, as another example, the differential encoding unit may perform the differential encoding only on the symbol block while leaving the pilot as in mathematical expression 43. In addition, when the differential encoding unit performs the differential encoding without including the pilot as in mathematical expression 34, the first sample to be differentially encoded may be poorly susceptible to hardware damage, so the last sample(s) of the preceding pilot 1 known to both the transmitter and receiver may be taken as the first sample. As described above, the differential encoding unit may not perform the differential encoding on all samples N ₁ of the corresponding symbol block, and may perform the differential encoding by resetting to a unit N ₁ smaller than N ₁ (for example, N ₁ =4N ₂ ) in order to reduce the quantization error mentioned above. However, since the differential encoding unit performs differential encoding in this way, there may be a burden of having to allocate known symbols because not every symbol that is reset is differentially encoded. To reduce this burden and improve channel estimation performance, the differential encoding unit can allocate known symbols as pilot symbols and utilize these distributed pilot symbols.

The real phase conversion unit can convert the differentially encoded signal into a phase value z(n) as in Equation 35 or Equation 36 through a real phase conversion process (S1805). In the case of applying the M-PSK and M-QAM modulation methods mentioned above, the real phase conversion unit can follow the real phase conversion process in the case of applying the M-BASK modulation method.

Here, n can be 0,1,…,N-1. Eun Yang's mistake It can be, can be. θ can be a positive real number.

Here, Eun Yang's mistake It can be, can be. θ can be a positive real number. If exists in (-1,1) or [-1,1), ambiguity occurring in the demodulation process at the receiver can be eliminated. Here, if the transmitter performs differential encoding and real-phase conversion process on the source bits, it can be robust to various phase distortions (e.g., phase distortions due to CFO, PhN, Doppler shift, Doppler spread, etc.) while minimizing implementation-side quantization errors, etc.

When applying the modulation method of M-BASK to the real phase-shifted signal z(n), the CP appended unit can generate the CP-added signal z'(n) by adding the CP (S1806). The TRP can perform digital baseband filtering by applying a size value A _c,t corresponding to the transmission power for enabling a desired communication according to the call quality of the TRP and the terminal to the CP-added real phase-shifted signal and then passing the signal through a digital baseband filter (S1807). Here, A _c,t can be a real number. Through such digital baseband filtering, the TRP can band-pass filter the CP-added real phase-shifted signal by a desired bandwidth. Thereafter, the TRP can wirelessly transmit the band-pass-filtered CP-added real phase-shifted signal through the antenna. Here, the CP period can be identical to the pilot 2 period embedded in the symbol period. Alternatively, the CP period can be smaller than the pilot 2 period embedded in the symbol period. A signal transmitted wirelessly from a TRP can pass through a wireless fading channel. A terminal can receive a signal that has passed through a wireless fading channel from a TRP.

FIG. 19 is a flowchart illustrating a third embodiment of a signal design method that is immune to phase distortion performed at a receiver.

Referring to FIG. 19, the terminal can perform analog bandpass filtering for a desired band by applying a bandpass filter to a signal received from the TRP. In addition, the terminal can convert the received signal into a baseband signal and perform digital baseband filtering using a digital baseband filter for a desired amount (S1901). Through this process, the terminal can add AWGN to the received signal.

Afterwards, the terminal can remove CP to remove ISI or ICI from a complex signal (in other words, a signal having both real and complex components) that has passed through a baseband filter through a CP removal unit (S1902). The complex signal r(n) from which CP has been removed in such a CP removal unit can be as shown in the following mathematical expression 37.

In equation 37, φ _l (n) is a real number. can be. f _c can be a carrier frequency, τ _l can be a delay value of channel path l, and β _l (n) can be a phase value of a channel coefficient. A _c,r can be a real number and can be a received power. This received power can be a magnitude value A _c,t corresponding to the transmit power in TRP and a magnitude value reflecting the path loss, etc. when received at the terminal. φ _d,l (n) can be a Doppler component occurring in channel path l, φ _c (n) can be a CFO component occurring in channel path l, φ _p (n) can be a PhN component occurring in channel path l, and φ _m (n) can be a phase distortion component that may occur randomly for n due to other hardware damage occurring in channel path l. In mathematical expression 37, w (n) can mean AWGN added to the terminal. In Equation 37, each of the phase components φ _d,l (n), φ _c (n), and φ _p (n) may generally have very high correlations between values in adjacent n. φ _m (n) may have random characteristics between values in adjacent n, but the influence of these components on the overall performance may not be that large.

In the present disclosure, TRP can perform differential encoding by considering these points. In other words, differential encoding can be performed at the transmitter so as to obtain the difference of values at adjacent n to remove Doppler, CFO, and phase components at the receiver, and to average the φ _m (n) component. Differential decoding at the receiver related to this can be described in detail below.

The terminal can effectively estimate a wireless fading channel, convert a time domain signal into a frequency domain signal for recovery (or equalization), estimate and recover in the frequency domain, and convert the recovered signal back into a time domain signal.

To this end, the terminal may be provided with a time-frequency conversion unit, a channel estimation unit, a frequency-domain equalization unit, a frequency-time conversion unit, etc. Here, the time-frequency conversion unit may perform a time-frequency conversion process for converting a time-domain signal from which a CP has been removed into a frequency-domain signal. In addition, the channel estimation unit may perform a channel estimation process for estimating a channel from a frequency-domain signal. The frequency-domain equalization unit may perform a frequency-domain equalization process for restoring a wireless channel using the frequency-domain signal and channel estimation. The frequency-time conversion unit may perform a frequency-time conversion process for converting the channel-restored frequency-domain signal back into a time-domain signal. The wireless channel environment experienced by the TRP and the terminal may only include a LoS path. Then, the time-frequency conversion process, the channel estimation process, the frequency-domain equalization process, and the frequency-time conversion process may not be required in the terminal.

The wireless channel environment experienced by the TRP and the terminal may also have an NLoS path. Then, the terminal may process the received signal through the processing of the four functional blocks mentioned above in order to overcome the performance degradation due to the multipath effect as shown in Equation 37. The processing of these functional blocks may be as follows.

First, the time-frequency conversion unit can convert a signal from the time domain to the frequency domain (S1903). This time-frequency conversion unit can extract a time-domain reception signal corresponding to a pilot component as in Equation 38 from a signal from which CP has been removed in Equation 37.

The time-frequency transform unit can perform N-point DFT or FFT on the extracted signal of Equation 38 to transform it into a signal R _p (k) of a pilot component in the frequency domain as in Equation 39.

In mathematical expression 39, W _p (k) may denote a noise component, H _p (k) may denote a frequency response, and Y _p (k) may denote a frequency domain signal corresponding to a pilot component converted from a known time domain pilot signal y _p (k) expressed by mathematical expression 40 into a frequency domain, and may be as in mathematical expression 41.

In addition, the time-frequency conversion unit can extract a time-domain reception signal corresponding to a symbol component from a reception signal from which CP has been removed, as in mathematical expression 42.

In mathematical expression 42, the time domain transmission signal corresponding to the time domain reception signal r _s (n) corresponding to the symbol component may be y _s (n). The time-frequency conversion unit may perform an N-point DFT or FFT on the extracted signal of mathematical expression 42 to convert it into a signal R _s (k) of a frequency domain symbol component such as mathematical expression 43.

In mathematical expression 43, W _s (k) may represent a noise component, H _s (k) may represent a frequency response, and Y _s (k) may represent a frequency domain signal that is a time domain symbol signal y _s (n) sent from a transmitter converted into a frequency domain.

The channel estimation unit of the terminal can estimate the wireless channel in the frequency domain (S1904). This channel estimation unit uses Y _p (k) of Equation 41, which is known to both the transmitter and receiver, to estimate the frequency response of the wireless channel for the signal R _p (k) of the pilot component of Equation 39 in the frequency domain using LS, MMSE, DFT or any other possible channel estimation methods. can be obtained.

The frequency domain equalizer of the terminal can perform frequency domain equalization by compensating for the wireless channel in the frequency domain (S1905). The channel estimation unit performs a channel estimation process to compensate for the frequency response H _s (k) reflected in the symbol component shown in Equation 43. can be estimated. The frequency domain equalizer converts the time domain signal into the frequency domain signal as in Equations 39 and 43, so that H _p (k) of Equation 43 is a value almost similar to H _s (k) of Equation 43, in other words, Considering this, it is estimated as above. The estimated value of the frequency response of mathematical expression 43 It can be assumed that the frequency domain equalizer is can be equalized in all possible ways for Equation 43. However, the frequency domain equalizer can perform a cyclic shift on H _p (k) to make it almost similar to H _s (k) by performing a transformation different from Equations 42 to 43. In addition, the frequency domain equalizer can perform a transformation different from Equations 42 to 43 by rotating an arbitrary phase of H _p (k) to make it almost similar to H _s (k). Alternatively, the frequency domain equalizer can perform a transformation different from Equations 42 to 43 when converting H _p (k) from the time domain to the frequency domain to make it almost similar to H _s (k). In this way, Any possible method of doing so may be included within the scope of the present disclosure.

The frequency-time transform unit is a signal equalized in the frequency domain. Perform N-point IDFT or IFFT on the time domain signal as in Equation 44. can be converted into (S1906).

Here, φ may denote a phase error component that may occur during frequency domain equalization. In the above, for the convenience of explanation, it is assumed that the time synchronization is perfect, but this is generally not the case, and the scope of the present disclosure can be applied without loss of generality under such imperfect time synchronization conditions.

Meanwhile, it can be assumed that the terminal has performed differential encoding on the symbol block in the TRP in the signal including the pilot as in Equation 35. In addition, the terminal can set P=B=2, _N1 =12, N=16. Then, the terminal uses the symbol extraction unit to convert the signal into the time domain of Equation 44. If symbol extraction is performed to extract only symbol components, it can be expressed as in mathematical expression 45 (S1907).

The time phase conversion part of the terminal extracts the symbol signal of mathematical expression 45. By performing time phase transformation on the signal, the time phase is transformed as in mathematical expression 46. can be printed (S1908).

Here, n can be n=2,3,…,13. The differential decoding unit of the terminal converts the signal into the phase component of mathematical expression 46. Differential decoding is performed in the differentially decoded signal as in Equation 47. (i.e., the restored M-BASK symbol) can be estimated (S1909).

The M-BASK symbol demapper of the terminal can perform M-BASK demapping on a signal differentially decoded in a differential decoding unit or on a signal symbol detected in a symbol detection unit (S1910). The terminal can generate a recovered bit string through this demapping process.

In the symbol extraction process, time phase conversion process, and differential decoding process, if it is assumed that the terminal has performed differential encoding on the symbol block including the pilot of Equation 33, the symbol extraction may not be performed first. Instead, the terminal may perform time phase conversion on the frequency-time converted signal and use it as an input of the differential decoding unit. Then, the differential decoding unit may perform differential decoding mentioned in Equation 47 on the time-phase converted signal and then perform a symbol extraction process of taking only the symbol components excluding the pilot components. Thereafter, the terminal may perform M-BASK symbol demapping on the extracted symbols.

Meanwhile, the symbol detection unit can obtain the interference component of mathematical expression 46 by subtracting the differentially decoded estimated M-BASK symbol from the output signal of the frequency-time conversion unit. At this time, the symbol detection unit can also obtain the interference component of mathematical expression 46 by processing the differentially decoded estimated M-BASK symbol and subtracting it from the output signal of the frequency-time conversion unit. The terminal can subtract the interference component calculated in this way from the output signal of the frequency-time conversion unit again to remove the interference, and then perform the symbol extraction process, time phase conversion process, and differential decoding process mentioned above once or repeatedly.

In addition, even if the positions of the symbol extractor and the time phase transformor are swapped, the consistency of the technology proposed in the present disclosure can be maintained. In other words, the time phase transformor can process the signal first, and then the symbol extractor can extract the symbol component excluding the pilot component. In the differential encoding-based PR-SC transmission method described with reference to the above Fig. 18, the transmitter can apply an M-PSK or M-QAM modulation scheme other than the M-BASK modulation scheme. Accordingly, the receiver can apply an M-PSK or M-QAM demapping process other than the M-BASK demapping process.

FIG. 20 is a flowchart illustrating a fourth embodiment of a signal design method that is immune to phase distortion performed at a transmitter.

Referring to Fig. 20, the TRP can generate a source bit stream to be transmitted to the terminal (S2001). Accordingly, the M-BASK symbol mapper of the TRP can perform M-BASK modulation on the generated source bit stream to generate a source symbol stream or source symbol block composed of elements of n=0,1,…,N ₁ having only integer components (S2002). Here, M and N ₁ can be positive integers.

Here, the integer signal constellation M-BASK mapping rules of the above bit/bits may be one example, and mapping rules of all other possible modulations may be included in the scope of the present disclosure. For example, another modulation mapping rule may be an M-PSK modulation mapping rule. An M-PSK mapper may perform M-PSK modulation on a generated source bit string to generate a symbol string or symbol block composed of elements of n=0,1,…,N ₁ having only a phase component. Another modulation mapping rule may be an M-QAM modulation mapping rule.

The M-QAM mapper can perform M-QAM modulation on the generated source bit stream to generate a symbol stream or symbol block composed of elements of n=0,1,…, _N1 having only a phase component. Other modulation mapping rules such as GMSK and DPSK may also be included in the scope of the present disclosure. In addition, the receiver can perform demodulation and restoration by considering the characteristics of the modulation methods mentioned above.

The pilot insertion unit can insert a pilot into the middle of a symbol block of a signal output from the M-BASK mapper (S2003). The pilot inserted in this way can simply and effectively restore or equalize a nonlinear wireless channel within each symbol of the symbol block at the receiver. The pilot insertion unit can insert a pilot into the middle of the symbol blocks generated from the M-BASK mapper. In this way, the pilot insertion unit can insert a pilot into the middle of the symbol blocks of N ₁ bits generated from the M-BASK mapper. Here, the pilot can be P bits. Here, P can be a positive integer. Accordingly, the output x(n) of the pilot insertion unit can be composed of N-1 bits. Here, N can be a positive integer. And, N can be P+2N _1. N can be 0,1,…,N-1.

The pilot insertion unit can assign a zero power value to the first and last time domain samples of the pilot to minimize interference caused by multipath when performing estimation of a wireless channel and frequency domain equalization at the receiver. N ₁ can mean the length of a time domain sample unit of a symbol block, and P can mean the length of a time domain sample unit of the pilot. The pilot length can be set by considering the length of the symbol block, frequency efficiency, etc.

The differential encoding unit can perform differential encoding on an M-BASK modulated symbol block signal with a pilot inserted (S2004). At this time, the output of the differential encoding unit can be modeled as in the following mathematical expression 48 or mathematical expression 49.

In mathematical expression 48, n is a positive real number, and can be n=0,1,…,N-1. Here, n' is a positive real number, and can be n'=N ₁ +P.

In mathematical expression 49, n can be a positive real number. Here, n' can be a positive real number such that n' = N ₁ + P. The differential encoding unit may perform differential encoding including a pilot as in mathematical expression 48 as an embodiment. Alternatively, the differential encoding unit may perform differential encoding only on symbol blocks while leaving the pilot as in mathematical expression 49 as another embodiment. In addition, when the differential encoding unit performs differential encoding without including a pilot as in mathematical expression 48, the first sample to be differentially encoded may be vulnerable to hardware damage, so the last sample(s) of the preceding pilot known to both the transmitter and the receiver may be taken as the first sample. In addition, when the differential encoding unit performs differential encoding without including a pilot as in mathematical expression 49, the first sample to be differentially encoded may be vulnerable to hardware damage, so the last sample(s) of the preceding pilot 1 known to both the transmitter and the receiver may be taken as the first sample. As described above, the differential encoding unit may not perform differential encoding on all samples N ₁ of the corresponding symbol block, and may perform differential encoding by resetting to a unit N ₁ smaller than N ₁ (for example, N ₁ =4N ₂ ) in order to reduce the quantization error mentioned above. However, if the differential encoding unit performs differential encoding in this way, there may be a burden of having to allocate known symbols because not every symbol that is reset is differentially encoded. In order to reduce this burden and improve channel estimation performance, the differential encoding unit may allocate known symbols as pilot symbols and may also utilize such distributed pilot symbols.

The real phase conversion unit can convert the differentially encoded signal into a phase value z(n) as in Equation 50 or Equation 51 through a real phase conversion process (S2005). In the case of applying the M-PSK and M-QAM modulation methods mentioned above, the real-phase conversion unit can follow the real-phase conversion process in the case of applying the M-BASK modulation method.

Here, n can be 0,1,…,N-1. Eun Yang's mistake It can be, can be. θ is a positive real number. It can be. α is a positive real number that can converge to 0.

Here, Eun Yang's mistake It can be, can be. θ is a positive real number. It can be. α is a positive real number that can converge to 0.

In equations 50 and 51 If it exists in (-1,1) or [-1,1), the ambiguity occurring in the demodulation process at the receiver can be eliminated. The above weighted value is much larger than the weighted value of the conventional FM-OFDM transmission method, and can significantly reduce the quantization error that causes performance degradation during actual implementation.

Here, if the transmitter performs differential encoding and real-phase conversion on the source bits, it can be robust to various phase distortions (e.g., phase distortions due to CFO, PhN, Doppler shift, Doppler spread, etc.) while minimizing implementation-side quantization errors, etc.

When applying the modulation method of M-BASK to the real phase-shifted signal z(n), the CP appended unit can add CP to generate the CP-added signal z'(n) (S2006). The TRP can perform digital baseband filtering by applying a size value A _c,t corresponding to the transmission power for enabling desired communication according to the call quality of the TRP and the terminal to the CP-added real phase-shifted signal and then passing the signal through a digital baseband filter (S2007). Here, A _c,t can be a real number. Through such digital baseband filtering, the TRP can band-pass filter the CP-added real phase-shifted signal by a desired bandwidth. Thereafter, the TRP can wirelessly transmit the band-pass-filtered CP-added real phase-shifted signal through the antenna. The signal wirelessly transmitted from the TRP can pass through a wireless fading channel. The terminal can receive the signal that has passed through the wireless fading channel from the TRP.

FIG. 21 is a flowchart illustrating a fourth embodiment of a signal design method that is immune to phase distortion performed at a receiver.

Referring to FIG. 21, the terminal can perform analog bandpass filtering for a desired band by applying a bandpass filter to a signal received from a TRP. In addition, the terminal can convert the received signal into a baseband signal and perform digital baseband filtering using a digital baseband filter for a desired amount (S2101). Through this process, the terminal can add AWGN to the received signal.

Afterwards, the terminal can remove CP to remove ISI or ICI from a complex signal (in other words, a signal having both real and complex components) that has passed through a baseband filter through a CP removal unit (S2102). The complex signal r(n) from which CP has been removed in such a CP removal unit can be as shown in mathematical expression 37.

In equation 37, φ _l (n) is a real number. can be. f _c can be a carrier frequency, τ _l can be a delay value of channel path l, and β _l (n) can be a phase value of a channel coefficient. A _c,r can be a real number representing received power. This received power can be a magnitude value reflecting the path loss, etc. when the magnitude value A _c,t corresponding to the transmit power in TRP is received at the terminal. φ _dl (n) can be a Doppler component occurring in channel path l, φ _c (n) can be a CFO component occurring in channel path l, φ _p (n) can be a PhN component occurring in channel path, and φ _m (n) can be a phase distortion component that may occur randomly for n due to other hardware damage occurring in channel path l. In mathematical expression 37, w (n) can mean AWGN added to the terminal. In Equation 37, each of the phase components φ _d,l (n), φ _c (n), and φ _p (n) may generally have very high correlations between values in adjacent n. φ _m (n) may have random characteristics between values in adjacent n, but the influence of these components on the overall performance may not be that large.

In the present disclosure, TRP can perform differential encoding by considering these points. In other words, differential encoding can be performed at the transmitter so as to obtain the difference of values at adjacent n to remove Doppler, CFO, and phase components at the receiver, and to average the φ _m (n) component. Differential decoding at the receiver related to this can be described in detail below.

The terminal can effectively estimate a wireless fading channel, convert a time domain signal into a frequency domain signal for recovery (or equalization), estimate and recover in the frequency domain, and convert the recovered signal back into a time domain signal.

To this end, the terminal may be provided with a time-frequency conversion unit, a channel estimation unit, a frequency-domain equalization unit, a frequency-time conversion unit, etc. Here, the time-frequency conversion unit may perform a time-frequency conversion process for converting a time-domain signal from which a CP has been removed into a frequency-domain signal. In addition, the channel estimation unit may perform a channel estimation process for estimating a channel from a frequency-domain signal. The frequency-domain equalization unit may perform a frequency-domain equalization process for restoring a wireless channel using the frequency-domain signal and channel estimation. The frequency-time conversion unit may perform a frequency-time conversion process for converting the channel-restored frequency-domain signal back into a time-domain signal. The wireless channel environment experienced by the TRP and the terminal may only include a LoS path. Then, the time-frequency conversion process, the channel estimation process, the frequency-domain equalization process, and the frequency-time conversion process may not be required in the terminal.

The wireless channel environment experienced by the TRP and the terminal may also have an NLoS path. Then, the terminal may process the received signal through the processing of the four functional blocks mentioned above in order to overcome the performance degradation due to the multipath effect as shown in Equation 37. The processing of these functional blocks may be as follows.

First, the time-frequency conversion unit can convert a signal from the time domain to the frequency domain (S2103). This time-frequency conversion unit can extract a time-domain reception signal corresponding to a pilot component from a reception signal from which CP has been removed, as in mathematical expression 52.

The time-frequency transform unit can perform N-point DFT or FFT on the extracted signal of Equation 52 to transform it into a signal Rp(k) of a pilot component in the frequency domain as in Equation 53.

In mathematical expression 53, W _p (k) may denote a noise component, H _p (k) may denote a frequency response, and Y _p (k) may denote a frequency domain signal corresponding to a pilot component converted from a known time domain pilot signal y _p (n) expressed by mathematical expression 54 into a frequency domain, and may be as in mathematical expression 55.

In addition, the time-frequency conversion unit can extract a time-domain reception signal corresponding to a symbol component from a reception signal from which CP has been removed, as in mathematical expression 56.

In mathematical expression 56, a time domain transmission signal corresponding to a time domain reception signal r _s (n) corresponding to a symbol component may be y _s (n). A time-frequency conversion unit may perform an N-point DFT or FFT on the extracted signal of mathematical expression 56 to convert it into a signal of a frequency domain symbol component such as mathematical expression 57.

In mathematical expression 57, W _s (k) may denote a noise component, H _s (k) may denote a frequency response, and Y _s (k) may denote a frequency domain signal converted from the time domain symbol signal y _s (n) sent from the transmitter into the frequency domain.

The channel estimation unit of the terminal can estimate the wireless channel in the frequency domain (S2104). This channel estimation unit uses Y _p (k) of Equation 55, which is known to both the transmitter and receiver, to estimate the frequency response of the wireless channel for the signal R _p (k) of the pilot component of Equation 53 in the frequency domain using LS, MMSE, DFT or any other possible channel estimation methods. can be obtained.

The frequency domain equalizer of the terminal can perform frequency domain equalization by compensating for the wireless channel in the frequency domain (S2105). The channel estimation unit performs a channel estimation process to compensate for the frequency response H _s (k) reflected in the symbol component shown in Equation 57. can be estimated. The frequency domain equalizer converts the time domain signal into the frequency domain signal as in Equations 53 and 55, so that H _p (k) in Equation 53 is a value almost similar to H _s (k) in Equation 55, in other words. Considering this, it is estimated as above. The estimated value of the frequency response of mathematical expression 55 It can be assumed that the frequency domain equalizer is can be equalized in all possible ways for Equation 55. However, the frequency domain equalizer can perform a cyclic shift on H _p (k) to make it almost similar to H _s (k) by performing a transformation different from Equations 54 to 55. In addition, the frequency domain equalizer can perform a transformation different from Equations 54 to 55 by rotating an arbitrary phase of H _p (k) to make it almost similar to H _s (k). Alternatively, the frequency domain equalizer can perform a transformation different from Equations 54 to 55 when converting H _p (k) from the time domain to the frequency domain to make it almost similar to H _s (k). In this way, Any possible method of doing so may be included within the scope of the present disclosure.

The frequency-time transform unit is a signal equalized in the frequency domain. Perform N-point IDFT or IFFT on the time domain signal as in Equation 58. can be converted into (S2106).

Here, φ may denote a phase error component that may occur during frequency domain equalization. In the above, for the convenience of explanation, it is assumed that the time synchronization is perfect, but this is generally not the case, and the scope of the present disclosure can be applied without loss of generality under such imperfect time synchronization conditions.

Meanwhile, it can be assumed that the terminal has performed TRP-based encoding on symbol blocks in a signal including a pilot as in Equation 49. In addition, the terminal can set B=2, P=2B, N ₁ =6, N=16. Then, the terminal can use the symbol extraction unit to convert the signal into the time domain of Equation 58. If symbol extraction is performed to extract only symbol components, it can be expressed as in mathematical expression 59 (S2107).

The time phase conversion part of the terminal extracts the symbol signal of mathematical expression 59. By performing time phase transformation on the signal, the time phase is transformed as in mathematical expression 60. can be output (S2108). Alternatively, the time phase conversion unit of the terminal can output the symbol extracted signal of mathematical expression 59. By performing time phase transformation on the signal, the time phase is transformed as in mathematical expression 61. can output.

The differential decoding unit of the terminal converts the signal into the phase component of mathematical expression 60. Differential decoding is performed in the differentially decoded signal as in mathematical expression 62. (i.e., the restored M-BASK symbol) can be estimated (S2109). Alternatively, the differential decoding unit of the terminal can estimate the signal converted into the phase component of mathematical expression 61. Differential decoding is performed in the differentially decoded signal as in mathematical expression 63. (i.e., the restored M-BASK symbol) can be estimated.

The M-BASK symbol demapper of the terminal can perform M-BASK demapping on a signal differentially decoded in a differential decoding unit or on a signal symbol detected in a symbol detection unit (S2110). The terminal can generate a recovered bit string through this demapping process.

In the above symbol extraction process, time phase conversion process, and differential decoding process, if it is assumed that the terminal has performed differential encoding on the symbol block including the pilot as in Equation 48, the symbol extraction may not be performed first. Instead, the terminal may perform time phase conversion on the frequency-time converted signal and use it as an input to the differential decoding unit. Then, the differential decoding unit may perform the differential decoding mentioned in Equation 59 on the time-phase converted signal and then perform a symbol extraction process that takes only the symbol components excluding the pilot components. Thereafter, the terminal may perform M-BASK symbol demapping on the extracted symbols.

The symbol detector can obtain the interference components of Equations 60 and 61 by subtracting the differentially decoded estimated M-BASK symbol from the output signal of the frequency-time converter. At this time, the symbol detector can also obtain the interference components of Equations 60 and 61 by processing the differentially decoded estimated M-BASK symbol and subtracting it from the output signal of the frequency-time converter. The terminal can subtract the interference component calculated in this way from the output signal of the frequency-time converter again to remove the interference, and then perform the symbol extraction process, time phase conversion process, and differential decoding process mentioned above once or repeatedly.

In addition, even if the positions of the symbol extractor and the time phase transformor are swapped, the consistency of the technology proposed in the present disclosure can be maintained. In other words, the time phase transformor can process the signal first, and then the symbol extractor can extract the symbol component excluding the pilot component. In the differential encoding-based PR-SC transmission method described with reference to the above Fig. 20, the transmitter can apply an M-PSK or M-QAM modulation scheme other than the M-BASK modulation scheme. Accordingly, the receiver can apply an M-PSK or M-QAM demapping process other than the M-BASK demapping process.

Fig. 22 is a block diagram showing a third embodiment of a signal design device immune to phase distortion.

Referring to FIG. 22, the transmitter may include a QAM symbol mapper (2201), a spreading part (2202), a centrally symmetric symbol mapper (2203), a subcarrier indexing part (2204), a time domain transform part (2205), a cumulative linear encoding part (2206), a real phase transform part (2207), a CP adding part (2208), and/or a baseband filter (2209). The receiver may include a baseband filter (2211), a CP removing part (2212), a time phase transform part (2213), a cumulative linear decoding part (2214), a frequency domain transform part (2215), a subcarrier deindexing part (2216), a centrally symmetric symbol demapper (2217), a despreading part (2218), a symbol detector part (2219), and/or a QAM symbol demapper (2220).

The transmitter may be a TRP, and the receiver may be a terminal. The transmitter and the receiver may use a PR-OFDM (phase distortion-resistant orthogonal frequency division multiplexing) wireless transmission method. Here, the PR-OFDM transmission method may have high resistance to phase distortion, and the PAPR may be zero.

The transmitter can generate a source bit stream to be transmitted to the receiver. The QAM symbol mapper can perform symbol mapping using QAM (Quadrature Amplitude Modulation) modulation method on the generated source bit stream to generate a complex signal block. Of course, the transmitter can perform symbol mapping using any type of modulation method other than QAM modulation method to generate a complex signal block.

FM-OFDM can have problems such as OOB (Out-Of-Band) problem, high PAPR problem, etc. To solve these problems of FM-OFDM, the spreading unit is used to spread the symbol-mapped signal sequence vector d (each element dk, k=0,1,…,M-1). The symbol sequence vector y can be generated by encoding and/or spreading with all possible kinds of spreading matrices S (M×M diagonal matrices and/or M×M off-diagonal matrices) as in Equation 63. In this way, the spreading unit can pass the symbol sequence vector as an input to the centrally symmetric symbol mapper.

A centrally symmetric symbol mapper can perform centrally symmetric symbol mapping in the frequency domain. To this end, the centrally symmetric symbol mapper can allocate complex signals of signal blocks generated in ascending order of subcarriers from DC (direct current) without including DC to an upper subcarrier group of available subcarriers centered on DC (direct current). In addition, the centrally symmetric symbol mapper can allocate conjugating signals of complex signals of signal blocks generated in ascending order of subcarriers from DC without including DC to a lower subcarrier group of available subcarriers centered on DC.

The subcarrier indexing unit can perform subcarrier indexing to transmit signal block information to be sent to a desired frequency band in the RF in the frequency domain. To this end, the subcarrier indexing unit can map signals allocated to a lower subcarrier group including DC as is to the first part of the input terminal of the time domain transform unit. In addition, the subcarrier indexing unit can map signals allocated to an upper subcarrier group as is to the last part of the input terminal of the time domain transform unit. The subcarrier indexing unit can map a zero value to other elements of the input terminal of the time domain transform unit.

The time domain transform unit can generate a signal in the inter-domain using a subcarrier-indexed N-point IDFT or IFFT. At this time, the generated signal can have only real components according to the characteristics of the centrally symmetric symbol mapping.

Next, the cumulative linear encoding unit can perform cumulative linear encoding on the generated time domain signal block. At this time, the cumulative linear encoding can be differential encoding or step-by-step encoding. The real phase transform unit can transform the cumulative linear encoded signal into a phase domain. And, the CP adding unit can add a CP to the signal transformed into the phase domain. The transmitter can upscale the baseband signal using a baseband filter and transmit it to the receiver. At this time, the transmitter can perform RF bandpass filtering.

A signal transmitted from a transmitter can be received by a receiver via a wireless channel. AWGN can be added to such a signal. The receiver can perform bandpass filtering on the received signal. And, the receiver can perform baseband filtering using a baseband filter. Thereafter, the receiver can perform a CP removal process using a CP remover. The receiver can perform time phase transformation using a time phase shifter. The receiver can perform cumulative linear decoding using a cumulative linear decoding unit. Here, the cumulative linear decoding can be differential decoding or pass-through decoding. The receiver can convert a time domain signal into a frequency domain signal using a frequency domain transform unit. The frequency domain transform unit can be an N-point DFT or FFT.

Next, the frequency domain transform unit can transfer the signal block that has undergone the N-point DFT or FFT process to the subcarrier deindexing unit. The subcarrier deindexing unit can perform a process corresponding to the reverse order of the process performed by the subcarrier indexing unit. For example, the subcarrier deindexing unit can move signals from element 0 to element N/2-1 of the output terminal of the N-point DFT or FFT to behind the signals from element N/2 to element N-1. In other words, the subcarrier deindexing unit can cyclically shift the received signal to the left by N/2. The subcarrier deindexing unit can transfer the subcarrier deindexed signal to a centrally symmetric symbol demapper.

The centrally symmetric symbol demapper can perform centrally symmetric symbol demapping by receiving a subcarrier-deindexed signal from a subcarrier deindexing unit. In other words, the centrally symmetric symbol mapper can perform conjugation on signals of a lower subcarrier group to which a signal mentioned at a transmitter is assigned, starting from element N/2+1 of an input terminal. In addition, the centrally symmetric symbol mapper can extract signals corresponding to a symbol block transmitted by combining signals of an upper subcarrier group to which a signal mentioned at a transmitter is assigned, starting from element N/2-1 of an input terminal, and conjugated signals corresponding to these signals.

The despreading unit can perform despreading on the extracted signal sequence vector r (each element d _k , k=0,1,…,M-1) as in mathematical expression 65 to solve problems such as OOB and high PAPR of FM-OFDM, and transmit the obtained symbol sequence vector d' to the QAM symbol demapper.

Here, S ^H denotes a Hermitian matrix of S , and S ^-1 may denote an inverse matrix of S. The symbol detector may detect a symbol in a signal including a pilot. Then, the QAM symbol demapper may perform QAM symbol demapping on the despread symbol sequence vector d' to restore the source bit sequence.

The operation of the method according to the embodiment of the present disclosure can be implemented as a computer-readable program or code on a computer-readable recording medium. The computer-readable recording medium includes all kinds of recording devices that store information that can be read by a computer system. In addition, the computer-readable recording medium can be distributed over network-connected computer systems so that the computer-readable program or code can be stored and executed in a distributed manner.

Additionally, the computer-readable recording medium may include hardware devices specifically configured to store and execute program instructions, such as ROM, RAM, flash memory, etc. The program instructions may include not only machine language codes produced by a compiler, but also high-level language codes that can be executed by the computer using an interpreter, etc.

While some aspects of the present disclosure have been described in the context of an apparatus, they may also represent a description of a corresponding method, wherein a block or device corresponds to a method step or a feature of a method step. Similarly, aspects described in the context of a method may also be described as a feature of a corresponding block or item or a corresponding device. Some or all of the method steps may be performed by (or using) a hardware device, such as, for example, a microprocessor, a programmable computer, or an electronic circuit. In some embodiments, at least one or more of the most significant method steps may be performed by such a device.

In embodiments, a programmable logic device (e.g., a field-programmable gate array) may be used to perform some or all of the functions of the methods described herein. In embodiments, a field-programmable gate array may operate in conjunction with a microprocessor to perform one of the methods described herein. In general, the methods are preferably performed by some hardware device.

Although the present disclosure has been described with reference to preferred embodiments thereof, it will be understood by those skilled in the art that various modifications and changes may be made to the present disclosure without departing from the spirit and scope of the present disclosure as set forth in the claims below.

Claims (20)

As a method of transmitter,
A step of generating a first symbol block by modulating a source bit string to be transmitted to a receiver;
A step of generating a second symbol block by inserting at least one pilot into the first symbol block;
A step of generating a third symbol block by cumulatively linearly encoding the second symbol block;
A step of performing a real phase transform on the third symbol block to generate a first signal converted into a phase value;
A step of generating a second signal by adding a CP (cyclic prefix) to the first signal; and
Comprising the step of wirelessly transmitting the second signal to the receiver,
Method of transmitter. In claim 1,
In the step of generating a first symbol block by modulating a source bit stream to be transmitted to the receiver, the transmitter generates the first symbol block from the source bit stream using any one of BASK (bipolar amplitude shift keying) modulation, PSK (phase shift keying) modulation, QAM (quadrature and amplitude modulation) modulation, GMSK (Gaussian filtered minimum-shift keying) modulation, or DPSK (differential phase-shift keying) modulation.
Method of transmitter. In claim 1,
In the step of inserting at least one pilot into the first symbol block, the transmitter inserts a first pilot into a preceding time domain of the first symbol block and inserts a second pilot into a succeeding time domain of the first symbol block.
Method of transmitter. In claim 3,
The last time domain samples of the first pilot and the first time domain samples of the second pilot have a power value of zero.
Method of transmitter. In claim 1,
In the step of inserting at least one pilot into the first symbol block, the transmitter inserts a third pilot in the middle of the first symbol block.
Method of transmitter. In claim 5,
The first and last time domain samples of the above third pilot have a power value of zero.
Method of transmitter. In claim 1,
The step of generating a third symbol block by accumulative linear encoding of the second symbol block is:
A step of dividing the second symbol block into first partial symbol blocks based on the cyclic period of the time domain sample unit; and
A step of generating a third symbol block that is cumulatively linearly encoded by performing differential encoding on each of the first partial symbol blocks,
Method of transmitter. In claim 7,
In the step of generating the third symbol block that is cumulatively linearly encoded by performing differential encoding on each of the first partial symbol blocks,
The transmitter performs differential encoding after adding the first sample value of the previous first partial symbol block to the first sample of each of the first partial symbol blocks.
Method of transmitter. In claim 1,
The step of generating a third symbol block by accumulative linear encoding of the second symbol block is:
A step of dividing the first symbol block within the second symbol block into second partial symbol blocks; and
A step of generating a third symbol block that is cumulatively linearly encoded by performing differential encoding on each of the second partial symbol blocks,
Method of transmitter. In claim 1,
The step of generating a third symbol block by accumulative linear encoding of the second symbol block is:
A step of generating a third symbol block that is cumulatively linearly encoded by differentially encoding the second symbol block,
Method of transmitter. As a method of receiver,
A step of converting a received signal from a transmitter into a baseband signal;
A step of removing CP from the above baseband signal;
A step of equalizing a wireless channel for a received signal from which the CP has been removed;
A step of performing time phase conversion on a received signal whose wireless channel is equalized;
A step of performing cumulative linear decoding on the above time phase converted signal; and
Comprising a step of demodulating the above accumulated linear decoded signal to generate a source bit stream,
Method of receiver. In claim 11,
The step of equalizing the wireless channel for the received signal from which the above CP has been removed is as follows:
A step of converting a time domain pilot signal of a received signal from which the CP has been removed into a frequency domain pilot signal;
A step of converting a time domain symbol signal of a received signal from which the CP has been removed into a frequency domain symbol signal;
A step of estimating a first wireless channel from the above frequency domain pilot signal;
a step of estimating a second wireless channel from the frequency domain symbol signal; and
Comprising a step of equalizing the second wireless channel using the first wireless channel,
Method of receiver. In claim 11,
The step of performing cumulative linear decoding on the above time phase converted signal is:
A step of dividing the time phase-converted signal into first partial symbol blocks based on the cyclic period of the time domain sample unit; and
A step of performing differential decoding on each of the first partial symbol blocks to generate the cumulative linear decoded signal,
Method of receiver. In claim 12,
The step of performing cumulative linear decoding on the above time phase converted signal is:
A step of extracting a symbol block of the above time phase transformed signal;
a step of dividing the above symbol block into second partial symbol blocks; and
A step of performing differential decoding on each of the second partial symbol blocks to generate the cumulative linear decoded signal,
Method of receiver. In claim 12,
The step of performing cumulative linear decoding on the above time phase converted signal is:
A step of extracting a symbol block from the above time phase converted signal;
comprising a step of differentially decoding the symbol block to generate the cumulative linearly decoded signal;
Method of receiver. In claim 11,
The step of demodulating the above-mentioned accumulated linear decoded signal to generate a source bit string is:
A step of demodulating the above accumulated linearly decoded signal to generate a first symbol block;
A step of generating a second symbol block by removing a pilot from the first symbol block; and
A step of generating the source bit string by performing symbol demapping on the second symbol block,
Method of receiver. As a transmitter,
Contains a processor,
The above processor is the transmitter,
Modulating a source bit string to be transmitted to a receiver to generate a first symbol block;
Generating a second symbol block by inserting at least one pilot into the first symbol block;
Generating a third symbol block by cumulatively linearly encoding the second symbol block;
Performing a real phase transform on the third symbol block to generate a first signal converted into a phase value;
Generating a second signal by adding a CP (cyclic prefix) to the first signal; and
causing the second signal to be wirelessly transmitted to the receiver;
transmitter. In claim 17,
In the step of inserting at least one pilot into the first symbol block, the processor causes the transmitter to insert a first pilot into a preceding time domain of the first symbol block and to insert a second pilot into a succeeding time domain of the first symbol block.
transmitter. In claim 17,
In the step of inserting at least one pilot into the first symbol block, the processor causes the transmitter to insert a third pilot in the middle of the first symbol block.
transmitter. In claim 17,
In the step of generating a third symbol block by accumulative linear encoding of the second symbol block, the processor causes the transmitter to:
Dividing the second symbol block into first partial symbol blocks based on the cyclic period of the time domain sample unit; and
Causing to generate the third symbol block which is cumulatively linearly encoded by performing differential encoding on each of the first partial symbol blocks.
transmitter.

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